JP2006179182A - Semiconductor device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce power and to increase an operating speed by evading the problem of leakage current caused by a subthreshold current during standby in a static memory operated by a low voltage such as about 1V power voltage, and to secure a voltage margin of the static memory decreased by the lowering of the power voltage. <P>SOLUTION: In a static memory cell composed of a MOS transistor having a cross-coupled and relatively high threshold voltage, a MOS transistor for controlling its power supply line voltage is disposed. To set a voltage difference between two storage nodes in an unselected state memory cell larger than that between two nodes when voltages corresponding to write information are applied from data pair lines DL and /DL to two nodes of a selected memory cell, the power supply line voltage control transistor is turned ON after a word line voltage is turned OFF to apply a high voltage VCH to the power supply line. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a semiconductor integrated circuit operating at a low voltage, and more particularly to a semiconductor integrated circuit having a static memory cell constituted by a MIS transistor or a MOS transistor (hereinafter simply abbreviated as a MOS or MOS transistor) as a memory cell. The present invention relates to a circuit suitable for high-speed and low-power (static random access memory).

  Since the breakdown voltage of a MOS transistor, which is a kind of field effect transistor having a gate insulating film, is reduced as it is miniaturized, its operating voltage must be lowered. Even in this case, in order to maintain high-speed operation, it is necessary to decrease the threshold voltage (VT) of the MOS transistor in accordance with the decrease in the operating voltage. This is because the operating speed is governed by the effective gate voltage of the MOS transistor, that is, the value obtained by subtracting VT from the operating voltage, and the larger this value, the higher the speed. However, in general, when VT is set to about 0.4 V or less, as is well known, a DC current called a subthreshold current that exponentially increases as VT decreases flows through a MOS transistor that should originally be cut off. . For this reason, in a semiconductor integrated circuit composed of a large number of MOS transistors, even if it is a CMOS circuit, the direct current increases remarkably. Therefore, it will be an essential problem in future semiconductor devices in which high speed, low power, and low voltage operation are important. That is, a subthreshold current is generated, and the entire chip becomes a large direct current. For this reason, VT of a transistor in the memory cell, particularly a cross-coupled transistor, cannot be set to about 0.4 V or less. However, the effective gate voltage becomes lower as the operating voltage decreases. For this reason, the operation margin (margin) of the memory cell is narrowed, the operation speed is lowered, or it is easily affected by manufacturing variations of VT.

  FIG. 2 shows a conventional memory cell and a waveform diagram for further explaining the above-mentioned problem.

  A CMOS type static memory (SRAM) is taken as an example of the memory cell. First, the memory cell is not selected, that is, the word line WL is at a low level such as 0V, the storage node N2 in the cell is at a high level such as 1V equal to the power supply voltage VCC, and the other storage node N1 is at a low level such as 0V. Let's consider the case of storing such information. Conventionally, since VT of all the transistors of the memory cell is 0.4 V or more, both the N channel MOS transistor QS2 and the P channel MOS transistor QC1 are non-conductive. This is because the gate-source voltage is 0 V in QS2 and QC1. Therefore, the current flowing through VCC is negligible. This is the reason why SRAM has low power. The voltage margin of this memory cell decreases as VCC-VT decreases. Therefore, as Vcc is lowered, VT must be lowered, but when VT is lowered to 0.4 V or less, subthreshold current begins to flow through the two transistors QS2 and QC1 which should be non-conductive. It increases with an exponential function as the value decreases. In general, VT varies due to variations in the manufacturing process, and the subthreshold current increases as the temperature rises. Therefore, in consideration of VT variation and an increase in junction temperature, this current further increases under worst conditions. Since this current flows through all the memory cells in the chip, a total current of about 10 mA or more may flow even in an SRAM of about 128 Kbits. This current is also the data holding current for the entire cell array. Considering that the data retention current of a normal SRAM using a MOS transistor having a relatively large threshold voltage so that a subthreshold current is not substantially generated can be reduced to 10 μA or less, this is a serious problem. Therefore, in terms of current, VT must be set to a relatively large value such as about 0.4V or more. Here, let us consider a case where VCC is lowered while VT is fixed at 0.5 V, for example. The demand for lowering VCC comes from the demand from lower power or the demand to drive with one battery in addition to the demand from lowering the breakdown voltage of the MOS transistor. For example, when the degree of miniaturization of the MOS transistor is such that the channel length is 0.5 μm or less or the thickness of the gate insulating film is 6 nm or less, the external power supply voltage VCC is reduced to about 1.5 to 1.0 V. However, since the transistor operates at a sufficiently high speed, VCC can be lowered to this level by giving priority to low power. However, when VCC is lowered, the voltage margin of the memory cell is significantly lowered. That is, the effective gate voltage of the conduction transistor QS1 is VCC-VT, and when VCC approaches VT, the effective gate voltage decreases and the rate of variation with respect to VT variation becomes significant. Also, the well-known soft error tolerance is reduced, and the margin for equivalent noise such as the threshold voltage difference (so-called offset voltage) between the cross-coupled transistors (QS1 and QS2, QC1 and QC2) in the memory cell is also reduced. descend.

  Even when a memory cell is selected, if VT is as high as 0.5 V and VCC is low, the speed is reduced and the operation margin is reduced. When VCC of 1V, for example, is applied to the word line WL, QT1 and QS1 become conductive, and a minute voltage change occurs in DL due to the current flowing therethrough and the load resistance (actually composed of a MOS transistor) connected to the data line DL. (0.2V) appears. On the other hand, QS2 is non-conductive because its gate voltage is sufficiently lower than VT, so that no voltage change appears on other data lines / DL. The storage information of the memory cell is discriminated and read out by the voltage polarity between the data pair lines. Here, the larger the voltage change appearing in DL, the more stable the discrimination. For this purpose, a current as large and constant as possible needs to flow through QS1 and QT1. Since the effective gate voltages of QS1 and QT1 are substantially equal to VCC-VT, this current decreases as VCC decreases, and is strongly influenced by variations in VT as described above.

From the above, in the conventional circuit and driving system, the direct current increases remarkably as the VCC decreases, the operating speed of the memory cell decreases or fluctuates, or the operating margin decreases. Therefore, the performance of the SRAM chip or the microprocessor chip incorporating the SRAM, for example, is significantly deteriorated as VCC decreases.
VLSI memory (Baifukan, published in November 1994), page 315 1995 Symposium on VLSI Circuits Digest of Techincal Papers, (1995), pp.75-76 JP 60-38796 A JP 02-108297 A

  An object of the present invention is to suppress an increase in subthreshold current and a decrease in voltage margin associated with a low voltage operation of a static memory cell composed of MOS transistors in a static memory or a semiconductor device incorporating a static memory.

  The purpose of the present invention is to provide a static memory cell in which MOS transistors are cross-coupled so that substantially no current flows between the drain and the source even when the gate and source voltages are equal. The voltage difference between the two storage nodes is greater than the voltage difference between the two storage nodes when the memory cell is selected and a voltage corresponding to write information is applied from the data pair line to the storage node of the memory cell. This is realized by controlling the voltage of at least one power supply line of the memory cell so as to increase. As a result, even when the main power supply voltage at the time of memory cell selection is low, the voltage between the two storage nodes in the memory cell can be sufficiently high, so that the memory cell can operate stably with a low power and a wide operation margin. .

  As is apparent from the embodiments described above, the present invention can realize a semiconductor device incorporating a high-speed static memory cell with a wide voltage margin without increasing current consumption even at low voltage operation.

  Embodiments of the present invention will be described below with reference to the drawings.

  FIGS. 1A to 1C are diagrams showing an embodiment of the present invention. FIG. 1A shows an example in which a transistor QP, which is a means for controlling the connection between the operating potential point VCH of the circuit and the memory cell, is added for each cell, and FIG. 1B shows an example in which QP is added for each row. c) shows an example in which QP is shared by all cells.

  For simplicity, the threshold voltage VT of all transistors in the memory cell is 0.5V. Therefore, if the gate and source voltages are substantially equal, the subthreshold current does not flow through the transistor. FIG. 1A is a conceptual diagram of the present invention which is the most basic. A P-channel MOS QP that functions as a switch is inserted between the common source of the P-channel MOSs QC1 and QC2, which are power supply nodes on the high potential side of the memory cell MC, and the power source VCH. A common source of N channel MOSs QS1 and QS2 which are power supply nodes on the low potential side of the memory cell MC is connected to a reference potential VSS (usually a ground potential of 0 V). The memory cell MC illustrated here is a static memory cell composed of MOS transistors whose gates and drains are cross-coupled to each other. More specifically, there are an inverter composed of QC1 and QS1, and an inverter composed of QC2 and QS2. A storage cell in which the output of one inverter is connected to the input of the other inverter, and one end of each of the storage nodes N1 and N2 respectively. Are composed of transfer MOS transistors QT1 and QT2. As for the operating environment of this memory cell MC, it is assumed that there is a power source having a sufficiently high current supply capability of the voltage VCC supplied from the outside, and the power source of VCH having a voltage higher than VCC is a power source having a low current supply capability. . A power supply with a low current supply capability has a problem that the voltage of the power supply temporarily decreases when a current exceeding the supply capability flows.

  When data is written to the memory cell from the data pair line, VCC is normally applied to one of the pair lines and 0 is applied to the other, with QP being non-conductive. At this time, if the word line voltage is set to VCC, VCC-VT lowered by VT of QT1 or QT2 is input to one of the storage nodes N1 and N2, and 0 is input to the other. However, the storage voltage (the difference voltage between the nodes N1 and N2) becomes VCC-VT if it remains as it is, for example, when VCC = 1V and VT = 0.5V, it becomes 0.5V, which is too low. Therefore, after writing, that is, after the word line voltage is turned off, QP is conducted, and a sufficiently high VCH (for example, 2 V) is applied to the common source. Then, the cross-coupled transistor in the memory cell is activated to act as a differential amplifier, and as a result, one of N1 and N2 is charged up to VCH and the other becomes zero. Eventually, the memory voltage increases from VCC-VT to VCH. The timing at which QP is turned on may not coincide with the word line selection timing, but in order to prevent unnecessary current from flowing from the power source VCH having a weak current supply capability to the data lines DL and / DL through the memory cell. It is desirable that the timing be after the word line voltage is turned off. Note that the write data of the nodes N1 and N2 in the period from when the word line is turned off to when QP is turned on is held by the parasitic capacitances of the nodes N1 and N2. If the QP is turned on and VCH is continuously supplied to the memory cell during the data holding period after activation of the memory cell or in the standby state as described above, the operation margin of the memory cell during this period is expanded. Also, the operation is fast and stable when reading from the memory cell. This is because the current driving capability of QS1 or QS2 is improved by the amount of increase in the gate voltage. Thus, since writing can be performed in a state where QP is non-conducting, any write data can be dynamically operated in the memory cell and can be written at low power and at high speed. If there is no QP and VCH is directly connected to the memory cell, or if QP is turned on in the middle of writing, a current flows for a long time through QP in the case of writing that reverses the stored information stored so far. Inconveniences such as high power and difficulty in reversing occur.

  FIG. 1B shows an embodiment in which QP is shared by a plurality of memory cells, and the memory cell becomes smaller as the number of transistors decreases. As described above, when QP1 is written in the non-conductive state as described above to MC1, 0 is applied to the gate of QC1 of MC1, for example, and VCC-VT (0.5V) is forcibly applied to the gate of QC2. It is done. Therefore, QC1 becomes conductive, and VCH that has been charged in the parasitic capacitance of the common power supply line PL1 is discharged to VCC-VT. At this time, the memory cell MC2 on the same word line WL1 is substantially subjected to a read operation, but the stored information of MC2 is not destroyed by the voltage drop of PL1 described above. The memory voltage of MC2 also only drops from the previous VCH to VCC-VT which is the voltage of PL1. The sensitivity of the differential amplifier in the memory cell is determined by the offset voltage of the paired transistors and the like, which is, for example, about 0.2 V or less, and information is not destroyed because VCC-VT is above this sensitivity. That is, if QP1 is turned on and VCH is again applied to PL1 after the writing is completed, the storage voltage of MC2 becomes VCH similarly to MC1. In this embodiment, only one power supply line corresponding to the selected word line needs to be charged to VCH. Since many other power supply lines (PL2 and the like) remain at VCH, charging operation does not occur even if the corresponding charging transistor (QP2) or the like is turned on. In other words, charging of the feeder line is localized and power consumption is reduced.

  FIG. 1C shows an embodiment in which the charge transistor QP is shared by all the memory cells, and the degree of integration is improved from the above-described example. However, in this case, since the voltage of all the power supply lines including the power supply lines of the memory cells on the non-selected word line side drops, the charge / discharge power for charging them to VCH increases or becomes slow. It is necessary to be careful. It should be noted that the area of the memory cell can be reduced by sharing the power supply line adjacent to the word line. For example, in the first (b), if PL1 and PL2 are made one common power supply line and controlled by one transistor, the number of memory cell wirings is effectively reduced.

  A circuit configuration in which a switch MOS transistor is provided at a power supply node on the high potential side of an SRAM memory cell is described in Japanese Patent Application Laid-Open Nos. 60-38796 and 02-108297. The connected potential is a power supply potential supplied to the device, and the idea is completely different from the present invention.

  In the following, a more specific embodiment including not only the write operation but also the read operation will be described mainly using FIG. 1B as an example.

  FIG. 3 is a circuit diagram of the embodiment of the present invention, FIG. 4 is a timing diagram at the time of reading, and FIG. 5 is a timing diagram at the time of writing.

  Taking a flip-flop type cell composed of a P-channel MOS transistor and an N-channel MOS transistor as an example of the memory cell, the threshold voltage VT of all the transistors in the cell is high enough to make the subthreshold current almost negligible, for example 0 .5V. For the sake of simplicity, a 4-bit cell array is taken up, and it is assumed that VCC = 1V and VCH = 2V on the assumption that the battery is driven by a VCC single power source. The features of the present invention are (1) switching the voltage of the cell power supply lines (PL1 and PL2) according to the operation timing of the cell. That is, when the cell is not selected, the information holding voltage (2 V in FIG. 3) is determined by the voltage applied from the power supply line of the cell, and the magnitude is larger than the write voltage written to the cell from the data line when the cell is selected. Thus, the voltage of the feeder line of the cell is controlled.

(2) The data lines (DL1, / DL1, DL2, / DL2) are based on the intermediate voltage (VCC / 2 = 0.5V) of the maximum voltage (VCC = 1V in FIG. 3) that the data line can take. Operate. This halves the charge / discharge power of the data line.

(3) The amplitude of the pulse voltage of the selected word line is larger than the maximum voltage that the data line can take. In order to eliminate the influence of the threshold voltage VT of the selection transistor connected to the word line, the amplitude of the pulse voltage is larger than the maximum voltage of the data line by a booster circuit in the chip (VCH). Set to Further, the current drive capability of QT1 and QT2 is improved by the boosted amount, and the speed is increased.

  Let us consider a case in which the SRAM portion incorporated in the microprocessor chip or the like, or the SRAM chip itself (hereinafter collectively referred to as SRAM) is inactivated by the SRAM activation signal CE. The main part in the SRAM is in a precharged state by a precharge signal ΦP. For example, the cell power supply lines (PL1, PL2) are precharged to a voltage VCH sufficiently boosted inside the chip based on the external power supply voltage (VCC). A decrease in the storage voltage in the cell due to a minute leak current in the cell is blocked by the compensation current from the P-MOSs QP1 and QP2, and thereby the storage state of each cell is maintained. Here, VCH is formed by the voltage conversion circuit VC2. VC2 is made by boosting VCC in the chip using a charge pump circuit for driving a capacitor, and its current driving capability is low accordingly. However, since the threshold voltage of the transistors in the cell is set to a sufficiently high value of 0.5 V or higher, the total leakage current of the cell can be sufficiently reduced to 10 μA or less at most even with a megabit class large capacity SRAM. Therefore, the compensation current can be supplied to all the memory cells from the VCH booster circuit. Details of the booster circuit are described in “VLSI LSI (Baifukan, published in November 1994), page 315”. The on-chip booster circuit that operates with a very low voltage power supply VCC of about 1V, which is the subject of this application, is described in “1995 Symposium on VLSI Circuits Digest of Techincal Papers, (1995), pp.75-76”. The The threshold voltage of the MOS transistor used in the booster circuit of this document is about 0.6 V. If a MOS transistor having a lower threshold voltage is used, a booster circuit that operates even at a lower power supply voltage VCC can be formed. it is conceivable that. When using a transistor with a low threshold voltage, attention must be paid to the above-mentioned subthreshold current. However, if the number of transistors is sufficient to form a booster circuit, the leakage current will not be practical enough. It is possible to do so. A circuit configuration in which a booster circuit for generating a boosted voltage by an external power supply is connected to a power supply node on the high potential side of an SRAM memory cell is described in JP-A-6-223581. Is connected to the potential of the booster circuit or an external power supply.

  In the precharge period inactivated by the SRAM activation signal CE, each data line (DL1, / DL1, DL2, / DL2 (in this specification, the inverted signal which is a pair of complementary signals is like / DL1). )) Is precharged to VCC / 2 by the precharge circuit PC. By doing so, the voltage amplitude of the data line is halved compared to the conventional VCC precharge, so that it is possible to halve the data line charge / discharge power which has been a problem at the time of simultaneous writing of multi-bit data. In this case, the VCC / 2 power source is formed from the VCC to the voltage conversion circuit VC1. Specifically, the circuit described in FIG. 4.60 on page 324 of the above-mentioned “VLSI LSI” can be used. Since VCC / 2 is produced inside the chip, the load current driving capability is generally low. Therefore, if one of the data pair lines is precharged directly from 0 to VCC / 2 with this VCC / 2 power supply during precharge, a sufficient charge current cannot be supplied and the level of VCC / 2 will fluctuate. Since the number of data pairs is usually as large as 64 or 128 or more, this variation is a particularly serious problem. Therefore, an amplifier AMP is provided for each data line. The role of the amplifier AMP is to amplify a minute differential voltage at the time of cell reading appearing on the data pair line to VCC at high speed. As a result, one of the data pair lines becomes 0 and the other becomes VCC. In the next precharge operation, QEQ is turned on, and the data pair lines are automatically balanced to VCC / 2. Therefore, it is not necessary to flow a large charging current from the VCC / 2 power source. When the precharge period is long, it is only necessary to pass a minute current that suppresses the level fluctuation of the data pair line due to the minute leakage current. Therefore, if AMP is used, a built-in VCC / 2 power supply circuit can be used.

  Hereinafter, a case where data is read from the memory cell will be described with reference to FIG. The SRAM is activated by the SRAM enable signal CE. When a certain word line, for example, WL1 is selected and a pulse of VCH is applied to WL1, all cells (MC1, MC2) on WL1 are activated. This word line selection signal pulse is generated by a row address decoder XDEC / driver DRV in response to a row address signal AX. If 0 and 2V (= VCH) are stored in the nodes N1 and N2 in the cell MC1, respectively, QT1 and QS1 become conductive, and the data line DL1 is gradually discharged toward 0V. On the other hand, since the gate voltages of QS2 and QC2 are almost zero, a current flows through QC2 and QT2, and the data line DL1 slightly rises from 0.5 V (= VCC / 2). Since it takes time for the minute differential voltage appearing on the data pair lines to be sufficiently large, a pulse is applied to the drive lines SP and SN of the amplifier AMP to amplify the data lines DL1 and / DL1 to 0 and 1 V, respectively, at high speed. To do. AMP does not determine the degree of SRAM integration and subthreshold current as much as the cell, so the size of the transistor in the AMP can be selected to be larger than that in the cell, and the threshold voltage can be lowered to about 0.2V, so high speed amplification. Is possible. Further, the AMP is activated when the memory cell is selected by the amplifier driving circuit SPG, and the drive lines SP and SN are kept at the same potential in the non-operating state (standby state), so that the subthreshold current becomes a problem. There is nothing. The AMP operates even when the data pair voltage is about 0.5V.

  The differential voltage of the data pair line amplified sufficiently large as described above is output onto the I / O pair line by the read selection signal ΦR1 of the column address decoder YDEC driver DRV, and passes through the read / write control circuit RWC for data Output DOUT. Here, QR1 and QR2 are circuits for converting the voltage of the data pair line into a current. When the threshold voltage of these transistors is 0.5V, the voltage of the data line DL1 is 0V, so that no current flows through the I / 0 line, while the current of / DL1 is 1V because it is 1V. Which large current flows can be detected in the RWC in the form of polarity discrimination of a differential current or a differential voltage (using the resistor R in the figure). If the threshold voltage VT of QR1 and QR2 is sufficiently low, for example, 0.2V, a minute voltage difference before amplification by the amplifier AMP can be detected, so that the speed is increased accordingly. This is because the mutual conductance increases as VT is lowered, and a larger current can flow.

  Let us examine the node voltage of the memory cell MC1 in detail in the above read operation (FIG. 4). If QP1 or QP2 is made conductive during this operation period, or if QP1 or QP2 is removed and VCH (2V) is forcibly applied to the feeder line PL1 or the like, a problem arises. When VCH is an external voltage having a large current driving capability, a large DC current continues to flow from all cells on PL1 during the period in which the voltage is applied to the word line, resulting in a large power. Alternatively, when the power supply voltage VCH boosted in the chip is used as in this embodiment, the current driving capability of the boosting circuit is insufficient, and the level of VCH is lowered. For this reason, the storage voltage of the non-selected cells on PL1 also decreases. Once the voltage on all the feeder lines is reduced, it takes a long time to restore the VCH level. This is because the total parasitic capacitance of the power supply line is large. This slows down the SRAM cycle time. Therefore, when the cell is inactive, all the power supply lines PL1 and PL2 are forcibly set to VCH (2 V) by the precharge signal ΦP, but each power supply line is disconnected from the VCH generation circuit during the activation period. Each feeder line is almost in a floating state, and the level of VCH is held in their parasitic capacitance. However, when the cell is activated (in this case, the read operation), the cell node N1 eventually becomes 0 and QC2 is strongly conducted. Since the sources of these transistors are connected to PL1, the floating voltage of PL1 drops from VCH, and as a result, N1 and N2 tend to be charged to a high level. However, N1 remains 0 because it is forcibly fixed to the voltage of DL1 (0V). On the other hand, since the voltage of the gate of QT2, that is, WL1, is 2V and / DL1 is also 1V, QT2 is conductive, and N2 continues to be charged by QC2 until the voltages of PL1 and N2 become equal, and eventually PL1 becomes 1V. Obviously, the feeder line discharged to 1V is localized. That is, it is only PL1 and PL2 corresponding to other non-selected word lines is not discharged but remains VCH. In an actual memory, there are a large number of power supply lines, and only one of them is discharged, so there is no wasteful charging / discharging power, and the power supply line to be charged by the built-in VCH generation circuit is localized as one. Therefore, the design of the VCH generation circuit becomes easy.

  As shown in FIG. 5, the write operation to the cell MC1 is performed by applying a differential voltage to the common I / O pair. An example of writing information opposite to the information stored in MC1 is taken as an example. Voltages of 1V and 0V are applied to the data pair lines DL1 and / DL1, respectively, and these voltages are applied as they are to the cell nodes N1 and N2. Therefore, the differential voltage 1V is written in the nodes N1 and N2. When the precharge operation is performed with ΦP after the word line WL1 is turned off from 2V to 0, the cell node differential voltage 1V is amplified to 2V by the amplification action of the cell itself. This is because the voltage of the cell power supply line PL1 is 2V. This high voltage becomes the subsequent information holding voltage. Here, also in the write operation, WL1 must be turned off, and the capacity to be charged by the VCH generation circuit must be minimized before applying VCH to PL1.

  The above operation does not destroy the stored information of other memory cells MC on the selected word line WL1, as described above. While the memory cell MC1 is being read or written and the data (data) is exchanged with the I / O pair line, a selection pulse is always applied to WL1 of MC2, so that the same reading as in FIG. The operation is performed between MC2 and data pair lines DL2 and / DL2. Therefore, even if PL1 changes from 2V to 1V, if 2V VCH is applied again, the two nodes in MC2 return to VCH, 0. Further, there is no adverse effect on the stored information of the memory cells MC3 and MC4 on the unselected word line WL2. This is because the sub-threshold current does not flow through the transistors in MC3 and MC4, and the junction leakage current is negligibly small even if it flows, so that the power supply line PL2 maintains the VCH during precharge. .

  The amplitude of the pulse voltage of the selected word line is VCC, and if the maximum value (VD) that can be taken by the data line is set to VCC-VT or less, the word voltage does not need to be generated from the boost power supply VCH, or when writing to a cell, etc. Further, since the influence of the threshold voltage VT of the transistors (QT1, QT2) in the memory cell can be eliminated, the design becomes easy. FIG. 6 shows an embodiment in that case, where (a) shows a circuit diagram and (b) shows a waveform diagram. FIG. 6 shows a portion related to the driving method of the memory cell in the entire SRAM of FIG. 3, and the difference when compared with FIG. 3 is the precharge circuit PC and the read / write circuit RWC. In this embodiment, the signal level of the word line is set to the reference potential of 0 V and the power supply potential VCC, and the power supply node on the high potential side of the memory cell when not selected is VCH (= 2 VCC), and the low potential side of the memory cell. The power supply node was set to 0 V which is the reference potential. Further, the precharge potential of the data line is set to a potential that is increased by at least the sensitivity voltage of the memory cell from the reference potential (0 V = VSS).

  The sensitivity voltage or sensitivity of the memory cell is the minimum potential difference necessary to invert the state of the memory cell that is a flip-flop circuit, for example, by the potential difference applied between DL and / DL in FIG. In order to make the potential difference applied between DL and / DL of the data line a sensitivity voltage, the precharge potential of the data line may be half or more of this sensitivity voltage. Since the sensitivity voltage of the normal memory cell is smaller than 0.2V, the reference voltage VR is set to 0.2V with a margin here, and the precharge potential of the data line is set to 0.2V. In other words, this embodiment is an example in which the maximum value of the voltage amplitude that can be taken by the data line is lowered to a low voltage VR which is lower than VT (0.5 V) and close to the sensitivity voltage of the memory cell itself. Since the voltage amplitude of the data line of the memory cell is minimized, the operation can be performed at higher speed and lower power accordingly. For this reason, the data pair lines can be precharged by a step-down power source composed of a comparator using QL1 and VR as reference voltages. The storage voltage of the memory cell can be made sufficiently high as VCH (2V).

  Hereinafter, the read operation will be described with reference to FIG. First, all the cell power supply lines are precharged to VCH (2 V) by the precharge signal ΦP. After the precharge is completed, a pulse with an amplitude VCC (1 V) is applied to the selected word line (WL1). Taking the case where the node N1 in the cell is 0 and N2 is VCH (2V) as an example, QT1 is conducted and the data line DL1 is discharged from 0.2V toward 0. The other data line / DL1 conducts QT2 but does not conduct QS2, so that the charge of the node N2 is distributed to / DL1 and the data line slightly rises from 0.2 V to υ. This increase is slight because the data line capacity is overwhelmingly 100 times larger than the in-cell node capacity. At this time, the voltage of N2 is discharged from 2V to υ. The differential voltage appearing on the data pair line in this way is taken out to the I / O pair line as cell read information through the read transistors QR1 and QR2. Here, in order to obtain a large gain, a P-channel MOS is used for QR1 and QR2. As a result of this series of operations, PL1 eventually drops to υ. However, when precharge operation starts next, υ is larger than the sensitivity of the cell itself, so that it is normally amplified to VCH by the cross-coupled P-channel MOSs QC1 and QC2. If the voltage difference υ between N2 and N1 is less than this sensitivity, it will not be amplified normally during precharging, and inverted information may be retained. As for the write operation, after applying a differential voltage of 0.2 V to one of the data pair lines selected from the I / O pair line and 0 to the other, PL1 is applied by the precharge operation as in the read operation. Is set to 2V.

  FIG. 7 shows an embodiment in which a large storage voltage is obtained by pulse driving the two power supply nodes on the high potential side and the low potential side of the memory cell during precharge. FIG. 7A is a circuit diagram thereof, and FIG. The waveform diagram is shown. FIG. 7 shows a portion related to the driving method of the memory cell in the entire SRAM of FIG. 3, and the difference when compared with FIG. 3 is that the potential on the low potential side of the memory cell is selected and not selected. That is, it can be changed accordingly. That is, the power supply node on the low potential side of the memory cell is set to the reference potential of 0 V when not selected, and to a potential that is lowered from VCC / 2 at least by the sensitivity voltage of the memory cell when selected. Further, in this embodiment, the signal level of the word line is set to the reference potential of 0 V and the power supply potential VCC, the precharge potential of the data line is VCC / 2, and the power supply node on the high potential side of the memory cell when not selected is VCH. (= 2 VCC).

  In FIG. 6, the precharge voltage of the data line is a low voltage in the vicinity of 0V, whereas this embodiment is characterized in that it is VCC / 2. Therefore, the read transistors QR1 and QR2 in FIG. 6 can be replaced with an N-channel MOS suitable for high-speed operation. In addition, since two types of amplifiers (QS1 and QS2, QC1 and QC2) in the cell are activated at the early stage of precharge, they are amplified at a higher speed. Assume now that VCH = 3V, VCC = 1.5V, VT = 0.5V, VR = 0.2V. Further, it is assumed that a VCC / 2 precharge circuit PC as shown in FIG. 3 is connected to each data pair line. During the precharge period, all data lines are set to 0.75V, power supply lines such as PL1 are set to 3V, and power supply lines connected to N-channel MOS in the cell such as PL1 'are set to 0V. This is because in the precharge period, QL2 is cut off by QL3, and PL1 'becomes 0 by QL4. The two nodes (N1, N2) in all the cells are 3V or 0 depending on the stored information. When the precharge is completed, PL1 is held at 3V. On the other hand, PL1 'starts to rise toward VCC by the resistor R', but when it becomes (VCC / 2) -VR, that is, 0.55V, the voltage limit created by the comparator and QL2 using (VCC / 2) -VR as a reference voltage. The circuit is activated and any further rise is suppressed. At the same time, for example, the node N1 on the low voltage side becomes 0.55V. Here, in order to suppress power consumption, R ′ is set to a relatively high resistance value, but a MOS transistor can be substituted. When the word voltage rises, since N1 is 3V and N1 is 0.55V, QT1 and QS1 are conducted and the data line DL1 is discharged. Since there is a difference of VR between DL1 and PL1 ', DL1 is eventually discharged to a voltage of 0.55V at PL1'. On the other hand, since QS2 is non-conductive, the charge at node N2 is discharged to / DL1 through QT2 as described above, and N2 and / DL1 become substantially equal voltage 0.75 V + υ. This differential voltage appearing on the data pair line is taken out to the I / O pair line through selection of a read circuit connected to each data line. Subsequent precharging amplifies the difference voltage of approximately 0.2V between the nodes N1 and N2 to 3V at high speed. When PL1 'becomes 0, N1 was 0.55V and N2 was slightly higher than 0.75V (υ) until then, so both QS1 and QS2 became conductive, and approximately 0.2V between N1 and N2 Is amplified by the cross-coupled amplifiers QS1 and QS2. This differential voltage is also amplified by the other cross-coupled amplifiers QC1 and QC2. In the example of FIG. 6, the amplifier composed of QS1 and QS2 is non-conductive at the initial stage of amplification in the cell at the start of precharge, and is a little slow because it is amplified only by the amplifier composed of QC1 and QC2. However, in this example, both amplifiers contribute to the amplification action at the initial stage of amplification, so that the speed is high. Obviously, the write operation may be performed by applying 0.75 V to one of the selected data pair lines and 0.55 V to the other according to the write data. Of course, PL1 'is controlled to be 0.55V at the time of cell selection as in the case of reading. In this example, since the voltage amplitude of the data line is as small as about 0.2, it can be driven by a VCC / 2 voltage generating circuit built in the chip. Therefore, the amplifier AMP of FIG. 3 can be removed in some cases, so that the chip becomes small. Further, since the data pair line always operates in the vicinity of VCC / 2, the stress voltage to the precharge circuit and read circuit (QR1, QR2) transistors on each data line is halved, so that the reliability is improved. Note that the precharge voltage of the data line does not necessarily have to be VCC / 2. Obviously, the precharge voltage of the data line may be set higher than the sensitivity of the in-cell amplifier with respect to the PL1 'voltage at the time of selection.

  Further, in this embodiment, an example is shown in which a power supply circuit composed of QL2, QL3 and a comparator is connected to each source drive line PL '(PL1', PL2 ') of the N-channel MOS in the cell. This is because the access time is increased by increasing the time for raising PL1 'to 0.55V. However, in order to reduce the chip area, this circuit can be shared with other feeder lines as shown in FIG. During the precharge period, the common power supply line PLC is always fixed to (VCC / 2) −VR by the common power supply circuit, but all the power supply lines (PL1 ′... PLn ′) are zero. Now, when PL1 'is selected, it is decoded by the external address, ΦX1 becomes 0, and PL1' is disconnected from the PLC. After that, / ΦP becomes VCC and PL1 is discharged to zero.

  FIG. 9 shows an example applied to a driving method in which the voltage of the data line takes a value near VCC at the time of reading. FIG. 9 shows a portion related to the driving method of the memory cell in the entire SRAM of FIG. 3, and the difference when compared with FIG. 3 is the precharge circuit PC and the read / write control circuit RWC. In this embodiment, the signal level of the word line is set to 0 V which is the reference potential and the power supply potential VCC, the power supply node on the high potential side of the memory cell when not selected is VCH (= 2 VCC), and the power supply on the low potential side of the memory cell. The node was set to a reference potential of 0V. The precharge potential of the data line was set to VCC.

  Each data line is connected to transistors QD1 and QD2 which are loads for the selected cell, and a transistor QEQ which balances the data line voltage. These circuits are the precharge circuit PC of this embodiment. The operation will be described below using the read operation timing of FIG.

  During the precharge period, the data pair line is VCC (1V) and PL1 is VCH (2V). Here, it is assumed that the data pair lines DL1, / DL1 are selected by the column address selection signal ΦRW1 (ΦRW1 is 1V to 0), the word line WL1 is selected, and a pulse of 0 to 1V is applied. If N2 is 2V, a direct current flows between QD1, QT1, and QS1, and as a result, a very small ratio voltage VS (about 0.2V) appears in DL1. On the other hand, N1 is almost 0, QS2 is non-conductive, and QT2 is non-conductive as apparent from the voltage relationship, so that no current flows through the paths QD2, QT2, and QS2. This is because the voltage of N1 slightly rises due to the ratio operation, because the size of the transistor in the cell is designed to be lower than VT. Therefore, a differential signal of only VS appears on the data pair line. Since this voltage is a ratio voltage, even if it does not go through a complicated readout circuit as shown in FIG. Here, since QS2 and QT2 are always non-conductive, the charge accumulated in the node of N2 is not lost. That is, the voltage of PL1 remains 2V. Therefore, even if the current drive capability of the VCH booster circuit built in the chip is not so much, no current flows through PL1 serving as a load. Therefore, in some cases, Qp1 can be removed and directly connected. However, this can only be done with a read operation. The fact that this is difficult in the write operation will be described with reference to FIG.

  When writing is performed so that 1V is applied to one DL1 of the data pair line from the I / O pair line and 0V is applied to the other one / DL1, the node N1 in the cell is changed from almost 0 to 0.5V. . This is because the threshold voltage of QT1 is 0.5V and the voltage of WL1 is 1V, so that the voltage dropped by the threshold voltage becomes the voltage of N1. On the other hand, N2 becomes 0 from the previous 2V. This is because QT2 conducts and N2 discharges to be equal to the voltage of / DL1. For this reason, QC1 is more conductive than QC2, and PL1 in the floating state is forcibly discharged from the data line to 0.5 V applied to N1. Therefore, PL1 must be charged again to 2V by the subsequent precharge.

  If the voltage drop of PL1 is large, the boosted voltage (VCH) generation circuit must supply charges corresponding to it to PL1, so the burden on the booster circuit becomes heavy. For this reason, the area of the VCH generating circuit itself increases and the power consumption increases. FIG. 12 shows a load circuit for suppressing the voltage drop to near VCC. In FIG. 12A, QP is made non-conductive in the time zone when the cell is selected, and QR is made conductive instead. Since the voltage of the power supply line is changed from VCH to VCC, one of the nodes in the cell (for example, N1) does not drop to 0.5V as shown in FIG. 11, and is suppressed to VCC (1V). In FIG. 12B, the precharge pulse / ΦP is removed and the design is simplified. An N channel MOS QR is used which has a threshold voltage of about 0.2 V, which is lower than that of other transistors. Since it is diode-connected, it conducts when the voltage of the feeder line is VCC-VT, that is, 0.8 V or less, so that a voltage drop below that can be prevented. That is, one of the cell nodes does not drop to 0.5V as shown in FIG. This transistor QR also prevents the voltage level of PL1, which is in a floating state, from being lowered too much by the diffusion layer leakage current in the cell and expands the cell voltage margin when Qp is off for a long time. To do.

  Assuming voltage application in FIGS. 10 and 11, the word lines WL1 and WL2 and the feed lines PL1 and PL2 are orthogonal to each other as shown in FIG. 13 in addition to the configuration in which the word lines and the feed lines are installed in parallel as shown in FIG. The configuration placed in can be taken. For example, when cells on WL1 are read, all the cells perform the same operation as in FIG. 10, and therefore the voltage (VCH) levels of all the feeder lines do not change. However, in the write operation, only the power supply line belonging to the selected data pair line changes. For example, if a pulse voltage of a combination of 1V and 0 corresponding to the write information is applied to the data pair lines DL1, / DL1 (not shown in the figure), the cell MC1 operates in the same manner as in FIG. It drops from 2V to 0.5V. Since the cell MC2 performs the same operation as in FIG. 10, the voltage VCH of PL2 does not change. Whether the mutual arrangement of the word lines and the feeder lines is parallel or orthogonal depends on the layout and area of the cell. In FIG. 9, since the feeder line and the data pair line cross each other, there is a disadvantage that the wiring layer must be laid out in different wiring layers, but there is an advantage of low noise. For example, consider a case where a large voltage change occurs in PL1 because a pulse is applied to WL1 and cell MC1 is written. At this time, since the read operation is effectively performed on the cell MC2, the signal appears on the data pair lines DL2 and / DL2. Since this signal is very small, the operation of MC2 is strongly affected by noise and easily affected. However, since the data pair line is orthogonal to PL1, the noise generated by the voltage change of PL1 through the coupling capacitance is canceled on the data pair line. FIG. 13 is contrary to FIG. 9 in terms of interests. For example, differential noise is generated in adjacent data pair lines (DL2, / DL2) due to voltage fluctuation of PL1. However, in this case, as is well known in dynamic memory, noise can be canceled by crossing the pair of data along the way.

  In the above embodiments, VCH has been premised on being generated from a power source obtained by boosting VCC in the chip. This is to realize a VCC single power supply operation that is easy for the user to use. However, in some cases, VCH may be the chip external power supply itself. For example, as shown in FIG. 14, the case of two external power sources (VCC1 and VCC2) can be considered. The chip includes an input / output interface circuit INTF and a core CORE such as a static memory SRAM and an arithmetic circuit (for example, a microprocessor MPU). INTF operates a relatively large element at a relatively high voltage (VCC1) to ensure the existing logic interface level. On the other hand, since CORE determines chip performance (speed, power) and chip area, the main part of this part is improved in performance by using fine elements that operate at a low voltage (VCC2). The elements in CORE are generally finer than the elements in INTF. In such a chip, VCC1 may be regarded as VCH in the previous embodiments. In this way, the entire chip operates with two power supplies, but problems such as output level fluctuations associated with the internal power supply operations are eliminated, and the design is facilitated. FIG. 15 shows an application example to a chip in which FIG. 14 is realized by a single power source. In a chip in which the main part of CORE is operated by an internal power supply (VCC2) obtained by stepping down an external single power supply (VCC1), VCC1 may be regarded as VCH in the above embodiments.

  In the above embodiment, the memory cell is assumed to be a CMOS type. However, since the present invention applies the differential amplifier function in the memory cell, there is at least one latch type amplifier that is cross-coupled in the memory cell. That's fine. Instead of the P-channel MOS (QC1, QC2), a well-known high resistance polysilicon load may be used. This is because the nodes N1 and N2 can be lifted toward VCH, so that they can eventually be amplified by cross-coupled N-channel MOSs (QS1 and QS2). Further, the VT of the N-channel transfer transistors QT1 and QT2 having a transfer (transfer) function in the memory cell is lower than the VT of other transistors in the memory cell, and may be 0.2, for example. The effective gate voltage of QT1 and QT2 is increased by the amount that VT is lowered at the time of selection, and the drive current is increased to enable high-speed operation. However, since the subthreshold current flows through QT1 or QT2 when not selected, the word lines in the unselected state, that is, the gates of QT1 and QT2 are made deeper than the negative voltage, for example, −0.2 V, from 0 so far. Must be biased. If the gate voltage and the source voltage are VG and VS, respectively, the effective gate voltage when QT1 or QT2 is not selected is VG−VS−VT, but VG, VS and VT are −0.2V or less, 0, At 0.2V, this effective gate voltage is −0.4V or less. On the other hand, if the minimum value of VT at which the sub-threshold current can be ignored is 0.4 V, the effective gate voltage of a transistor having VT of 0.4 V under normal bias conditions is 0, 0, VG, VS, VT respectively. Since it is 0.4V, it becomes -0.4V. Therefore, in the above-described method in which the low VT and the negative voltage gate are combined, a lower effective gate voltage is applied, so that the subthreshold current does not flow. In this case, the selected word voltage is a pulse rising from -0.2 V in the non-selected state to VCC or higher.

  In the past, it has been assumed that VT of the P-channel and N-channel transistors in the memory cell is equal to 0.5 V, but this is not always necessary. Since the N-channel transistor is an important transistor that determines the read current to the data line, this VT is set to a VT as low as possible, for example, 0.4 V so that the subthreshold current does not become a problem. However, since the P-channel transistor has a main role of charging a minute capacity in the memory cell and may be a little slow, its absolute value may be set to 0.4V or more, for example, 0.6V. Further, for the sake of simplicity, it has been assumed that VCH is twice VCC, but VCH may be equal to or higher than VCC as long as it is lower than the breakdown voltage of the transistor, for example, the gate breakdown voltage.

  There is also a method of charging the power supply line at a high speed while increasing the sensitivity in the memory cell. As described above, a circuit in which transistors are cross-coupled in a memory cell can be regarded as a differential amplifier. In addition to the offset voltage, a capacitance difference between nodes N1 and N2 also affects the sensitivity of the differential amplifier. Depending on the layout of the memory cell, there may be a difference in capacity when priority is given to high density, but if this value is large, the sensitivity will deteriorate. That is, a larger voltage difference is required between the nodes N1 and N2 immediately before amplification. The sensitivity due to this capacitance difference becomes worse as the speed at which the feed line (for example, PL1) is raised to VCH is faster. This problem can be solved by two-stage amplification as shown in FIG. That is, two transistors whose channel widths are significantly different (for example, 10 times) from each other are connected in parallel to each power supply line (PL1 or the like). Applying ΦP, the transistor with a small channel width (QP1) is turned on to charge the feed line little by little. After amplification to a large voltage difference between the nodes N1 and N2, ΦP 'is applied to A large transistor (QP1 ') is turned on to charge at high speed.

  FIG. 17 is a cross-sectional view of an embodiment of the present invention. As shown in this embodiment, the switch MOS (QP) and the PMOS transistor of the memory cell are formed in an n-well. However, since the source or drain electrode of each transistor is increased to VCH, the potential of these wells is also increased. It is necessary to keep VCH. At this time, if the potential of the n-well forming the PMOS transistor of the peripheral circuit is set to VCC, the substrate may be made P-type.

  FIG. 18 is another cross-sectional embodiment of the present invention. In this embodiment, since a large voltage VCH is applied to the switch MOS and the PMOS transistor of the memory cell, the breakdown voltage is increased by making the gate oxide film of these MOSs thicker than the peripheral circuit. The MOS transistor in the peripheral circuit has an effect that the transconductance increases because the oxide film pressure remains thin, and it can operate at high speed.

  FIG. 19 is a cross-sectional view of another embodiment of the present invention. In this embodiment, as shown in FIG. 1A, the switch MOS and the memory cell PMOS are not separated as in the case where the switch MOS is attached to each memory cell. In such a case, the well in which both MOS transistors are formed should be set to the potential of VCH.

  FIG. 20 is a cross-sectional view of another embodiment of the present invention, in which the present invention is formed on an N-type substrate. When the present invention is applied to an N-type substrate, the peripheral circuit, the switch MOS, and the PMOS of the memory cell cannot be separated. Accordingly, as shown in the present embodiment, a common deep P well can be formed in the switch MOS and the PMOS of the memory cell, and an N well can be formed therein to change the potential of the peripheral circuit.

  In order to make the best use of the present invention, it is desirable that the memory array and the peripheral circuit be further devised. FIG. 21 shows an embodiment in which the present invention is applied to an SRAM portion in a chip or a one-chip SRAM. The memory portion is divided into a plurality of memory arrays (MA1, MA2,...). Global word lines are wired across a plurality of memory arrays. Within the memory array, m in the direction of sub-word lines (WL11,..., WLn1, WL12,..., WLn2,...) And n in the direction of data lines (DL11, / DL11,..., DL12, / DL12,. , Composed of a plurality of m × n memory cells MC arranged in a matrix. Sub-feed lines (PL11,...,...) To which a boosted voltage VCH is applied to power supply nodes to the high potential side of the plurality of memory cells via switch MOS transistors (QPL11,..., QPLn1, QPL12,. PLn1, PL12,..., PLn2,... Are wired so as to be paired with the sub-word lines described above. The sub word line can be simply read as a word line in correspondence with the above-described embodiment.

  Now, in the system based on FIG. 9, as shown in FIG. 22, VT of the MOS transistors (QC1, QC2, QS1, QS2) forming the memory cells of the memory cell MC is 0.5 V, and the transfer MOS transistors (QT1,. The VT of QT2) is 0.2V. That is, the MOS transistor included in the memory cell is set to a threshold voltage at which the subthreshold current does not matter as a whole SRAM, and conversely, the transfer MOS transistor is set to a threshold voltage that requires attention. The power supply VCC supplied from the outside to the SRAM is 1V, the boosted voltage VCH formed from the VCC by the voltage conversion circuit VC2 is 2V (= 2VCC), and the negative voltage -VWB formed from the VCC by the voltage conversion circuit VC3. Was 0.2V.

  For example, in order to select one sub word line WL11, that is, to apply the cell activation pulse rising from the negative voltage -VWB (for example, -0.2V) to VCC (1V) to WL11, control is performed with the global word line GL1. The line RX1 may be selected by an address signal. In order to select RX1, a memory array selection signal Фsr1, which is formed by using YDEC / DRV and the timing control circuit TC and substantially selects the memory array MA1, is used. That is, a pulse rising from −VWB to VCC by the LCB receiving Фsr1 may be applied to RX1, and a pulse rising from VCC to −VWB may be applied to GL1 by another level converter LCB connected to GL1. The global word line GL1 is selected from the row address AX by the row address decoder / driver XDEC / DRV. At this time, the other GL lines (global word lines) and the other RX lines remain at VCC and -VWB, respectively. On the other hand, only ФP1 rises from 0 to VCH in the switch MOS selection signal group (ФP1, ＬＣP2...) By the other level converter LCA, and the others remain at 0V. Therefore, the switch MOS connected to PL11,..., PLn1 is turned off, and the corresponding switch MOS group of the non-selected memory array remains on. In order to raise ФP1 from 0V to VCH, a memory array selection signal Фsp1 which is formed using YDEC · DRV and the timing control circuit TC2 and substantially selects the memory array MA1 is used. In this way, the memory cell (MC) group on WL11 is activated and operates as described above.

  Here, Q'D1 and Q'D2 on each data pair line are acceleration transistors for precharging the voltage of the data pair line to VCC at high speed. The RWC is a read / write circuit selected by a column read selection signal (ФRY1) similar to that in FIG. 2, and all use a low VT for speeding up. In order to perform a write operation from the I / O line to the data line at high speed, an N channel and a P channel MOS selected by a column write selection signal (ФWY1, / ФWY1) are connected in parallel.

  As described above, by dividing and partially driving the word line and the feeder line, the burden on the built-in VCH and -VWB generation circuit can be reduced, and the single power supply design becomes easier. This is because a power supply line or a word line that must supply power to VCH or −VWB because the voltage fluctuates with the operation is localized to the sub power supply line subword line WL11. In this embodiment, since one switch MOS may be added for each power supply line, there is an advantage that an increase in area due to division is small. However, for example, since ФP1 is a high voltage (VCH) pulse, the power for charging and discharging the gate capacitances of a large number of switch MOSs connected to this line becomes relatively large.

  FIG. 23 shows the calculated operating voltage margin of the memory cell of FIG. The horizontal axis in this figure is the power supply voltage VCC supplied from the outside, and the vertical axis is the time from when the word line WL is selected (from 0 V to VCC) until the potential difference between the data lines DL and / DL reaches 100 mV. Represents the signal rise time τ defined in (1). The smaller the signal rise time τ, the better. Conventional is a memory cell of FIG. 22 in which all six MOS transistors have the same threshold voltage V T = 0.75 V, and the source side power supply node (the high potential side power supply node of the memory cell) of QC1 and QC2 is the power source. The characteristic of a conventional memory cell directly connected to the voltage VCC is shown. In this conventional configuration, since the VT of the MOS transistor is large, the subthreshold current is not substantially a problem. However, it can be seen that in the conventional configuration, when the power supply voltage becomes 0.8 V or less, the signal rise time τ increases abruptly and substantially does not operate. In other words, when the power supply voltage VCC falls below the threshold voltage VT of the MOS transistor used, the memory cell does not substantially operate due to an increase in the rise time τ.

  On the other hand, when the memory cell of FIG. 22 of the present application is used, it operates up to a lower power supply voltage. The curve shown by This work in FIG. 23 indicates that the threshold voltages of QC1, QC2, QS1, and QS2 constituting the memory cells in the memory cell of FIG. 22 are 0.75 V, and the threshold voltages of the transfer MOS transistors QT1 and QT2. Is calculated as 0.2V. Further, the boosted voltage VCH is calculated for two cases of 2 VCC and 3 VCC, and the calculation points are indicated by circles and squares, respectively. In this example, it can be seen that even if the power supply voltage becomes lower than the threshold voltage of the MOS transistor of the memory cell, it operates at about τ = 10 ns and operates up to about 0.5V. In other words, according to the present application, the threshold voltage of the MOS transistor of the memory cell cannot be reduced to a certain value (for example, 0.5 V) due to the subthreshold current restriction, but the SRAM operates at this threshold voltage or less. The law was shown. In FIG. 22, since the threshold voltage of QT1 and QT2 is set to 0.2 V where the subthreshold current becomes a problem, the signal level on the low potential side of the word line is set to −VWB, and when the memory cell is in the non-selected state, QT1 and QT2 Prevented subthreshold current from flowing. In order to prevent the subthreshold current from becoming a problem in QT1 and QT2, for example, when a MOS transistor having a threshold voltage of 0.5 V is used, the signal level on the high potential side of the word line is sufficiently set so as to increase its driving capability. What is necessary is just to raise the pressure. Further, a sufficiently low voltage operation is possible if VT such as a load MOS on the data line shown in FIG. 21 or a MOS in the read / write circuit RWC is made sufficiently small (for example, 0.2 V or less). The other peripheral drive / logic circuits are effective at a sufficiently low VT, that is, at a sufficiently low VCC by using a subthreshold current low-frequency circuit as described in the above-mentioned single-line book “VLSI LSI”. Therefore, the chip as a whole operates even at VCC below VT of the in-cell cross-coupled MOS.

  The present application is particularly advantageous in a device that operates at a low power supply voltage such as a battery. That is, although the power supply voltage of the solar cell is about 0.5V, an SRAM that can operate even with this solar cell is made possible for the first time. Moreover, since the voltage can be reduced, the effect of reducing power consumption is remarkable.

  FIG. 24 shows another embodiment for reducing the power consumption although the area is slightly larger. For simplicity, only WL11 and PL11 in FIG. 21 are extracted. The MOS transistors PL11 to PLn1 for switching VCH in FIG. 21 are simultaneously controlled by one signal ФP1, whereas in FIG. 24, a level converter for controlling the switch MOS and its gate is added to each divided feeder line. It is. For example, when WL11 is selected and an activation pulse is applied, the gate of QPL1 is changed from 0 to VCH, and QPL1 is turned off. Therefore, the gate capacitance driven by the high voltage (VCH) becomes one and the power is reduced. At this time, the gates of the other switch MOSs remain zero.

It is a figure which shows the concept of this invention which controls the feeder line voltage of a static memory cell. It is a conventional static memory cell and its operation waveform diagram. This is an embodiment applied to a static memory cell array. FIG. 4 is a read operation timing chart of FIG. 3. FIG. 4 is a timing diagram of the write operation in FIG. 3. This is an embodiment applied to a static memory cell array. This is an embodiment applied to a static memory cell array. It is the Example which shared the power supply circuit for electric power feeding. This is an embodiment applied to a static memory cell array. FIG. 10 is a read operation timing chart of FIG. 9. FIG. 10 is a timing diagram of the write operation in FIG. 9. It is a voltage drop prevention circuit diagram of a feeder. This is an embodiment in which a feeder line and a word line are orthogonal. This is an application example to an external dual power supply chip. This is an application example to an external single power supply chip. It is an Example of the drive system of a feeder. Sectional drawing of the Example of this invention. Sectional drawing of another Example of this invention. Sectional drawing of another Example of this invention. Sectional drawing of another Example of this invention. This is an embodiment applied to a divided memory cell array. FIG. 22 is an embodiment of a memory cell internal circuit of FIG. 21. FIG. 23 is a characteristic diagram of the example of the memory cell of FIG. 22. It is an Example of the drive system of the divided | segmented feeder line.

Explanation of symbols

QC1, QC2, QT1, QT2, QS1, QS2,. . . . Transistors in memory cells, N1, N2,. . . . Memory node in memory cell, DL, / DL, DL11, / DL11, DL12, / DL12. . . . Data lines WL1, WL2, WL11, WL12, WLn1, WLn2,. . . Word lines, PL1, PL2, PL1 ′, PLm ′, PL11, PL12, PLn1, PLn2. . . Feed line, PLC. . . Common feeder, MC, MC1 to MC4. . . Memory cell, VSS. . . Reference potential, VCC. . . Supply voltage, VCH. . . Supply voltage or boosted supply voltage, QP1, QP2, QP, QP1 '. . . Switch transistor, CE. . . Chip activation signal, PC. . . Precharge circuit, ΦP, / ΦP ', ФP1, ФP2, ФP1'. . . Precharge signal, AMP. . Amplifier, SP, SN. . . Amplifier drive line, QEQ. . . Balancing transistors, ΦR1, ΦR2. . . Read selection symbol, ΦW1, ΦW2. . . Write selection symbol, ΦRW1. . . Read / write selection symbol, AX, AY. . . Row and column addresses, Din, Dout. . . Data input and data output, / WE. . . Write control signal, QR1, QR2. . . Read transistor, QW1, QW2. . . Write transistor, SPG. . . Amplifier drive circuit, XDEC, DRV. . . Row decoder and driver, YDEC, DRV. . . Column decoder and driver, I / O, / I / O. . . Data input / output line, RWC. . . Read / write control circuit, QL1, QL2, QL3, QL4. . . Internal voltage control transistor, ΦX1, ΦXn. . . Feed line selection signal, INTF. . . Chip input / output interface circuit, CORE. . . The main circuit of the chip, VDC. . . Built-in step-down circuit, VCC1, VCC2. . . Supply voltage VC1, VC2, VC3,. . . Voltage conversion circuit, PCG. . . Precharge signal generation circuit, LCA, LCB. . . Level converter, RX1, RX2. . . Control lines, GL1, GLn. . . Global word line, ФRY1. . . Column readout selection signal, ФWY1, ФWY1. . . Column write selection signal, QPL1, QPL2. . . Switch transistor, VWB. . . Word line bias voltage, MA1, MA2. . . Memory array, Фsr1, Фsr2. . . Memory array selection signal, Фsp1, Фsp2. . . Memory array selection signals TC1, TC2,. . . Timing control circuit, GA11, GAn1, GA12, GAn2. . . NAND gate.

Claims (10)

A semiconductor device using a static memory cell having a MOS transistor in which a gate and a drain are cross-coupled to each other and connected to a word line and a data line,
The cross-coupled MOS transistor is configured such that substantially no current flows between the drain and the source even when the gate and source voltages are equal,
The power supply node of the static memory cell is connected to a first power supply voltage via a power supply voltage supply means provided outside the static memory cell,
The first power supply voltage is greater than the maximum voltage of the data line;
A semiconductor device characterized in that when the static memory cell is selected, a power supply line of the selected static memory cell is brought into a floating state. In claim 1,
The semiconductor device according to claim 1, wherein the operation of the selected static memory cell is a write operation. A semiconductor device using a static memory cell having a MOS transistor in which a gate and a drain are cross-coupled to each other and connected to a word line and a data line,
The cross-coupled MOS transistor is configured such that substantially no current flows between the drain and the source even when the gate and source voltages are equal,
The power supply node of the static memory cell is connected to a first power supply voltage via a power supply voltage supply means provided outside the static memory cell,
The first power supply voltage is greater than the maximum voltage of the data line;
A semiconductor device, wherein a data line pair of the static memory cell is connected to a circuit precharged to a value approximately half of the maximum voltage that can be taken by the data line and a cross-coupled sense amplifier. A semiconductor device using a static memory cell having a MOS transistor in which a gate and a drain are cross-coupled to each other and connected to a word line and a data line,
The cross-coupled MOS transistor is configured such that substantially no current flows between the drain and the source even when the gate and source voltages are equal,
The power supply node of the static memory cell is connected to a first power supply voltage via a power supply voltage supply means provided outside the static memory cell,
The first power supply voltage is greater than the maximum voltage of the data line;
A semiconductor device, wherein a voltage difference between two storage nodes when the static memory cell is not selected is larger than a voltage difference between two storage nodes when the static memory cell performs a write operation. In claim 4,
The semiconductor device, wherein a maximum voltage of the data line when performing a write operation of the static memory cell is smaller than a power supply voltage of the static memory cell when the static memory cell is not selected. In claim 4,
A semiconductor device, wherein a voltage higher than the maximum voltage of the data line is applied to a power supply terminal of the static memory cell after a write voltage from the data line is determined at the storage node of the static memory cell. . A semiconductor device using a static memory cell having a MOS transistor in which a gate and a drain are cross-coupled to each other and connected to a word line and a data line,
The cross-coupled MOS transistor is configured such that substantially no current flows between the drain and the source even when the gate and source voltages are equal,
The power supply node of the static memory cell is connected to a first power supply voltage via a power supply voltage supply means provided outside the static memory cell,
The first power supply voltage is greater than the maximum voltage of the data line;
A semiconductor device comprising a circuit capable of controlling a power supply voltage of the static memory cell every cycle at a voltage larger than a maximum voltage of the data line. In claim 7,
A semiconductor device characterized in that a threshold voltage of a MOS transistor provided in the static memory cell is larger than a threshold voltage of a MOS transistor provided in a peripheral circuit of the static memory cell. In any one of Claims 1 thru | or 8,
The static memory cell uses a MOS transistor having a channel length of 0.5 μm or less and a gate oxide film thickness of 6 nm or less. The maximum voltage of the data line to which the static memory cell is connected is 0.3 to 1.5 V. A semiconductor device characterized by being in the range. A semiconductor device, wherein at least one of two kinds of power supply voltages is controlled when a static memory cell is selected.

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