JP2006074848A - Non-contact power transmission system - Google Patents
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Abstract
Description
本発明は非接触電力伝送装置において２次側における電圧および電流を１次側にフィードバックし、２次側出力電圧、電流を安定化制御する非接触電力伝送装置に関する。 The present invention relates to a non-contact power transmission device that feeds back a voltage and current on a secondary side to a primary side in a non-contact power transmission device and stabilizes and controls a secondary output voltage and current.
非接触電力伝送装置において、２次側の安定化を計る場合、２次側に独立した安定化回路を設ける方法がある。（例えば、特許文献１、２，３参照。）。
２つ目の方法として、２次側共振回路の共振用コンデンサ−の値を変える事により、共振周波数を変えて、２次側で受電出来る電力を可変する方法を用い安定化する方法がある。（例えば、特許文献４参照。）。
従来方式の課題として以下のものがあげられる。すなわち２次側に独立した安定化回路を設ける方法は２次側の安定化回路の損失が有り、全体の効率を下げてしまう。
２次側が軽負荷の場合においても、結合トランスの１次側コイルの電流は少なくならず１次側コイルの消費電力が低減しない。
The following are problems of the conventional method. That is, the method of providing an independent stabilization circuit on the secondary side has a loss of the stabilization circuit on the secondary side, and lowers the overall efficiency.
Even when the secondary side is lightly loaded, the current of the primary side coil of the coupling transformer is not reduced and the power consumption of the primary side coil is not reduced.
２次側の共振回路の共振周波数を変える方法は、２次側が軽負荷の場合においても、結合トランスの１次コイルの電流は小さくならず１次側コイルの消費電力が低減しない。 In the method of changing the resonance frequency of the secondary side resonance circuit, even when the secondary side is lightly loaded, the current of the primary coil of the coupling transformer is not reduced and the power consumption of the primary side coil is not reduced.
以上に述べた従来の非接触電力伝送装置における２次側出力電圧、電流を安定化制御する方法では、全体の効率が悪い。また２次側負荷が軽負荷の場合における消費電力を低減出来ない。 The method for stabilizing and controlling the secondary output voltage and current in the conventional non-contact power transmission apparatus described above has poor overall efficiency. Moreover, power consumption cannot be reduced when the secondary load is light.
本発明は非接触電力伝送装置における２次側出力電圧、電流の安定化制御を高効率で実現することを目的とするものである。 It is an object of the present invention to realize the stabilization control of the secondary output voltage and current in the non-contact power transmission apparatus with high efficiency.
上記目的を達成するため、本発明の非接触電力伝送装置は、２次側における電圧および電流に関する信号を１次側ユニットへ非接触の電磁結合コイルを用いて１次側に送信し、１次側ユニットでは２次側から送られてきた信号を受信して、１次側ユニットのインバータの駆動周波数を可変し２次側の出力電圧及び電流を安定化制御する手段を設けたものである。 In order to achieve the above object, the non-contact power transmission apparatus of the present invention transmits a signal related to voltage and current on the secondary side to the primary side using a non-contact electromagnetic coupling coil to the primary side unit. The side unit is provided with means for receiving a signal sent from the secondary side, varying the drive frequency of the inverter of the primary side unit, and stabilizing and controlling the output voltage and current on the secondary side.
本発明の非接触電力伝送装置によれば、非接触で２次側の電圧電流情報を１次側に伝送しインバータの駆動周波数を変え出力電圧、電流を制御出来るため、閉磁路コアを用いたスイッチング電源と同じ安定化電源を実現出来る事にある。
このため２次側ユニットに独立した安定化のためのチョッパ回路やシリーズレギュレータが不要であり、２次側の安定化回路の為の損失が無いので、高効率の非接触電力伝送装置が実現出来る。２次側ユニットに独立した安定化回路が不要なのでこの回路の実装面積も不要となり小型化が可能に成りコストも低減する。
According to the non-contact power transmission device of the present invention, since the voltage / current information on the secondary side is transmitted to the primary side in a non-contact manner, the drive frequency of the inverter can be changed and the output voltage and current can be controlled. It is possible to realize the same stabilized power supply as the switching power supply.
This eliminates the need for a separate chopper circuit or series regulator for stabilization on the secondary side unit, and there is no loss for the secondary side stabilization circuit, so a highly efficient non-contact power transmission device can be realized. . Since an independent stabilization circuit is not required for the secondary unit, the mounting area of this circuit is not required, and miniaturization is possible and costs are reduced.
２次側負荷が軽負荷もしくは無負荷時において、インバータの駆動周波数を上げる制御を行う事により、駆動される結合トランスの１次側直列共振回路の共振周波数から離調するため１次側コイルの高周波電流も減る事になり、２次側負荷が軽負荷もしくは無負荷時において交流電源の消費電力を低減できる効果がある。 When the secondary load is light or no load, the control of increasing the drive frequency of the inverter is performed to detune from the resonance frequency of the primary series resonance circuit of the coupled transformer to be driven. The high-frequency current is also reduced, and the power consumption of the AC power supply can be reduced when the secondary load is light or no load.
図１は非接触電源装置に応用した回路図である。
以下本発明の実施の形態を図１から図６に基づいて説明する。
FIG. 1 is a circuit diagram applied to a non-contact power supply device.
Hereinafter, embodiments of the present invention will be described with reference to FIGS.
図１において１次側ユニットのインバータは１次側整流回路の直流出力を電源とする。前記インバータはハーフブリッジ接続され発振周波数可変２相発振器により駆動される。結合トランスの１次側コアに巻き線を巻き回し、１次側コイルを作る。この１次側コイルをＬ１とする。Ｌ１と直列に共振コンデンサＣ1を接続する。
一方、２次側ユニットでは、２次側コアに巻き線を巻き回し、２次側コイルを得る。この２次側コイルをＬ２とする。前記Ｌ２を用いてＬ１の磁力線から高周波電力を取り出す。Ｌ２にＣ２を直列に接続して２次側直列共振回路を得る。Ｃ２の後のブリッジ整流回路とＣ６から構成する２次側整流回路により整流した後、負荷に電力を供給する。
In FIG. 1, the inverter of the primary unit uses the DC output of the primary rectifier circuit as a power source. The inverter is half-bridge connected and driven by a variable oscillation frequency two-phase oscillator. A winding is wound around the primary core of the coupling transformer to form a primary coil. Let this primary side coil be L1. A resonant capacitor C1 is connected in series with L1.
On the other hand, in the secondary unit, a winding is wound around the secondary core to obtain a secondary coil. Let this secondary coil be L2. The high frequency power is extracted from the magnetic field lines of L1 using L2. A secondary side series resonance circuit is obtained by connecting C2 in series to L2. After rectification by the bridge rectifier circuit after C2 and the secondary side rectifier circuit composed of C6, power is supplied to the load.
図１のインバータは電界効果トランジスタＱ1のドレインＤを１次側整流回路のVCC側に接続し、Ｑ1のソースＳとＱ２のドレインＤを直列に接続し、Ｑ２のソースＳを１次側整流回路のGND側に接続する。
Ｑ1のソースＳとＱ２のドレインＤの接続点に、１次側直列共振回路のＣ１側を接続する。Ｌ１側をＱ２のソースに接続する。
Ｑ１のドレインＤとソースＳに並列にＣ３を、Ｑ２のドレインＤとソースＳに並列にＣ４を接続する。
Ｄ１、Ｄ２は電界効果トランジスタで有るＱ１、Ｑ２のボディダイオードである。
In the inverter of FIG. 1, the drain D of the field effect transistor Q1 is connected to the VCC side of the primary side rectifier circuit, the source S of Q1 and the drain D of Q2 are connected in series, and the source S of Q2 is connected to the primary side rectifier circuit. Connect to the GND side.
The C1 side of the primary side series resonance circuit is connected to the connection point between the source S of Q1 and the drain D of Q2. Connect the L1 side to the source of Q2.
C3 is connected in parallel to the drain D and source S of Q1, and C4 is connected in parallel to the drain D and source S of Q2.
D1 and D2 are body diodes of Q1 and Q2, which are field effect transistors.
図１の発振周波数可変２相発振器は、Ｑ１の Ｇ−Ｓ 間 及び Ｑ２の Ｇ−Ｓ 間にデットタイムを設けて交互に電圧駆動する発振周波数可変の発振器で有る。 The oscillation frequency variable two-phase oscillator of FIG. 1 is an oscillator with variable oscillation frequency that is driven by voltage alternately by providing a dead time between GS of Q1 and GS of Q2.
図５のタイミングチャートにおいてＱ１の Ｇ−Ｓ 間を駆動する電圧をＱ１ ＶＧＳ と表し、Ｑ2の Ｇ−Ｓ 間を駆動する電圧をＱ2 ＶＧＳ と表して有る。
同様に図５のタイミングチャートにおいてＱ１ ＶＧＳとＱ2 ＶＧＳ 間のデットタイムをＤtと表して有る。
In the timing chart of FIG. 5, the voltage for driving the GS of Q1 is expressed as Q1 VGS, and the voltage for driving the GS of Q2 is expressed as Q2 VGS.
Similarly, in the timing chart of FIG. 5, the dead time between Q1 VGS and Q2 VGS is represented as Dt.
図４−Ａは結合トランスの１次側直列共振回路の共振周波数と２次側直列共振回路の共振周波数対２次側負荷出力電圧の関係をグラフで表したもので有る。
縦軸は２次側負荷出力電圧を示し、横軸はインバータの駆動周波数を示す。
１次側直列共振回路の共振周波数をｆpに設定し、２次側直列直列共振回路の共振周波数をｆsに設定して有る。
ｆp ＜ ｆs の関係に設定する。
駆動周波数の使用範囲は図４−Ａの ｆｓからｆ１までとする。
FIG. 4A is a graph showing the relationship between the resonance frequency of the primary side series resonance circuit of the coupling transformer and the resonance frequency of the secondary side series resonance circuit versus the secondary side load output voltage.
The vertical axis represents the secondary load output voltage, and the horizontal axis represents the inverter drive frequency.
The resonance frequency of the primary side series resonance circuit is set to fp, and the resonance frequency of the secondary side series resonance circuit is set to fs.
The relation of fp <fs is set.
The use range of the drive frequency is from fs to f1 in FIG.
図４−Ｂは駆動周波数対、１次側直列共振回路の共振周波数特性と２次側直列共振回路の共振周波数特性を合成した２次側整流電圧のカーブで有る。 FIG. 4-B is a curve of the secondary side rectified voltage obtained by synthesizing the drive frequency versus the resonance frequency characteristic of the primary side series resonance circuit and the resonance frequency characteristic of the secondary side series resonance circuit.
図４−Ｂのグラフにおいて、ｆｓの点を使用範囲の最大出力電圧の点なるよう１次、２次の共振回路の共振周波数を設定しインバータの最低駆動周波数をｆｓに設定する。
ＡＣ入力電圧の変動や負荷の変動に応じて駆動周波数の変化範囲をｆ０からｆ１の範囲に設定する。
負荷に最大出力電圧を供給できる駆動周波数はｆｓの点になる。
定格負荷時は周波数をｆ０に設定し、ＡＣ入力電圧が低下した場合や、負荷が重たくなった場合など、周波数fｓに設定する。ＡＣ入力電圧が高くなった場合や軽負荷時は、周波数をｆ1設定する。
ｆp、ｆs、f０、ｆ1 の各周波数の１例を下記に示す。
ｆp ≒６０ＫＨｚ、 ｆs ≒７４ＫＨｚ 、ｆ0 ≒７７ＫＨｚ、ｆ1 ≒１００ＫＨｚ
In the graph of FIG. 4-B, the resonance frequency of the primary and secondary resonance circuits is set so that the point of fs becomes the point of the maximum output voltage in the use range, and the minimum drive frequency of the inverter is set to fs.
The change range of the drive frequency is set in the range of f0 to f1 according to the fluctuation of the AC input voltage and the fluctuation of the load.
The drive frequency at which the maximum output voltage can be supplied to the load is at the point fs.
At the rated load, the frequency is set to f0, and is set to the frequency fs when the AC input voltage drops or the load becomes heavy. When the AC input voltage becomes high or when the load is light, the frequency is set to f1.
An example of each frequency of fp, fs, f0, f1 is shown below.
fp ≒ 60KHz, fs ≒ 74KHz, f0 ≒ 77KHz, f1 ≒ 100KHz
図４−Ｂのグラフからは、インバータの駆動周波数を変化させることで出力電圧を可変出来ることが判る。２次側出力電圧が設定より下回った場合その情報を、信号制御用送信コイルを用いて非接触で１次側ユニットに送り、１次側ユニットのインバータの駆動周波数を下げる方向でフィードバック制御を行う。また、軽負荷時等において、出力電圧が設定より上回った場合は、その情報を、制御用送信コイルを用いて非接触で１次側ユニットに送り、１次側ユニットのインバータの駆動周波数を上げる方向でフィードバック制御を行う。 From the graph of FIG. 4-B, it can be seen that the output voltage can be varied by changing the drive frequency of the inverter. When the secondary output voltage falls below the setting, the information is sent to the primary unit in a non-contact manner using a signal control transmission coil, and feedback control is performed in a direction to lower the drive frequency of the primary unit inverter. . Also, when the output voltage exceeds the setting at light load, etc., the information is sent to the primary unit in a non-contact manner using the control transmission coil, and the drive frequency of the primary unit inverter is increased. Perform feedback control in the direction.
図１において２次側整流回路の出力電圧を誤差増幅器に接続し、この電圧と基準電圧を比較する。
誤差増幅器の出力を変調器に接続し変調器の出力を制御用発振回路に接続する。
制御用発振器はＬ４とＣ７の並列共振周波数で自励発振を行う。
In FIG. 1, the output voltage of the secondary side rectifier circuit is connected to an error amplifier, and this voltage is compared with a reference voltage.
The output of the error amplifier is connected to the modulator, and the output of the modulator is connected to the control oscillation circuit.
The control oscillator performs self-excited oscillation at the parallel resonance frequency of L4 and C7.
図１においてＬ３とＣ８の並列共振回路を検波回路に接続し、検波回路の出力を復調器に接続する。
復調回路の出力を発振周波数可変２相発振器に接続する。
In FIG. 1, the parallel resonant circuit of L3 and C8 is connected to the detector circuit, and the output of the detector circuit is connected to the demodulator.
The output of the demodulation circuit is connected to the oscillation frequency variable two-phase oscillator.
図３は誤差増幅器、変調器、制御用発振器、復調器、発振周波数可変２相発振器の具体例で有る。
出力電圧をＲ２０１とＲ２０２で分割しＵ２０２のオペアンプのインバータ端子（−端子）に接続する。
Ｕ２０２のオペアンプのノンインバータ端子（＋端子）には基準電圧を印可しておく。
Ｕ２０２の出力を、Ｕ２０１のコンパレータのインバータ端子（−端子）に接続する。
三角波発振器の出力をＵ２０１のノンインバータ端子（＋端子）に接続する。
三角波発振器の発信周波数を１０ＫＨｚ程度で有る。
制御用発振器はＬ４，Ｃ７の並列共振回路を用いたＬＣ発振器で有り、８ＭＨｚ程度の発振周波数である。Ｌ４は制御用送信コイルである。
Ｕ２０１の出力を制御用発振器に接続し、制御用発振器の発振を断続させる。
FIG. 3 shows specific examples of an error amplifier, a modulator, a control oscillator, a demodulator, and a variable oscillation frequency two-phase oscillator.
The output voltage is divided by R201 and R202 and connected to the inverter terminal (− terminal) of the operational amplifier U202.
A reference voltage is applied to the non-inverter terminal (+ terminal) of the operational amplifier U202.
The output of U202 is connected to the inverter terminal (− terminal) of the comparator of U201.
The output of the triangular wave oscillator is connected to the non-inverter terminal (+ terminal) of U201.
The transmission frequency of the triangular wave oscillator is about 10 KHz.
The control oscillator is an LC oscillator using a parallel resonant circuit of L4 and C7, and has an oscillation frequency of about 8 MHz. L4 is a control transmission coil.
The output of U201 is connected to the control oscillator, and the oscillation of the control oscillator is interrupted.
一方１次側ユニットにおいて、Ｌ３はＬ４と電磁結合された制御用受信コイルで有る。Ｌ３，Ｃ８の並列共振回路は制御用発振器の発振周波数に同調させる値とする。
Ｌ３，Ｃ８の並列共振回路をＤ３、Ｃ９で構成された検波回路に接続する。
検波回路の出力をＵ１０１のコンパレータのノンインバータ端子（＋端子）に接続する。Ｕ１０１のコンパレータのインバータ端子（−端子）には、＋１２ＶをＲ１０３とＲ１０４で分割した電圧を印可する。
検波回路の出力をＲ１０３とＲ１０４で分割した電圧と比較しＵ１０１の出力とする。Ｕ１０１の出力をＮＰＮトランジスタＱ１０３のベースへ抵抗を介して接続する。
Ｑ１０３のコレクタをＰＮＰトランジスタＱ１０２のベースへ抵抗を介して接続する。Ｑ１０２のエミッタを＋１２Ｖ電源に接続する。
Ｑ１０２のエミッタにＤ１０１，Ｌ１０１，Ｃ１０２から構成するパルス幅−電圧変換回路に接続する。
パルス幅−電圧変換回路の出力をＮＰＮトランジスタＱ１０1のベースへ抵抗を介して接続する。
Ｑ１０1のエミッタにＲ１０２を接続し回路のＧＮＤに接続する。
２相発振回路はＣ１０１とＲ１０１のＣＲ発振器を構成して有る。
Ｑ１０1のコレクタをＲ１０１に接続する。Ｒ１０１はＲ１０１とＱ１０1のコレクタ−エミッタを通して並列接続されている。
２相発振回路の出力はＡ相とＢ相の２出力を持ち、それぞれ逆相で出力される。
２相発振回路の出力をデットタイム生成回路に接続し、Ａ相とＢ相間に僅かなデッドタイムを設ける。
デットタイム生成回路の出力をハーフブリッジドライバーに接続する。
ハーフブリッジドライバーに接続されているＤ１０１とＣ１０２はハイサイドスイッチのＱ１の駆動電源を作る目的でブーストアップ回路を構成してある。
ハーフブリッジドライバーの出力はＱ１のゲート、ソース間とＱ２のゲート、ソース間に接続する。
Ｑ１とＱ２のゲートソース間電圧波形は図５のＱ１ＶGSとＱ２ＶGSになる。
On the other hand, in the primary unit, L3 is a control receiving coil that is electromagnetically coupled to L4. The parallel resonant circuit of L3 and C8 is a value that is tuned to the oscillation frequency of the control oscillator.
The parallel resonance circuit of L3 and C8 is connected to the detection circuit composed of D3 and C9.
The output of the detection circuit is connected to the non-inverter terminal (+ terminal) of the comparator of U101. A voltage obtained by dividing +12 V by R103 and R104 is applied to the inverter terminal (− terminal) of the comparator of U101.
The output of the detection circuit is compared with the voltage divided by R103 and R104 to obtain the output of U101. The output of U101 is connected to the base of NPN transistor Q103 via a resistor.
The collector of Q103 is connected to the base of PNP transistor Q102 via a resistor. Connect the emitter of Q102 to the + 12V power supply.
The emitter of Q102 is connected to a pulse width-voltage conversion circuit composed of D101, L101, C102.
The output of the pulse width-voltage conversion circuit is connected to the base of the NPN transistor Q101 via a resistor.
R102 is connected to the emitter of Q101 and connected to the GND of the circuit.
The two-phase oscillation circuit constitutes a CR oscillator of C101 and R101.
Connect the collector of Q101 to R101. R101 is connected in parallel through the collector-emitter of R101 and Q101.
The output of the two-phase oscillation circuit has two outputs, the A phase and the B phase, which are output in opposite phases.
The output of the two-phase oscillation circuit is connected to the dead time generation circuit, and a slight dead time is provided between the A phase and the B phase.
Connect the output of the dead time generation circuit to the half bridge driver.
D101 and C102 connected to the half-bridge driver constitute a boost-up circuit for the purpose of creating a driving power source for the high-side switch Q1.
The output of the half bridge driver is connected between the gate and source of Q1 and between the gate and source of Q2.
The gate-source voltage waveforms of Q1 and Q2 are Q1VGS and Q2VGS in FIG.
図２にＵ字型コアを用いた結合トランスの一例をしめす。
１次側ユニットに設けたＵ字型コアの胴部に１次巻き線 Ｌ１を巻き回する。
２次側ユニットに設けたＵ字型コアの胴部に２次巻き線 Ｌ2を巻き回する。
結合トランスの１次２次間は、１次側ユニットの樹脂製外郭と２次側ユニットの樹脂製外郭で隔てられている。
結合トランスの１次側Ｕ字型コアの脚部の先端と２次側Ｕ字型コアの脚部の先端とは、数mmの間隔を隔てて対向している。
１次側Ｕ字コアの両脚の中央と１次側ユニットの樹脂製外郭の内側の境界面にＬ３を配置し、２次側Ｕ字コアの両脚の中央と２次側ユニットの樹脂製外郭の内側の境界面にＬ４を配置する。
Ｌ３、Ｌ４は空芯コイルを用いる。
FIG. 2 shows an example of a coupling transformer using a U-shaped core.
The primary winding L1 is wound around the body of the U-shaped core provided in the primary unit.
The secondary winding L2 is wound around the trunk of the U-shaped core provided in the secondary unit.
The primary and secondary sides of the coupling transformer are separated by a resin outer shell of the primary unit and a resin outer shell of the secondary unit.
The tip of the leg part of the primary U-shaped core of the coupling transformer and the tip of the leg part of the secondary U-shaped core are opposed to each other with an interval of several mm.
L3 is arranged at the boundary between the center of both legs of the primary U-shaped core and the resin outer shell of the primary unit, and between the center of both legs of the secondary U-shaped core and the resin outer casing of the secondary unit. L4 is arranged on the inner boundary surface.
L3 and L4 use air-core coils.
フィードバック制御の説明を図３、図４、図６を用いて説明する。
１次側ユニットから２次側ユニットへ非接触で電力を伝送している状態において２次側ユニットの出力電圧が設定値より高い場合、（図６の左側部分）オペアンプＵ２０２のインバータ端子（―端子）の電圧である、Ｒ２０１とＲ２０２で出力電圧を分割した電圧とノンインバータ端子（＋端子）に接続してある基準電圧の比較でインバータ端子（―端子）の方が高いため、Ｕ２０２の出力電圧は低い方へ変化する。
Ｕ２０２の出力電圧はコンパレータのＵ２０１のインバータ端子（―端子）に接続してあり、Ｕ２０１のノンインバータ端子（＋端子）には三角波を印加してある為、三角波の振幅が、Ｕ２０１のインバータ端子より高い期間、Ｈiレベル出力電圧になる。三角波は固定周波数で有るためその周期も固定である。一定周期の期間においてＵ２０１の出力がＵ２０１のインバータ端子の電圧に応じてＵ２０１のパルス幅が変化する為、パルス幅変調の動作といえる。
２次側ユニットの出力電圧が設定値より高い場合、Ｕ２０１のＯＮ幅が広がる結果になる。
The feedback control will be described with reference to FIGS. 3, 4, and 6. FIG.
When the output voltage of the secondary unit is higher than the set value in the state where power is transmitted from the primary unit to the secondary unit in a non-contact manner (left side in FIG. 6), the inverter terminal (-terminal) of the operational amplifier U202 ), The voltage obtained by dividing the output voltage by R201 and R202 and the reference voltage connected to the non-inverter terminal (+ terminal) is higher at the inverter terminal (− terminal). Changes to lower.
Since the output voltage of U202 is connected to the inverter terminal (-terminal) of U201 of the comparator and a triangular wave is applied to the non-inverter terminal (+ terminal) of U201, the amplitude of the triangular wave is greater than that of the inverter terminal of U201. During a high period, the output voltage becomes Hi level. Since the triangular wave has a fixed frequency, its period is also fixed. Since the pulse width of U201 changes in the output of U201 in accordance with the voltage of the inverter terminal of U201 in the period of a fixed period, it can be said to be an operation of pulse width modulation.
When the output voltage of the secondary unit is higher than the set value, the result is that the ON width of U201 is widened.
Ｕ２０１の出力で制御用発振器の発振を断続する制御を行う。
Ｕ２０１の出力がＨiレベルの状態で制御用発振器が発振し、Ｕ２０１の出力がＬowレベルの状態で制御用発振器の発振が停止する。
Ｕ２０１の出力がＯＮの期間制御用発振器が発振し、出力がＯＦＦの期間制御用発振器が発振を停止する。
２次側ユニットの出力電圧が設定値より高い場合、制御用発振器が発振している期間が長くなる。
Control to intermittently oscillate the control oscillator is performed by the output of U201.
The control oscillator oscillates when the output of U201 is at the Hi level, and the oscillation of the control oscillator stops when the output of U201 is at the Low level.
The control oscillator oscillates while the output of U201 is ON, and the control oscillator oscillates while the output is OFF.
When the output voltage of the secondary unit is higher than the set value, the period during which the control oscillator is oscillating becomes long.
１次側ユニットに設けられたＬ３は２次側ユニットのＬ４と電磁結合しているためＬ３、Ｃ８の並列共振回路の高周波誘導電圧は、制御用発振器の発振の断続に応じて断続する。 Since L3 provided in the primary unit is electromagnetically coupled to L4 of the secondary unit, the high frequency induced voltage of the parallel resonant circuit of L3 and C8 is intermittent according to the intermittent oscillation of the control oscillator.
Ｌ３、Ｃ８に接続された検波回路の出力はＵ２０１の出力がＯＮの時ＯＮ、Ｕ２０１の出力がＯＦＦの時、ＯFＦとなり、検波回路のＯＮ幅は、Ｕ２０１の出力のＯＮ幅に同期する。 The output of the detection circuit connected to L3 and C8 is ON when the output of U201 is ON, and is OFF when the output of U201 is OFF, and the ON width of the detection circuit is synchronized with the ON width of the output of U201.
復調器のＵ１０１は検波回路電圧を増幅する目的のコンパレータで有る。
検波回路の出力電圧をＵ１０１のノンインバータ端子（＋端子）に接続して有るためＲ１０３とＲ１０４で生成した基準電圧よりＵ１０１のインバータ端子（―端子）の電圧が高いと、Ｕ１０１の出力はＯＮ、ＮＰＮタイプのＱ１０３がＯＮ、ＰＮＰタイプのＱ１０２がＯＮとなる。
Ｑ１０２のＯＮ幅はＵ2０１のＯＮ幅と同じになる。
The demodulator U101 is a comparator for the purpose of amplifying the detection circuit voltage.
Since the output voltage of the detection circuit is connected to the non-inverter terminal (+ terminal) of U101, if the voltage of the inverter terminal (−terminal) of U101 is higher than the reference voltage generated by R103 and R104, the output of U101 is ON, The NPN type Q103 is ON and the PNP type Q102 is ON.
The ON width of Q102 is the same as the ON width of U201.
Ｌ２０１、Ｄ１０１、Ｃ１０２を用いたパルス幅―電圧変換回路はフィードフォワードタイプのスイッチング電源回路の２次側整流回路に用いられるものと同じ構成である。
従ってＱ１０２のＯＮ幅に比例した出力電圧が得られる。
結果Ｕ２０１のＯＮ幅に比例した出力電圧が得られる。
以上から２次側ユニットの出力電圧が設定値より高い場合、パルス幅―電圧変換回路の出力電圧も高くなる。
The pulse width-voltage conversion circuit using L201, D101, and C102 has the same configuration as that used in the secondary rectifier circuit of the feedforward type switching power supply circuit.
Therefore, an output voltage proportional to the ON width of Q102 is obtained.
An output voltage proportional to the ON width of the result U201 is obtained.
From the above, when the output voltage of the secondary unit is higher than the set value, the output voltage of the pulse width-voltage conversion circuit also becomes high.
パルス幅―電圧変換回路の出力を、抵抗を介してＮＰＮのＱ１０１のベースに接続して有るため、パルス幅―電圧変換回路の出力電圧が高くなるとＱ１０１のコレクタ電流が増える。
Ｑ１０１のコレクタ電流が増える事は、等価的にＲ１０１の抵抗値が減る事になる。
Since the output of the pulse width-voltage conversion circuit is connected to the base of the NPN Q101 via a resistor, the collector current of the Q101 increases as the output voltage of the pulse width-voltage conversion circuit increases.
Increasing the collector current of Q101 equivalently decreases the resistance value of R101.
２相発振器はＣＲ発振器で構成されているため、Ｒ１０１の抵抗値が減ると２相発振器の発振周波数は高くなる。２相発振器の発振周波数が高くなるとインバータの駆動周波数が高くなる。図４−Ｂから駆動周波数が高くなると２次側ユニットの出力電圧が低くなる事が判る。 Since the two-phase oscillator is composed of a CR oscillator, the oscillation frequency of the two-phase oscillator increases as the resistance value of R101 decreases. As the oscillation frequency of the two-phase oscillator increases, the drive frequency of the inverter increases. FIG. 4B shows that the output voltage of the secondary unit decreases as the drive frequency increases.
２次側ユニットの出力電圧が設定値より低くい場合は図６の右側に示すとおりＵ２０１のコンパレータのＯＮ幅が狭まり、制御用発振器の発振幅が狭まりＬ４の高周波電流の発振幅が狭まる。 When the output voltage of the secondary unit is lower than the set value, the ON width of the comparator of U201 is narrowed as shown on the right side of FIG. 6, the emission amplitude of the control oscillator is reduced, and the emission amplitude of the high-frequency current of L4 is reduced.
制御用発振器の発振幅が狭まりＬ４の高周波電流の発振幅が狭まると、検波回路の出力のＯＮ幅が狭まり、パルス幅―電圧変換回路の出力電圧が低くなり、Ｑ１０１のコレクタ電流が減る。Ｑ１０１のコレクタ電流が減ると等価的にＲ１０１の抵抗値に近づく事になる。 When the oscillation amplitude of the control oscillator is narrowed and the oscillation amplitude of the high frequency current of L4 is narrowed, the ON width of the output of the detection circuit is narrowed, the output voltage of the pulse width-voltage conversion circuit is lowered, and the collector current of Q101 is reduced. When the collector current of Q101 is reduced, the resistance value of R101 is equivalently approached.
２相発振器はＣＲ発振器で構成されているため、２相発振器の発振周波数は低くなる。２相発振器の発振周波数が低くなるとインバータの駆動周波数が低くなる。
図４−Ｂから駆動周波数が低くなると２次側ユニットの出力電圧が高くなる事が判る。
Since the two-phase oscillator is composed of a CR oscillator, the oscillation frequency of the two-phase oscillator is low. When the oscillation frequency of the two-phase oscillator is lowered, the drive frequency of the inverter is lowered.
It can be seen from FIG. 4-B that the output voltage of the secondary unit increases as the drive frequency decreases.
以上の様に２次側電圧情報が１次側にフィードバックされ、インバータの駆動周波数を変え、２次側の出力電圧を安定化する。 As described above, the secondary side voltage information is fed back to the primary side, and the drive frequency of the inverter is changed to stabilize the secondary side output voltage.
同様な手法を用いて２次側の電流情報を１次側にフィードバックされ、インバータの駆動周波数を変え、２次側の出力電流を安定化することが出来る。 By using the same method, the secondary side current information is fed back to the primary side, and the drive frequency of the inverter can be changed to stabilize the secondary side output current.
Ｌ１：結合トランスの１次側巻き線インダクタンス。
Ｌ２：結合トランスの２次側巻き線インダクタンス。
Ｃ１：１次側直列共振用コンデンサ。
Ｃ２：２次側直列共振用コンデンサ。
Ｌ３：制御用受信コイル。
Ｌ４：制御用送信コイル。
Ｃ７：Ｌ４の並列共振用コンデンサ−。
Ｃ８：Ｌ３の並列共振用コンデンサ−。
ｆｐ：１次側直列共振回路の共振周波数。
ｆs ：２次側共振回路の共振周波数。
ｆ1 ：定格負荷時のインバータ駆動周波数。
ｆ０：重負荷時のインバータ駆動周波数。
ｆ２：軽負荷時のインバータ駆動周波数。
L1: Primary winding inductance of the coupling transformer.
L2: Secondary winding inductance of the coupling transformer.
C1: Capacitor for primary side series resonance.
C2: Secondary side series resonance capacitor.
L3: Control receiving coil.
L4: Transmitter coil for control.
C7: L4 parallel resonance capacitor.
C8: L3 parallel resonance capacitor.
fp: resonance frequency of the primary side series resonance circuit.
fs: resonance frequency of the secondary side resonance circuit.
f1: Inverter drive frequency at rated load.
f0: inverter drive frequency under heavy load.
f2: inverter drive frequency at light load.
Claims (2)
A non-contact power transmission apparatus including a primary side unit and a secondary side unit configured to separately accommodate a primary side and a secondary side of a coupling transformer so as to be separable from each other. The resonance frequency and the driving frequency of the resonance circuit are set to be detuned, and the current of the primary side series resonance circuit is limited by the degree of detuning of the resonance frequency and the driving frequency of the primary side series resonance circuit, and the secondary In the state where the resonance frequency of the primary and secondary series resonance circuits is set so that the resonance frequency of the side series resonance circuit is substantially equal to the primary side drive frequency, the secondary side drive frequency is varied to change the secondary side. A non-contact power transmission device characterized by varying the side output voltage or current.
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