JP2005287196A - Brushless dc motor driving apparatus - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To smoothly rotate a three-phase brushless DC motor, and stably and rapidly activate and stop it without providing an expensive and large sensor such as a resolver and an incremental encoder and an expensive and large circuit such as a phase lock loop circuit (a PLL circuit) having a large circuit scale. <P>SOLUTION: An upper limit modulation rate of each phase is set based on each peak level in position detection signals V21-V23 of a rotation detecting section 1 sinusoidally changing in accordance with the rotation of the three-phase brushless DC motor M. Three-phase PWM signals V91-V93 have pulse widths sinusoidally changing in accordance with level changes of the signal V21-V23 and are outputted from a PWM modulation section 6. Currents carried to phases of the motor M are controlled by drive gate signals V41-V43 of a drive section 5 based on the signals V91-V93. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a brushless DC motor driving apparatus that drives a three-phase brushless DC motor.

  Conventionally, when a three-phase brushless DC motor is driven, a position detection sensor represented by a Hall element is used to detect the rotational position (rotation angle) of the rotor (see, for example, Patent Document 1).

  A conventional motor drive circuit using this position detection sensor is configured as shown in FIG. 10 and has three holes arranged around the rotor of the three-phase brushless DC motor M, for example, at intervals of 120 degrees or 60 degrees. As shown in FIGS. 11 (a), 11 (b), and 11 (c), the elements 11, 12, and 13 are signals V11a, V12a, and V13a that change in a sine waveform according to a magnetic field change accompanying the rotation of the rotor, Negative-phase negative signals V11b, V12b, and V13b are output.

  The positive signals V11a to V13a and the negative signals V11b to V13b of the hall elements 11 to 13 are input to the error amplifiers 21, 22, and 23. The error amplifiers 21 to 23 are V11a to V11b and V12a to V12b. , V13a-V13b are calculated, and the binary signals of the error outputs thereof are three-phase signals V21 *, V22 having a rectangular waveform shifted by 120 degrees as shown in (d), (e), and (f) of FIG. *, V23 *, and signals V21 * to V23 * and signals V31, V32, and V33 obtained by inverting the signals V21 * to V23 * by signal inverters (inverter gates) 31, 32, and 33 4 *.

  The drive gate unit 4 * includes upper switching elements 51a, 52a, 53a in series connection of the arms 51, 52, 53 of the drive unit 5 of the subsequent three-phase full-bridge inverter configuration, and a lower switching element 51b, In order to control the switching of 52b and 53b, the AND gates 41a and 41b for supplying the gate signals V41a and V41b of (g) and (h) of FIG. 11 to the switching elements 51a and 51b and the switching elements 52a and 52b of FIG. The AND gates 42a and 42b for supplying the gate signals V42a and V42b of (i) and (j), and the AND gate 43a for supplying the gate signals V43a and V43b of (k) and (l) of FIG. 11 to the switching elements 53a and 53a. 43b.

  Note that each of the switching elements 51a to 53b as the energization current switch is composed of a semiconductor switch such as an IGBT or a power FET. Further, + V in the figure is a positive voltage power supply terminal.

  Also, the gate signals V41a, 42a, 43a are 120 degree energized rectangular wave signals shifted by 120 degrees, and the gate signals V41b, 42b, 43b delayed 180 degrees from the gate signals V41a, 42a, 43a are also shifted by 120 degrees. It is a rectangular wave signal energized 120 degrees.

  Then, by the switching of the switching elements 51a to 53b, the rectangular wave-shaped drive current Iu of each phase shown in (m), (n), and (o) of FIG. Iv and Iw flow, and the Y-connected motor M is driven and rotated by a 120-degree energization rectangular wave drive system.

  In addition, based on the change of the logic gate of the drive gate unit 4 *, etc., the motor M is also driven by energizing 180 degrees by switching of the switching elements 51a to 53b based on the signals V41a to V43b.

  By the way, in the case of the motor drive device of FIG. 10, each signal 41a-43b of the drive gate section 4 * is a rectangular wave signal that is continuously turned on during a 120-degree energization period. Since the motor M is driven by the rectangular wave driving method based on it, the energization switching of each phase of the motor M at the start and end of energization is performed steeply with the large drive currents Iu, Iv and Iw. The noise during rotation increases, and fluctuations in rotational torque occur, resulting in uneven rotation.

  In order to eliminate these inconveniences and drive the motor M quietly and smoothly, in this kind of brushless DC motor driving device, instead of the rectangular wave driving method, the driving currents Iu, Iv, Iw are set. Adopting a sine wave driving method for changing to a sine wave shape is performed (for example, see Non-Patent Document 1).

  In addition, a trapezoidal wave drive method (soft switching method) is employed in which drive currents Iu, Iv, and Iw before and after switching of energization of each phase are gradually increased or decreased in a sine wave shape.

JP 61-277396 A (paragraph [0002], page 2, FIG. 1) Naoki Mijo and Shigenobu Nagamori “New Brushless Motor”, General Electronic Publishing Company, June 1, 2001, P.A. 63-64

  When a three-phase brushless DC motor such as the motor M is driven by a conventional silent driving method such as the sine wave driving method or the trapezoidal wave driving method, the rotational position (phase angle) of the rotor is accurately detected with high resolution. In the non-patent document 1, P.I. 67, P75-76, etc., as the 6-step position detection of the three Hall elements 11 to 13 in FIG. 10 is insufficient and difficult to detect, an electromagnetic resolver or optical type, It is necessary to provide a high-resolution position sensor such as a magnetic incremental encoder.

  In this case, the resolver, the incremental encoder, etc. are expensive and large, and there is a problem that the apparatus cannot be reduced in price and size.

  In addition, it is conceivable to provide a phase-locked loop (PLL circuit) to achieve quietness of the three-phase brushless DC motor by synchronizing the phase during driving. In this case, it is detected except when the motor rotates at a constant speed. It takes time to stabilize the phase synchronization of the motor, and it is difficult to control sufficiently when the motor is started and stopped, so that it cannot be sufficiently quieted. Moreover, because of the PLL circuit configuration, the circuit scale becomes enormous. Expensive and large.

  The present invention has been made in consideration of the above-mentioned points, and without providing an expensive and large sensor such as the resolver or the incremental encoder, and a circuit such as the phase synchronization circuit (PLL circuit). The three-phase brushless DC motor can be smoothly rotated to obtain sufficient silence without providing a large, expensive and large circuit, and can be started and stopped stably and quickly. For the purpose.

  In order to solve the above-described problems, a brushless DC motor driving device of the present invention has three position detection sensors arranged at regular intervals around the rotor of a three-phase brushless DC motor, A rotation detection unit that outputs a three-phase position detection signal that changes sinusoidally according to rotation, and an upper limit modulation rate for each phase is set based on the peak level of each position detection signal. A PWM modulation unit that outputs a three-phase PWM signal whose pulse width changes in a sine wave shape according to the level change, and a three-phase drive gate signal based on each PWM signal, for each phase of the three-phase brushless DC motor And a drive unit having a three-phase inverter circuit configuration for controlling energization (claim 1).

  In the brushless DC motor driving device of the present invention, the driving unit is provided with a driving gate unit that forms positive or negative half cycles of the driving gate signal of each phase by the PWM signal of each phase. Item 2), the PWM modulation unit calculates the errors of the U-phase and V-phase position detection signals, the errors of the V-phase and W-phase position detection signals, and the errors of the W-phase and U-phase position detection signals. An arithmetic circuit unit is provided for adjusting the phase shift of the position detection signal phase from the energization phase of the three-phase brushless DC motor (Claim 3), and the upper limit modulation rate of each phase is the position detection signal. It is also characterized in that it is set to a modulation rate of 100% corresponding to the peak level (Claim 4), and in accordance with the peak level of the correction detection signal of each phase in which the level of each position detection signal is varied in the PWM modulator, The upper limit modulation rate is set to the peak of each position detection signal. Providing the signal level adjusting means for variably from 100% of the modulation factor corresponding to Kleber also characterized (claim 5).

  Furthermore, the brushless DC motor driving device of the present invention includes a PWM modulator that includes a peak hold unit that holds a peak of the position detection signal of each phase, and a charge based on the peak level signal of each phase that is held in the peak hold unit. By repeating the discharge, a frequency signal of each phase that changes in a sawtooth wave shape or a triangular wave shape in a range of the upper limit modulation rate level to the lower limit modulation rate level is formed, and each position detection signal is equal to or higher than each frequency signal. And a PWM circuit section for forming a PWM signal for each phase having a pulse width corresponding to a level period (Claim 6), and a position detection signal for each phase is provided to the PWM modulator section. Sets the phase angle range for PWM control based on the peak hold unit for peak hold and the peak level signal of each phase held in the peak hold unit. A modulation rate setting signal having a level of two phase angles of a phase is formed, and a sawtooth wave is generated in the range of the modulation rate of both modulation rate setting signals by repeating charge and discharge based on a difference signal between the both modulation rate setting signals of each phase. A PWM circuit unit that forms a frequency signal of each phase that changes into a shape or a triangular wave shape, and forms a PWM signal of each phase of a pulse width in a period in which each position detection signal is at a level equal to or higher than each frequency signal (7).

  First, according to the configuration of the first aspect, the three-phase is changed into a sine wave shape according to the rotation of the motor of the rotation detection unit having the three position detection sensors arranged around the rotor of the three-phase brushless DC motor. Based on the position detection signal, a three-phase PWM whose pulse width changes in a sinusoidal shape in accordance with the rotation of the three-phase brushless DC motor from the PWM modulator without providing a large and expensive resolver or incremental encoder. A signal is output, and energization of each phase of the drive unit is controlled by a three-phase drive gate signal based on each PWM signal, and a three-phase brushless DC motor is smoothly rotated by a sine wave drive method to obtain sufficient silence. be able to.

  Further, since the three-phase position detection signal of the rotation detector changes into a sine wave shape according to the rotation of the three-phase brushless DC motor, the circuit scale is enormous and expensive, and a large phase synchronization circuit is not required. Based on the phase position detection signal, the 3-phase brushless DC motor is output stably and quickly with sufficient silence, even when the 3-phase brushless DC motor is started and stopped. Can be started and stopped.

  Therefore, it is possible to smoothly rotate a three-phase brushless DC motor without providing an expensive and large sensor such as a resolver or an incremental encoder, and without a large and expensive circuit such as a phase synchronization circuit. In addition, sufficient silence can be obtained, and the motor can be started and stopped stably and quickly with sufficient silence.

  According to the second aspect of the present invention, since the positive or negative half cycle of the drive gate signal of each phase is set to the PWM signal by the drive gate unit, each arm of the three-phase inverter of the drive unit is PWM controlled by half cycle. Therefore, the energization of each phase of the three-phase brushless DC motor can be controlled in a sine wave shape, and the configuration and control are simplified compared to the case where both positive and negative cycles of each phase drive gate signal are PWM signals. The effect of claim 1 can be achieved.

  Furthermore, according to the configuration of the third aspect, the shift between the phase of the position detection signal of each phase based on the arrangement of the position detection sensors and the energization phase of the three-phase brushless DC motor can be easily performed by phase shift adjustment of the arithmetic circuit unit. According to the configuration of claim 4, the range of the modulation rate of the PWM signal of each phase can be set to 0 to 100%, and the effect of claim 1 can be obtained. Therefore, by changing the level of each position detection signal, the upper limit modulation rate of the PWM signal of each phase can be variably set from 100%, and the speed control of the three-phase brushless DC motor can be easily performed. There are also advantages.

  Next, according to the configuration of claim 6, it is possible to provide a specific configuration of the PWM modulation unit provided with the peak hold unit and the PWM circuit unit.

  According to the configuration of claim 7, since the PWM signal of each phase of the PWM circuit unit is formed only between the rotational positions of the two phase angles set for each phase, each of the three-phase brushless DC motors Based on the setting of the two phase angles of each phase, for example, the phase energization increases or decreases in a sinusoidal shape only before and after switching of the energized phases, and the trapezoidal wave driving method provides the same effect as in the first aspect.

  An embodiment of the present invention will be described with reference to FIGS.

(First embodiment)
First, a first embodiment of the present invention will be described with reference to the block diagram of FIG. 1 and the flowchart of FIG.

  In FIG. 1, the same reference numerals as those in FIG. 10 denote the same components, and the rotation detection unit 1 includes Hall elements 11 to 13 and error amplifiers 21 to 23 as position detection sensors, and the arrangement of the Hall elements 11 to 13 and the motor M In order to correct the 30 ° deviation from the energization phase by adjusting the phase, the three-phase position detection signals V21, V22, and V23 of U, V, and W shown in FIG. Without being binarized, the non-inverting input terminals (+) of the arithmetic circuits 71, 72, 73 of each phase of the arithmetic circuit unit 7 provided in the modulation circuit unit 6, and the inverting inputs of the arithmetic circuits 72, 73, 71 Supply to terminal (-).

  Then, the arithmetic circuit 71 outputs a U-phase sinusoidal signal V71 (= V21−V22) and a signal V71n of the opposite phase shown in FIG. 2B, and the arithmetic circuit 72 has the same V-phase. The sine wave shaped signal V72 (= V22-V23) and its opposite phase signal V72n are output, and the arithmetic circuit 73 outputs the W phase signal V73 (= V23-V21) and its opposite phase signal V63n. At this time, by taking the difference, there is also an advantage that odd-numbered harmonics, particularly third-order harmonics are removed.

  Then, these three-phase signals V71 to V73 are input to the peak hold circuits 81, 82 and 83 of the respective phases of the peak hold unit 8 to detect the peak levels of the signals V71 to V73, as shown in FIG. A hold signal V81 having a peak level of the signal V71 and hold signals V82 and V83 having a peak level of similar signals V72 and V73 are formed.

  Furthermore, even if the sine wave drive of the motor M does not perform PWM control of both of the two switching elements 51a to 53b of the arms 51 to 53 of the driving unit 5, any one of the two switching elements 51a to 53b is possible. Since one PWM control can be used, in this embodiment, the switching elements 51a, 52a, 53a on the upper side of the arms 51 to 53 are controlled by the PWM control in order to reduce the number of parts and simplify the control. The switching elements 51b, 52b, and 53b below the arms 51 to 53 are switched every half cycle according to the positive and negative of each phase as in the prior art.

  That is, based on the three-phase signals V71 to V73 and the peak level hold signals V81 to V83, the upper limit modulation rate is set to 100% by the PWM circuits 81, 82, and 83 of each phase of the PWM circuit unit 9, and the signal V81 Three-phase PWM signals V91, V92, and V93 having a peak level of .about.V83 as a 100% modulation level are formed.

  At this time, as shown in FIG. 2 (c), the PWM signal V91 is a PWM waveform signal whose pulse width changes in a sine wave shape in the range of the modulation rate of 0 to 100%, and the PWM signals V92 and 93 are also shown. The same PWM waveform signal.

  By the way, as shown in the PWM circuit 91 of FIG. 3, the PWM circuits 91 to 93 have the same circuit configuration by a constant current source circuit CCS, comparators CMP1, CMP2, a capacitor C for time constant, and a switching element SW for discharge control. Formed.

  The operation of the PWM circuit 91 in FIG. 3 will be described with reference to FIG. 4 showing the waveforms of the respective parts in FIG. 3. The current of the constant current source circuit CCS is varied according to the level (voltage) of the peak level signal V81. The capacitor C is charged by the current of the constant current source circuit CCS.

  Further, the comparator CMP1 compares the charging voltage Va of the capacitor C with the voltage of the signal V81, and generates a discharge control pulse signal Vb each time the charging voltage Va reaches the voltage of the signal V81 (peak level voltage). The switching element SW is instantly turned on by this signal, the capacitor C is discharged, and the charging voltage Va is reset to zero.

  The charging voltage Va is changed to a sawtooth waveform by repeatedly charging and discharging the capacitor C, and a sawtooth waveform frequency signal having an amplitude value corresponding to the level of the signal V81 is formed. The level of this frequency signal and the signal V71 (Voltage) is compared, and the signal V71 is modulated by the frequency signal of the charging voltage Va to form a PWM signal V91 having a pulse width of a period of V71 ≧ Va and a period of the frequency signal.

  The PWM signal V91 is a PWM waveform signal in which the voltage (peak level voltage) of the signal V81 is 100% modulation level and the reference potential of the signal V81, for example, 0V is 0% modulation level, and the signal V71 is the peak level. This is a sine wave shape in which the pulse width becomes the largest at the rotational position of the electrical angle of 120 degrees.

  The PWM circuits 92 and 93 form the PWM signals V92 and V93 in the same manner, via the PWM signals V91, V92 and V93, and the level adjusters 100a, 101a and 102a for analog level / digital level matching. The three-phase signals V71 to V73 are input to the AND gates 41, 42, and 43 of the drive gate unit 4.

  The drive gate signals V41 to V43 of the AND gates 41 to 43 change, for example, as shown in FIG. 2D, and the PWM signal V91 is sent from the AND gate 41 to the switching element 51a in the positive half cycle of the signal V71. The switching element 51a is switched according to the PWM signal V91, the PWM signal V92 is output from the AND gate 42 to the switching element 52a in the positive half cycle of the signal V72, and the switching element 52a is switched according to the PWM signal V92. Then, the PWM signal V93 is output from the AND gate 43 to the switching element 53a in the positive half cycle of the signal V73, and the switching element 53a is switched according to the PWM signal V93.

  Further, the three-phase signals V71n to V73n via the level adjusters 100b, 101b, and 102b are output to the lower switching elements 51b, 52b, and 53b of the arms 51, 52, and 53, and the negative half cycle of the U-phase is output. Then, the switching element 51b is turned on, the switching element 52b is turned on in the negative half cycle of the V phase, and the switching element 53b is turned on in the negative half cycle of the W phase.

  The switching elements 51b to 53b are switched according to the PWM signals V91 to V93 in each of the three-phase positive half cycles shifted by 120 degrees in electrical angle, and the switching elements 51b to 53b are switched in each of the three-phase negative half cycles. 2 is turned on, the currents Iu, Iv, and Iw in FIGS. 2E, 2F, and 2G flow through the U, V, and W phases of the motor M, respectively.

  At this time, as the switching cycle of the PWM signals V91 to V93 is shortened, the currents Iu, Iv, and Iw of the U, V, and W phases of the motor M become sine waveform signals corresponding to the rotation of the motor M, The motor M rotates smoothly by 180-degree energization drive.

  Then, based on the three-phase position detection signals V21 to V23 that change in a sine wave shape according to the rotation of the motor of the rotation detection unit 1, the PWM modulation unit 6 can perform 3 to 3 without providing a large and expensive resolver or incremental encoder. Three-phase PWM signals V91 to V93 whose pulse width changes in a sine wave shape in accordance with the rotation of the phase brushless DC motor M are output, and the three-phase drive gate signals V41 to V43 based on the PWM signals V91 to V93 are output. The energization of each phase of the drive unit 5 is controlled, and the motor M is smoothly rotated by a sine wave drive method, so that sufficient silence can be obtained.

  Further, the circuit scale is enormous, expensive, and a large phase synchronization circuit is unnecessary, and the three-phase position detection signals V21 to V23 of the rotation detection unit 1 change into a sine wave shape as the motor M rotates. Based on the signals V21 to V23, three-phase PWM signals V91 to V93 corresponding to the rotational position of the motor M are output even when the motor M is started and stopped, so that the motor M can be started and stopped stably and quickly.

  Therefore, the three-phase brushless DC motor M can be made smooth without providing an expensive and large sensor such as a resolver or an incremental encoder, or without a large and expensive circuit such as a phase synchronization circuit. The motor M can be rotated to obtain sufficient silence, and the motor M can be started and stopped quickly with sufficient silence.

  Further, in the case of this embodiment, the arithmetic circuit unit 7 of the PWM modulation unit 6 easily corrects the shift between the phase of the position detection signals V21 to V23 of each phase and the energization phase of the motor M by performing phase shift adjustment. There is an advantage that the drive control of the power supply can be performed.

  In addition, since the positive half cycle of the drive gate signals V41 to V43 of each phase is converted into a PWM signal by the drive gate unit 4, each arm 51 to 53 of the three-phase inverter of the drive unit 5 is PWM-controlled by positive half cycle. Thus, the energization of each phase of the motor M can be controlled in a sinusoidal shape with a modulation rate of 0 to 100%, and both positive and negative cycles of the drive gate signals V41 to V43 of each phase are converted to PWM signals. The configuration and control of the drive gate unit 4 and the like can be simplified.

(Second Embodiment)
Next, a second embodiment will be described with reference to FIGS.

  5 is a block diagram of a PWM circuit 91α that replaces the PWM circuit 91 of FIGS. 1 and 3, and FIG. 6 is a waveform diagram for explaining the operation. In these drawings, the same reference numerals as those in FIGS. Show things.

  In this embodiment, the PWM circuit unit 9 is provided with PWM circuits 91α, 92α, and 93α having the configuration shown in FIG. 5 instead of the PWM circuits 91 to 93, and PWM modulation is performed based on the speed control of the motor M or the like. The upper limit modulation rate is variably set.

  Therefore, for example, in the PWM circuit 91α of FIG. 5, the signal V71 of the arithmetic circuit 71 is supplied to the comparator CMP1 via the variable gain amplifier VGA as the signal level adjusting means, and the gain of the amplifier VGA is set to the speed of the motor M, for example. A signal (correction detection signal) that is varied according to the gain control signal (attenuation amount control signal) Vcnt of the control terminal Gin whose level changes based on the control, and is corrected to increase or decrease in proportion to the control speed of the motor M. V71α is formed.

  Then, the signal V71α is supplied to the comparator CMP1 instead of the signal V71, and the PWM signal V91 having a pulse width during the period of V71α ≧ charge voltage Va is output from the comparator CMP1.

  In this case, the upper limit modulation factor of the PWM signal V91 is not fixed to 100% of the peak level of the signal V81, as is apparent from FIG. 6, but based on the change of the peak level of the signal V71α, for example, An arbitrary modulation rate is set according to speed control.

  For the PWM circuits 92α and 93α (not shown), the upper limit modulation rate of the PWM signals V92 and 93 is based on the change in the peak levels of the signals V72α and V73α corresponding to the signal V71α, as in the PWM circuit 91α. An arbitrary modulation rate is set according to the speed control of the motor M.

  Therefore, in the case of this embodiment, the upper limit modulation rate of the PWM signals V91 to V93 is varied according to, for example, the speed control of the motor M, and the motor is controlled by PWM control of an arbitrary upper limit modulation rate according to the control speed of the motor M and the like. M can be driven, the motor M is driven in a sine wave, and the average driving current is controlled, and for example, the motor M can be smoothly rotated at a speed corresponding to the speed control.

(Third embodiment)
Next, a third embodiment will be described with reference to FIGS.

  7 is a block diagram of a PWM circuit 91β in place of the PWM circuit 91 in FIGS. 1 and 3, and FIG. 8 is a waveform diagram for explaining the operation. In these drawings, the same reference numerals as those in FIGS. The equivalent is shown.

  In this embodiment, PWM circuits 91β, 92β, 93Ββ having the configuration shown in FIG. 7 are provided in the PWM circuit unit 9 shown in FIG. Only in the range of electrical angle), control equivalent to conventional trapezoidal wave control is performed by switching to PWM control.

  Therefore, for example, in the PWM circuit 91β of FIG. 5, for example, a manual operation is performed based on the setting of the control condition from the data of the sine wave value at each phase angle of the sine wave drive preset in the memory (not shown). Alternatively, data D (Φ1) and D (Φ2) of two phase angles Φ1 and Φ2 in order from the smallest selected by automatic control are given to the phase angle data terminals D1in and D2in, and these data D (Φ1) and D (Φ2) is converted into an analog level (voltage) by the digital / analog converters DAC1 and DAC2, and then multiplied by a signal (peak level signal) V81 supplied to the reference terminal ref of both converters DAC1 and DAC2. Modulation rate setting signals of two phase angles Φ1 and Φ2 of sine wave drive, that is, control voltages Vdca1 and Vdca2 (Vdca1 <Vdca2) of the two phase angles Φ1 and Φ2. 2) is formed.

  Further, a difference signal Vdif (= Vdca2−Vdca1) between the control voltages Vdca1 and Vdca2 is calculated by the comparator CMP3, and the current of the constant current control circuit CCS is controlled by the difference signal Vdif and the reference level of the control voltage Vdca1. Then, the charging voltage Vaβ of the capacitor C is controlled to a voltage corresponding to the difference between the rotational phase angles Φ1 and Φ2.

  Then, the comparator CMP2 compares the voltage Vdc2 as the PWM control peak voltage with the charging voltage Vaβ, for example, forms a reset signal Vbβ every 10 degrees, and instantaneously resets the capacitor C by the signal bβ. By repeating charging and discharging of the capacitor C, a frequency signal Vbβ that changes in a sawtooth shape in the range of the voltage Vdca1 to the voltage Vdca2 shown in FIG. 8 is formed.

  Further, the comparator CMP1 compares the signals V71 and Vbβ, and a PWM drive signal having a pulse width during which the signal V71 is equal to or higher than the frequency signal Vbβ is formed as the PWM signal V91.

  At this time, as is apparent from FIG. 8, the U-phase PWM signal V91 changes in the modulation rate to 0 to 100% during the phase angle (electrical angle) of 0 degrees to 90 degrees, and during the other periods. 100% modulation state is maintained.

  Since the current of the constant current control circuit CCS is controlled by the difference signal Vdif, the PWM control cycle is constant regardless of the voltages Vdca1 and Vdca2.

  The PWM circuits 92β and 93β (not shown) operate in the same manner as the PWM circuit 91β. As a result, the V-phase and W-phase PWM signals V92 and 93 have rotational phase angles (0 to 9 degrees corresponding to the U-phase). Electrical angle) The modulation factor changes to 0 to 100% during the period of 120 degrees to 210 degrees and 240 degrees to 330 degrees, and is maintained in the 100% modulation state during the other periods.

  Therefore, in the case of this embodiment, the motor M can be smoothly rotated and driven by the drive control equivalent to so-called trapezoidal wave driving, and the same effect as in the first embodiment can be obtained.

  In the third embodiment, the upper limit modulation rate is fixed at 100%. However, the upper limit modulation rate may be set to an arbitrary modulation rate, and the upper limit modulation rate may be set as in the second embodiment. For example, the modulation rate may be variably set according to the rotation speed of the motor M.

  The present invention is not limited to the above-described embodiment, and various modifications other than those described above can be made without departing from the spirit thereof. For example, the signals V71n to V73n in FIG. The PWM signals shown in FIGS. 2, 4, 6, and 8 are supplied from the AND gates 41 to 43 of the drive gate unit 4 to the hold circuits 81 to 83 and the PWM circuits 91 to 93 and in the negative half cycle of the three phases. The same PWM signal may be output, and the lower switching elements 51b, 52b, 53b of the arms 51, 52, 53 of the drive unit 5 may be switched by PWM control. As shown in FIG. 9 which is a block diagram of the fourth embodiment, absolute value circuits 110, 111, 112 and AND gates 41n, 42n, 43n are added to the configuration of FIG. The absolute value circuits 110 to 112 form absolute value waveform signals V71 to V73, and the drive gate signals V41 to V43 and V41n to V43n provide the upper and lower sides of the arms 51, 52 and 53 of the drive unit 5. The switching elements 51a, 51b, 52a, 52b, 53a, and 53b may be switched by PWM control.

  Next, for example, in FIG. 1, arithmetic circuits 61 to 63 are provided in order to perform a 180-degree energization drive by correcting a 30-degree shift caused by the mounting positions of the Hall elements 11 to 13, but this correction is unnecessary. In such a case, the arithmetic circuits 61 to 63 can be omitted.

  Next, in each embodiment, when the motor M is started, the PWM control drive is prohibited until at least the peak hold circuits 71 to 73 first detect the peak value and output the signals V71 to V73. By performing constant current driving corresponding to 100% modulation, unnecessary control at the initial stage of startup may be prevented to achieve smoother rotation.

  The position detection sensor is not limited to the Hall element, and the arrangement interval of the position detection sensors is not limited to 120 degrees and 60 degrees.

  Further, each PWM circuit 91 to 93 of the PWM circuit unit 9 may form a triangular wave-shaped frequency signal instead of the sawtooth wave-shaped frequency signal.

It is a block diagram of a 1st embodiment of this invention. It is a wave form diagram for operation | movement description of FIG. FIG. 2 is a detailed block diagram of a part of FIG. 1. It is a wave form diagram for operation | movement description of FIG. It is a block diagram of 2nd Embodiment of this invention. FIG. 6 is a waveform diagram for explaining the operation of FIG. 5. It is a block diagram of 3rd Embodiment of this invention. FIG. 8 is a waveform diagram for explaining the operation of FIG. 7. It is a block diagram of 4th Embodiment of this invention. It is a block diagram of a prior art example. It is a wave form diagram for operation | movement description of FIG.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Rotation detection part 11, 12, 13 Hall element 5 Drive part 6 PWM modulation part 7 Arithmetic circuit part 8 Peak hold part 9 PWM circuit part M 3 phase brushless DC motor V21-V23 Position detection signal V41-V43, V41n-V43n drive Gate signal V91 to V93 PWM signal

Claims (7)

Rotation detection that has three position detection sensors arranged at regular intervals around the rotor of a three-phase brushless DC motor and outputs a three-phase position detection signal that changes sinusoidally according to the rotation of the rotor. And
The upper limit modulation rate of each phase is set based on the peak level of each position detection signal, and a three-phase PWM signal whose pulse width changes in a sine wave shape according to the level change of each position detection signal is output. A PWM modulator;
A brushless DC motor drive apparatus comprising: a drive unit having a three-phase inverter circuit configuration for controlling energization of each phase of the three-phase brushless DC motor by a three-phase drive gate signal based on each PWM signal.   2. The brushless DC motor driving apparatus according to claim 1, wherein the driving unit is provided with a driving gate unit that forms positive or negative half cycles of the driving gate signal of each phase by a PWM signal of each phase.   The PWM modulation unit calculates the errors of the U-phase and V-phase position detection signals, the errors of the V-phase and W-phase position detection signals, and the errors of the W-phase and U-phase position detection signals. The brushless DC motor driving device according to claim 1, further comprising an arithmetic circuit unit that adjusts a phase shift of a phase difference between the phase and the energization phase of the three-phase brushless DC motor.   4. The brushless DC motor driving apparatus according to claim 1, wherein the upper limit modulation rate of each phase is set to a modulation rate of 100% corresponding to the peak level of each position detection signal.   A signal for varying the upper limit modulation rate from a modulation rate of 100% corresponding to the peak level of each position detection signal according to the peak level of the correction detection signal of each phase whose level is changed for each position detection signal. 4. The brushless DC motor driving apparatus according to claim 1, further comprising a level adjusting unit.   The PWM modulation unit has a peak hold unit for peak-holding the position detection signal of each phase, and charging and discharging are repeated based on the peak level signal of each phase held in the peak hold unit. Each frequency signal having a pulse width corresponding to a period in which each position detection signal is higher than each frequency signal is formed by forming a frequency signal of each phase that changes in a sawtooth waveform or a triangular waveform in the range of the modulation factor level. 6. A brushless DC motor driving apparatus according to claim 1, further comprising a PWM circuit section for generating a PWM signal.   Based on the peak hold unit for peak-holding the position detection signal of each phase in the PWM modulation unit and the signal of the peak level of each phase held in the peak hold unit, the phase angle range of the PWM control is set. A modulation rate setting signal having a level of two phase angles is formed, and by repeating charge / discharge based on a difference signal between the two modulation rate setting signals of each phase, a sawtooth waveform or A PWM circuit unit is provided that forms a frequency signal of each phase that changes to a triangular wave shape, and forms a PWM signal of each phase with a pulse width in a period in which each position detection signal is at a level equal to or higher than each frequency signal. The brushless DC motor drive device according to any one of claims 1 to 5.

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