JP2005244860A - High frequency switching module and communication apparatus using the same - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a high frequency switching module and communication apparatus using the same in which phase balance can be adjusted without damaging flatness of amplitude balance and phase balance. <P>SOLUTION: In a high frequency switching module wherein a switching element, a capacitor, an inductor and at least one or more SAW filters are integrally formed on a dielectric substrate, at least one of the SAW filters is an unbalanced/balanced output type SAW filter, and at least one of two transmission lines connected, respectively between two balanced output side terminals of the unbalanced/balanced output type SAW filter and two output terminals of a reception circuit formed on a surface of the dielectric substrate, and a ground electrode formed inside the dielectric substrate are arranged so as not to overlap with each other when the dielectric substrate is viewed in the direction of lamination. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a high-frequency switch module that is used in a high-frequency band such as a microwave band and handles a transmission / reception system with a single antenna, and a communication device using the same.

  In recent years, ceramic RF devices have attracted much attention as contributing greatly to the miniaturization of high-frequency radio equipment such as mobile phones. Mobile phones such as EGSM and DCS, which are widely used in Europe, have recently used balanced RF devices with two signal terminals in the RF circuit to lower the noise figure and increase the reception sensitivity. Has been proposed. The balanced circuit is more resistant to noise than the unbalanced circuit with one signal terminal. FIG. 9 shows a circuit block diagram of an EGSM and DCS dual-band mobile phone. Demultiplexer 120, high frequency switches 121 and 122, low pass filters 123 and 124, RF interstage filters (SAW filters) 125 and 126, low noise amplifiers (LNA) 127 and 128, directional couplers 131 and 132, amplifiers (PA) ) 133, 134 etc. RF devices are provided. Due to the above demand, recently, the balanced input LNA is being used instead of the unbalanced input LNA. In this case, in order to connect the balanced input type LNAs 127 and 128 and the interstage filters (SAW filters) 125 and 126 arranged in the preceding stage, a conventional balanced type that converts between an unbalanced type circuit and a balanced type circuit is used. -The unbalance conversion transformers 129 and 130 were used. Each of these RF devices is composed of independent components, and the RF devices are mounted on a printed circuit board and connected to each other using a connection line such as a microstrip line.

  However, using a balanced-unbalanced conversion transformer inevitably increases the number of components. For example, dual-band mobile phones that support the EGSM and DCS systems must be equipped with a balanced-unbalanced conversion transformer in each transmission / reception system, and can only meet the demands for miniaturization and price reduction. In addition, there is a problem that the mounting area increases due to an increase in the number of components.

  For this problem, for example, a high-frequency switch module as disclosed in Patent Document 1 is disclosed. This high-frequency switch module is an unbalanced input / balanced output type SAW filter, which is a monolithic unit, a duplexer, a high frequency switch for switching transmission and reception, a low-pass filter, and an unbalanced input / balanced output type SAW filter. Reduces the number of parts, and by mounting the SAW filter on the laminated dielectric substrate, the conductor line between the antenna and the SAW filter is shortened, and downsizing and improved insertion loss are realized. .

JP 2003-152590 A

  When using an unbalanced input / balanced output type SAW filter as in Patent Document 1, if the amplitude balance and phase balance of the signal output from the SAW filter are not good, they are input to a low noise amplifier connected to the subsequent stage. As a result, the low-noise amplifier becomes more susceptible to external noise, causing problems such as transmission. Therefore, it is preferable that the amplitude balance is within ± 1 dB and the phase balance is within 180 ± 10 °. When the phase balance characteristic of the output signal of the SAW filter deviates from 180 ° due to the characteristics of the SAW filter or due to the influence of the circuit upstream of the SAW filter, the high frequency switch module of Patent Document 1 adjusts the phase balance. Is difficult.

  The high-frequency switch module of the present invention solves this problem, and provides a high-frequency switch module capable of adjusting the phase balance without impairing the flatness of the amplitude balance and phase balance, and a communication device using the same. With the goal.

  The high frequency switch module of the present invention is a high frequency switch module in which a switching element, a capacitor, an inductor, and at least one SAW filter are integrally formed on a dielectric substrate, wherein at least one of the SAW filters is an unbalanced input Two transmissions connected to a balanced output type SAW filter between two balanced output side terminals of the unbalanced input / balanced output type SAW filter and two receiving terminals formed on the dielectric substrate, respectively. At least one of the transmission lines and a ground electrode formed in the dielectric substrate are arranged so as not to overlap each other when the dielectric substrate is viewed in the stacking direction.

  The high-frequency switch module of the present invention is a high-frequency switch module that appropriately switches the connection between a plurality of transmission / reception system transmission circuits or reception circuits having different passbands and sharing the antenna, and the signals of the plurality of transmission / reception systems A demultiplexing circuit composed of a low-pass filter and a high-pass filter, a first switch circuit that is connected to the low-pass filter of the demultiplexing circuit and switches between a transmission path and a reception path; A second switch circuit that is connected to the wide-band filter of the wave circuit and switches between a transmission path and a reception path; a low-pass filter provided in the transmission path of the first switch circuit and the second switch circuit; Each receiving path from the switch circuit and the second switch circuit to the receiving circuit has a SAW filter, and the demultiplexing circuit and the low-pass filter are LC circuits. The switch circuit is mainly composed of a diode and a transmission line, the branching circuit and the LC circuit of the low-pass filter, and at least a part of the transmission line of the switch circuit is a laminated substrate of an electrode pattern and a dielectric layer In the antenna switch module mounted on the dielectric substrate, the chip element that is configured by the electrode pattern and forms part of the SAW filter, the diode, and the LC circuit, at least one of the SAW filters is: An unbalanced input / balanced output type SAW filter, between two balanced output side terminals of the unbalanced input / balanced output type SAW filter and two output terminals of a receiving circuit formed on the dielectric substrate. At least one of the two transmission lines connected to each other and a ground electrode formed in the dielectric substrate include the induction electrode. Characterized by being arranged so as not to overlap each other when viewed body substrate in the stacking direction.

  The high-frequency switch module of the present invention is a high-frequency switch module that appropriately switches the connection between a plurality of transmission / reception system transmission circuits or reception circuits having different passbands and sharing the antenna, and the signals of the plurality of transmission / reception systems A demultiplexing circuit composed of a low-pass filter and a high-pass filter, a first switch circuit that is connected to the low-pass filter of the demultiplexing circuit and switches between a transmission path and a reception path; Connected to the high-pass filter of the wave circuit, connected to the second switch circuit for switching the transmission path and the reception path and the second switch circuit, for switching two reception paths among the plurality of transmission / reception system reception circuits A third switch circuit; a low-pass filter provided in a transmission path of the first switch circuit and the second switch circuit; and the first switch circuit. Each receiving path connected from the third switch circuit to the receiving circuit has a SAW filter, the branching circuit and the low-pass filter are composed of LC circuits, the switch circuit is mainly composed of a diode and a transmission line, and the branching At least a part of the circuit and the LC circuit of the low-pass filter and the transmission line of the switch circuit are configured by the electrode pattern in the laminated substrate of the electrode pattern and the dielectric layer, and one of the SAW filter, the diode, and the LC circuit. In the antenna switch module mounted on the dielectric substrate, at least one of the SAW filters is an unbalanced input / balanced output SAW filter, and the unbalanced input / balanced output Between the two balanced output side terminals of the SAW filter and the two output terminals of the receiving circuit formed on the dielectric substrate. At least one of the two transmission lines to be transmitted and a ground electrode formed in the dielectric substrate are arranged so as not to overlap each other when the dielectric substrate is viewed in the stacking direction. And

  The high-frequency switch module of the present invention is characterized in that the two transmission lines have different lengths.

  According to the present invention, in a high-frequency switch module in which a balanced input / unbalanced output type SAW filter, a switching element, a capacitor, and an inductor are integrally formed on a dielectric substrate, two balanced output side terminals of the SAW filter and the dielectric By arranging two transmission lines respectively connected between two output terminals formed on the substrate surface and a ground electrode formed in the dielectric substrate without overlapping each other, an amplitude balance, It is possible to provide a high frequency switch module capable of outputting a reception signal with a stable phase balance. Further, by making the lengths of the two transmission lines different, the level of the phase balance can be adjusted without impairing the stability of the amplitude balance and the phase balance. Furthermore, since chip components such as switch elements, resistors, capacitors, inductors, and SAW filters are mounted on the laminated dielectric substrate, a smaller and less expensive high-frequency switch module can be provided.

  A first embodiment according to the present invention is shown in FIGS. This embodiment is a triple band type high-frequency switch module compatible with EGSM, DCS, and PCS. FIG. 1 is a circuit block diagram of this embodiment, FIG. 2 is an equivalent circuit diagram, and FIG. The dielectric sheet and electrode pattern figure to be shown are shown. FIG. 4 shows the arrangement of the transmission line electrode pattern and the ground electrode on the balanced output side of the unbalanced input / balanced output SAW filter, and is a view transparently seen from the top surface of the dielectric substrate. FIG. 5 is a perspective view showing the appearance of the dielectric substrate.

  The demultiplexer DIP includes an EGSM system (transmission frequency: 880 to 915 MHz, reception frequency: 925 to 960 MHz), an 880 to 960 MHz band signal, a DCS system (transmission frequency 1710 to 1785 MHz, reception frequency 1805 to 1880 MHz), and a PCS system ( A signal from an antenna in a 1710 to 1990 MHz band having a transmission frequency of 1850 to 1910 MHz and a reception frequency of 1930 to 1990 MHz is demultiplexed. The switch circuit SW1 switches the EGSM system signal demultiplexed by the duplexer DIP to the transmission terminal EGMS TX and the reception terminals EGSM RX1 and EGSM RX2. The switch circuit SW2 switches the DCS system and PCS system signals demultiplexed by the duplexer DIP to the transmission terminal DCS / PCS TX, the reception terminals DCS RX1, DCS RX2, and the reception terminals PCS RX1, PCSRX2. The first low-pass filter LPF1 passes the EGSM transmission signal and suppresses the higher-order harmonic distortion contained in the transmission signal input from the power amplifier on the EGSM side, and has a frequency more than twice that of the EGSM transmission signal. A filter having a characteristic that sufficiently attenuates is used. Similarly, the second low-pass filter LPF2 sufficiently attenuates a frequency twice or more that of the DCS and PCS transmission signals in order to suppress high-order harmonic distortion included in the transmission signals from the DCS and PCS sides. A filter having such characteristics is used. Thereby, the amount of harmonics generated by the power amplifier and radiated from the antenna ANT can be reduced. The SAW filter SAW1 is a bandpass filter that uses the EGSM reception frequency as a passband, and blocks an unnecessary frequency band signal of the reception signal received from the antenna ANT. Similarly, SAW filters SAW2 and SAW3 are bandpass filters that use DCS and PCS reception frequencies as passbands, respectively, and block signals in unnecessary frequency bands. SAW1, SAW2, and SAW3 are all unbalanced input / balanced output SAW filters, and SAW1 outputs two signals that are 180 degrees out of phase to the receiving terminals EGSM RX1 and EGSM RX2, respectively. The same applies to SAW2 and SAW3. SAW2 outputs signals to DCS RX1 and DCS RX2, and SAW3 outputs signals to PCS RX1 and PCS RX2.

  FIG. 2 shows an equivalent circuit of this embodiment. The duplexer DIP includes transmission lines lf1 to lf3 and capacitors cf1 to cf4. It is desirable that the transmission line lf2 and the capacitor cf1 form a series resonance circuit and be designed to have a resonance frequency in the DCS and PCS frequency bands. In this embodiment, the resonance frequency is adjusted to 1800 MHz. Further, it is desirable that the transmission line lf3 and the capacitor cf3 form a series resonance circuit and be designed to have a resonance frequency in the EGSM frequency band. In this embodiment, the resonance frequency is set to 900 MHz. With this circuit, it is possible to demultiplex and synthesize the EGSM signal and the DCS and PCS signals. The transmission line lf1 is preferably set to a certain length so as to have a high impedance for the frequency of the DCS and PCS signals. This makes it difficult for DCS and PCS signals to be transmitted to EGSM circuits. On the other hand, the capacitors cf2 and cf4 are preferably set to relatively small capacitance values so as to have a high impedance for the frequency of the EGSM signal. This makes it difficult to transmit EGSM signals to DCS and PCS circuits.

  The first switch circuit SW1 includes capacitors cg1, cg5, cg6 transmission lines lg2, lg3, PIN diodes Dg1, Dg2, and a resistor Rg. The lengths of the transmission lines are set so that the transmission lines lg2 and lg3 become λ / 4 resonators in the EGSM transmission frequency band. However, the transmission line lg2 may be a choke coil whose ground level appears to be open (high impedance state) at the EGSM transmission frequency. In this case, the inductance value is desirably about 10 to 100 nH. The resistor Rg determines the current flowing through the PIN diodes Dg1 and Dg2 when a voltage is applied to the control terminal Vc1. In this embodiment, 100 to 200Ω was used. Capacitors cg1, cg5, and cg6 are necessary for DC cutting of the control terminal. In a state where a voltage is applied to the control terminal Vc1, a parasitic inductance such as a connection wire exists in the PIN diode Dg2. For this reason, series resonance with the capacitor cg5 is performed so as to cancel this. The capacitance value of the capacitor cg5 is set as appropriate. As described above, when a voltage is applied to the control terminal Vc1, the PIN diodes Dg1 and Dg2 are both turned on, and the connection point between the PIN diode Dg2 and the transmission line lg3 becomes the ground level, and the transmission line is a λ / 4 resonator. The impedance on the opposite side of lg3 becomes infinite. Therefore, when a voltage is applied to the control terminal Vc1, no signal is transmitted through the path between the duplexer DIP and the EGSM reception terminal EGSM RX1 and EGSM RX2, and the path between the duplexer DIP and the EGSM transmission terminal EGSM. It becomes easy to transmit a signal. On the other hand, when no voltage is applied to the control voltage Vc1, the PIN diode Dg1 is also turned off, no signal is transmitted through the path between the duplexer DIP and the EGSM transmission terminal EGSM TX, and the PIN diode Dg2 is also turned off. Therefore, the signal is easily transmitted through the path between the duplexer DIP and the EGSM reception terminals EGSM RX1 and EGSMRX2. With the above configuration, transmission / reception of the EGSM signal can be switched.

  The second switch circuit SW2 includes capacitors cd4, cp1 to cp3, transmission lines ld3, lp1, lp2, PIN diodes Dd1, Dd2, Dp1, Dp2, and resistors Rd, Rp. The lengths of the transmission lines are set so that the transmission lines ld3, lp1, and lp2 become λ / 4 resonators at the frequency of the DCS to PCS signals. However, the transmission line ld2 can be replaced by a choke coil whose ground level appears to be open (high impedance state) at the DCS and PCS transmission frequencies. In this case, the inductance value is preferably about 5 to 60 nH. The resistor Rd determines the current that flows through the PIN diodes Dd1 and Dd2 when a voltage is applied to the control terminal Vc2. In this example, 100Ω to 200Ω was used. The resistor Rp determines the current that flows through the PIN diodes Dp1 and Dp2 when a voltage is applied to the control terminal Vc3. In this example, 100 to 2000Ω was used. Capacitors cd4, cp1, and cp2 are necessary for DC cutting of the control terminal. Further, when a voltage is applied to the control terminal Vc2, a parasitic inductance such as a connection wire exists in the PIN diode Dd2, so that the capacitance value of the capacitor cd4 is set so as to resonate in series with the capacitor cd4. As described above, when a voltage is applied to the control terminal Vc2, the PIN diodes Dd1 and Dd2 are both turned on, and the connection point between the PIN diode Dd2 and the transmission line ld3 becomes the ground level, and the transmission line is a λ / 4 resonator. The impedance on the opposite side of ld3 becomes infinite. Therefore, when a voltage is applied to the control terminal Vc2, no signal is transmitted through the path between the duplexer DIP to the DCS reception terminals DCS RX1, DCS RX2, and the PCS reception terminals PCS RX1, PCS RX2. Signals are easily transmitted on the path between DIP to DCS and PCS transmission terminals DCS / PCS TX. On the other hand, when no voltage is applied to the control terminal Vc2, the PIN diode Dd1 is turned off, and no signal is transmitted between the duplexers DIP to DCS and the PCS transmission terminal DCS / PCS TX, and the PIN diode Dd2 Since the signal is also in the OFF state, signals are easily transmitted through the paths between the duplexers DIP to DCS reception terminals DCS RX1 and DCS RX2 and the PCS reception terminals PCS RX1 and PCS RX2.

  Further, when a voltage is applied to the control terminal Vc3, since the parasitic inductance such as a connection wire exists in the PIN diode Dp2, the capacitance value of the capacitor cp2 is set so as to resonate in series with the capacitor cp2. As a result, when a voltage is applied to the control terminal Vc3, both the PIN diodes Dp1 and Dp2 are turned on, the connection point between the PIN diode Dp2 and the transmission line lp2 becomes the ground level, and the transmission is a λ / 4 resonator. The impedance on the opposite side of the line lp2 becomes infinite. Accordingly, in a state where a voltage is applied to the control terminal Vc3, signals are not transmitted to the DCS reception terminals DCS RX1 and DCS RX2, and signals are easily transmitted to the PCS reception terminals PCS RX1 and PCS RX2. Conversely, when no voltage is applied to the control terminal Vc3, the PIN diode Dp1 is in the OFF state, no signal is transmitted to the PCS reception terminals PCS RX1 and PCS RX2, and the PIN diode Dp2 is also in the OFF state. Therefore, the signal is easily transmitted to the DCS receiving terminals DCS RX1 and DCS RX2. With the above configuration, when the voltage is applied to the control terminal Vc2, the DCS and PCS transmission terminals DCS / PCS TX are not applied to the control terminal Vc2, and when the voltage is applied to Vc3, the PCS reception is performed. When no voltage is applied to any of the control terminals Vc2 and Vc3 to the terminals PCS RX1 and PCS RX2, switching to the DCS reception terminals DCS RX1 and DCS RX2 is possible.

  The first low-pass filter LPF1 is a π-type low-pass filter composed of a transmission line lg1 and capacitors cg2 to cg4. The transmission line lg1 and the capacitor cg2 constitute a parallel resonance circuit, and the resonance frequency is set to twice or three times the EGSM transmission frequency. In this embodiment, it is set to 3700 times 2700 MHz. With the above configuration, harmonic distortion included in the transmission signal on the EGSM side input from the power amplifier can be removed. In FIG. 2, the first low-pass filter LPF1 is arranged between the PIN diode Dg1 of the first high-frequency switch SW1 and the transmission line lg2, which is composed of the duplexer DIP and the first high-frequency switch. Or between the transmission line lg2 and the EGSM transmission terminal EGSM TX. If a capacitor connected to the ground of the first low-pass filter LPF1 is arranged in parallel with the transmission line lg2, a parallel resonance circuit is formed, and the line length of the transmission line lg2 is made shorter than λ / 4. In addition, the inductance value of the choke coil can be reduced.

  The second low-pass filter LPF2 is a π-type low-pass filter including the transmission line ld1 and the capacitors cd1 to cd3. Here, the transmission line ld1 and the capacitor cd1 constitute a parallel resonance circuit, and the resonance frequency is set to a frequency twice or three times the DCS to PCS transmission frequency. In this embodiment, it is set to 3600 MHz which is double. With the above configuration, harmonic distortion included in the DCS and PCS side transmission signals input from the power amplifier can be removed. Similarly to the first low-pass filter LPF1, the second low-pass filter LPF2 may be disposed between the duplexer DIP and the second high-frequency switch SW2, or the transmission line ld2, DCS, and PCS. You may arrange | position between transmission terminals DCS / PCS TX. In the embodiment of FIG. 2, the first and second low-pass filters LPF1 and LPF2 are configured between the PIN diode Dg1 and the transmission line lg2, and between the PIN diode Dd1 and the transmission line ld2, and thus are switched circuits. Is provided. This is preferable in terms of circuit design, but is not essential. The low-pass filter may be provided at any position on the transmission path between the duplexer through which the transmission signal passes and the transmission terminal.

  A SAW filter SAW1, an inductor ls1, and transmission lines ls2, ls3 are connected between the capacitor cg6 and the EGSM reception terminals EGSM RX1, EGSM RX2. SAW filter SAW1 is an unbalanced input / balanced output SAW filter, and transmission lines ls2 and ls3 are provided between two balanced output terminals of SAW1 and EGSM reception terminals EGSM RX1 and EGSM RX2, respectively. If the phase balance at the EGSM reception frequency is about 180 °, the transmission lines ls2 and ls3 are as short as possible and preferably the same length, but if the phase balance deviates from 180 °, one of the transmission lines ls2 and ls3 By making one longer than the other, the phase balance can be adjusted to around 180 °. In this embodiment, the lengths of ls2 and ls3 are almost the same, and are set to 0.5 to 2.0 mm. The inductor ls1 is provided between one end of the transmission lines ls2 and ls3 and the EGSM reception terminals EGSM RX1 and EGSM RX2, and is connected in parallel between the EGSM RX1 and EGSM RX2. The inductor ls1 can adjust the impedance matching viewed from the EGSM receiving terminal to an optimum state. The inductor ls1 can be easily adjusted for impedance matching later by using an external element such as a chip inductor. In this example, 10 to 100 nH was used.

  A SAW filter SAW2, an inductor ls4, and transmission lines ls5 and ls6 are connected between the anode terminal of the PIN diode Dp2, the transmission line lp2, the contact of the capacitor cp3 and the DCS reception terminals DCS RX1 and DCS RX2. The SAW filter SAW2 is an unbalanced input / balanced output SAW filter, and transmission lines ls5 and ls6 are provided between two balanced output terminals of the SAW2 and the DCS reception terminals DCS RX1 and DCS RX2, respectively. When the phase balance at the DCS reception frequency is about 180 °, the transmission lines ls5 and ls6 are as short as possible and preferably the same length, but when the phase balance is shifted from 180 °, one of the transmission lines ls5 and ls6 By making one longer than the other, the phase balance can be adjusted to around 180 °. In this embodiment, the transmission line ls6 is longer than ls5, ls6 is set to 2.0 to 10.0 mm, and ls5 is set to 0.5 to 1.0 mm. The inductor ls4 is provided between one end of the transmission lines ls5 and ls6 and the DCS reception terminals DCS RX1 and DCS RX2, and is connected in parallel between the DCS RX1 and DCS RX2. The inductor ls4 can adjust the impedance matching viewed from the DCS receiving terminal to an optimum state. The inductor ls4 can easily adjust the impedance matching later by using an external element such as a chip inductor. In this example, 4 to 20 nH was used.

  A SAW filter SAW3, an inductor ls7, and transmission lines ls8 and ls9 are connected between the anode terminal of the PIN diode Dp1, the contact of the transmission line lp1, and the PCS reception terminals PCS RX1 and PCS RX2. SAW filter SAW3 is an unbalanced input / balanced output SAW filter, and transmission lines ls8 and ls9 are provided between two balanced output terminals of SAW3 and PCS reception terminals PCS RX1 and PCS RX2, respectively. When the phase balance at the PCS reception frequency is about 180 °, the transmission lines ls8 and ls9 are as short as possible and preferably the same length, but when the phase balance deviates from 180 °, one of the transmission lines ls8 and ls9 By making one longer than the other, the phase balance can be adjusted to around 180 °. In this embodiment, the lengths of ls8 and ls9 are almost the same, and are set to 0.5 to 2.0 mm. The inductor ls7 is provided between one end of the transmission lines ls8 and ls9 and the PCS reception terminals PCS RX1 and PCS RX2, and is connected in parallel between the PCS RX1 and the PCS RX2. The inductor ls7 can adjust the impedance matching viewed from the PCS receiving terminal to an optimum state. The inductor ls7 can be easily adjusted for impedance matching later by using an external element such as a chip inductor. In this example, 4 to 20 nH was used.

  In the present invention, a duplexer, a switch circuit, a transmission line constituting a low-pass filter, etc., and a part of a capacitor are built in a dielectric laminated substrate, and a PIN diode, a resistor, a capacitor constituting a part of the switch circuit, By mounting a chip component such as an inductor on a dielectric multilayer substrate, a small and inexpensive multiband high-frequency switch module can be obtained.

FIG. 3 is a view showing a green sheet and an electrode pattern constituting the dielectric laminated substrate of the high-frequency switch module shown in the equivalent circuit of FIG. The green sheets 1-15 are laminated | stacked in order from the top. The green sheet 16 is the back surface of the green sheet 15. The green sheet 1 is printed with a land electrode 17 for mounting a PIN diode, a chip resistor, a chip inductor, and a chip capacitor, and a land electrode 18 for mounting a metal shield (metal case). In addition, a via-hole electrode 19 (indicated by a black circle in the figure) that connects electrode patterns formed on different green sheets is formed.
The green sheet 16 includes ground terminals 99 and 100, antenna terminals 101, EGSM transmission terminals 102, DCS / PCS transmission terminals 103, EGSM reception terminals 104 and 105, DCS reception terminals 106 and 107, PCS reception terminals 108 and 109, and a power source. Terminals 110 to 112 are formed.
The green sheets 6 to 11 are printed with line electrode patterns mainly serving as transmission lines, and the green sheets 3, 4, 5, 12, 13, and 14 mainly form capacitors, and are electrode patterns for capacitors. Is printed. Further, ground electrodes 20 to 23 are printed on the green sheets 5, 12, 13, and 15.

In the following, the correspondence with the equivalent circuit of FIG. 2 will be described. In FIG. 3, reference numerals 24 to 33 denote transmission lines constituting the duplexer DIP. 24 to 29 form lf1, 30 to 33 form lf2, and 34 to 38 form lf3. Reference numerals 72 to 79 correspond to the electrode patterns for capacitors constituting the duplexer DIP. 72 is cf1, 73 to 75 is cf2, 76 is cf3, and 77 to 79 is cf4. Reference numerals 39 to 47 denote transmission lines constituting the switch circuit SW1, and lg2 is formed from 39 to 44, and lg3 is formed from 45 to 47. 82 forms the capacitor electrode cg5 of the switch circuit SW1. Reference numerals 53 to 68 denote transmission lines constituting the switch circuit SW2, and ld2 is formed by 53 to 58, ld3 is formed by 59 to 61, lp1 is formed by 62 to 65, and lp2 is formed by 66 to 68. 86 to 88 correspond to the capacitor electrode pattern of the switch circuit SW2, and 86 and 22, 23 form cd4, 87 and 22, 23 form cp1, and 88 and 20 form cp2. 48 to 52 form a transmission line lg1 constituting the first low-pass filter LPF1. 80 and 81 correspond to the capacitor electrode pattern of the first low-pass filter LPF1. 80 and 81 form cg2, 80 and 21 form cg3, and 81 and 23 form cg4. 48 to 52 form a transmission line ld1 of the second low-pass filter LPF2. Reference numerals 83 to 85 correspond to the capacitor electrode pattern of the second low-pass filter LPF2. 83 and 85 form cd1, 84 and 20 form cd2, and 85 and 23 form cd3.
89 and 90 correspond to the transmission line between the SAW filter SAW1 and the EGSM receiving terminal, 89 forms ls3, and 90 forms ls2. 91 to 96 correspond to the transmission line between the SAW filter SAW2 and the DCS receiving terminal, 91 forms ls5, and 92 to 96 form ls6. Reference numerals 97 and 98 correspond to transmission lines between the SAW filter SAW3 and the PCS reception terminal, and 98 forms ls8 and 97 forms ls9. The through-hole electrode 19 performs electrical connection between the green sheets.

When the electrode patterns 92 to 96 forming the transmission line overlap with the ground electrode, the electrode patterns 92 to 96 are formed on different green sheets, respectively, and the layer distance to the ground electrode is different and the characteristic impedance is changed. In addition, since the distance between the electrode patterns 92 to 96 and the ground electrode is as narrow as 0.2 to 0.4 mm, the impedance change is large. For this reason, the change in the impedance of the transmission line with respect to the frequency of the received signal also increases, and the flatness of the phase balance within the received frequency band is impaired.
FIG. 4 is a diagram showing the positional relationship between the electrode pattern 94 printed on the green sheet 7 and the ground electrode 23 printed on the green sheet 15. In the figure, the dielectric multilayer substrate is seen transparently from above the mounting surface, and the electrode pattern 94 of the green sheet 7 is drawn on the green sheet 15. Thus, the electrode pattern 94 and the ground electrode 23 are arranged at positions that do not overlap. Although not shown, the other electrode patterns 92, 93, 95, and 96 that form ls6 are similarly arranged so as not to overlap with the ground electrodes 20 to 23. As a result, it is possible to eliminate a decrease in characteristic impedance due to interference with the ground electrode of the transmission line and obtain a stable phase balance characteristic of the received signal. At this time, a part of the electrode pattern 91 forming ls5 is arranged at a position overlapping with the ground electrode 23, but ls5 is shorter in length than ls6 and has little influence on the stability of the phase balance characteristic due to the impedance reduction. Therefore, the stability of the phase balance is not impaired. In this embodiment, part of the electrode patterns 89, 90, 97, and 98 that form the other transmission lines ls2, ls3, ls8, and ls9 also overlap with the ground electrode 23, but all have a length of 2.0 mm or less. Since it is short, it does not impair the phase balance characteristics of the received signal. The overlapping area between each transmission line and the ground electrode is preferably 50% or less of the area of the electrode pattern forming the transmission line.

  The green sheet used in this example uses a ceramic dielectric material that can be fired at a low temperature of 950 ° C. or lower, and has a sheet thickness of 40 to 200 μm so that a transmission line and a capacitor can be easily formed. The ceramic green sheets 1 to 12 are laminated and pressure-bonded, and then fired at 950 ° C. to obtain a laminated body of the high-frequency switch module. Further, as shown in FIG. 5, by mounting a diode 114, a chip resistor 115, a chip inductor 116, a chip capacitor 117, and a SAW filter 118 on the top surface of the multilayer body 113, the high-frequency switch module shown in the equivalent circuit of FIG. can get.

FIG. 6 and FIG. 7 show the evaluation results of the phase balance and amplitude balance at the DCS reception frequency of the high frequency switch module configured as described above. As shown in FIG. 6, the phase balance characteristic (thick line in FIG. 6) in the present embodiment has a phase balance level of nearly 180 ° without substantially changing the waveform of the characteristic before adjustment of the phase balance (thin line in FIG. 6). It was possible to adjust to. On the other hand, the lengths of the transmission lines ls5 and ls6 are the same as in this embodiment, but the phase balance characteristic (dotted line in FIG. 6) when overlapping with the ground electrode is the waveform of the characteristic before phase balance adjustment. It turns out not to maintain. Further, as shown in FIG. 7, the amplitude balance characteristic before the phase balance adjustment (thin line in FIG. 7) and the amplitude balance characteristic in this embodiment (thick line in FIG. 7) are substantially the same at the DCS reception frequency. The lengths of the transmission lines ls5 and ls6 are the same as in this embodiment, but the amplitude balance characteristic (dotted line in FIG. 7) when overlapping with the ground electrode is greatly different, and the flatness of the amplitude balance is significantly lost. I found out.
FIG. 8 shows the phase balance characteristics when the transmission line ls5 is made longer than ls6, in contrast to the present embodiment, before the phase balance adjustment, and contrary to the present embodiment. As shown in FIG. 8, the phase balance characteristic (dotted line in FIG. 8) in the case where the transmission line ls5 is longer than ls6 shifts the phase balance level as a whole lower than the characteristic before the phase balance adjustment. I understand. That is, it has been found that it is effective to make ls5 longer than ls6 when adjusting the phase balance level to a lower level.
From the above results, it can be seen that according to the configuration of the present embodiment, the level of the phase balance can be adjusted without losing the flatness of the amplitude balance and the phase balance characteristics. It can also be seen that the phase balance level can be adjusted in a desired direction by appropriately changing the length or thickness relationship of the transmission lines ls5 and ls6.

  In this embodiment, the phase balance level in the DCS received signal is adjusted. However, it is apparent that the phase balance level can be adjusted for other received signals such as EGSM and PCS.

  The present invention is not limited to the embodiments described above. In the above-described embodiment, the triple band type high frequency switch module corresponding to EGSM, DCS, and PCS is used. However, a dual band type and a four band type high frequency switch module may be used. Further, the present invention is not limited to the above-described high-frequency switch module, and the present invention is effective for other electronic components that integrally form a dielectric multilayer substrate and an unbalanced input / balanced output SAW filter.

  The high-frequency switch module of the present invention can be used for mobile communication terminal devices such as mobile phones.

It is a figure which shows the circuit block diagram of the high frequency switch module which concerns on this invention. It is a figure which shows the equivalent circuit schematic of the high frequency switch module which concerns on this invention. It is a figure which shows the electrode pattern of the green sheet which comprises the laminated body of the high frequency switch module shown by the equivalent circuit of FIG. It is a figure which shows the positional relationship of the transmission line connected between a ground electrode, a DCS SAW filter, and a DCS receiving terminal in the electrode pattern of the green sheet of FIG. It is a perspective view of the high frequency switch module shown by the equivalent circuit schematic of FIG. It is a figure which shows the phase balance characteristic of the high frequency switch module of a present Example. It is a figure which shows the amplitude balance characteristic of the high frequency switch module of a present Example. It is a figure which shows the phase balance characteristic of the high frequency switch module of a present Example. It is a figure which shows RF circuit block of a dual band portable machine.

Explanation of symbols

Dip: Diplexer
SW: Switch circuit
LPF: Low-pass filter circuit
SAW: SAW filter
SL: Transmission lines Cg, Cd, Cp: Capacitors Dg, Dd, Dp: Diodes

Claims (5)

In a high frequency switch module in which a switching element, a capacitor, an inductor, and at least one SAW filter are integrally formed on a dielectric substrate, at least one of the SAW filters is an unbalanced input / balanced output type SAW filter. At least one of the two transmission lines connected between the two balanced output terminals of the unbalanced input / balanced output type SAW filter and the two receiving terminals formed on the dielectric substrate. And a ground electrode formed in the dielectric substrate so as not to overlap each other when the dielectric substrate is viewed in the stacking direction. A high-frequency switch module that switches an appropriate connection between a transmission circuit or a reception circuit of a plurality of transmission / reception systems that share an antenna and have different passbands,
A demultiplexing circuit comprising a low-pass filter and a high-pass filter for demultiplexing the signals of the plurality of transmission and reception systems;
A first switch circuit connected to the low pass filter of the branching circuit, for switching a transmission path and a reception path;
A second switch circuit connected to the high-pass filter of the branching circuit, for switching a transmission path and a reception path;
A low-pass filter provided in a transmission path of the first switch circuit and the second switch circuit;
A SAW filter in each reception path connected to the reception circuit from the first switch circuit and the second switch circuit;
The branching circuit and the low-pass filter are configured by an LC circuit, and the switch circuit mainly includes a diode and a transmission line, and the branching circuit and the LC circuit of the low-pass filter, at least a part of the transmission line of the switch circuit is In the laminated substrate of the electrode pattern and the dielectric layer, constituted by the electrode pattern,
The chip element that constitutes a part of the SAW filter, the diode, and the LC circuit is an antenna switch module mounted on the dielectric substrate.
At least one of the SAW filters is an unbalanced input / balanced output SAW filter, and is a receiving circuit formed on two balanced output terminals of the unbalanced input / balanced output SAW filter and the dielectric substrate. At least one of the two transmission lines connected to each of the two output terminals and a ground electrode formed in the dielectric substrate when the dielectric substrate is viewed in the stacking direction. A high-frequency switch module characterized by being arranged so as not to overlap each other. A high-frequency switch module that switches an appropriate connection between a transmission circuit or a reception circuit of a plurality of transmission / reception systems that share an antenna and have different passbands,
A demultiplexing circuit comprising a low-pass filter and a high-pass filter for demultiplexing the signals of the plurality of transmission and reception systems;
A first switch circuit connected to the low pass filter of the branching circuit, for switching a transmission path and a reception path;
A second switch circuit connected to the high-pass filter of the branching circuit, for switching a transmission path and a reception path;
A third switch circuit that is connected to the second switch circuit and switches between two reception paths among a plurality of transmission / reception system reception circuits,
A low-pass filter provided in a transmission path of the first switch circuit and the second switch circuit;
A SAW filter in each reception path connected from the first switch circuit and the third switch circuit to the reception circuit;
The branching circuit and the low-pass filter are configured by an LC circuit, and the switch circuit mainly includes a diode and a transmission line, and the branching circuit and the LC circuit of the low-pass filter, at least a part of the transmission line of the switch circuit is In the laminated substrate of the electrode pattern and the dielectric layer, constituted by the electrode pattern,
The chip element that constitutes a part of the SAW filter, the diode, and the LC circuit is an antenna switch module mounted on the dielectric substrate.
At least one of the SAW filters is an unbalanced input / balanced output SAW filter, and is a receiving circuit formed on two balanced output terminals of the unbalanced input / balanced output SAW filter and the dielectric substrate. At least one of the two transmission lines connected to each of the two output terminals and a ground electrode formed in the dielectric substrate when the dielectric substrate is viewed in the stacking direction. A high-frequency switch module characterized by being arranged so as not to overlap each other. The high-frequency switch module according to claim 1, wherein the two transmission lines have different lengths. A communication device using the high-frequency switch module according to claim 1.

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