JP2005094193A - Wireless communication transmitter and wireless communication receiver - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a wireless communication transmitter and a wireless communication receiver wherein a receiver configuration of a direct conversion system is simplified so as to attain a low cost and low power consumption. <P>SOLUTION: In the wireless communication transmitter, a mixer 11 generates a modulation signal 3 subjected to spread spectrum processing by a pseudo random code 2 and a baseband signal 1 and a VCO 12 uses the modulation signal 3 to apply frequency spread processing to a carrier signal with a prescribed frequency thereby generating and transmitting a frequency modulation signal 4. In the wireless communication receiver, a mixer applies quadrature detection to a received signal, and an FSK demodulator demodulates the modulation signal on the basis of a phase difference between obtained I, Q modulation signals, and a despreading demodulator to apply despreading processing to the demodulated signal to obtain the original baseband signal. <P>COPYRIGHT: (C)2005,JPO&NCIPI

Description

  The present invention relates to a wireless communication transmitter and a wireless communication receiver, and in particular, short-range weak wireless that performs wireless communication using a signal obtained by frequency-modulating a local oscillation signal of a predetermined frequency with a modulation signal composed of a desired baseband signal. The present invention relates to a wireless communication transmitter and a wireless communication receiver suitable for communication.

  Conventionally, in a modulation method in which radio communication is performed using a signal obtained by modulating a carrier signal of a predetermined frequency with a modulation signal composed of a desired baseband signal, a frequency deviation with respect to the predetermined frequency is used in a radio communication transmitter and a radio communication receiver. A highly accurate local oscillation signal and carrier signal with a small amount of data were used.

  FIG. 15 shows a configuration of a conventional wireless communication transmitter using a phase modulation method using spread spectrum communication. In this wireless communication transmitter, a mixer (multiplier) 101 performs spread processing on a desired baseband signal including transmission information using a spread code (pseudorandom code) from a spread code generator 100, and spread spectrum. The modulated signal is generated. Next, a phase modulation signal is generated by the mixer 103 from this modulation signal and the carrier signal subjected to frequency lock control from the frequency synthesizer 102. The phase modulation signal is amplified by the power amplifier PA 104 and then transmitted from the antenna 105. At this time, an operation for performing information modulation using transmission information on a carrier signal is referred to as primary modulation, and an operation for performing spread spectrum by multiplying a carrier signal by a spreading code is referred to as secondary modulation. As in the configuration example described above, it is common to perform phase modulation (PSK: Phase Shift Keying) for both primary modulation and secondary modulation.

  FIG. 16 shows a configuration of a conventional phase modulation type radio communication receiver using spread spectrum communication. In this wireless communication receiver, the phase modulation signal received from the antenna 110 is amplified by the low noise amplifier 111, and then the frequency is directly converted to the baseband signal band by multiplying the local oscillation signals having different 90 ° phases by the mixers 114I and 114Q. Direct conversion). Then, high-frequency components are removed from the frequency-converted received signal by the low-pass filters 115I and 115Q, and the amplitude is limited by the limiters 116I and 116Q to obtain I and Q modulated signals. The baseband demodulator 117 performs despreading processing on the obtained I and Q modulation signals, and then performs BPSK (Binary Phase Shift Keying) demodulation to obtain the original baseband signal.

Non-patent documents 1 to 3 describe the phase modulation type radio communication transmitter using spread spectrum communication and the phase modulation type radio communication transmitter / receiver using spread spectrum communication which are the conventional techniques. ing.
The applicant has not yet found prior art documents related to the present invention by the time of filing other than the prior art documents specified by the prior art document information described in this specification.
Marubayashi Gen, Nakagawa Masao, Kono Ryuji, "Spread Spectrum Communication and its Applications", IEICE, pp19-260, ISBN 4-88552-153-X A Direct-Conversion Receiver for 900 MHz (ISM Band) Spread-Spectrum Digital Cordless Telephone, Christopher Dennis Hull, Joo Leong Tham, IEEE Journal of Solid-State Circuits, vol.31, no.12, pp1955-1963, Dec. 1996 Yoshikazu Nishimura, "Data modulation / demodulation technology by radio", CQ Publisher, pp42-43, pp133-134, ISBN 4-7898-3349-6

In recent years, as a transceiver configuration, in place of the conventional superheterodyne method using an intermediate frequency, a direct conversion method that is a signal processing for a system LSI, in which an external intermediate frequency filter or the like can be omitted, is often used. In particular, in spread spectrum communication, a configuration using BPSK modulation for spread modulation, which is secondary modulation, is common.
In such a configuration, when demodulating a phase-modulated signal, the reference phase becomes important. In the direct conversion system of FIGS. 15 and 16 described above, it is difficult to completely match the frequencies of the carrier signal and the local oscillation signal used on the transmission side and the reception side, and a frequency deviation is likely to occur.

In the conventional radio communication transmitter and radio communication receiver, when such a frequency deviation occurs, the reference phase is not determined due to the residual frequency error caused by the deviation, and this frequency error is generated when demodulating the phase modulation signal. Since this may cause erroneous demodulation, it is necessary to perform an operation for correcting frequency and phase shift called AFC (Auto Frequency Control) or APC (Auto Phase Control) with a frequency synthesizer so as to automatically cancel this influence. . Therefore, there is a problem that the signal processing on the receiver side becomes complicated to perform such control, and the hardware amount and power consumption of the receiver increase.
The present invention solves such problems, and provides a wireless communication transmitter and a wireless communication receiver capable of simplifying a direct conversion receiver configuration, reducing costs, and reducing power consumption. The purpose is to do.

  In order to achieve such an object, a wireless communication transmitter according to the present invention uses a frequency modulation signal obtained by frequency modulation processing of a modulation signal including a baseband signal to be transmitted and a carrier signal of a predetermined frequency. In a wireless communication transmitter for transmission, spreading means for generating a spectrum-spread modulated signal from a predetermined spreading code and a baseband signal, and frequency-spreading a carrier signal having a predetermined frequency using the modulation signal from the spreading means Thus, a modulation means for generating a frequency modulation signal and an amplifier for amplifying and outputting the frequency modulation signal from the modulation means are provided.

At this time, a band pass filter that outputs only a signal component of a predetermined frequency band to the amplifier among the frequency modulation signals generated by the modulation means may be further provided.
The band-pass filter may have a pass band with a bandwidth narrower than twice the amount of frequency displacement modulated by the modulation means, centering on the frequency of the carrier signal.

A frequency synthesizer that performs frequency lock control of the carrier signal used by the modulation unit may be further provided, and the modulation unit may generate the frequency modulation signal using the carrier signal that is frequency locked by the frequency synthesizer.
At this time, the spreading means may generate the modulation signal with a chip interval smaller than the feedback time constant of the frequency lock control in the frequency synthesizer.
Also, the carrier signal may be frequency-modulated by a modulation means using a modulation index larger than 1.

  The radio communication receiver according to the present invention receives a radio frequency modulation signal generated from a modulation signal including a baseband signal to be transmitted and a carrier signal having a predetermined frequency, and demodulates the original baseband signal. In this machine, quadrature detection means for generating a quadrature modulation signal by performing quadrature detection on the received frequency modulation signal, and demodulation of frequency modulation by detecting the phase difference between the quadrature modulation signals from this quadrature detection means And a despreading demodulator that demodulates and outputs a baseband signal by despreading the modulated signal from the frequency demodulating unit.

At this time, a capacitive element that is connected in series to the output stage of the quadrature detection means and that removes the DC component contained in the reception signal quadrature detected by the quadrature detection means and outputs the signal to the frequency demodulation means side may be further provided.
As the despreading demodulator, an asynchronous despreading demodulator that performs a despreading process using a spreading code whose phase or frequency is not synchronized with the spreading code at the time of spectrum spreading may be used.

According to the present invention, in a wireless communication transmitter, a spreader performs spectrum spread (primary modulation) on a baseband signal, and a carrier signal is frequency-modulated (secondary modulation) based on the obtained modulated signal and transmitted. Therefore, compared with the conventional case of using PSK or BPSK, it is not necessary to perform control for canceling frequency errors such as AFC and APC on the receiving side.
Further, in the wireless communication receiver, I and Q modulation signals are generated from the orthogonal reception signals obtained by orthogonal detection of the reception signals, and the phase delay relationship is determined by comparing the phases of these I and Q modulation signals. Since the determination result is output as a modulation signal that has been FSK demodulated, the reference phase is not a problem unlike the case of demodulating a phase-modulated signal as in the prior art. Even if an error occurs to some extent in the generated carrier frequency, it can be demodulated, and the receiving side does not need to perform control for canceling the frequency error such as AFC or APC.
Therefore, it is possible to greatly simplify the configuration of wireless communication using the direct conversion spread spectrum communication, and to reduce the cost and power consumption of the wireless communication receiver.

[First Embodiment]
A wireless communication transmitter according to a first embodiment of the present invention will be described with reference to FIG. FIG. 1 is a block diagram showing a configuration of a wireless communication transmitter according to a first embodiment of the present invention.
This radio communication transmitter is a frequency modulation type radio communication transmitter using spread spectrum, and includes a spread code generator 10, a mixer 11, a variable control oscillator (hereinafter referred to as VCO) 12, a power amplifier (hereinafter referred to as PA). ) 13 and the antenna 14.

The spreading code generator 10 is a circuit unit that outputs a predetermined pseudorandom code 2.
The mixer 11 is a circuit unit (multiplier) that outputs a spectrum-spread modulated signal 3 by multiplying the digital baseband signal 1 including transmission information by the pseudo-random code 2 from the spread code generator 10. is there.
The VCO 12 is a circuit unit that outputs the frequency modulation signal 4 by variably controlling the oscillation frequency based on the modulation signal 3 from the mixer 11.
The PA 13 is an amplifier that amplifies the frequency modulation signal 4 from the VCO 12 and outputs the amplified signal to the antenna 14.

  In this wireless communication transmitter, a baseband signal 1 to be transmitted is input to a mixer 11 and multiplied by a pseudo-random code from a spread code generator 10 so that the baseband signal 1 is directly spread spectrum over a wide band. At this time, the spreading code generator 10 and the mixer 11 function as spreading means for performing spread spectrum by multiplying the baseband signal to be transmitted by the spreading code.

The modulation signal 3 is input to the VCO 12, and the frequency of the carrier signal is controlled based on the modulation signal 3, and the carrier signal is frequency-modulated. At this time, one of the two subcarrier signals fc ± Δf provided above and below the reference oscillation signal frequency of the VCO 12, that is, the carrier signal frequency fc, is selected based on the binary modulation signal 3. Therefore, the VCO 12 functions as a modulation unit that modulates the frequency of the carrier signal using the modulation signal 3.
The frequency modulation signal 4 output from the VCO 12 is amplified by the PA 13 and wirelessly transmitted from the antenna 14.

  As described above, since the baseband signal 1 is spectrum spread (primary modulation) by the mixer 11 and the carrier signal is frequency-modulated (secondary modulation) based on the obtained modulation signal 3, it is transmitted. As compared with the case where PSK or BPSK is used, it is not necessary to perform control for canceling the frequency error such as AFC or APC on the receiving side. In particular, frequency-modulated signals can be demodulated with a very simple circuit configuration, and even when frequency modulation is used for spread modulation, process gain can be obtained by performing despreading after frequency demodulation. Even if some errors are sometimes included, the transmitted digital signal can be recovered. Therefore, according to the present embodiment, the configuration of the spread spectrum communication wireless communication receiver using the direct conversion method can be simplified, and the cost and power consumption can be reduced.

[Second Embodiment]
Next, a wireless communication transmitter according to the second embodiment of the present invention will be described with reference to FIG. FIG. 2 is a block diagram showing the configuration of a wireless communication transmitter according to the second embodiment of the present invention, which is the same as or equivalent to the wireless communication transmitter (see FIG. 1) according to the first embodiment described above. The parts are given the same reference numerals.
The radio communication transmitter according to the present embodiment is a frequency modulation type radio communication transmitter using spread spectrum communication. In addition to the configuration of the radio communication transmitter of FIG. 1, a band pass filter (hereinafter referred to as BPF) is used. ) 15 is provided. The BPF 15 is a filter that outputs only a predetermined frequency band component of the frequency modulation signal generated by the VCO 12 and amplified by the PA 13 to the antenna 14.

FIG. 3 shows the relationship between the frequency spectrum of the frequency-modulated wave and the passband of the bandpass filter. This is a frequency spectrum when the modulation signal is subjected to frequency modulation with a modulation factor of 1 with respect to the reference frequency fc.
When performing frequency modulation, although depending on the degree of modulation to some extent, when the amount of frequency displacement is Δf, a relatively large spectral peak appears at the frequency fc ± Δf of the two frequency-shifted subcarriers.

In normal wireless communication, a spectrum mask for transmission power is defined according to wireless standards. For example, in the case of the weak wireless standard, a peak value of radio wave power that can be transmitted as 500 mV / m at a distance of 3 m is defined for a band below 322 MHz.
When the spectrum of FIG. 3 is transmitted, since a large spectrum peak appears in fc ± Δf, the spectrum power of fc ± Δf is applied to the spectrum mask relating to transmission power. As a result, the spectrum mask is reached at the spectrum peak, and the spectrum power in the frequency band other than fc ± Δf that can be transmitted becomes relatively small.

  In this embodiment, the BPF 15 is used to suppress the peak signal appearing at fc ± Δf and output the transmission signal. The pass band 15A is a pass band of the BPF 15 added in the present embodiment. By setting the center frequency of the BPF 15 as fc and making its pass bandwidth narrower than twice the frequency deviation amount Δf (−Δf to + Δf), the peak signal appearing at fc ± Δf is suppressed, and from fc−Δf A signal in a band within fc + Δf is transmitted. In this case, since a large spectrum peak does not appear in the band of the band pass filter, a signal can be transmitted with power corresponding to a value defined by the spectrum mask for most signal spectrum components in the band. As a result, it is possible to increase the total signal power that can be transmitted compared to a transmitter having no filter, and it is possible to increase the communication capacity such as the communication distance and data rate of the radio.

[Third Embodiment]
Next, a wireless communication transmitter according to the third embodiment of the present invention will be described with reference to FIG. FIG. 4 is a block diagram showing a configuration of a wireless communication transmitter according to the third embodiment of the present invention, which is the same as or equivalent to the wireless communication transmitter (see FIG. 1) according to the first embodiment described above. The parts are given the same reference numerals.
The radio communication transmitter according to the present embodiment is a frequency modulation type radio communication transmitter using spread spectrum communication. In the configuration of the radio communication transmitter of FIG. 1, a frequency synthesizer 20 is used instead of the VCO 12. It is what was used.

The frequency synthesizer 20 is a circuit unit that performs frequency modulation as secondary modulation and performs frequency lock control of the carrier signal.
In the wireless communication transmitter according to the present embodiment, the frequency synthesizer 20 generates a frequency-modulated signal 4A that is frequency-modulated based on the modulated signal 3 from the mixer 11 using a carrier signal that is subjected to frequency lock control. And output to PA13.

FIG. 5 shows a configuration example of the frequency synthesizer 20. The frequency synthesizer 20 includes a vibrator 21, a phase comparator 22, a low-pass filter (hereinafter referred to as LPF) 23, a variable control oscillator (hereinafter referred to as VCO) 24, and a frequency divider 25.
The vibrator 21 is a circuit unit that includes a crystal vibrator or the like and outputs a reference signal 30 having a predetermined frequency. The phase comparator 22 is a circuit unit that compares the phase of the reference signal 30 from the vibrator 21 and the feedback signal 33 from the frequency divider 25 and outputs the comparison result as a carrier signal 31. The LPF 23 is a circuit unit that removes the high-frequency component of the carrier signal 31 from the phase comparator 22 and outputs the carrier signal 32.

The VCO 24 is a circuit unit that modulates the frequency of the carrier signal 32 based on the modulation signal 3 and outputs a frequency modulation signal 4A. The frequency divider 25 is a circuit unit that divides the frequency modulation signal 4 </ b> A and outputs the frequency division result as a feedback signal 33.
In this frequency synthesizer 20, since the phase comparator 22 compares the phase of the reference signal 30 and the feedback signal 33 from the frequency divider 25, the carrier signal is locked to a predetermined frequency by operating this feedback loop. Is done.

In the case of a configuration that does not use such a frequency synthesizer, the carrier signal frequency fc fluctuates every time frequency modulation is performed, so that a frequency error when performing direct conversion to the original modulated signal on the receiver side is large. As shown in the first embodiment, frequency demodulation can be performed even if there is a certain frequency error. However, if the frequency error is large, the sensitivity characteristic deteriorates. Therefore, the frequency error should be as small as possible. desirable.
In the present embodiment, the frequency error is reduced by providing a frequency synthesizer in the transmitter, performing lock control on the carrier signal frequency, and performing frequency modulation. Therefore, in this embodiment, the sensitivity characteristic of the receiver can be improved as compared with the first embodiment.

However, in this case, since the modulation is performed while being locked to the carrier signal, the carrier signal shifted to fc ± Δf is subjected to a feedback operation to return to the frequency of the carrier signal frequency fc, and the deviation frequency Δf may not be determined. Conceivable.
In such a case, by making the chip interval of the spreading code used for modulation smaller than the feedback time constant of the frequency lock control of the frequency synthesizer 20, the shift amount of the carrier signal frequency shifted to fc ± Δf can be obtained. It can be prevented from changing. When a signal of the same polarity continues for a long time in the modulation signal 3, there is a possibility that the influence of lock control may appear. However, in this embodiment, since the modulation signal is spread by a spreading code, the modulation signal of the same polarity The pattern does not last long. Therefore, as described above, it is possible to prevent changes in the frequency shift amount by making the chip interval of the spread code smaller than the feedback time constant to some extent.

Further, in common with all the above-described embodiments, if the modulation index for frequency modulation is increased, the sensitivity characteristic when frequency demodulation is performed on the receiver side can be improved. In spread spectrum communication, multipath tolerance can be obtained by expanding the signal spectrum over a wide band.
When spread modulation, which is secondary modulation, is used as frequency modulation as in these embodiments, if the modulation index is small, the transmission signal spectrum becomes narrower than when ASK or PSK is used for secondary modulation. For this reason, when the transmission signal spectrum is viewed on the frequency axis, the diffused effect is hardly seen, and the multipath resistance is greatly reduced.

  Therefore, it is desirable that the modulation index when performing frequency modulation is large. By increasing the modulation index, the sensitivity characteristic at the time of frequency demodulation on the receiver side can be improved, and the transmission signal spectrum seen on the frequency axis can be widened to improve multipath tolerance. The modulation index is preferably as large as possible, and in the case of each of the above-described embodiments, it is desirable that the modulation index be at least about 1 (100%).

[Fourth Embodiment]
Next, a wireless communication receiver according to the fourth embodiment of the present invention will be described with reference to FIG. FIG. 6 is a block diagram showing a configuration of a wireless communication receiver according to the fourth embodiment of the present invention.
The radio communication receiver according to the present embodiment is a frequency modulation type radio communication receiver using spread spectrum communication, and includes an antenna 50, a low noise amplifier (hereinafter referred to as LNA) 51, an oscillator 52, and a π / 2 phase shift. And a low-pass filter (hereinafter referred to as LPF) 55I and 55Q, limiters 56I and 56Q, an FSK (Frequency Shift Keying) demodulator 57, and a despreading demodulator 58.

  The LNA 51 is an amplifier that amplifies the reception signal received by the antenna 50 with low noise. The oscillator 52 is a circuit unit that generates a local oscillation signal 6 having a predetermined frequency substantially equal to the carrier signal frequency. The π / 2 phase shifter 53 is a circuit unit that shifts the local oscillation signal 6 by 90 °. The mixers 54I and 54Q multiply the reception signal 5 from the LNA 51 and the local oscillation signal 6 orthogonal to each other, thereby converting (frequency shifting) the reception signal 5 to a frequency band lower than the carrier signal frequency (multiplication). ). LPFs 55I and 55Q are filters that pass signal components in a frequency band lower than the chip rate of the spread code among the orthogonally received signals 7I and 7Q after the frequency conversion obtained by the mixers 54I and 54Q.

  The limiters 56I and 56Q are circuit units that limit the amplitude of the received signals from the LPFs 55I and 55Q. The FSK demodulator 57 is a circuit unit that demodulates the frequency modulation of the I modulation signal 8I and the Q modulation signal 8Q obtained from the limiters 56I and 56Q, and outputs the original modulation signal 8. The despread demodulator 58 is a circuit unit that demodulates and outputs the original baseband signal 9 by despreading the modulated signal 8 from the FSK demodulator 57.

  In this wireless communication receiver, the reception signal received by the antenna 50 is amplified by the LNA 51 and input to the mixers 54I and 54Q as the reception signal 5. The mixer 54I multiplies the local oscillation signal 6 from the oscillator 52 and the mixer 54Q multiplies the local oscillation signal 6 rotated by 90 ° from the π / 2 phase shifter 53. The frequency is converted to a frequency band that is lower than the band and is used for spread spectrum processing. These mixers 54I and 54Q function as quadrature detection means.

The orthogonal reception signals 7I and 7Q subjected to frequency conversion in this way pass through the LPFs 55I and 55Q and become signal components in a frequency band lower than the chip rate of the spreading code used at the time of transmission. Then, after being limited in amplitude by the limiters 56I and 56Q, the I-modulation signal 8I and the Q-modulation signal whose phases are orthogonal to each other and whose phase-delay relationship is inverted depending on the frequency shift of the received frequency signal. 8Q is output to the FSK demodulator 57.
The I modulation signal 8I and the Q modulation signal 8Q are compared in phase with each other in the FSK demodulator 57, the phase delay relationship is determined, and the modulation signal 8 is demodulated. Then, the despreading demodulator 58 performs despreading processing using a spreading code equivalent to that on the transmission side, and outputs the original baseband signal 9.

There are several methods for comparing the phases of the modulated signals of the I channel and Q channel used in the FSK demodulator 57, for example, a lag lead type phase shift comparator as shown in FIG.
The FSK demodulator 57 is not limited to the demodulation method as long as it has a function of outputting the result of phase comparison. The phase comparator that determines and outputs such a phase delay relationship is a frequency detector. It can be regarded as a frequency demodulator that demodulates the modulated signal.

As described above, in this embodiment, I and Q modulation signals are generated from quadrature reception signals obtained by quadrature detection of reception signals, the phases of these I and Q modulation signals are compared, and the phase delay relationship Since the determination result is output as the FSK demodulated modulated signal 8, unlike the conventional case where the phase modulated signal is demodulated, the reference phase is not a problem.
Therefore, even if a certain amount of error occurs in the carrier frequency generated on the transmitting side and the receiving side, it can be demodulated, and there is no need to perform control for canceling the frequency error such as AFC and APC on the receiving side as in the prior art. .

Further, as described above, the frequency-modulated signal can be demodulated with a very simple circuit configuration. Even when frequency modulation is used for spread modulation, process gain can be obtained by performing despreading processing after frequency demodulation, so the transmitted digital signal can be restored even if some errors are included during frequency demodulation. Can do.
Therefore, it is possible to greatly simplify the configuration of wireless communication using the direct conversion spread spectrum communication, and to reduce the cost and power consumption of the wireless communication receiver.

[Fifth Embodiment]
Next, a radio communication receiver according to the fifth embodiment of the present invention will be described with reference to FIG. FIG. 8 is a block diagram showing a configuration of a wireless communication receiver according to the fifth embodiment of the present invention.
The radio communication receiver according to the present embodiment is a frequency modulation type radio communication receiver using spread spectrum communication, and an orthogonal reception signal is provided at the output terminals of the mixers 54I and 54Q in the radio communication receiver of FIG. Except that capacitive elements 60I and 60Q for removing DC (direct current) components of 7I and 7Q are provided, this is the same as the wireless communication receiver of FIG.

  In general, direct conversion wireless communication receivers that directly convert the received signal to the frequency band of the baseband signal without frequency conversion to the intermediate frequency band do not require an external channel selection filter or intermediate frequency converter. The receiver can be downsized. However, in such a direct conversion method, since the frequency conversion is performed in the frequency band of the baseband signal, that is, near 0 Hz, the DC component generated in the received signal due to the signal mixed in the received signal inside the receiver, A phenomenon that the midpoint potential of the signal is shifted up and down, that is, a DC offset occurs, and the signal is deteriorated in the circuit section after the frequency conversion section. For example, when a received signal having a DC offset is input to the amplifier, the output of the amplifier is saturated and a desired signal cannot be amplified. Therefore, it is usually necessary to take measures against DC offset and manage the level of the received signal.

Conventionally, as a countermeasure against such DC offset, a configuration in which a received signal is passed through a high-pass filter has been taken, but there are the following problems.
In the direct conversion method, a signal peak exists in the vicinity of 0 Hz in the modulated signal spectrum after frequency conversion. That is, since a lot of energy (information) is included in the vicinity of DC, it is necessary to set the cutoff frequency of the high-pass filter to an extremely low value. However, in this case, a very large capacitance element is required, and the merit realized by the integrated circuit is reduced, contrary to the advantage of the direct conversion method that is an architecture for the system LSI. Further, if the capacitive element is made larger, the offset voltage that changes at a high speed cannot be followed, so that offset removal may be insufficient. On the other hand, when the capacitance element is reduced, the cutoff frequency of the high-pass filter increases and the peak component of the modulation signal is cut off, so that the bit error rate (BER) is greatly reduced.

  On the other hand, a method of performing DC free coding so that the peak of the modulation signal does not come near DC is conceivable, but in this case, an encoder and a decoder are newly required for the transceiver. In addition, this method can achieve a certain effect when the frequency band of the modulation signal is wide, but cannot achieve the effect when the frequency band is not so wide. That is, even if the coding is performed so that the signal peak of the signal having the original frequency band of about 100 KHz is moved from the vicinity of DC to the vicinity of 50 KHz, the effect is small.

  As another DC offset countermeasure, a method of removing the DC offset using a time during which data is not transmitted can be considered. In the case of TDMA communication, since transmission and reception are intermittently repeated, the cutoff frequency of the HPF can be virtually set to 0 Hz in the reception slot by holding the DC offset component in the capacitive element between transmission and reception. it can. However, in this case, it is necessary to provide a time zone during which no data signal is transmitted at the time of reception, and there is a problem that control of such time slots and communication methods are limited.

In addition, since thermal noise due to the switch that switches the capacitive elements is actually accumulated in the capacitive element, it is necessary to use an element with a large capacitance value, and the receiver cannot be miniaturized. Further, when a disturbing signal is received at the time of DC offset removal, the signal is also held, and there is a problem that accurate offset removal cannot be performed. Furthermore, if the charging / discharging of the capacitive element is not completed in a time sufficiently shorter than the transmission / reception switching time, the reception quality immediately after switching to the receiving side deteriorates.
According to this countermeasure method, as described above, an element having a large capacitance value is required as the capacitor element, so that the charge / discharge time is also increased, and the influence on the reception quality cannot be ignored.

  In the present embodiment, as shown in FIG. 8, since the capacitive elements 60I and 60Q are connected in series between the output terminals of the mixers 54I and 54Q, here, the LPFs 55I and 55Q, the mixers 54I and 54Q The DC offset generated by the DC component included in the frequency-converted orthogonal reception signals 7I and 7Q is removed, and the desired modulation signal can be normally demodulated by the FSK demodulator 57.

FIG. 9 shows a frequency spectrum in spread spectrum communication. FIG. 9A shows the frequency spectrum of a baseband signal that has not been subjected to spread spectrum processing, and FIG. 9B shows the frequency spectrum of a spread modulation signal that has been subjected to spread spectrum processing.
Generally, in spread spectrum communication, a baseband signal is spread over a wide band using a spreading code. Usually, the spectrum band 72 of the spread modulation signal is several MHz to several tens of MHz because it is spread to a bandwidth several to several tens of times that of the spectrum band 71 of the baseband signal.

  In such a frequency band, the capacitive elements 60I and 60Q for removing the DC offset frequency band 70 do not require a very large capacitance value, and thus can be mounted on an integrated circuit. In addition, it is possible to quickly cope with a DC offset that changes at high speed. In spread spectrum communication, there is a process gain due to despreading, and even if a signal near DC is lost, the modulated signal can be demodulated without error if the S / N ratio after despreading is good.

[Sixth Embodiment]
Next, a radio communication receiver according to the sixth embodiment of the present invention will be described with reference to FIG. FIG. 10 is a block diagram showing a configuration of a wireless communication receiver according to the sixth embodiment of the present invention.
The radio communication receiver according to the present embodiment is a frequency modulation type radio communication receiver using spread spectrum communication that uses a signal obtained by spectrum-spreading a desired baseband signal as a modulation signal. Compared with the wireless communication receiver according to the embodiment (see FIG. 6), an asynchronous despreading demodulator 58 is used as the despreading demodulator 58, and a capacitive element 61 is provided at the input end of the despreading demodulator 58. Different.

  FIG. 11 shows a configuration example of an asynchronous despreading demodulator. This asynchronous despread demodulator includes a sample hold circuit (hereinafter referred to as S / H circuit) 81a to 81g, a sample hold control circuit (hereinafter referred to as S / H control circuit) 82, a flip-flop circuit (hereinafter referred to as FF circuit). ) 83a to 83f, a spread code generation circuit 84, a multiplier 85, an adder 86, and a peak detector 87.

  The S / H circuits 81a to 81g sample and hold the input signal. The S / H control circuit 82 receives a clock f1 having substantially the same frequency as the clock used for spreading the input signal, divides the frequency by N (N = 7 in this embodiment), and sets the S / H circuits 81a to 81g. Generate a signal to control. The FF circuits 83a to 83f constitute a shift register that shifts the output signal from the S / H control circuit 82 with the clock f1. The spread code generation circuit 84 generates the same spread code as that used at the time of spectrum spread based on the clock f2. Multipliers 85a to 85g multiply the signals output from S / H circuits 81a to 81g and the spread codes output from spread code generation circuit 84, respectively. The adder 86 adds the output signals of the multipliers 85a to 85g. The peak detector 87 detects a peak value from the output signal of the adder 86.

The input signal is sampled and held by the S / H circuits 81a to 81g and input to one input terminal of the multipliers 85a to 85g. At this time, new signals received by the S / H control circuit 82 and the FF circuits 83a to 83f are sent to the multipliers 85a to 85g at the same clock number intervals as the number of these multipliers. Updated at 81g and held.
On the other hand, a spread code is generated from the spread code generation circuit 84 based on the clock f2, and is input to the other input terminal of each of the multipliers 85a to 85g. In the multipliers 85a to 85g, the signals from the S / H circuits 81a to 81g and the spread code from the spread code generation circuit 84 are multiplied for each chip, and the multiplication results are added by the adder 86 to be output as an output signal. Is output.

As a result, a peak value is generated in the output signal at a period determined by the frequency of the clock f1 of the input signal, the frequency of the clock f2 of the spreading code, and the spreading code length, and this output signal includes the input signal, the spreading code, Shows a high value when synchronized, and almost zero when not synchronized.
Therefore, by adjusting the timing of the spreading code generated by the spreading code generation circuit 84 according to this peak value before and after the input signal, the input signal and the spreading code are synchronized, and each spreading code length is synchronized. A peak value can be obtained continuously.

  FIG. 12 shows a configuration example of the spread code generation circuit 84. The spreading code generation circuit 84 shifts the spreading code in the forward direction according to the clock f2 and outputs the spreading code to the multipliers 85a to 85g, and reverses the spreading code according to the clock f2. A second spreading code generation circuit 90b that shifts in the direction and outputs it to the multipliers 95a to 95g, and outputs either a forward control signal 95a or a backward control signal 95b based on the control signal from the peak detector 87. Thus, it comprises a spreading code control circuit 95 that operates either the first or second spreading code generation circuit 90a, 90b.

The first spreading code generation circuit 90a outputs to the outputs of the FF circuits 93a to 93g and the FF circuits 93a and 93c that constitute a shift register that shifts the spreading code in the forward direction (from the multiplier 85a to the multiplier 85g) by the clock f2. Based on the exclusive OR circuit 91 that generates a spread code based on the FF circuits 93a to 93g and the switches 94a to 94g and 94o for turning on / off the outputs of the exclusive OR circuit 91.
The second spreading code generation circuit 90b outputs to the outputs of the FF circuits 93h to 93n and FF circuits 93i and 93j that constitute a shift register that shifts the spreading code in the reverse direction (from the multiplier 85g to the multiplier 85a) by the clock f2. The circuit includes an exclusive OR circuit 92 that generates a spread code, and switches 94h to 94n and 94p that turn on / off the outputs of the FF circuits 93h to 93n and the exclusive OR circuit 92.

When the forward control signal 95a is output from the spreading code control circuit 95, the switches 94a to 94g and 94o are turned on and the switches 94h to 94o are turned off. As a result, the FF circuits 93a to 93g are connected in series and the exclusive OR circuit 91 is connected, and spreading codes that shift in the forward direction according to the clock f2 are output to the multipliers 85a to 85g.
On the other hand, when the reverse direction control signal 95b is output from the spreading code control circuit 95, the switches 94a to 94g and 94o are turned off and the switches 94h to 94o are turned on. As a result, the FF circuits 93h to 93n are connected in series and the exclusive OR circuit 92 is connected, and spreading codes that are shifted in the reverse direction according to the clock f2 are output to the multipliers 85a to 85g.

At this time, the inputs of the FF circuits 93b to 93g are connected to the inputs of the FF circuits 93h to 93m, the output of the FF circuit 93g is connected to the input of the FF circuit 93h via the switch 94g, and the output of the FF circuit 93n is switched to 94n is connected to the input of the FF circuit 93a, and when switching between the forward direction and the reverse direction, switching is performed in a state where the spreading code output at that time is held.
Therefore, the spread code control circuit 95 switches between the forward control signal 95a and the reverse control signal 95b and outputs the signal every time a peak value is detected based on the control signal from the peak detector 87, for example. The phase between the signal and the spread code is substantially maintained at the timing when the peak value is obtained.

  As a result, a signal waveform as shown in FIG. 13 is obtained as an output signal from the adder 86 and a received signal from the peak detector 87. The output signal from the adder 86 becomes a positive value when the input signal and the polarity of the spreading code are synchronized in the same state, and becomes a negative value when the polarity is synchronized with the opposite polarity. At this time, a peak value having the same polarity as that of the baseband signal is obtained by switching the polarity of the spread code to positive / negative in correspondence with the bit value 0/1 of the baseband signal on the transmission side. Therefore, a received signal, that is, a desired baseband signal 9 is obtained from the envelope of these peak values.

  In this embodiment, the case of spreading code length = 7 (PN7) has been described as an example. However, the present invention is not limited to this, and an arbitrary spreading code length N (N is an integer of 2 or more) is used. be able to. At that time, the S / H circuits 81a to 81g, the FF circuits 83a to 83g, the multipliers 85a to 85g, the FF circuits 93a to 93n, and the switches 94a to 94p may be provided in a number corresponding to the spreading code length N. . Also, with respect to the spreading code to be used, codes of other series may be used by selecting the inputs of the exclusive OR circuits 91 and 92.

In a general synchronous despreading demodulator, it is necessary to completely synchronize the frequency and phase of the spreading code used for the despreading processing with respect to the spreading code at the time of spread spectrum on the transmission side. Is used to hold (lock) the clock frequency used to generate the spreading code with high accuracy before starting the despreading process.
Therefore, in such a synchronous despreading demodulator, the circuit configuration becomes complicated and the scale increases, and it takes time until synchronization is obtained while maintaining the clock frequency with high accuracy.

On the other hand, as described above, the asynchronous despreading demodulator basically does not need to completely synchronize the frequency and phase of the spread code at the time of spread spectrum and the spread code at the time of despread. Since the whole operates asynchronously, desired data can be demodulated in a relatively short time by selecting the clock f2 used to generate the spread code.
Therefore, by using an asynchronous despreading demodulator, desired data can be demodulated in a relatively short time without requiring a complicated circuit configuration such as a PLL circuit, and the power consumption of the entire wireless communication receiver can be reduced.

  For multipliers 95a to 95g, a multiplier circuit as shown in FIG. 14 may be used. In this multiplication circuit, NM1 to NM7 are MOS transistors, and are constituted by a two-stage vertical and horizontal differential circuit. The spread code from the spread code generation circuit 84 and the signals from the S / H circuits 81a to 81g are differential signals, and the spread codes are input in opposite phases to the upper two differential circuits. A signal from 81 is input to the two differential circuits in the lower stage. Thereby, both signals are multiplied, and the multiplication result is output in the current mode.

  Such an analog signal system multiplier has a DC level (direct current bias) that operates properly with respect to an input signal as compared with a digital signal system multiplier. Therefore, when the DC level of the modulated signal 8 output from the FSK demodulator 57 is different from the DC level of the asynchronous despread demodulator 58, the input stage of the asynchronous despread demodulator 58 as shown in FIG. The capacitive element 61 is provided to the FSK demodulator 57 and the asynchronous despread demodulator 58, and the center potential of the modulation signal 8 may be set to an appropriate DC level by a resistor divider circuit or the like.

Although the fourth embodiment has been described as an example in the present embodiment, it is applied to the fifth embodiment (see FIG. 8) using a despread demodulator in the same manner as described above. And similar effects can be obtained.
In particular, the functions of the capacitive elements 60I and 60Q used in the fifth embodiment can be realized by the capacitive element 61, and the orthogonal received signal 7I frequency-converted by the mixers 54I and 54Q with one capacitive element 61. , 7Q can remove the DC offset generated by the DC component and can set an appropriate DC level for the input signal of the asynchronous despreading demodulator. Therefore, it is possible to reduce the capacity elements that require a relatively large area when mounted on an integrated circuit, and to easily realize a chip for a wireless communication receiver.

It is a block diagram which shows the structure of the radio | wireless communication transmitter concerning the 1st Embodiment of this invention. It is a block diagram which shows the structure of the radio | wireless communication transmitter concerning the 2nd Embodiment of this invention. It is explanatory drawing which shows the relationship between the frequency spectrum of a frequency modulation wave, and the pass band of a band pass filter. It is a block diagram which shows the structure of the radio | wireless communication transmitter concerning the 3rd Embodiment of this invention. 5 is a configuration example of the frequency synthesizer of FIG. It is a block diagram which shows the structure of the radio | wireless communication receiver concerning the 4th Embodiment of this invention. 7 is a configuration example of the FSK demodulator of FIG. 6. It is a block diagram which shows the structure of the radio | wireless communication receiver concerning the 5th Embodiment of this invention. It is a frequency spectrum in spread spectrum communication. It is a block diagram which shows the structure of the radio | wireless communication receiver concerning the 6th Embodiment of this invention. It is an example of a structure of the asynchronous despreading demodulator of FIG. It is an example of a structure of the spreading code generator of FIG. It is a signal waveform diagram which shows each part signal of the asynchronous despreading demodulator of FIG. 12 is a configuration example of a multiplier in FIG. It is a block diagram which shows the structure of the conventional radio | wireless communication transmitter. It is a block diagram which shows the structure of the conventional radio | wireless communication receiver.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Baseband signal, 2 ... Pseudorandom code, 3 ... Modulation signal (after spread spectrum), 4, 4A ... Frequency modulation signal, 5 ... Reception signal, 6 ... Local oscillation signal, 7I, 7Q ... Orthogonal reception signal, 8 , 8I, 8Q ... modulated signal (after spread spectrum), 9 ... baseband signal, 10 ... spreading code generator, 11 ... mixer, 12 ... VCO, 13 ... PA, 14 ... antenna, 15 ... BPF, 20 ... frequency synthesizer , 21 ... vibrator, 22 ... phase comparator, 23 ... LPF, 24 ... VCO, 25 ... frequency divider, 30 ... reference signal, 31, 32 ... carrier signal, 33 ... feedback signal, 50 ... antenna, 51 ... LNA 52... Oscillator 53... Π / 2 phase shifter 54I and 54Q Mixer 55I and 55Q LPF 56I and 56Q Limiter 57 FSK demodulator 58 Despread recovery 60I, 60Q, 61 ... capacitance element, 81a-81g ... S / H circuit, 82 ... S / H control circuit, 83a-83f ... FF circuit, 84 ... spread code generation circuit, 85a-85g ... multiplier, 86 , Adder, 87, peak detector, 90a, first spreading code generation circuit, 90b, second spreading code generation circuit, 91, 92, exclusive OR circuit, 93a to 93n, FF circuit, 94a to 94p ... switch, 95 ... spread code control circuit, 95a ... forward control signal, 95b ... reverse control signal.

Claims (9)

In a wireless communication transmitter that transmits a frequency modulation signal obtained by frequency modulation processing of a modulation signal including a baseband signal to be transmitted and a carrier signal of a predetermined frequency,
Spreading means for generating a spectrum-spread modulated signal from a predetermined spreading code and the baseband signal;
Modulation means for generating a frequency modulation signal by frequency spreading a carrier signal of a predetermined frequency using the modulation signal from the spreading means;
A wireless communication transmitter comprising: an amplifier for amplifying and outputting a frequency modulation signal from the modulation means. The wireless communication transmitter according to claim 1, wherein
A radio communication transmitter, further comprising: a band pass filter that outputs only a signal component of a predetermined frequency band to the amplifier among the frequency modulation signal generated by the modulation means. The wireless communication transmitter according to claim 2, wherein
The wireless communication transmitter, wherein the band-pass filter has a pass band with a bandwidth narrower than twice the amount of frequency displacement modulated by the modulation means, centered on the frequency of the carrier signal. The wireless communication transmitter according to claim 1, wherein
A frequency synthesizer that performs frequency lock control of a carrier signal used in the modulation unit;
The radio communication transmitter characterized in that the modulation means generates the frequency modulation signal using a carrier signal that is frequency-lock controlled by the frequency synthesizer. The wireless communication transmitter according to claim 4,
The wireless communication transmitter, wherein the spreading means generates the modulation signal at a chip interval smaller than a feedback time constant of frequency lock control in the frequency synthesizer. The wireless communication transmitter according to claim 1, wherein
The radio communication transmitter, wherein the modulation means frequency-modulates the carrier signal using a modulation index larger than 1. In a wireless communication receiver that receives a frequency modulation signal generated from a modulation signal including a baseband signal to be transmitted and a carrier signal of a predetermined frequency, and demodulates the original baseband signal,
Orthogonal detection means for generating an orthogonal modulation signal by performing orthogonal detection on the received frequency modulation signal;
A frequency demodulation means for generating the modulation signal obtained by demodulating the frequency modulation by detecting a phase difference between the orthogonal modulation signals from the orthogonal detection means;
A radio communication receiver comprising: a despreading demodulator that demodulates and outputs the baseband signal by despreading the modulated signal from the frequency demodulating means. The wireless communication receiver according to claim 7, wherein
A capacitance element that is connected in series to the output stage of the quadrature detection unit and that removes a direct current component included in the reception signal quadrature detected by the quadrature detection unit and outputs the signal to the frequency demodulation unit side; Wireless communication receiver. The wireless communication receiver according to claim 7 or 8,
The despreading demodulator comprises an asynchronous despreading demodulator that performs despreading processing using a spreading code whose phase or frequency is not synchronized with a spreading code at the time of spread spectrum.

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