JP2004236383A

JP2004236383A - Method for estimating pole position of permanent magnet type synchronous motor and controller for same

Info

Publication number: JP2004236383A
Application number: JP2003019107A
Authority: JP
Inventors: Sukeatsu Inazumi; 祐敦稲積
Original assignee: Yaskawa Electric Corp
Current assignee: Yaskawa Electric Corp
Priority date: 2003-01-28
Filing date: 2003-01-28
Publication date: 2004-08-19
Anticipated expiration: 2023-01-28
Also published as: JP4273775B2

Abstract

<P>PROBLEM TO BE SOLVED: To improve the estimating accuracy of a pole position by compensating the change of an induced voltage constant due to a temperature in the controller of a permanent magnet type synchronous motor. <P>SOLUTION: An induced voltage regulator 11 inputs a γ phase induced voltage estimated value eγ<SB>est</SB>and a δ phase induced voltage estimated value eδ<SB>est</SB>calculated by a γ-δ phase current/induced voltage estimating unit 8, finds the induced voltage constant Ke, and regulates the induced voltage constant Ke so that the quotient of the γ phase induced voltage estimated value eγ<SB>est</SB>and the δ phase induced voltage estimated value eδ<SB>est</SB>becomes "0". Even if the induced voltage constant Ke is changed in association with the temperature change of a permanent magnet, the temperature change of the induced voltage constant Ke is compensated by the induced voltage regulator 11, and the estimating accuracy of the pole position is improved. <P>COPYRIGHT: (C)2004,JPO&NCIPI

Description

【０００１】
【発明の属する技術分野】
本発明は、永久磁石を回転子とした永久磁石型同期電動機を制御するための永久磁石型同期電動機の制御装置に関し、特に永久磁石型同期電動機の磁極位置を推定する磁極位置推定方法に関する。
【０００２】
【従来の技術】
永久磁石を回転子とするブラシレスＤＣモータを同期電動機として運転する場合、回転子の絶対位置を得て、正確な運転を行う必要がある。回転子の絶対位置を得るためには、エンコーダやレゾルバ等の回転子位置検出器を用いることが一般的であるが、配線や構造の複雑さ、価格や使用環境等について問題があるため、回転位置検出器を用いないで回転子の磁極位置を求める方法が提案されている。
【０００３】
従来の永久磁石型同期発電機の磁極位置検出方法としては、モータに供給される電流、電圧を入力とし、誘起電圧をモータが回転していない時の電流応答に対する外乱として推定する外乱オブザーバを用いた磁極位置検出方法の提案がなされている（例えば、特許文献１参照。）。
【０００４】
しかし、この従来の永久磁石型同期発電機の磁極位置検出方法は、モータの電機モデルを規範とした推定機構であるため、推定精度はモデルに使用する電機定数の測定精度に影響を受ける。推定機構に必要な電機定数の１つである誘起電圧定数は、温度により変化する特性があるので、環境の変化やモータの温度上昇により推定精度が劣化するという問題があった。
【０００５】
温度による変化特性を補償して永久磁石型同期電動機の制御精度の向上を図った磁極位置推定方法としては、永久磁石の温度に対する減磁特性を磁束テーブルに予め記憶しておき、温度センサにより測定した温度に対応する磁束をこの磁束テーブルから読み出すことにより減磁特性を補償したトルク電流指令を得るようにした方法が開示されている（例えば、特許文献２参照。）。しかし、この従来の磁極位置推定方法では、永久磁石の温度を測定するための温度センサを設ける必要があり、配線や構造の複雑さを招くことにもなる。また、この従来の磁極位置推定方法によっては、誘起電圧定数の温度による変化を補償することはできないため、さらなる推定精度の向上が求められている。
【０００６】
【特許文献１】
特開平９−１９１６９８号公報
【特許文献２】
特開平９−５１７００号公報
【０００７】
【発明が解決しようとする課題】
上述した従来の永久磁石型同期電動機の制御装置では、永久磁石の温度変化により誘起電圧定数が変化してしまうため、磁極位置の推定精度が劣化してしまうという問題点があった。
【０００８】
本発明の目的は、誘起電圧定数の温度による変化を補償することにより磁極位置の推定精度を向上させることができる永久磁石型同期電動機の磁極位置推定方法および制御装置を提供することである。
【０００９】
【課題を解決するための手段】
上記目的を達成するために、本発明の永久磁石型同期電動機の制御装置は、外部から入力された角速度指令およびγ相電流指令に基づいて、永久磁石型同期電動機の制御を行うための、永久磁石型同期電動機の制御装置であって、
外部から入力された角速度指令、導出された角速度推定値に基づいてδ相電流指令を算出する速度コントローラと、
前記速度コントローラにより算出されたδ相電流指令と、算出されたδ相電流推定値とに基づいてδ相電圧指令を算出するδ相電流コントローラと、
外部から入力されたγ相電流指令と、算出されたγ相電流推定値とに基づいてγ相電圧指令を算出するγ相電流コントローラと、
前記δ相電流コントローラにより算出されたδ相電圧指令および前記γ相電流コントローラにより算出されたγ相電圧指令と、算出された磁極位置とに基づいて、電圧値絶対値とγ軸からの電圧出力方向の位相である電圧指令位相とを算出して出力するベクトル制御回路と、
前記ベクトル制御回路により算出された電圧値絶対値および電圧指令位相に基づいて制御対象である同期電動機に駆動電流を出力しているインバータ回路と、
前記同期電動機の少なくとも２相のステータ電流をγ−δ軸座標系に変換することにより、γ相電流およびδ相電流を算出する相変換器と、
前記相変換器により算出されたγ相電流、δ相電流と、導出された磁極位置と、前記δ相電流コントローラおよび前記γ相電流コントローラによりそれぞれ算出されたδ相電圧指令、γ相電圧指令に基づいてγ相電流推定値、γ相電流推定値、γ相誘起電圧推定値、およびδ相誘起電圧推定値を出力するγ−δ相電流・誘起電圧推定器と、
前記γ−δ相電流・誘起電圧推定器によって算出されたγ相誘起電圧推定値およびδ相誘起電圧推定値を用いて誘起電圧定数を導出し、前記γ相誘起電圧推定値と前記δ相誘起電圧推定値の商が０となるように前記誘起電圧定数を調整する誘起電圧調整器と、
前記誘起電圧調整器により導出された誘起電圧定数、γ相誘起電圧推定値、およびδ相誘起電圧推定値に基づいて角速度推定値を導出している角速度導出器と、
前記角速度導出器により導出された角速度推定値に基づいて磁極位置を導出して、前記ベクトル制御回路、前記インバータ回路、前記相変換器、前記γ−δ軸電流・誘起電圧推定器に出力する磁極位置導出器とを備えている。
【００１０】
本発明によれば、誘起電圧調整器は、γ−δ相電流・誘起電圧推定器で推定されたδ相誘起電圧推定値、δ相誘起電圧推定値の商に基づいて、誘起電圧定数の温度変化を補償するようにしているので、永久磁石の温度変化に伴い誘起電圧定数が変化した場合でも、この誘起電圧定数の温度変化を補償することにより磁極位置の推定精度を向上させることができる。
【００１１】
【発明の実施の形態】
先ず、本実施形態の永久磁石型同期電動機の磁極位置推定方法の概要について説明する。
【００１２】
上記の特許文献１に記載されている同期電動機のセンサレスベクトル制御方法では、先ず、回転子の磁軸上に設定したγ−δ軸座標系に変換されたステータ電流ｉγ、ｉδと、前回推定された電流値ｉγ_ｅｓｔ、ｉδ_ｅｓｔとの差および、γ−δ軸座標系に変換された電圧指令Ｖγ^＊、Ｖδ^＊を入力とし、γ−δ軸座標系における電流ｉγ_ｅｓｔ、ｉδ_ｅｓｔと誘起電圧ｅγ_ｅｓｔ、ｅδ_ｅｓｔおよび、回転子の角速度推定値ωｒ_ｅｓｔを算出していた。
【００１３】
これに対して本実施形態の永久磁石型同期電動機の磁極位置推定方法は、上記の同期電動機のセンサレスベクトル制御方法では、γ相誘電推定値ｅγ_ｅｓｔ、δ相誘起電圧推定値ｅδ_ｅｓｔを用いて、誘起電圧定数Ｋｅの温度変化を補償することにより推定される磁極位置の精度を向上させるものである。そこで、γ相誘電推定値ｅγ_ｅｓｔ、δ相誘起電圧推定値ｅδ_ｅｓｔを用いて誘起電圧定数Ｋｅの温度変化を補償する方法について下記に説明する。
【００１４】
時間ｋ・Ｔｓ秒時（但し、ｋ＝０、１、２、３、・・・、Ｔｓはサンプリングタイム）に同期電動機に供給される少なくとも２相分のステータ電流を検出し、このステータ電流を回転子上に設定したγ−δ座標系に変換することにより、γ相電流ｉγ（ｋ）、δ相電流ｉδ（ｋ）を導出する。そして、γ相電流ｉγ（ｋ）、δ相電流ｉδ（ｋ）と、前回導出したγ相電流推定値ｉγ_ｅｓｔ（ｋ）、δ相電流推定値ｉδ_ｅｓｔ（ｋ）と、γ相電圧指令Ｖγ^＊（ｋ）、δ相電圧指令Ｖδ^＊（ｋ）を用い、同期電動機のγ−δ軸座標系における状態方程式を離散値系に展開することにより下記の式（１）が得られる。
【００１５】
【数１】

但し、ここで、Ｒｓ：ステータ側抵抗、Ｌｑ：ｑ軸インダクタンス、Ｌｄ：ｄ軸インダクタンス、ωｒ：回転子角速度である。
【００１６】
この式（１）によって、時間（ｋ＋１）Ｔｓ秒時の電流推定値ｉγ_ｅｓｔ（ｋ＋１）、ｉδ_ｅｓｔ（ｋ＋１）、γ相誘起電圧推定値ｅγ_ｅｓｔ（ｋ＋１）、およびδ相誘起電圧推定値ｅδ_ｅｓｔ（ｋ＋１）が求まる。
【００１７】
なお、「^＊」は指令値であることを示し、「_ｅｓｔ」は推定値であることを示す。
【００１８】
また、γ相誘起電圧推定値ｅγ_ｅｓｔ（ｋ＋１）、およびδ相誘起電圧推定値ｅδ_ｅｓｔ（ｋ＋１）は、下記の式（２）、（３）のように表すことができる。
【００１９】
ｅγ_ｅｓｔ（ｋ＋１）＝−ｓｉｎθｅ（ωｒ_ｅｓｔ（ｋ＋１）・Ｋｅ）・・・（２）
ｅδ_ｅｓｔ（ｋ＋１）＝ｃｏｓθｅ（ωｒ_ｅｓｔ（ｋ＋１）・Ｋｅ）・・・（３）
ここで、Ｋｅは、誘起電圧定数、θｅは、γ−δ軸とｄ−ｑ軸とのずれ角、であるから、θｅが小さいと考え、ωｒ_ｅｓｔ（ｋ＋１）の符号を、
ｓｉｇｎ（ωｒ_ｅｓｔ（ｋ＋１））＝−ｓｉｇｎ（ｅδ_ｅｓｔ（ｋ＋１））・・・（４）
とし、式（２）、（３）の２乗和と式（４）の結果より、下記の式（５）でωｒ_ｅｓｔ（ｋ＋１）を求める。
【００２０】
【数２】

ここで、誘起電圧定数Ｋｅに正確な値Ｋｅ＿ｔが設定されているとすると、磁極位置とγ軸が一致し、上記の式（２）、（３）は、下記の式（６）、（７）となる。
【００２１】
ｅγ_ｅｓｔ（ｋ＋１）＝０・・・（６）
ｅδ_ｅｓｔ（ｋ＋１）＝ωｒ_ｅｓｔ（ｋ＋１）・Ｋｅ＿ｔ・・・（７）
しかし、正しい誘起電圧定数Ｋｅ＿ｔと設定値に誤差ΔＫｅがある場合、上記の式（５）は、
【００２２】
【数３】

となり、誤差ΔＫｅ分だけωｒ_ｅｓｔ（ｋ＋１）に誤差が生じる。回転子の位置θ_ｅｓｔ（ｋ＋１）はωｒ_ｅｓｔ（ｋ＋１）の積分により求めているので、ωｒ_ｅｓｔ（ｋ＋１）の誤差だけ回転子の位置θ_ｅｓｔ（ｋ＋１）にも誤差が生じる。そして、回転子の位置θ_ｅｓｔ（ｋ＋１）に誤差が存在すると、磁極位置とγ軸がずれθｅが生じる。つまり、誘起電圧定数誤差Δθが発生するとγ−δ軸とｄ−ｑ軸のずれθｅが生じ、γ相誘起電圧推定値ｅγ_ｅｓｔ（ｋ＋１）とδ相誘起電圧推定値ｅδ_ｅｓｔ（ｋ＋１）は、それぞれ上記の式（２）、（３）となる。よって、γ相誘起電圧推定値ｅγ_ｅｓｔとδ相誘起電圧推定値ｅδ_ｅｓｔの商はこの式（２）、（３）より下記の式（９）のように表される。
【００２３】
ｅγ_ｅｓｔ／ｅδ_ｅｓｔ＝ｓｉｎθｅ／ｃｏｓθｅ＝ｔａｎθｅ・・・（９）
そして、この式（９）より下記の式（１０）が求められる。
【００２４】
θｅ＝ａｔａｎ^−１（ｅγ_ｅｓｔ／ｅδ_ｅｓｔ）・・・（１０）
ｔａｎ^−１（アークタンジェント：逆正接関数）の計算をソフト化して処理するとソフト負荷率が高くなるので、θｅが零に近い値であると仮定して上記の式（１０）を下記の式（１１）により近似する。
【００２５】
θｅ＝ｅγ_ｅｓｔ／ｅδ_ｅｓｔ・・・（１１）
故に、誘起電圧定数Ｋｅに誤差がある場合、磁極位置とγ軸の間にずれθｅが生じ、そのずれθｅの量はγ相誘起電圧推定値ｅγ_ｅｓｔとδ相誘起電圧推定値ｅδ_ｅｓｔの商に相当する事がわかる。
【００２６】
よって、下記の式（１２）でｅγ_ｅｓｔ（ｋ＋１）／ｅδ_ｅｓｔ（ｋ＋１）＝０となるように誘起電圧定数Ｋｅを調整すれば、Ｋｅ＝Ｋｅ＿ｔとなる。
【００２７】
【数４】

但し、Ｋｐは比例ゲイン、Ｔは積分定数である。
【００２８】
次に、本発明の実施の形態について図面を参照して詳細に説明する。図１は本発明の一実施形態の永久磁石型同期電動機の制御装置の構成を示すブロック図である。
【００２９】
本実施形態の永久磁石型同期電動機の制御装置は、図１に示されるように、速度コントローラ１と、δ相電流コントローラ２と、γ相電流コントローラ３と、ベクトル制御回路４と、インバータ回路５と、相変換器７と、γ−δ相電流・誘起電圧推定器８と、角速度導出器９と、磁極位置導出器１０と、誘起電圧調整器１１とから構成され、外部から入力された角速度指令ωｒ^＊およびγ相電流指令ｉγ^＊に基づいて、永久磁石型のモータである同期電動機６の制御を行っている。
【００３０】
速度コントローラ１は、外部から入力された角速度指令ωｒ^＊、および角速度導出器９により導出された角速度推定値ωｒ_ｅｓｔに基づいてδ相電流指令ｉδ^＊を算出して出力する。
【００３１】
δ相電流コントローラ２は、速度コントローラ１により算出されたδ相電流指令ｉδ^＊と、δ−γ相電流・誘起電圧推定器８により算出されたδ相電流推定値ｉδ_ｅｓｔとに基づいてδ相電圧指令Ｖδ^＊を算出して出力する。
【００３２】
γ相電流コントローラ３は、外部から入力されたγ相電流指令ｉγ^＊と、δ−γ相電流・誘起電圧推定器８により算出されたγ相電流推定値ｉγ_ｅｓｔとに基づいてγ相電圧指令Ｖγ^＊を算出して出力する。
【００３３】
ベクトル制御回路４は、δ相電流コントローラ２により算出されたδ相電圧指令Ｖδ^＊およびγ相電流コントローラ３により算出されたγ相電圧指令Ｖγ^＊と、磁極位置導出器１０により算出された磁極位置θ_ｅｓｔに基づいて、電圧値絶対値（Ｖδ^２＋Ｖγ^２）^１／２とγ軸からの電圧出力方向の位相である電圧指令位相ｔａｎ^−１（Ｖδ／Ｖγ）を算出してインバータ回路５に出力している。
【００３４】
インバータ回路５は、ベクトル制御回路４により算出された電圧値絶対値（Ｖδ^２＋Ｖγ^２）^１／２および電圧指令位相ｔａｎ^−１（Ｖδ／Ｖγ）に基づいて同期電動機６に駆動電流を出力している。
【００３５】
相変換器７は、同期電動機６のステータ電流ｉｕ、ｉｖをγ−δ軸座標系に変換することにより、γ相電流ｉγおよびδ相電流ｉδを算出して出力している。
【００３６】
γ−δ相電流・誘起電圧推定器８は、相変換器７により算出されたγ相電流ｉγ、δ相電流ｉδと、磁極位置θ_ｅｓｔと、δ相電流コントローラ２およびγ相電流コントローラ３によりそれぞれ算出されたδ相電圧指令Ｖδ^＊、γ相電圧指令Ｖγ^＊を入力し、上記の式（１）の演算を実施し、γ相電流推定値ｉγ_ｅｓｔ、γ相電流推定値ｉδ_ｅｓｔと、γ相誘起電圧推定値ｅγ_ｅｓｔ、δ相誘起電圧推定値ｅδ_ｅｓｔを出力する。
【００３７】
誘起電圧調整器１１はγ−δ相電流・誘起電圧推定器８によって算出されたγ相誘起電圧推定値ｅγ_ｅｓｔおよびδ相誘起電圧推定値ｅδ_ｅｓｔ入力し、上述した式（１２）を実行することにより誘起電圧定数Ｋｅを導出し、γ相誘起電圧推定値ｅγ_ｅｓｔとδ相誘起電圧推定値ｅδ_ｅｓｔの商が０となるように誘起電圧定数Ｋｅを調整する。
【００３８】
角速度導出器９は、誘起電圧調整器１１により導出された誘起電圧定数Ｋｅと、γ相誘起電圧推定値ｅγ_ｅｓｔ、δ相誘起電圧推定値ｅδ_ｅｓｔを入力し、上述した式（４）、（５）を実行することにより、角速度推定値ωｒ_ｅｓｔを導出している。
【００３９】
磁極位置導出器１０は、角速度導出器９により導出された角速度推定値ωｒ_ｅｓｔに基づいて磁極位置θ_ｅｓｔを導出して、ベクトル制御回路４、インバータ回路５、相変換器７、γ−δ軸電流・誘起電圧推定器８に出力している。
【００４０】
次に、本実施形態の永久磁石型同期電動機の制御装置の動作について図２のフローチャートを参照して詳細に説明する。
【００４１】
先ず、速度コントローラ１は、角速度指令ωｒ^＊と角速度推定値ωｒ_ｅｓｔを入力して、δ相電流指令ｉδ^＊を出力する。δ相電流コントローラ２は、δ相電流指令ｉδ^＊とδ相電流推定値ｉδ_ｅｓｔとを入力し、δ相電圧指令Ｖδ^＊を出力する。
【００４２】
一方、γ相電流指令ｉγ^＊とγ相電流推定値ｉγ_ｅｓｔが、γ相電流コントローラ３に入力され、γ相電流コントローラ３はγ相電圧指令Ｖγ^＊を出力する。電圧指令Ｖδ^＊、Ｖγ^＊と、誘起電圧調整器１１から出力されるγ−δ軸位置がベクトル制御回路４に入力され、電圧値絶対値（Ｖδ^２＋Ｖγ^２）^１／２とγ軸からの電圧出力方向の位相ｔａｎ^−１（Ｖδ／Ｖγ）がインバータ回路５に入力され点弧が実施される。
【００４３】
一方、γ−δ相電流・誘起電圧推定器８は、同期電動機６のステータ電流ｉｕとｉｖを、相変換器７を介して得られるγ相電流ｉγ、δ相電流ｉδと、γ−δ軸の位置と、電圧指令Ｖδ^＊、Ｖγ^＊を入力し、上記の式（１）の演算を実施し、γ−δ相電流推定値ｉγ_ｅｓｔ、ｉδ_ｅｓｔと、γ−δ相誘起電圧ｅγ_ｅｓｔ、ｅδ_ｅｓｔを出力する。ｅγ_ｅｓｔが誘起電圧調整器１１に入力され、式（１２）を実行することにより、誘起電圧定数Ｋｅが導出される。この誘起電圧定数Ｋｅと、γ−δ相誘起電圧ｅγ_ｅｓｔ、ｅδ_ｅｓｔが角速度導出器９に入力され、上述した式（４）、（５）を実行することにより、角速度推定値ωｒ_ｅｓｔが導出される。この角速度推定値ωｒ_ｅｓｔが磁極位置導出器１０に入力され、磁極位置θ_ｅｓｔを出力する。
【００４４】
次に、制御動作を図２のフローチャートを参照して説明する。
【００４５】
相変換器７は、ｋ・Ｔｓ秒の時点で同期電動機６に供給されている少なくとも２相分の電流、例えばｉｕ（ｋ）、ｉｖ（ｋ）を検出し（ステップ１０１）、前回ループで補正されたγ−δ軸座標系に変換し、γ相電流ｉγ（ｋ）、δ相電流ｉδ（ｋ）を導出する（ステップ１０２）。次に、γ−δ相電流・誘起電圧推定器８では、γ相電流ｉγ（ｋ）、δ相電流ｉδ（ｋ）およびγ−δ座標系に変換された電圧指令Ｖγ^＊（ｋ）、Ｖδ^＊（ｋ）を入力し（ステップ１０３）、上述した式（１）により（ｋ＋１）・Ｔｓ秒後の推定値ｉγ_ｅｓｔ（ｋ＋１）、ｉδ_ｅｓｔ（ｋ＋１）、ｅγ_ｅｓｔ（ｋ＋１）、ｅδ_ｅｓｔ（ｋ＋１）を導出する（ステップ１０４）。
【００４６】
次に、誘起電圧調整器１１では、導出されたδ相誘起電圧推定値ｅδ_ｅｓｔ（ｋ＋１）が入力され、上述した式（９）を用いて誘起電圧定数Ｋｅの導出が行われる（ステップ１０５）。そして、角速度導出器９では、上述した式（４）を用いて、γ相誘起電圧推定値ｅγ_ｅｓｔ（ｋ＋１）の符号より、角速度推定値ωｒ_ｅｓｔ（ｋ＋１）の符号判断を行う（ステップ１０６）。そして、角速度導出器９では、上述した式（５）に基づいて、この符号と、γ相誘起電圧推定値ｅγ_ｅｓｔ（ｋ＋１）とδ相誘起電圧推定値ｅδ_ｅｓｔ（ｋ＋１）の２乗和を誘起電圧定数Ｋｅで除算してωｒ_ｅｓｔ（ｋ＋１）を導出する（ステップ１０７）。最後に、磁極位置導出器１０では、導出されたωｒ_ｅｓｔ（ｋ＋１）を積分してθ_ｅｓｔ（ｋ＋１）を導出する（ステップ１０８）。
【００４７】
本実施形態の永久磁石型同期電動機の制御装置によれば、γ−δ相電流・誘起電圧推定器８で推定されたγ相誘起電圧推定値ｅγ_ｅｓｔとδ相誘起電圧推定値ｅδ_ｅｓｔの商に基づいて、誘起電圧定数Ｋｅの温度変化を補償する手法をソフトウェアで構成することにより、高速で正確にパラメータを同定することができ、高性能な電動機制御が実現できる。
【００４８】
【発明の効果】
以上説明したように、本発明では、誘起電圧調整器は、γ相誘起電圧推定値とδ相誘起電圧推定値の商が０となるように誘起電圧定数を調整するようにしているので、永久磁石の温度変化に伴い誘起電圧定数が変化した場合でも、この誘起電圧定数の温度変化を補償することにより磁極位置の推定精度を向上させることができるという効果を得ることができる。
【図面の簡単な説明】
【図１】本発明の一実施形態の磁極位置、速度推定方法が適用された同期電動機の制御システムの構成を示すブロック図である。
【図２】図１の制御システムのデジタル制御動作を示すフローチャートである。
【符号の説明】
１速度コントローラ
２ δ相電流コントローラ
３ γ相電流コントローラ
４ベクトル制御回路
５インバータ回路
６同期電動機
７相変換器
８ γ−δ相電流・誘起電圧推定器
９角速度導出器
１０磁極位置導出器
１１誘起電圧調整器[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a permanent magnet type synchronous motor control device for controlling a permanent magnet type synchronous motor using a permanent magnet as a rotor, and more particularly to a magnetic pole position estimating method for estimating a magnetic pole position of a permanent magnet type synchronous motor.
[0002]
[Prior art]
When a brushless DC motor using a permanent magnet as a rotor is operated as a synchronous motor, it is necessary to obtain the absolute position of the rotor and perform accurate operation. In order to obtain the absolute position of the rotor, it is common to use a rotor position detector such as an encoder or a resolver.However, there are problems with the wiring and structure, the price, the usage environment, etc. There has been proposed a method of determining a magnetic pole position of a rotor without using a position detector.
[0003]
As a conventional method of detecting the magnetic pole position of a permanent magnet type synchronous generator, a disturbance observer is used in which a current and a voltage supplied to a motor are input and an induced voltage is estimated as a disturbance to a current response when the motor is not rotating. There has been proposed a magnetic pole position detecting method (for example, see Patent Document 1).
[0004]
However, since the conventional method of detecting the magnetic pole position of the permanent magnet type synchronous generator is an estimating mechanism based on an electric machine model of the motor, the estimation accuracy is affected by the measurement accuracy of the electric constant used in the model. Since the induced voltage constant, which is one of the electrical constants required for the estimating mechanism, has a characteristic that changes with temperature, there has been a problem that the estimation accuracy is deteriorated due to a change in environment or an increase in motor temperature.
[0005]
A magnetic pole position estimating method that compensates for the change characteristics due to temperature and improves the control accuracy of the permanent magnet type synchronous motor is as follows. The demagnetization characteristics with respect to the temperature of the permanent magnet are stored in advance in a magnetic flux table and measured by a temperature sensor. A method is disclosed in which a magnetic flux corresponding to the set temperature is read from the magnetic flux table to obtain a torque current command in which demagnetization characteristics are compensated (for example, see Patent Document 2). However, in this conventional magnetic pole position estimation method, it is necessary to provide a temperature sensor for measuring the temperature of the permanent magnet, which leads to complexity of wiring and structure. Further, the conventional magnetic pole position estimating method cannot compensate for a change in the induced voltage constant due to temperature, and therefore, a further improvement in estimation accuracy is required.
[0006]
[Patent Document 1]
Japanese Patent Application Laid-Open No. 9-191698 [Patent Document 2]
JP-A-9-51700
[Problems to be solved by the invention]
The above-described conventional permanent magnet type synchronous motor control device has a problem that the estimation accuracy of the magnetic pole position is deteriorated because the induced voltage constant changes due to the temperature change of the permanent magnet.
[0008]
An object of the present invention is to provide a magnetic pole position estimating method and a control device of a permanent magnet type synchronous motor that can improve the accuracy of magnetic pole position estimation by compensating for a change in induced voltage constant due to temperature.
[0009]
[Means for Solving the Problems]
In order to achieve the above object, a permanent magnet synchronous motor control device according to the present invention includes a permanent magnet synchronous motor for controlling a permanent magnet synchronous motor based on an externally input angular velocity command and a γ-phase current command. A control device for a magnet type synchronous motor,
An angular velocity command input from the outside, a speed controller that calculates a δ-phase current command based on the derived angular velocity estimated value,
Δ-phase current command calculated by the speed controller, a δ-phase current controller that calculates a δ-phase voltage command based on the calculated δ-phase current estimated value,
A γ-phase current command input from the outside, and a γ-phase current controller that calculates a γ-phase voltage command based on the calculated γ-phase current estimated value,
Based on the δ-phase voltage command calculated by the δ-phase current controller and the γ-phase voltage command calculated by the γ-phase current controller, and the calculated magnetic pole position, the absolute value of the voltage value and the voltage output from the γ axis are calculated. A vector control circuit that calculates and outputs a voltage command phase that is a phase in the direction,
An inverter circuit that outputs a drive current to the synchronous motor to be controlled based on the voltage value absolute value and the voltage command phase calculated by the vector control circuit,
A phase converter that calculates a γ-phase current and a δ-phase current by converting at least two-phase stator current of the synchronous motor into a γ-δ axis coordinate system;
The γ-phase current and δ-phase current calculated by the phase converter, the derived magnetic pole position, and the δ-phase voltage command and the γ-phase voltage command calculated by the δ-phase current controller and the γ-phase current controller, respectively. A γ-δ phase current / induced voltage estimator that outputs a γ-phase current estimated value, a γ-phase current estimated value, a γ-phase induced voltage estimated value, and a δ-phase induced voltage estimated value,
The induced voltage constant is derived using the estimated γ-phase induced voltage and the estimated δ-phase induced voltage calculated by the γ-δ phase current / induced voltage estimator, and the estimated γ-phase induced voltage and the δ-phase induced voltage are derived. An induced voltage regulator that adjusts the induced voltage constant such that the quotient of the voltage estimation value is 0,
An induced voltage constant derived by the induced voltage regulator, a γ-phase induced voltage estimate, and an angular velocity deriving unit that derives an angular velocity estimate based on the δ-phase induced voltage estimate,
A magnetic pole to derive a magnetic pole position based on the angular velocity estimated value derived by the angular velocity deriving device, and to output the vector control circuit, the inverter circuit, the phase converter, and the γ-δ axis current / induced voltage estimating device. A position deriving device.
[0010]
According to the present invention, the induced voltage regulator determines the temperature of the induced voltage constant based on the quotient of the estimated δ phase induced voltage and the estimated δ phase induced voltage estimated by the γ-δ phase current / induced voltage estimator. Since the change is compensated, even when the induced voltage constant changes due to the temperature change of the permanent magnet, the accuracy of estimating the magnetic pole position can be improved by compensating for the temperature change of the induced voltage constant.
[0011]
BEST MODE FOR CARRYING OUT THE INVENTION
First, the outline of the method for estimating the magnetic pole position of the permanent magnet type synchronous motor of the present embodiment will be described.
[0012]
In the sensorless vector control method for a synchronous motor described in Patent Document 1 described above, first, the stator currents iγ and iδ converted to a γ-δ axis coordinate system set on the magnetic axis of the rotor are estimated previously. The difference between the current values iγ _est and iδ _est and the voltage commands Vγ ^* and Vδ ^* converted into the γ-δ axis coordinate system are input, and the currents iγ _est and iδ _est and the induced voltage in the γ-δ axis coordinate system are input. eγ _est , eδ _est and the estimated angular velocity ωr _est of the rotor were calculated.
[0013]
On the other hand, in the method of estimating the magnetic pole position of the permanent magnet type synchronous motor according to the present embodiment, the sensorless vector control method of the synchronous motor described above uses the γ-phase dielectric estimated value eγ _est and the δ-phase induced voltage estimated value eδ _est. In addition, the accuracy of the estimated magnetic pole position is improved by compensating for the temperature change of the induced voltage constant Ke. Therefore, a method for compensating the temperature change of the induced voltage constant Ke using the estimated γ-phase dielectric value eγ _est and the estimated δ-phase induced voltage eδ _est will be described below.
[0014]
At time k · Ts seconds (where k = 0, 1, 2, 3,..., Ts is a sampling time), a stator current for at least two phases supplied to the synchronous motor is detected. By converting into a γ-δ coordinate system set on the rotor, γ-phase current iγ (k) and δ-phase current iδ (k) are derived. Then, the γ-phase current iγ (k), the δ-phase current iδ (k), the previously derived γ-phase current estimated value iγ _est (k), the δ-phase current estimated value iδ _est (k), and the γ-phase voltage command Vγ ^{The following} equation (1) is obtained by expanding the state equation in the γ-δ axis coordinate system of the synchronous motor into a discrete value system using ^* (k) and δ-phase voltage command Vδ ^* (k).
[0015]
(Equation 1)

Here, Rs: stator-side resistance, Lq: q-axis inductance, Ld: d-axis inductance, and ωr: rotor angular velocity.
[0016]
According to the equation (1), the current estimated values iγ _est (k + 1), iδ _est (k + 1), the γ-phase induced voltage estimated value eγ _est (k + 1), and the δ-phase induced voltage estimated value eδ at time (k + 1) Ts seconds _est (k + 1) is obtained.
[0017]
Note that “ ^* ” indicates a command value, and “ _est ” indicates an estimated value.
[0018]
Further, the estimated γ-phase induced voltage eγ _est (k + 1) and the estimated δ-phase induced voltage eδ _est (k + 1) can be represented by the following equations (2) and (3).
[0019]
eγ _est (k + 1) = − sin θe (ωr _est (k + 1) · Ke) (2)
eδ _est (k + 1) = cos θe (ωr _est (k + 1) · K e) (3)
Here, Ke is the induced voltage constant, and θe is the deviation angle between the γ-δ axis and the dq axis. Therefore, θe is considered to be small, and the sign of ωr _est (k + 1) is
sign (ωr _est (k + 1)) = − sign (eδ _est (k + 1)) (4)
From the sum of the squares of Equations (2) and (3) and the result of Equation (4), ωr _est (k + 1) is obtained by the following Equation (5).
[0020]
(Equation 2)

Here, assuming that an accurate value Ke_t is set for the induced voltage constant Ke, the magnetic pole position matches the γ axis, and the above equations (2) and (3) are replaced by the following equations (6) and (7). ).
[0021]
eγ _est (k + 1) = 0 (6)
eδ _est (k + 1) = ωr _est (k + 1) · Ke_t (7)
However, if there is an error ΔK e between the correct induced voltage constant Ke_t and the set value, the above equation (5) becomes
[0022]
[Equation 3]

Next, an error occurs in the error ΔKe amount corresponding ωr _est (k + 1). Since the position of the rotor θ _est (k + 1) is determined by integration of ωr _est (k + 1), an error occurs in ωr _est (k + 1) by an error of the rotor position θ _est (k + 1). Then, if there is an error in the position θ _est (k + 1) of the rotor, a deviation θe occurs between the magnetic pole position and the γ axis. That is, when the induced voltage constant error Δθ occurs, a shift θe between the γ-δ axis and the dq axis occurs, and the estimated γ-phase induced voltage eγ _est (k + 1) and the estimated δ-phase induced voltage eδ _est (k + 1) The above equations (2) and (3) are obtained, respectively. Accordingly, the quotient of the estimated value of the γ-phase induced voltage eγ _est and the estimated value of the δ-phase induced voltage eδ _est is represented by the following expression (9) from the expressions (2) and (3).
[0023]
eγ _est / eδ _est = sin θe / cos θe = tan θe (9)
Then, the following equation (10) is obtained from the equation (9).
[0024]
θe = atan ⁻¹ (eγ _est / eδ _est ) (10)
If the calculation of tan ^-1 (arc tangent: arc tangent function) is softened and processed, the soft load factor becomes high. Therefore, assuming that θe is a value close to zero, the above equation (10) is replaced with the following equation ( 11).
[0025]
θe = eγ _est / eδ _est (11)
Therefore, when there is an error in the induced voltage constant Ke, a shift θe occurs between the magnetic pole position and the γ-axis, and the amount of the shift θe is the quotient of the estimated γ-phase induced voltage eγ _est and the estimated δ-phase induced voltage eδ _est . It is understood that it corresponds to.
[0026]
Therefore, if the induced voltage constant Ke is adjusted so that eγ _est (k + 1) / eδ _est (k + 1) = 0 in the following equation (12), Ke = Ke_t.
[0027]
(Equation 4)

Here, Kp is a proportional gain, and T is an integration constant.
[0028]
Next, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 1 is a block diagram showing a configuration of a control device for a permanent magnet type synchronous motor according to one embodiment of the present invention.
[0029]
As shown in FIG. 1, the control device for a permanent magnet type synchronous motor of the present embodiment includes a speed controller 1, a δ-phase current controller 2, a γ-phase current controller 3, a vector control circuit 4, an inverter circuit 5, , A phase converter 7, a γ-δ phase current / induced voltage estimator 8, an angular velocity deriving device 9, a magnetic pole position deriving device 10, and an induced voltage adjusting device 11. The synchronous motor 6 which is a permanent magnet type motor is controlled based on the command ωr ^* and the γ-phase current command iγ ^* .
[0030]
The speed controller 1 calculates and outputs a δ-phase current command iδ ^* based on the angular speed command ωr ^* input from the outside and the estimated angular speed ωr _est derived by the angular speed deriving device 9.
[0031]
The δ-phase current controller 2 determines the δ-phase current based on the δ-phase current command iδ ^* calculated by the speed controller 1 and the estimated δ-phase current iδ _est calculated by the δ-γ phase current / induced voltage estimator 8. The voltage command Vδ ^* is calculated and output.
[0032]
The γ-phase current controller 3 outputs a γ-phase voltage command based on the γ-phase current command iγ ^* input from the outside and the γ-phase current estimated value iγ _est calculated by the δ-γ phase current / induced voltage estimator 8. Vγ ^* is calculated and output.
[0033]
The vector control circuit 4 includes a δ-phase voltage command Vδ ^* calculated by the δ-phase current controller 2, a γ-phase voltage command Vγ ^* calculated by the γ-phase current controller 3, and a magnetic pole position calculated by the magnetic pole position deriving unit 10. Based on θ _est , a voltage command phase tan ⁻¹ (Vδ / Vγ), which is a voltage value absolute value (Vδ ² + Vγ ² ) ^1/2 and a phase in a voltage output direction from the γ-axis, is calculated. Output.
[0034]
Inverter circuit 5 outputs a drive current to synchronous motor 6 based on voltage value absolute value (Vδ ² + Vγ ² ) ^1/2 calculated by vector control circuit 4 and voltage command phase tan ⁻¹ (Vδ / Vγ). ing.
[0035]
The phase converter 7 calculates and outputs the γ-phase current iγ and the δ-phase current iδ by converting the stator currents iu and iv of the synchronous motor 6 into a γ-δ axis coordinate system.
[0036]
The γ-δ phase current / induced voltage estimator 8 calculates the γ-phase current iγ, δ-phase current iδ calculated by the phase converter 7, the magnetic pole position θ _est , the δ-phase current controller 2 and the γ-phase current controller 3. The calculated δ-phase voltage command Vδ ^* and γ-phase voltage command Vγ ^* are input, and the calculation of the above equation (1) is performed, and the γ-phase current estimated value iγ _est , the γ-phase current estimated value iδ _est , The estimated γ-phase induced voltage eγ _est and the estimated δ-phase induced voltage eδ _est are output.
[0037]
The induced voltage adjuster 11 inputs the estimated value of the γ-phase induced voltage eγ _est and the estimated value of the δ-phase induced voltage eδ _est calculated by the γ-δ phase current / induced voltage estimator 8 and executes the above-described equation (12). Thus, the induced voltage constant Ke is derived, and the induced voltage constant Ke is adjusted so that the quotient of the estimated γ-phase induced voltage eγ _est and the estimated δ-phase induced voltage eδ _est becomes zero.
[0038]
The angular velocity deriving unit 9 inputs the induced voltage constant Ke derived by the induced voltage adjuster 11, the estimated value of the γ-phase induced voltage eγ _est , and the estimated value of the δ-phase induced voltage eδ _est, and obtains the above-described equations (4) and (4). By performing 5), the estimated angular velocity ωr _est is derived.
[0039]
The magnetic pole position deriving unit 10 derives the magnetic pole position θ _est based on the estimated angular velocity ωr _est derived by the angular velocity deriving unit 9, and the vector control circuit 4, the inverter circuit 5, the phase converter 7, and the γ-δ axis It outputs to the current / induced voltage estimator 8.
[0040]
Next, the operation of the control device for the permanent magnet type synchronous motor of the present embodiment will be described in detail with reference to the flowchart of FIG.
[0041]
First, the speed controller 1 receives the angular velocity command ωr ^* and the estimated angular velocity ωr _est and outputs a δ-phase current command iδ ^* . The δ-phase current controller 2 receives the δ-phase current command iδ ^* and the estimated δ-phase current value iδ _est and outputs a δ-phase voltage command Vδ ^* .
[0042]
On the other hand, the γ-phase current command iγ ^* and the estimated γ-phase current value iγ _est are input to the γ-phase current controller 3, and the γ-phase current controller 3 outputs the γ-phase voltage command Vγ ^* . The voltage commands Vδ ^* and Vγ ^* and the γ-δ axis position output from the induced voltage regulator 11 are input to the vector control circuit 4, and the absolute value of the voltage value (Vδ ² + Vγ ² ) ^1/2 and the value from the γ axis The phase tan ⁻¹ (Vδ / Vγ) in the voltage output direction is input to the inverter circuit 5 and firing is performed.
[0043]
On the other hand, the γ-δ phase current / induced voltage estimator 8 converts the stator currents iu and iv of the synchronous motor 6 into γ-phase currents iγ and δ-phase currents iδ obtained through the phase converter 7 and γ-δ axes. , And the voltage commands Vδ ^* and Vγ ^* are input, and the calculation of the above equation (1) is performed, and the γ-δ phase current estimated values iγ _est , iδ _est and the γ-δ phase induced voltage eγ _est , e _est is output. eγ _est is input to the induced voltage adjuster 11, and the induced voltage constant Ke is derived by executing equation (12). The induced voltage constant Ke and the γ-δ phase induced voltages eγ _est and eδ _est are input to the angular velocity deriving unit 9 and the above-described equations (4) and (5) are executed to derive the angular velocity estimated value ωr _est. Is done. The estimated angular velocity ωr _est is input to the magnetic pole position deriving device 10 to output the magnetic pole position θ _est .
[0044]
Next, the control operation will be described with reference to the flowchart of FIG.
[0045]
The phase converter 7 detects at least two phases of current, for example, iu (k) and iv (k), supplied to the synchronous motor 6 at the time of k · Ts seconds (step 101), and corrects the current in the previous loop. Γ-δ axis coordinate system, and γ-phase current iγ (k) and δ-phase current iδ (k) are derived (step 102). Next, in the γ-δ phase current / induced voltage estimator 8, the γ-phase current iγ (k), the δ-phase current iδ (k), and the voltage commands Vγ ^* (k), Vδ converted into the γ-δ coordinate system. ^* (k) inputted (steps 103), by the above-mentioned formula (1) (k + 1) · Ts seconds estimates _{iγ est (k + 1),} iδ est (k + 1), eγ est (k + 1), eδ est ( k + 1) is derived (step 104).
[0046]
Next, the induced voltage regulator 11 receives the derived δ-phase induced voltage estimated value eδ _est (k + 1), and derives the induced voltage constant Ke using the above-described equation (9) (step 105). . Then, the angular velocity deriving unit 9 determines the sign of the estimated angular velocity ωr _est (k + 1) from the sign of the estimated γ-phase induced voltage eγ _est (k + 1) using the above equation (4) (step 106). . Then, the angular velocity deriving unit 9 calculates the sign and the sum of squares of the estimated γ-phase induced voltage eγ _est (k + 1) and the estimated δ-phase induced voltage eδ _est (k + 1) based on the above equation (5). Divide by the induced voltage constant Ke to derive ωr _est (k + 1) (step 107). Finally, to derive the in pole position deriving unit 10, derived ωr _est (k + 1) the integrated and θ _est (k + 1) (step 108).
[0047]
According to the control device for the permanent magnet type synchronous motor of the present embodiment, the quotient of the estimated γ-phase induced voltage eγ _est and the estimated δ-phase induced voltage eδ _est estimated by the γ-δ phase current / induced voltage estimator 8. By using software to compose a method for compensating the temperature change of the induced voltage constant Ke based on the above, parameters can be identified accurately at high speed, and high-performance motor control can be realized.
[0048]
【The invention's effect】
As described above, in the present invention, the induced voltage adjuster adjusts the induced voltage constant so that the quotient of the estimated value of the γ-phase induced voltage and the estimated value of the δ-phase induced voltage becomes zero. Even when the induced voltage constant changes with the magnet temperature change, it is possible to obtain an effect that the accuracy of estimating the magnetic pole position can be improved by compensating for the temperature change of the induced voltage constant.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a configuration of a control system for a synchronous motor to which a magnetic pole position / speed estimation method according to an embodiment of the present invention is applied.
FIG. 2 is a flowchart showing a digital control operation of the control system of FIG. 1;
[Explanation of symbols]
Reference Signs List 1 speed controller 2 δ-phase current controller 3 γ-phase current controller 4 vector control circuit 5 inverter circuit 6 synchronous motor 7 phase converter 8 γ-δ phase current / induced voltage estimator 9 angular velocity deriver 10 magnetic pole position deriver 11 induced voltage Moderator

Claims

A control device for a permanent magnet type synchronous motor for controlling a permanent magnet type synchronous motor based on an angular velocity command and a γ-phase current command input from outside,
An angular velocity command input from the outside, a speed controller that calculates a δ-phase current command based on the derived angular velocity estimated value,
Δ-phase current command calculated by the speed controller, a δ-phase current controller that calculates a δ-phase voltage command based on the calculated δ-phase current estimated value,
A γ-phase current command input from the outside, and a γ-phase current controller that calculates a γ-phase voltage command based on the calculated γ-phase current estimated value,
Based on the δ-phase voltage command calculated by the δ-phase current controller and the γ-phase voltage command calculated by the γ-phase current controller, and the calculated magnetic pole position, the absolute value of the voltage value and the voltage output from the γ axis are calculated. A vector control circuit that calculates and outputs a voltage command phase that is a phase in the direction,
An inverter circuit that outputs a drive current to a synchronous motor to be controlled based on the voltage value absolute value and the voltage command phase calculated by the vector control circuit; a phase converter that calculates a γ-phase current and a δ-phase current by converting to a δ-axis coordinate system;
The γ-phase current and δ-phase current calculated by the phase converter, the derived magnetic pole position, and the δ-phase voltage command and the γ-phase voltage command calculated by the δ-phase current controller and the γ-phase current controller, respectively. A γ-δ phase current / induced voltage estimator that outputs a γ-phase current estimated value, a γ-phase current estimated value, a γ-phase induced voltage estimated value, and a δ-phase induced voltage estimated value,
The induced voltage constant is derived using the estimated γ-phase induced voltage and the estimated δ-phase induced voltage calculated by the γ-δ phase current / induced voltage estimator, and the estimated γ-phase induced voltage and the δ-phase induced voltage are derived. An induced voltage regulator that adjusts the induced voltage constant such that the quotient of the voltage estimation value is 0,
An induced voltage constant derived by the induced voltage regulator, a γ-phase induced voltage estimate, and an angular velocity deriving unit that derives an angular velocity estimate based on the δ-phase induced voltage estimate,
A magnetic pole to derive a magnetic pole position based on the angular velocity estimated value derived by the angular velocity deriving device, and to output the vector control circuit, the inverter circuit, the phase converter, and the γ-δ axis current / induced voltage estimating device. A control device for a permanent magnet type synchronous motor, comprising: a position deriving device.

A magnetic pole position estimation method for a permanent magnet type synchronous motor, which estimates a magnetic pole position in a control device that controls a permanent magnet type synchronous motor based on an angular velocity command and a γ-phase current command input from outside,
Angular velocity command input from the outside, a step of calculating a δ-phase current command based on the derived angular velocity estimated value,
Calculating the δ phase current command based on the calculated δ phase current command and the calculated δ phase current estimated value;
Calculating a γ-phase voltage command based on the γ-phase current command input from the outside and the calculated γ-phase current estimated value;
Based on the calculated δ-phase voltage command and the calculated γ-phase voltage command, and the calculated magnetic pole position, a voltage value absolute value and a voltage command phase that is a phase in a voltage output direction from the γ axis are calculated. Calculating and outputting;
Outputting a drive current to a synchronous motor to be controlled based on the calculated voltage value absolute value and the voltage command phase;
Calculating a γ-phase current and a δ-phase current by converting at least two-phase stator current of the synchronous motor into a γ-δ axis coordinate system;
The calculated γ-phase current, the δ-phase current, the derived magnetic pole position, the calculated δ-phase voltage command, the γ-phase current estimated value based on the γ-phase voltage command, the γ-phase current estimated value, outputting a γ-phase induced voltage estimated value and a δ-phase induced voltage estimated value;
An induced voltage constant is derived using the calculated estimated value of the γ-phase induced voltage and the estimated value of the δ-phase induced voltage, such that the quotient of the estimated value of the γ-phase induced voltage and the estimated value of the δ-phase induced voltage is 0. Adjusting the induced voltage constant to;
Deriving an angular velocity estimated value based on the derived induced voltage constant, the γ-phase induced voltage estimated value, and the δ-phase induced voltage estimated value,
A magnetic pole position estimating method for a permanent magnet type synchronous motor, comprising a step of deriving a magnetic pole position based on the derived angular velocity estimation value.