JP2004236188A - Multicarrier diversity demodulation method and multicarrier diversity demodulator - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a multicarrier diversity demodulator having a matching filter and an interference component elimination process. <P>SOLUTION: Quadrature demodulation of a received signal is performed by a quadrature demodulation part 1-i (i=1, 2), N complex digital signals are selected in an effective symbol extraction part 2-i and furthermore, M (M<N) complex digital signals without an influence of delayed wave are extracted by output of a time response estimation part of a multipath propagation environment estimation part 4-i and processed by a matching filter part 3-i. Maximum ratio composition by amplitude is performed to signals of two sets of L subcarriers separated by two matching filter parts 3-1 and 3-2 on adjusting phases by every frequency in a signal composition part 5. To the L complex signals as output of the maximum ratio composition, estimation of interference components between carriers, elimination of the interference component from the signals of two sets of L subcarriers separated by the matching filter parts 3-1 and 3-2 and diversity composite of the signals of two sets of L subcarriers from which the interference components between carriers are eliminated are repeated in three stages of interference component elimination parts 101, 102, 103. <P>COPYRIGHT: (C)2004,JPO&NCIPI

Description

【００１３】
以下、煩雑さを避けるため、行列の定数倍、ベクトルの定数倍については適宜省略し、例えばＮ点フーリエ逆変換の行列を単にＦ^＊などと示す。送信側において、Ｎ個のサブキャリアの複素信号（ヌルキャリアに対応するものは０）をＸ（ｋ）（ｋは０からＮ−１までの整数）とおくと、有効シンボル長の離散波形ｘ（ｎ）（ｎは０からＮ−１までの整数）は次の通りとなる。
【数２】

【００１４】
ここで、Ｎ点フーリエ逆変換の行列Ｆ^＊の要素を、第１列、第２列、第Ｎ列ごとに縦ベクトルｆ_０、ｆ_１、…、ｆ_Ｎ−１とおくと、各縦ベクトルｆ_０、ｆ_１、…、ｆ_Ｎ−１に対応する横ベクトル^ｔｆ_０、^ｔｆ_１、…、^ｔｆ_Ｎ−１は、各々第０、第１、第Ｎ−１のサブキャリアの波形になっている。Ｎ点フーリエ逆変換の行列Ｆ^＊の共役転置行列Ｆは、当該横ベクトル^ｔｆ_０、^ｔｆ_１、…、^ｔｆ_Ｎ−１の複素共役を第１行、第２行、…、第Ｎ行に有するものである。有効シンボル長の離散波形ｘ（ｎ）（ｎは０からＮ−１までの整数）をＮ点フーリエ変換の行列Ｆの右から乗ずることは、Ｎ本のサブキャリアを抽出する整合フィルタ群に当該有効シンボル長の離散波形ｘ（ｎ）を作用させてキャリア成分の複素信号（複素振幅）を求めることと等価である。
【００１５】
ここで、有効シンボル長から先頭のＮ−Ｍ個の欠けた、Ｍ個の複素ディジタル信号を整合フィルタに作用させる場合は次のように考えることができる。上記Ｎ次の各横ベクトル^ｔｆ_０、^ｔｆ_１、…、^ｔｆ_Ｎ−１の複素共役について、左からＮ−Ｍ個の欠けたＭ次の横ベクトルを、有効シンボル長から先頭のＮ−Ｍ個の欠けたＭ個の複素ディジタル信号からなる縦ベクトルに左から乗ずることで達成できることは明らかである。
【００１６】
本願発明においてはＮ点フーリエ逆変換、Ｎ点フーリエ変換を用いるＯＦＤＭ通信方式には限定されない。Ｌ個の複素信号で変調されたＬ個のサブキャリアの波形を合成した、Ｎ個の離散波形からＭ個を取り出した、時間軸上のＭ個の複素信号からＬ個のサブキャリアのシンボルを求める。変調側がＬ本のサブキャリアに対応するＬ個の複素シンボルからなるベクトルを、Ｍ行Ｌ列の行列Ａに右から乗ずることにより達成されるならば、復調側の整合フィルタの演算としては、Ｍ個の離散波形を示すＭ個の複素信号からなるベクトルを、Ｍ行Ｌ列の行列Ａの共役転置行列であるＬ行Ｍ列の行列Ａ^＊に右から乗ずることで達成できる。
【００１７】
勿論、ＯＦＤＭ復調側において、Ｎ点フーリエ変換の行列Ｆを用いる場合は、有効シンボル長から先頭のＮ−Ｍ個の欠けたＭ個の複素ディジタル信号の先頭にＮ−Ｍ個の０を加えたＮ次のベクトルをＮ点フーリエ変換の行列Ｆの右から乗じて、必要なＬ本のサブキャリア番号に対応する複素信号を取り出すことで達成できる。
【００１８】
１有効シンボル長のＮ個の複素ディジタル信号からなるベクトルをＮ点フーリエ変換の行列Ｆの右から乗じた場合は、各キャリア間の干渉が生じず、得られるＮ個の複素信号はヌルキャリアを含めたＮ本のサブキャリアのシンボルである。しかし、本願は有効シンボル長から先頭のＮ−Ｍ個の欠けたＭ個の複素ディジタル信号に対して整合フィルタを作用させているので、必要なＬ本のサブキャリア番号に対応する複素信号はキャリア間干渉を有する。これを次のように、ダイバーシチ合成をしながらキャリア間干渉を除去していく。
【００１９】
まず、非特許文献１の技術内容を説明する。複数個の整合フィルタの出力のうち、１組のＬ本のサブキャリア番号に対応する複素信号をＬ次のベクトルＸ_０で示す。まず、仮判定により、Ｌ次のベクトルＸ_０からＬ次のベクトルＸ_１を仮判定する。即ち、ＱＰＳＫ、１６ＱＡＭその他の変調方式にあった、複素信号のいずれかに当てはめる。この時、Ｌ次のベクトルＸ_０はキャリア間干渉を有するのでＬ次のベクトルＸ_１は送信されたＬ本のサブキャリアの複素信号とは一部異なっている可能性が高い。そこで仮判定されたＬ次のベクトルＸ_１から、ヌルキャリア番号の位置を０とおいたＮ次のベクトルを構成する。次に、当該Ｎ次のベクトルを変調した時間軸上の波形に変換する。この際、伝送経路特性（伝送経路の伝達関数）を乗じておく。次にＮ個の時間軸上の複素ディジタル信号先頭のＮ−Ｍ個を０と置き換えて、再度周波数軸上の値に変換するためにＮ点フーリエ変換する。次に周波数軸上の値においてＬ本のサブキャリアに対応するＬ個の複素信号を選択する。この際、伝送経路特性（伝送経路の伝達関数）で除する。これは行列Ｒを用いてＲＸ_１とおくことができる。
【００２０】
前記仮判定されたＬ次のベクトルＸ_１に対し、Ｌ次のベクトルＲＸ_１がえられたので、これらに差があれば、その差ＲＸ_１−Ｘ_１は、上記整合フィルタの出力Ｘ_０の有するキャリア間干渉成分である可能性が高い。そこで、整合フィルタの出力Ｘ_０から、ＲＸ_１−Ｘ_１を減じ、その結果をもとにＬ次のベクトルＸ_２を仮判定する。仮判定されたＬ次のベクトルＸ_２は、一段前の仮判定されたＬ次のベクトルＸ_１よりもキャリア間干渉が減っていることが下記実施例のシミュレーションで確かめられている。そこで同様に、仮判定されたＬ次のベクトルＸ_２から、ヌルキャリア番号の位置を０とおいたＮ次のベクトルを構成する。次に、当該Ｎ次のベクトルを変調した時間軸上の波形に変換する。この際、伝送経路特性（伝送経路の伝達関数）を乗じておく。次にＮ個の時間軸上の複素ディジタル信号先頭のＮ−Ｍ個を０と置き換えて、再度周波数軸上の値に変換するためにＮ点フーリエ変換する。次に周波数軸上の値においてＬ本のサブキャリアに対応するＬ個の複素信号を選択する。この際、伝送経路特性（伝送経路の伝達関数）で除する。こうして出力ＲＸ_２を求め、整合フィルタの出力Ｘ_０からＲＸ_２−Ｘ_２を減じ、その結果をもとにＬ次のベクトルＸ_３を仮判定する。このように、Ｌ本のサブキャリアの複素信号を順次仮判定し、整合フィルタにおけるキャリア間干渉を見積もってＬ個の複素信号をＬ本のサブキャリアの複素信号に近づけていくことができる。これが非特許文献１の技術内容である。
【００２１】
本願発明では、ダイバーシチ合成を各アンテナごとに非特許文献１の技術を適用した最終結果に適用するのではなく、信号合成をシンボル仮判定の前において各段の干渉成分除去工程のシンボル仮判定を一元化する。各段の仮判定されたシンボルはアンテナに対応した複数個の整合フィルタごとに設けられたキャリア間干渉推定小工程、干渉成分減算小工程を通して、元の複数個の整合フィルタの出力よりも確度の高い複数組のＬ本のサブキャリア信号となる。これを次段においてダイバーシチ合成し、その結果である１組のＬ本の信号からＬ本のサブキャリア信号を仮判定する。これを複数個の整合フィルタごとに設けられたキャリア間干渉推定小工程、干渉成分減算小工程を通して、アンテナに対応した元の複数個の整合フィルタの出力よりも確度の高い複数組のＬ本のサブキャリア信号となる。
【００２２】
このような干渉成分除去工程を１乃至複数段有することで、単に各アンテナごとに非特許文献１の技術を適用した最終結果にダイバーシチ合成を適用するよりも、格段にエラービットレートを抑制することが可能となる。
【００２３】
本願発明の復調方法又は復調方式は、ヌルキャリアでないＬ本のサブキャリアの複素信号は常に正確に得られるものではないが、後述する通り、通信方式として十分有効性のあるエラービットレートに押えることができる。これは復調に用いる時間軸上のシンボル長を短くすることができることを意味し、ガードインターバルを越える遅延波が到来しても、当該遅延波にあわせた、本来の有効シンボル長よりも短い時間軸上の離散波形からヌルキャリアでないＬ本のサブキャリアの複素信号を得ることが可能となる。
【００２４】
上記の通り、Ｌ次のベクトルＸ_１から、ＲＸ_１−Ｘ_１を求めるには、予め行列Ｒを求めておいてＲＸ_１−Ｘ_１と計算することも可能である。また、ＯＦＤＭの場合には、ヌルキャリア番号の位置を０とおいたＮ次のベクトルを構成して、伝送経路特性（伝送経路の伝達関数）を乗じた上でＮ点フーリエ逆変換し、得られるＮ個の時間軸上の複素ディジタル信号先頭のＮ−Ｍ個を０と置き換えて、Ｎ点フーリエ変換し、周波数軸上の値においてＬ本のサブキャリアに対応するＬ個の複素信号を選択した上で伝送経路特性（伝送経路の伝達関数）で除して、ＲＸ_１を求めてＸ_１を減じてＲＸ_１−Ｘ_１を計算しても良い。
【００２５】
【発明の実施の形態】
本発明は、遅延波の電力が小さい場合は、周波数歪みによる補正を行わなくても有効であるが、その場合はダイバーシチを用いた効果も小さい。そこで下記実施例においては当該周波数伝搬特性による複素信号の周波数歪みを修正する部分を省略してまず説明し、シミュレーションで用いた構成は別の図で示すものとする。
【００２６】
〔実施例〕
図１は本願の具体的な第１の実施例に係るマルチキャリアダイバーシチ復調装置１００の構成の概略を示すブロック図、図２は構成の詳細を示すブロック図である。マルチキャリアダイバーシチ復調装置１００は、２つのアンテナに接続され、第ｉ（本実施例ではｉは１又は２）アンテナに対応して各々直交復調部１−ｉ、有効シンボル抽出部２−ｉ、整合フィルタ部３−ｉ、マルチバス伝搬環境推定部４−ｉを有する。また、第１段の干渉成分除去部１０１として、シンボル仮判定部１１１、整合フィルタ３−ｉに対応して設けられた干渉推定部１２１−ｉ及び減算器１３１−ｉ、並びに信号合成部１４１を有する。第２段及び第３段の干渉成分除去部１０２及び１０３には同様に、それぞれシンボル仮判定部１１２と１１３が、整合フィルタ３−ｉに対応して設けられた干渉推定部１２２−ｉと１２３−ｉ及び減算器１３２−ｉと１３３−ｉが、並びに信号合成部１４２と１４３が設けられている。また、第１段の干渉成分除去部１０１のシンボル仮判定部１１１の前段には信号合成部５が、第３段の干渉成分除去部１０３の後段にはシンボル判定部６と並直列変換器７が設けられている。
【００２７】
図１のマルチキャリアダイバーシチ復調装置１００のマルチバス伝搬環境推定部４−１は、図２に示すように２つの部分である、時間応答推定部４１−１と周波数応答推定部４２−１で構成される。尚、図２では第２アンテナに対する構成を略しているが、マルチバス伝搬環境推定部４−２も、２つの部分である、時間応答推定部４１−２と周波数応答推定部４２−２で構成される。同様に各段の干渉推定部１２１−ｉ、１２２−ｉ、１２３−ｉは、各々２つの部分である、伝搬路特性再現部１５１−ｉ、１５２−ｉ、１５３−ｉと、キャリア間干渉推定部１６１−ｉ、１６２−ｉ、１６３−ｉとで構成される。
【００２８】
本実施例は請求項１乃至４の具体的な実施例に当たる。シンボル仮判定部１１１、１１２、１１３が仮判定器に、干渉推定部１２１−ｉ、１２２−ｉ、１２３−ｉが干渉成分推定器に、減算器１３１−ｉ、１３２−ｉ、１３３−ｉが干渉成分減算器に、干渉成分除去部１０１、１０２、１０３が干渉成分除去部に当たる。また、直交復調部１−ｉが直交復調及びサンプリング部に、マルチバス伝搬環境推定部４−ｉが遅延時間差推定部に、有効シンボル抽出部２−ｉが使用シンボル抽出部に当たる。
【００２９】
以下、本実施例ではガードインターバルを有するマルチキャリアダイバーシチ変調波からデータを復調するものとする。キャリア数はＮ本、うち有効キャリアをＬ本（Ｌ＜Ｎ）とする。
【００３０】
マルチキャリアダイバーシチ復調装置１００においては、受信信号が直交復調部１−ｉでいわゆる同相成分Ｉ及び直交成分Ｑの信号列が形成されたのち、有効シンボル抽出部２−ｉにおいて、シンボルタイミングにより１有効シンボル長ごとにＮ個の複素ディジタル信号が選択され、更にマルチバス伝搬環境推定部４−ｉの時間応答推定部が出力に基づき、遅延波の影響を受けないＭ個（Ｍ＜Ｎ）の複素ディジタル信号が抽出される。ここでは本来抽出すべきＮ個の複素ディジタル信号のうち、先頭のＮ−Ｍ個が遅延波の前シンボルの影響を受けたものとする。ガードインターバルを有している場合はここで同時にガードインターバルも除去される。Ｍ個の複素ディジタル信号が整合フィルタ部３−ｉに出力される。
【００３１】
整合フィルタ部３−ｉにおいては、まずＭ個の複素ディジタル信号の先頭に、Ｎ−Ｍ個の０を加えてＮ個の複素信号とする。次に高速フーリエ変換（ＦＦＴ）を行う。この出力の内、ヌルキャリアでないサブキャリア番号に対応するものを選択して出力する。尚、Ｌ個のヌルキャリアでないサブキャリアによるＭ個の複素ディジタル信号を生成するためのＭ行Ｌ列の行列Ａの共役転置行列Ａ^＊を用意し、右からＭ個の複素ディジタル信号からなるベクトルを乗ずることによりＬ個の複素信号を求めても良い。マルチキャリアダイバーシチ復調装置１００の整合フィルタ部３−ｉの出力であるＬ個の複素信号は、各サブキャリアに整合したフィルタにより分離することで、ノイズの影響を最小限に抑えたものであるが、他のサブキャリアからの干渉成分（キャリア間干渉）を含んでいる。
【００３２】
２個の整合フィルタ部３−１及び３−２で分離された２組のＬ本のサブキャリアの信号は、信号合成部５においてダイバーシチ合成される。この時、マルチバス伝搬環境推定部４−１及び４−２の周波数応答推定部４２−１、４２−２から各々伝搬路特性が各サブキャリアごとに複素係数として出力される。よってこの複素係数を用いて２組のＬ本のサブキャリアの信号を周波数ごとに位相を調整した上、振幅による最大比合成を行う。尚、ダイバーシチ合成の方法は任意の方法をとり得る。この出力であるＬ個の複素信号が第１段の干渉成分除去部１０１のシンボル仮判定部１１１に出力される。
【００３３】
第１段の干渉成分除去部１０１のシンボル仮判定部１１１にて仮判定が行われ、その出力は、整合フィルタ３−１に対応した干渉推定部１２１−１の伝搬路特性再現部１５１−１と、整合フィルタ３−２に対応した干渉推定部１２１−２の伝搬路特性再現部１５１−２とに出力される。伝搬路特性再現部１５１−１及び後段のキャリア間干渉推定部１６１−１、減算器１３１−１においては、マルチバス伝搬環境推定部４−１の周波数応答推定部４２−１の出力する伝搬路係数に基づき次のようにして受信信号を再現する。
【００３４】
まず伝搬路特性再現部１５１−１において、Ｌ個の複素信号に対し、ヌルキャリアの番号に対応してＮ−Ｌ個の０が挿入され、Ｎ個のサブキャリア信号が形成される。これは、そもそもの受信信号とは若干異なるレプリカであり、上述の通り、キャリア間干渉成分による影響が含まれている。これをマルチバス伝搬環境推定部４−１の周波数応答推定部４２−１の出力する伝搬路係数に基づき周波数歪みをかける（伝送路の伝達関数を乗ずる）。次に逆高速フーリエ変換（ＩＦＦＴ）してＮ個の時間軸上の複素信号を得る。次にキャリア間干渉推定部１６１−１において、マルチバス伝搬環境推定部４−１の時間応答推定部４１−１の出力する有効シンボル長に基づき、有効シンボル区間抽出ウインドウで先頭のＮ−Ｍ個の複素信号を０に置換したＮ個の複素信号とする。これを高速フーリエ変換して周波数軸上のＮ個の複素信号を得る。次にマルチバス伝搬環境推定部４−１の周波数応答推定部４２−１の出力する伝搬路係数に基づき周波数歪みを除く（伝送路の伝達関数で除する）。ヌルキャリアでないサブキャリア番号に対応するＬ個を選択する。この出力は、Ｌ個の複素数からなり、整合フィルタ部３−１の出力とは若干異なるレプリカがである。ここからＬ個の複素数であるシンボル仮判定部１１１の出力を減算すると、Ｌ個のサブキャリア各々のキャリア間干渉成分を概算することができる。尚、シンボル仮判定部１１１の出力するＬ個の複素信号に対して周波数歪みを加えたのちの、ヌルキャリアの挿入、ＩＦＦＴ処理及び有効シンボル長の選択を、周波数歪みが加わったＬ個の複素信号に対し１の行列Ａを左から乗ずる演算により行って時間軸上のＭ個の複素信号を得ても良い。また、ＦＦＴ処理に変えて、その時間軸上のＭ個の複素信号、又は時間軸上のＮ個の複素信号からＭ個を選択したのち、当該行列Ａの共役転置行列Ａ^＊を左から乗ずる演算によってＬ個の複素信号を得ても良く、そもそも周波数歪みが加わったＬ個の複素信号に対し行列Ａ^＊Ａを乗じて新たなＬ個の複素信号を求めても良い。このＬ個の干渉成分を減算器１３１−１でＬ個の複素数からなる整合フィルタ部３−１の出力から減じると、整合フィルタ部３−１の出力よりもより確度の高い各サブキャリアの信号が得られる。
【００３５】
全く同様にして、シンボル仮判定部１１１の出力するＬ個の複素信号に対して伝搬路特性再現部１５１−２、後段のキャリア間干渉推定部１６１−２、減算器１３１−２において、マルチバス伝搬環境推定部４−２の周波数応答推定部４２−２の出力する伝搬路係数に基づき、整合フィルタ部３−２の出力よりもより確度の高い各サブキャリアの信号が得られる。こうして、信号合成部１４１において、減算器１３１−１と１３１−２の出力である２組のＬ個の複素信号を、マルチバス伝搬環境推定部４−１及び４−２の周波数応答推定部４２−１、４２−２から各々出力される伝搬路特性に基づき、最大比合成によりダイバーシチ合成を行う。尚、ダイバーシチ合成は公知の任意の方法を採用できる。
【００３６】
第１段の干渉成分除去部１０１の信号合成部１４１の出力であるＬ個の複素信号は、整合フィルタ３−１、３−２の出力よりも、また、それらをダイバーシチ合成した信号合成部５の出力よりも確度の高い各サブキャリアの信号である。このような干渉成分除去を第２段の干渉成分除去部１０２、第３段の干渉成分除去部１０３で行う。最終段である第３段の干渉成分除去部１０３の信号合成部１４３の出力であるＬ個の複素信号は、整合フィルタ３−１、３−２の出力よりも、また、それらをダイバーシチ合成した信号合成部５の出力よりも、第１段の干渉成分除去部１０１の信号合成部１４１又は第２段の干渉成分除去部１０２の信号合成部１４２の出力よりも確度の高い各サブキャリアの信号である。これをシンボル判定部６で最終判定し、並直列変換器（Ｐ／Ｓ）７で並直列変換して復調データを得る。
【００３７】
図３にマルチキャリアダイバーシチ復調装置１００のシミュレーションを示す。シミュレーション条件は、無線ＬＡＮ（８０２．１１ａ）に準拠し、ＤＣキャリアを含む１２本をヌルとしたＬ＝５２のサブキャリアを用い、ガードインターバル長Ｔ_ＧＩを有効シンボル長Ｔの１／４、サブキャリア変調をＱＰＳＫ、２波レイリーフェージングモデル、遅延時間差を２Ｔ_ＧＩとした。本発明によれば以下に示す比較例に比べ格段にビット誤り率の低いマルチキャリアダイバーシチ復調装置を構成することができる。
【００３８】
〔第１の比較例〕
例えば特開平１１−２０５２０８（東芝）の出願内容である。図４は、図３で比較例１とした、高速フーリエ変換器を２台有するマルチキャリアダイバーシチ復調装置９００の構成を示すブロック図である。マルチキャリアダイバーシチ復調装置９００はアンテナを２つ接続するものであり、各々のアンテナｉに対し、直交復調部１−ｉ、シンボルタイミング検出部９４１−ｉ、周波数応答推定部９４２−ｉ、ガードインターバル（ＧＩ）除去部９２−ｉ、高速フーリエ変換器（ＦＦＴ）９３−ｉを有する。また、信号合成部９５、シンボル判定部９６、並直列変換器（Ｐ／Ｓ）７を有する。この構成のうち、直交復調部１−ｉと並直列変換器（Ｐ／Ｓ）７の作用は図１、図２のマルチキャリアダイバーシチ復調装置１００の直交復調部１−ｉと並直列変換器（Ｐ／Ｓ）７の作用と同一であり、信号合成部９５、シンボル判定部９６は入出力するデータの大きさが異なるものの実質的な作用は図１、図２のマルチキャリアダイバーシチ復調装置１００の信号合成部５、シンボル判定部６と変わらない。また周波数応答推定部９４２−ｉは、図１、図２のマルチキャリアダイバーシチ復調装置１００の周波数応答推定部４２−ｉと比較して出力先のみが異なるだけである。このような構成の通常のＯＦＤＭ受信部を２台設けて１回のダイバーシチ合成を行った場合、図３に示す通り、遅延時間差が２Ｔ_ＧＩに及ぶとビット誤り率は０．０４程度と非常に大きな値となる。
【００３９】
〔第２の比較例〕
図５は、図３で比較例２とした、非特許文献１の復調部を２台有するマルチキャリアダイバーシチ復調装置９９０の構成を示すブロック図である。マルチキャリアダイバーシチ復調装置９９０はアンテナを２つ接続するものであり、各々のアンテナｉに対し、直交復調部１−ｉ、有効シンボル抽出部２−ｉ、整合フィルタ部３−ｉ、時間応答推定部４１−ｉ、周波数応答推定部４２−ｉ、第１段の干渉成分除去部９０１−ｉとして、伝搬路特性補償部９７１−ｉ、シンボル仮判定部９１１−ｉ、伝搬路特性再現部９５１−ｉ、キャリア間干渉推定部９６１−ｉ及び減算器９３１−ｉを有する。第２段及び第３段の干渉成分除去部９０２−ｉ及び９０３−ｉには同様に、それぞれ伝搬路特性補償部９７２−ｉと９７３−ｉ、シンボル仮判定部９１２−ｉと９１３−ｉ、伝搬路特性再現部９５２−ｉと９５３−ｉ、キャリア間干渉推定部９６２−ｉと９６３−ｉ、減算器９３２−ｉと９３３−ｉが設けられている。また、最終段である第３段の干渉成分除去部９０３−１及び９０３−２の後段には信号合成部９９５とシンボル判定部６と並直列変換器７が設けられている。この構成のうち、図１、図２のマルチキャリアダイバーシチ復調装置１００と同一の符号を付したものはその作用は図１、図２のマルチキャリアダイバーシチ復調装置１００の同一の符号のブロックの作用と同一であり、信号合成部９９５はその入力及び出力がＬ個であること以外は図４の信号合成部９５と実質同一である。図１、図２のマルチキャリアダイバーシチ復調装置１００において信号合成部５、１４１、１４２においてダイバーシチ合成を周波数歪みを除去したのち行うのに替えて、伝搬路特性補償部９７１〜３−ｉを各段のシンボル仮判定部９１１〜３−ｉの前段に設けて周波数歪みを除去する構成とし、シンボル仮判定部９１１〜３−ｉをアンテナごとに設けた。このような構成のマルチキャリアダイバーシチ復調装置９９０は、図３に示す通り、遅延時間差が２Ｔ_ＧＩに及ぶとビット誤り率は０．００３程度である。これは図４のマルチキャリアダイバーシチ復調装置９００に比して１／１０以上改善されているが、ダイバーシチ合成を１回しか用いていないため、各段の干渉成分除去部でダイバーシチ合成を行う図１のマルチキャリアダイバーシチ復調装置１００のビット誤り率には及ばない。
【００４０】
〔変形例〕
図６は、図１のマルチキャリアダイバーシチ復調装置１００を一部変形した、本発明に係るマルチキャリアダイバーシチ復調装置２００の構成を示すブロック図である。本変形例は、図１の構成においてアンテナを２、干渉成分推定部を３段としていたものを、アンテナを３、干渉成分推定部を２段としたものである。符号ｉが１、２又は３になった点を除けば図１と図６とで同一の符号を付してあるものは同一の作用をするブロックである。また、信号合成部２０５、２４１、２４２は、３組のＬ個の複素信号を３つのマルチパス伝搬環境推定部４−１、４−２、４−３からの各周波数の係数を基に最大比合成することを除けば図１の信号合成部５、１４１、１４２と同一の作用をするものである。
【００４１】
上述の各実施例では、最も早い到来波と最も遅い遅延波との遅延時間差を考慮し、且つ最も早い到来波により同期が確立することを前提としているが、遅延時間差推定部の働きとして次のような遅延時間差を各構成要素に出力するとしても良い。即ち、各遅延波の強度も考慮した遅延時間差、或いは、予め定められた閾値以上の電力を持つ最も遅い遅延波と最も早い到来波との遅延時間差が挙げられる。
【００４２】
上述の各実施例では、遅延波の有効シンボル先頭或いはガードインターバル末尾から最も早い到来波の有効シンボル末尾までを使用シンボル部としたが、使用シンボル部は当該区間の内部であれば良い。
【図面の簡単な説明】
【図１】本願の実施例に係るマルチキャリアダイバーシチ復調装置１００の構成の概略を示すブロック図。
【図２】本願の実施例に係るマルチキャリアダイバーシチ復調装置１００の構成の詳細を示すブロック図。
【図３】本願の実施例に係るマルチキャリアダイバーシチ復調装置１００の平均ビット誤り率を示すグラフ図。
【図４】第１の比較例に係るマルチキャリアダイバーシチ復調装置９００の構成を示すブロック図。
【図５】第２の比較例に係るマルチキャリアダイバーシチ復調装置９９０の構成を示すブロック図。
【図６】実施例の変形例に係るマルチキャリアダイバーシチ復調装置２００の構成の概略を示すブロック図。
【符号の説明】
１−ｉ 第ｉアンテナに対応した直交復調部
２−ｉ 第ｉアンテナに対応した有効シンボル抽出部
３−ｉ 第ｉアンテナに対応した整合フィルタ部
４−ｉ 第ｉアンテナに対応したマルチパス伝搬環境推定部
５ 信号合成部
１０１、１０２、１０３、２０１、２０２ 干渉成分除去合成部
１１１、１１２、１１３、２１１、２２１ シンボル仮判定部
１２１−ｉ、１２２−ｉ、１２３−ｉ、２２１−ｉ、２２２−ｉ 干渉推定部
１３１−ｉ、１３２−ｉ、１３３−ｉ、２３１−ｉ、２３２−ｉ 減算器
１５１−ｉ、１５２−ｉ、１５３−ｉ キャリア間干渉推定部
１６１−ｉ、１６２−ｉ、１６３−ｉ 伝搬路特性再現部
１４１、１４２、１４３ 信号合成部
６ シンボル判定部
７ 並直列変換器（Ｐ／Ｓ）
９３−１、９３−２ 高速フーリエ変換器（ＦＦＴ）[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a multicarrier diversity demodulation method and a multicarrier diversity demodulation device. The present invention is particularly effective for OFDM reception in a multipath environment.
[0002]
[Prior art]
For example, in the OFDM modulation system, a waveform called a guard interval is added before an effective symbol in order to prevent the orthogonality from being broken due to the superposition of a delayed wave. The guard interval is, for example, added with the last 1/4 of an effective symbol and makes 1 symbol 5/4 times so that a delay wave having a delay time equal to or less than the guard interval length is not affected at the time of demodulation. It is. At this time, a waveform corresponding to the effective symbol length is used for demodulation by applying a “window” of the effective symbol length.
[0003]
However, when a delayed wave having a delay time difference exceeding the guard interval arrives due to the influence of multipath, there is a problem that the error rate is greatly deteriorated. In such a case, it is necessary to increase the guard interval length, but this increases the redundancy of communication and results in lower communication efficiency.
[0004]
By the way, OFDM often has many null carriers such as guard bands even when a band of N carriers is used. In view of the above, the applicant of the present invention in Japanese Patent Application No. 2002-281868 based on Japanese Patent Application No. 2001-298078 pays attention to the fact that all effective carriers can be demodulated from a shorter symbol length in OFDM communication having a null carrier. The present invention discloses a demodulation method and a demodulation device in which an error rate is not significantly deteriorated even under the influence of a multi-bus in which a delayed wave having a delay time difference exceeding a guard interval arrives by separating and demodulating an effective carrier without using a signal portion that is present. did. The inventor has posted and announced this.
[0005]
[Non-patent document 1]
N. Suzuki, et al. , IEICE Trans. Fundamentals, Vol. E85-A, No. 12 Dec. 2002, p. 2859
[0006]
[Problems to be solved by the invention]
In a multipath environment, a plurality of demodulation units based on the above technique may be prepared, and the error bit rate may be suppressed by diversity using the same number of antennas. However, the present inventor has found that the error bit rate can be further suppressed by incorporating the diversity technology into the above technology, and completed the present invention.
[0007]
[Means for Solving the Problems]
The invention according to claim 1 of the present application is directed to a multi-channel communication system in which the effective symbol length is T, the frequency interval between adjacent N subcarriers is 1 / T, and NL (L <N) subcarriers are null carriers. In a multicarrier diversity demodulation method in which a carrier modulated signal is received by a plurality of antennas and separated and demodulated into subcarriers, a multipath propagation environment on a time axis and a frequency axis provided for each of a plurality of antennas is estimated. And a step provided for each of the plurality of antennas, based on a delay time difference between the delayed waves estimated in the step of estimating the multipath propagation environment, so as not to include a portion where waveform distortion due to the delayed wave occurs. A used symbol portion having a length TM / N (M <N) is determined from the effective symbol length T, and the used symbol portion is orthogonally demodulated at a sampling interval T / N. A step of extracting the symbol portion M points; and a step of separating the L subcarriers by using a matched filter using the used symbol portion M points provided for each of the steps of extracting the plurality of used symbol portion M points. A step of diversity combining signals of a plurality of sets of L subcarriers, which is an output of the step of separating by the plurality of matched filters, and a small step of temporarily determining a signal of the diversity combined L subcarriers, An interference component between each of the L subcarriers, which is provided for each of the plurality of matched filters and is separated from the temporarily determined L symbols by the corresponding matched filter, included in each signal of the L subcarriers Is determined in consideration of the frequency characteristics of the multipath propagation environment, and L subcarriers are separated from the L subcarriers separated by the corresponding matched filter. One or more stages including a small step of reducing the interference component between carriers and a small step of diversity combining signals of a plurality of sets of L subcarriers which are outputs of the small step of reducing the plurality of interference components. And a multi-carrier diversity demodulation method.
[0008]
According to a second aspect of the present invention, in the multi-carrier diversity demodulation method according to the first aspect, the Marciaria modulation is OFDM, and the matched filter includes the used symbol portion M points and NM 0s. In the small step of inputting the N complex digital signals and using the N-point fast Fourier transform to reduce the interference component, the frequency distortion calculated in the step of estimating the multipath propagation environment is An N-point inverse fast Fourier transform is performed on N complex digital signals of the tentatively determined L symbols and NL null symbols, and M points corresponding to the used symbol portion M points are selected from the result. After replacing the NM symbols with 0, N-point fast Fourier transform is performed to select L symbols corresponding to the subcarriers, and the L symbols are obtained from the difference from the tentatively determined L symbols. And it features.
[0009]
Further, the invention according to claim 3 is a multi-carrier communication system in which the effective symbol length is T, the frequency interval between adjacent N subcarriers is 1 / T, and NL (L <N) subcarriers are null carriers. In a multi-carrier diversity demodulator that receives a carrier modulated signal with a plurality of antennas and separates and demodulates the received signals into subcarriers, N number of orthogonally demodulated sampling intervals T / N provided for each of the plurality of antennas are provided. A quadrature demodulation and sampling unit for obtaining a complex digital signal, a delay time difference estimating unit provided for each of the plurality of antennas, for estimating a delay time difference of a delayed wave and a frequency distortion of a received wave, From the delay time difference of the delay time difference estimating unit provided, the N complex digital signals are included so as not to include a portion where the waveform distortion due to the delayed wave occurs. A symbol extraction unit for extracting M (M <N) complex digital signals as symbols to be used among the signals, and the M complex signals output from the symbol extraction unit provided for each of the plurality of antennas A matched filter unit that separates L subcarriers by a matched filter using a digital signal, a signal combining unit that diversity-combines a plurality of sets of L subcarrier signals output from the plurality of matched filters, A provisional decision unit for provisionally determining L symbols from the signals of L subcarriers output from the signal combining unit, and a provisional decision unit corresponding to the provisionally determined L symbols provided for each of the plurality of matched filters. An interference component estimator for obtaining an interference component between the L subcarriers separated by the matched filter unit and other subcarriers included in each signal; An interference component subtractor provided for each matched filter for subtracting the interference component between L subcarriers from the signal of L subcarriers output from the corresponding matched filter unit, and subtracting the plurality of interference components A multi-carrier demodulation device comprising: a signal combining unit for diversity combining signals of a plurality of sets of L subcarriers output from a device; and one or more interference component removing units.
[0010]
According to a fourth aspect of the present invention, in the multi-carrier diversity demodulator according to the third aspect, the Marciaria modulation is OFDM, and the matched filter unit is configured to output the M complex digital signals output from the used symbol extracting unit. A signal and N complex digital signals of N-M 0 are input and formed by using an N-point fast Fourier transform. The interference component estimator calculates a frequency distortion calculated by a multipath propagation environment estimating unit. A null carrier inserter for adding NL null symbols to the L symbols temporarily determined, and an inverse fast Fourier for N-point inverse fast Fourier transform of the N complex digital signals output from the null carrier inserter. Selecting M points corresponding to the M complex digital signals output by the used symbol extraction unit from the outputs of the converter and the inverse fast Fourier transformer. An effective symbol interval extraction window unit that replaces the other NM with 0, a fast Fourier transformer for N-point fast Fourier transform of N complex digital signals output from the effective symbol interval extraction window unit, and a fast Fourier transformer And a subtracter for calculating a difference between the L complex digital signals output from the carrier selector and the L symbols provisionally determined. It is characterized by the following.
[0011]
[Action and effect of the invention]
The multi-carrier diversity demodulation method and the multi-carrier diversity demodulation device of the present invention do not use N complex digital signals having one effective symbol length, but use N-M missing, M complex digital signals. Is calculated for the L subcarriers that are not the same. Here, the matched filter works to estimate the signal on the frequency axis based on the signal on the time axis partially missing, instead of each subcarrier having one effective symbol length. That is, L matched filters corresponding to a waveform on the time axis partially missing from one effective symbol length of each of the L subcarriers are realized as one matched filter by, for example, one matrix operation. First, this will be described. The missing part will be described as the head of the effective symbol length.
[0012]
For example, in the OFDM communication system using the N-point Fourier inverse transform and the N-point Fourier transform, if the N-point Fourier transform is defined as a matrix F, the matrix of the N-point Fourier inverse transform is (1 / N) F ^* Can be saved. Where the matrix F ^* Denotes the conjugate transpose of the matrix F, and the matrix F of the N-point Fourier transform is a symmetric matrix, and the (k + 1) rows and (n + 1) columns of the matrix F are as follows.
(Equation 1)

[0013]
Hereinafter, in order to avoid complexity, the constant multiplication of the matrix and the constant multiplication of the vector are omitted as appropriate. ^* And so on. On the transmitting side, if a complex signal of N subcarriers (0 corresponding to a null carrier) is set to X (k) (k is an integer from 0 to N−1), a discrete waveform x of an effective symbol length is obtained. (N) (n is an integer from 0 to N-1) is as follows.
(Equation 2)

[0014]
Here, the matrix F of the inverse N-point Fourier transform ^* Of the vertical vector f for each of the first column, the second column, and the Nth column. ₀ , F ₁ , ..., f _N-1 In other words, each vertical vector f ₀ , F ₁ , ..., f _N-1 Horizontal vector corresponding to ^t f ₀ , ^t f ₁ ,…, ^t f _N-1 Are the waveforms of the 0th, 1st, and N-1st subcarriers, respectively. N-point Fourier inverse matrix F ^* Is the conjugate transposed matrix F of ^t f ₀ , ^t f ₁ ,…, ^t f _N-1 , In the first, second,..., Nth rows. Multiplying a discrete waveform x (n) (n is an integer from 0 to N-1) of an effective symbol length from the right of a matrix F of an N-point Fourier transform corresponds to a matched filter group for extracting N subcarriers. This is equivalent to obtaining a complex signal (complex amplitude) of a carrier component by applying a discrete waveform x (n) having an effective symbol length.
[0015]
Here, when the first NM missing M complex digital signals from the effective symbol length are applied to the matched filter, it can be considered as follows. Each of the above-mentioned N-order horizontal vectors ^t f ₀ , ^t f ₁ ,…, ^t f _N-1 By multiplying from the left, NM missing M-order horizontal vectors from the left with the vertical vector consisting of the leading NM missing M complex digital signals from the effective symbol length for the complex conjugate of Clearly what can be achieved.
[0016]
The present invention is not limited to the OFDM communication method using the N-point Fourier inverse transform and the N-point Fourier transform. Symbols of L subcarriers are extracted from M complex signals on the time axis by extracting M from N discrete waveforms by combining the waveforms of L subcarriers modulated by L complex signals. Ask. If the modulation side is achieved by multiplying a matrix A of M rows and L columns from the right by a vector composed of L complex symbols corresponding to L subcarriers, the operation of the matched filter on the demodulation side is M A vector composed of M complex signals indicating the discrete waveforms is transformed into a matrix A of L rows and M columns, which is a conjugate transposed matrix of a matrix A of M rows and L columns. ^* From the right.
[0017]
Of course, when the matrix F of the N-point Fourier transform is used on the OFDM demodulation side, NM 0s are added to the head of the NM missing M complex digital signals at the head from the effective symbol length. This can be achieved by multiplying the N-order vector from the right of the matrix F of the N-point Fourier transform and extracting the complex signal corresponding to the required L subcarrier numbers.
[0018]
When a vector composed of N complex digital signals having one effective symbol length is multiplied from the right of the matrix F of the N-point Fourier transform, no interference occurs between the carriers, and the obtained N complex signals have null carriers. This is a symbol of N subcarriers including the subcarrier. However, in the present application, a matching filter is applied to the first NM missing complex digital signals from the effective symbol length, so that the complex signal corresponding to the required L subcarrier numbers is a carrier signal. Interference. As described below, the interference between carriers is removed while performing diversity combining.
[0019]
First, the technical content of Non-Patent Document 1 will be described. Of the outputs of the plurality of matched filters, a complex signal corresponding to a set of L subcarrier numbers is converted into an L-order vector X ₀ Indicated by First, an L-order vector X is determined by tentative determination. ₀ To the L-th order vector X ₁ Is temporarily determined. That is, it is applied to any of the complex signals corresponding to QPSK, 16QAM and other modulation methods. At this time, the L-order vector X ₀ Has an inter-carrier interference, so the L-order vector X ₁ Is likely to be partially different from the transmitted complex signal of L subcarriers. Then, the L-order vector X temporarily determined ₁ , An Nth-order vector in which the position of the null carrier number is set to 0 is formed. Next, the N-order vector is converted into a modulated waveform on the time axis. At this time, the transmission path characteristic (transfer function of the transmission path) is multiplied. Next, the top N-M complex digital signals on the N time axes are replaced with 0, and N-point Fourier transform is performed again to convert them to values on the frequency axis. Next, L complex signals corresponding to the L subcarriers in the value on the frequency axis are selected. At this time, it is divided by the transmission path characteristic (transfer function of the transmission path). This is done using the matrix R as RX ₁ Can be saved.
[0020]
The tentatively determined L-order vector X ₁ , The L-order vector RX ₁ If there is a difference between them, the difference RX ₁ -X ₁ Is the output X of the matched filter ₀ It is highly possible that this is the inter-carrier interference component of Therefore, the output X of the matched filter ₀ From RX ₁ -X ₁ And an L-order vector X based on the result ₂ Is temporarily determined. L-order vector X temporarily determined ₂ Is the tentatively determined L-order vector X of the previous stage ₁ It has been confirmed by the simulation of the following example that the inter-carrier interference is reduced as compared with the above. Then, similarly, the L-order vector X temporarily determined ₂ , An Nth-order vector in which the position of the null carrier number is set to 0 is formed. Next, the N-order vector is converted into a modulated waveform on the time axis. At this time, the transmission path characteristic (transfer function of the transmission path) is multiplied. Next, the top N-M complex digital signals on the N time axes are replaced with 0, and N-point Fourier transform is performed again to convert them to values on the frequency axis. Next, L complex signals corresponding to the L subcarriers in the value on the frequency axis are selected. At this time, it is divided by the transmission path characteristic (transfer function of the transmission path). Thus, the output RX ₂ And the output X of the matched filter ₀ To RX ₂ -X ₂ And an L-order vector X based on the result ₃ Is temporarily determined. In this way, it is possible to tentatively determine the complex signals of the L subcarriers sequentially, estimate the inter-carrier interference in the matched filter, and bring the L complex signals closer to the complex signals of the L subcarriers. This is the technical content of Non-Patent Document 1.
[0021]
In the present invention, instead of applying diversity combining to the final result of applying the technique of Non-Patent Document 1 for each antenna, signal combining is performed before symbol temporary determination in each stage by symbol temporary determination in the interference component removal step before symbol temporary determination. Centralize. The tentatively determined symbols of each stage are more accurate than the outputs of the original plurality of matched filters through an inter-carrier interference estimation small step and an interference component subtraction small step provided for each of the plurality of matched filters corresponding to the antennas. A plurality of high sets of L subcarrier signals are obtained. This is subjected to diversity combining in the next stage, and L subcarrier signals are provisionally determined from a set of L signals obtained as a result. This is passed through an inter-carrier interference estimation sub-step and an interference component subtraction sub-step provided for each of a plurality of matched filters, through a plurality of L sets of L sets having higher accuracy than the outputs of the original plurality of matched filters corresponding to the antennas. It becomes a subcarrier signal.
[0022]
By having one or more stages of such interference component removal steps, it is possible to significantly reduce the error bit rate rather than simply applying diversity combining to the final result of applying the technique of Non-Patent Document 1 for each antenna. Becomes possible.
[0023]
According to the demodulation method or the demodulation method of the present invention, a complex signal of L subcarriers which are not null carriers is not always obtained accurately, but as described later, an error bit rate which is sufficiently effective as a communication method is required. Can be. This means that the symbol length on the time axis used for demodulation can be shortened. Even if a delayed wave exceeding the guard interval arrives, the time axis shorter than the original effective symbol length is adjusted to the delayed wave. From the above discrete waveform, it is possible to obtain a complex signal of L subcarriers that are not null carriers.
[0024]
As described above, the L-order vector X ₁ From RX ₁ -X ₁ Is determined by calculating a matrix R in advance and RX ₁ -X ₁ It is also possible to calculate In the case of OFDM, an Nth-order vector in which the position of a null carrier number is set to 0 is formed, multiplied by a transmission path characteristic (transfer function of the transmission path), and then subjected to an N-point Fourier inverse transform. The N-M complex digital signals at the beginning of the N time axes are replaced with 0, and N-point Fourier transform is performed, and L complex signals corresponding to L subcarriers in the value on the frequency axis are selected. Above divided by the transmission path characteristic (transfer function of the transmission path) ₁ X in search of ₁ RX ₁ -X ₁ May be calculated.
[0025]
BEST MODE FOR CARRYING OUT THE INVENTION
The present invention is effective without performing correction by frequency distortion when the power of the delayed wave is small, but in that case, the effect of using the diversity is also small. Therefore, in the following embodiments, a part for correcting the frequency distortion of a complex signal due to the frequency propagation characteristic will be omitted and described first, and the configuration used in the simulation will be shown in another figure.
[0026]
〔Example〕
FIG. 1 is a block diagram schematically showing the configuration of a multicarrier diversity demodulator 100 according to a first specific example of the present application, and FIG. 2 is a block diagram showing the details of the configuration. The multi-carrier diversity demodulation device 100 is connected to two antennas, and corresponds to an i-th (i is 1 or 2 in this embodiment) antenna, respectively, a quadrature demodulation unit 1-i, an effective symbol extraction unit 2-i, a matching It has a filter unit 3-i and a multi-bus propagation environment estimation unit 4-i. Further, as the first-stage interference component removing section 101, a symbol temporary determining section 111, an interference estimating section 121-i and a subtractor 131-i provided corresponding to the matched filter 3-i, and a signal combining section 141 are included. Have. Similarly, the second-stage and third-stage interference component elimination units 102 and 103 respectively include provisional symbol determination units 112 and 113 respectively provided with interference estimation units 122-i and 123 provided corresponding to the matched filter 3-i. -I and subtractors 132-i and 133-i, and signal combining units 142 and 143 are provided. Further, the signal combining section 5 is provided before the symbol provisional determining section 111 of the first-stage interference component removing section 101, and the symbol determining section 6 and the parallel / serial converter 7 are provided downstream of the third-stage interference component removing section 103. Is provided.
[0027]
The multi-bus propagation environment estimating section 4-1 of the multi-carrier diversity demodulation apparatus 100 in FIG. 1 is composed of two parts, a time response estimating section 41-1 and a frequency response estimating section 42-1 as shown in FIG. Is done. Although the configuration for the second antenna is omitted in FIG. 2, the multi-bus propagation environment estimation unit 4-2 also includes two parts, a time response estimation unit 41-2 and a frequency response estimation unit 42-2. Is done. Similarly, the interference estimating units 121-i, 122-i, and 123-i of each stage are respectively composed of two parts, a propagation path characteristic reproducing unit 151-i, 152-i, and 153-i, and an inter-carrier interference estimating unit. It is composed of sections 161-i, 162-i and 163-i.
[0028]
This embodiment corresponds to a specific embodiment of claims 1 to 4. The symbol tentative decision units 111, 112, and 113 are tentative deciders, the interference estimating units 121-i, 122-i, and 123-i are interference component estimators, and the subtractors 131-i, 132-i, and 133-i are The interference component subtractors 101, 102 and 103 correspond to the interference component subtractor. The quadrature demodulation unit 1-i corresponds to a quadrature demodulation and sampling unit, the multi-bus propagation environment estimation unit 4-i corresponds to a delay time difference estimation unit, and the effective symbol extraction unit 2-i corresponds to a used symbol extraction unit.
[0029]
Hereinafter, in the present embodiment, it is assumed that data is demodulated from a multicarrier diversity modulated wave having a guard interval. The number of carriers is N, and the number of effective carriers is L (L <N).
[0030]
In the multi-carrier diversity demodulator 100, after the received signal is formed into a signal sequence of the so-called in-phase component I and the quadrature component Q in the quadrature demodulation unit 1-i, the effective symbol extraction unit 2-i activates one signal according to the symbol timing. N complex digital signals are selected for each symbol length, and the time response estimating unit of the multi-bus propagation environment estimating unit 4-i is based on the output and outputs M (M <N) complex signals unaffected by delayed waves. A digital signal is extracted. Here, it is assumed that of the N complex digital signals to be originally extracted, the first NM signals are affected by the preceding symbol of the delayed wave. If there is a guard interval, the guard interval is also removed here. M complex digital signals are output to matched filter section 3-i.
[0031]
In the matched filter unit 3-i, first, NM zeros are added to the head of the M complex digital signals to obtain N complex signals. Next, fast Fourier transform (FFT) is performed. Among these outputs, those corresponding to subcarrier numbers that are not null carriers are selected and output. Note that the conjugate transpose matrix A of the matrix A of M rows and L columns for generating M complex digital signals by L non-null carrier subcarriers ^* May be prepared, and L complex signals may be obtained by multiplying by a vector composed of M complex digital signals from the right. The L complex signals output from the matching filter unit 3-i of the multicarrier diversity demodulation apparatus 100 are separated by a filter matched to each subcarrier to minimize the influence of noise. , And interference components from other subcarriers (inter-carrier interference).
[0032]
Two sets of L subcarrier signals separated by the two matched filter units 3-1 and 3-2 are subjected to diversity combining in the signal combining unit 5. At this time, the propagation path characteristics are output as complex coefficients for each subcarrier from the frequency response estimation units 42-1 and 42-2 of the multi-bus propagation environment estimation units 4-1 and 4-2. Therefore, using the complex coefficients, the phases of the two sets of L subcarrier signals are adjusted for each frequency, and then the maximum ratio combining based on the amplitude is performed. In addition, the method of diversity synthesis can take any method. The output L complex signals are output to the provisional symbol determination section 111 of the first-stage interference component removal section 101.
[0033]
A provisional decision is made in the provisional symbol decision section 111 of the first-stage interference component removal section 101, and the output is output to the propagation path characteristic reproduction section 151-1 of the interference estimation section 121-1 corresponding to the matched filter 3-1. And the propagation path characteristic reproducing unit 151-2 of the interference estimating unit 121-2 corresponding to the matched filter 3-2. In the propagation path characteristic reproducing section 151-1, the inter-carrier interference estimating section 161-1 and the subtractor 131-1 in the subsequent stage, the propagation path output from the frequency response estimating section 42-1 of the multi-bus propagation environment estimating section 4-1. The received signal is reproduced based on the coefficient as follows.
[0034]
First, in the propagation path characteristic reproducing unit 151-1, NL 0s are inserted into the L complex signals corresponding to the null carrier numbers, and N subcarrier signals are formed. This is a replica slightly different from the received signal in the first place, and includes the influence of the inter-carrier interference component as described above. This is subjected to frequency distortion (multiplied by the transfer function of the transmission path) based on the propagation path coefficient output from the frequency response estimation section 42-1 of the multi-bus propagation environment estimation section 4-1. Next, inverse fast Fourier transform (IFFT) is performed to obtain N complex signals on the time axis. Next, in the inter-carrier interference estimating unit 161-1, based on the effective symbol length output from the time response estimating unit 41-1 of the multi-bus propagation environment estimating unit 4-1, the first NM number of symbols in the effective symbol section extraction window are displayed. Are replaced with 0 complex signals as N complex signals. This is subjected to fast Fourier transform to obtain N complex signals on the frequency axis. Next, frequency distortion is removed based on the propagation path coefficient output from the frequency response estimation section 42-1 of the multibus propagation environment estimation section 4-1 (divided by the transfer function of the transmission path). The L number corresponding to the subcarrier number which is not a null carrier is selected. This output is composed of L complex numbers, and is a replica slightly different from the output of the matched filter unit 3-1. By subtracting the output of the symbol temporary determination unit 111, which is L complex numbers, the inter-carrier interference component of each of the L subcarriers can be roughly estimated. It should be noted that insertion of a null carrier, IFFT processing, and selection of an effective symbol length after applying frequency distortion to the L complex signals output from the symbol temporary determination section 111 are performed on the L complex signals to which frequency distortion has been added. The signal may be multiplied by one matrix A from the left to obtain M complex signals on the time axis. Also, instead of FFT processing, after selecting M complex signals on the time axis or M complex signals on the time axis, the conjugate transpose matrix A of the matrix A is selected. ^* May be obtained by multiplying the L complex signals from the left, and the matrix A is calculated for the L complex signals to which frequency distortion is added in the first place. ^* A may be multiplied by A to obtain new L complex signals. When the L interference components are subtracted by the subtractor 131-1 from the output of the matched filter unit 3-1 composed of L complex numbers, the signal of each subcarrier having higher accuracy than the output of the matched filter unit 3-1 is obtained. Is obtained.
[0035]
In exactly the same way, the multipath of the L complex signals output from the provisional symbol determination section 111 is output to the propagation path characteristic reproduction section 151-2, the inter-carrier interference estimation section 161-2 and the subtractor 131-2 at the subsequent stage. Based on the propagation path coefficient output from the frequency response estimation section 42-2 of the propagation environment estimation section 4-2, a signal of each subcarrier with higher accuracy than the output of the matched filter section 3-2 is obtained. Thus, in the signal combining section 141, the two sets of L complex signals output from the subtracters 131-1 and 131-2 are combined with the frequency response estimating sections 42 of the multi-bus propagation environment estimating sections 4-1 and 4-2. Based on the channel characteristics output from -1 and 42-2, diversity combining is performed by maximum ratio combining. Note that any known method can be used for diversity synthesis.
[0036]
The L complex signals output from the signal combining unit 141 of the first-stage interference component removing unit 101 are output from the matched filters 3-1 and 3-2, and the signal combining unit 5 that diversity-combines them. Is a signal of each subcarrier having higher accuracy than the output of the subcarrier. Such interference component removal is performed by the second-stage interference component removal unit 102 and the third-stage interference component removal unit 103. The L complex signals output from the signal synthesis unit 143 of the third-stage interference component elimination unit 103, which is the final stage, are subjected to diversity synthesis with respect to the outputs of the matched filters 3-1 and 3-2. The signal of each subcarrier having higher accuracy than the output of the signal combining unit 5 is higher than the output of the signal combining unit 141 of the first-stage interference component removing unit 101 or the output of the signal combining unit 142 of the second-stage interference component removing unit 102. It is. This is finally determined by a symbol determination unit 6 and parallel-to-serial converted by a parallel / serial converter (P / S) 7 to obtain demodulated data.
[0037]
FIG. 3 shows a simulation of the multicarrier diversity demodulator 100. The simulation conditions are based on the wireless LAN (802.11a), using 12 subcarriers including DC carriers as null and L = 52, and using a guard interval length T _GI Is 1/4 of the effective symbol length T, subcarrier modulation is QPSK, 2-wave Rayleigh fading model, delay time difference is 2T _GI And According to the present invention, a multicarrier diversity demodulator having a significantly lower bit error rate than the comparative example described below can be configured.
[0038]
[First Comparative Example]
For example, this is the content of an application filed in Japanese Patent Application Laid-Open No. H11-205208 (Toshiba). FIG. 4 is a block diagram showing a configuration of a multicarrier diversity demodulator 900 having two fast Fourier transformers, which is a comparative example 1 in FIG. The multi-carrier diversity demodulation device 900 connects two antennas. For each antenna i, a quadrature demodulation unit 1-i, a symbol timing detection unit 941-i, a frequency response estimation unit 942-i, a guard interval ( GI) It has a removing unit 92-i and a fast Fourier transformer (FFT) 93-i. Further, it has a signal synthesizing unit 95, a symbol determining unit 96, and a parallel / serial converter (P / S) 7. In this configuration, the operations of the quadrature demodulation unit 1-i and the parallel / serial converter (P / S) 7 are the same as those of the quadrature demodulation unit 1-i and the parallel / serial converter (P / S) of the multicarrier diversity demodulation device 100 shown in FIGS. P / S) 7 is the same as that of the multi-carrier diversity demodulator 100 shown in FIG. 1 and FIG. There is no difference from the signal combining unit 5 and the symbol determining unit 6. The frequency response estimator 942-i differs from the frequency response estimator 42-i of the multicarrier diversity demodulator 100 in FIGS. 1 and 2 only in the output destination. In the case where two ordinary OFDM receivers having such a configuration are provided and one diversity combination is performed, as shown in FIG. _GI , The bit error rate becomes a very large value of about 0.04.
[0039]
[Second comparative example]
FIG. 5 is a block diagram showing a configuration of a multi-carrier diversity demodulator 990 having two demodulators of Non-Patent Document 1, which is Comparative Example 2 in FIG. The multi-carrier diversity demodulation device 990 connects two antennas. For each antenna i, a quadrature demodulation unit 1-i, an effective symbol extraction unit 2-i, a matched filter unit 3-i, a time response estimation unit 41-i, a frequency response estimating unit 42-i, a first-stage interference component removing unit 901-i, a channel characteristic compensating unit 971-i, a tentative symbol determining unit 911-i, and a channel characteristic reproducing unit 951-i. , An inter-carrier interference estimator 961-i and a subtractor 931-i. Similarly, the second-stage and third-stage interference component elimination units 902-i and 903-i respectively include propagation path characteristic compensation units 972-i and 973-i, temporary symbol determination units 912-i and 913-i, Propagation path characteristic reproducing units 952-i and 953-i, inter-carrier interference estimating units 962-i and 963-i, and subtractors 932-i and 933-i are provided. Further, a signal combining section 995, a symbol determining section 6, and a parallel-to-serial converter 7 are provided downstream of the third-stage interference component removing sections 903-1 and 903-2, which are final stages. In this configuration, those having the same reference numerals as those of the multicarrier diversity demodulation device 100 shown in FIGS. 1 and 2 have the same effects as those of the blocks having the same reference numerals of the multicarrier diversity demodulation device 100 shown in FIGS. The signal synthesizing unit 995 is substantially the same as the signal synthesizing unit 95 of FIG. 4 except that the number of inputs and outputs is L. In the multi-carrier diversity demodulator 100 of FIGS. 1 and 2, the signal combining units 5, 141, and 142 perform the diversity combining after removing the frequency distortion. Are provided in front of the symbol temporary determination units 911 to 3-i to remove the frequency distortion, and the symbol temporary determination units 911 to 3-i are provided for each antenna. As shown in FIG. 3, the multi-carrier diversity demodulator 990 having such a configuration has a delay time difference of 2T. _GI , The bit error rate is about 0.003. This is improved by 1/10 or more compared to the multi-carrier diversity demodulator 900 in FIG. 4, but since diversity combining is used only once, the diversity combining unit in each stage performs diversity combining. Does not reach the bit error rate of the multi-carrier diversity demodulation device 100 of FIG.
[0040]
(Modification)
FIG. 6 is a block diagram showing a configuration of a multicarrier diversity demodulation apparatus 200 according to the present invention, in which the multicarrier diversity demodulation apparatus 100 of FIG. 1 is partially modified. In this modified example, the configuration in FIG. 1 has two antennas and three interference component estimating units, but has three antennas and two interference component estimating units. Except for the point where the symbol i is 1, 2, or 3, the same reference numerals in FIGS. 1 and 6 denote blocks that perform the same operation. Further, the signal combining units 205, 241, and 242 convert the three sets of L complex signals to the maximum based on the coefficients of the respective frequencies from the three multipath propagation environment estimating units 4-1, 4-2, and 4-3. Except for the ratio synthesis, the signal synthesizing sections 5, 141, and 142 in FIG. 1 operate in the same manner.
[0041]
In each of the above-described embodiments, the delay time difference between the earliest arriving wave and the latest delay wave is taken into consideration, and it is assumed that synchronization is established by the earliest arriving wave. Such a delay time difference may be output to each component. That is, the delay time difference considering the strength of each delay wave, or the delay time difference between the slowest delayed wave having the power equal to or more than a predetermined threshold value and the earliest arriving wave can be cited.
[0042]
In each of the above embodiments, the used symbol portion is from the beginning of the effective symbol of the delayed wave or the end of the guard interval to the end of the effective symbol of the earliest arriving wave. However, the used symbol portion may be inside the section.
[Brief description of the drawings]
FIG. 1 is a block diagram schematically illustrating a configuration of a multicarrier diversity demodulation device 100 according to an embodiment of the present application.
FIG. 2 is a block diagram showing details of a configuration of a multicarrier diversity demodulation device 100 according to the embodiment of the present application.
FIG. 3 is a graph showing an average bit error rate of the multicarrier diversity demodulator 100 according to the embodiment of the present application.
FIG. 4 is a block diagram showing a configuration of a multicarrier diversity demodulator 900 according to a first comparative example.
FIG. 5 is a block diagram showing a configuration of a multicarrier diversity demodulator 990 according to a second comparative example.
FIG. 6 is a block diagram schematically illustrating a configuration of a multicarrier diversity demodulation device 200 according to a modification of the embodiment.
[Explanation of symbols]
1-i Quadrature demodulation unit corresponding to i-th antenna
2-i Effective symbol extraction unit corresponding to the ith antenna
3-i Matching filter unit corresponding to i-th antenna
4-i Multipath propagation environment estimating unit corresponding to the ith antenna
5 Signal synthesis section
101, 102, 103, 201, 202 Interference component removal / combination unit
111, 112, 113, 211, 221 Symbol temporary determination unit
121-i, 122-i, 123-i, 221-i, 222-i Interference estimation unit
131-i, 132-i, 133-i, 231-i, 232-i Subtractor
151-i, 152-i, 153-i Inter-carrier interference estimation unit
161-i, 162-i, 163-i Propagation path characteristic reproduction unit
141, 142, 143 Signal synthesis unit
6 Symbol judgment section
7 Parallel-serial converter (P / S)
93-1 and 93-2 Fast Fourier Transformer (FFT)

Claims (4)

A multi-carrier modulation signal in which the effective symbol length is T, the adjacent frequency interval of N subcarriers is 1 / T, and NL (L <N) subcarriers are null carriers is received by a plurality of antennas. Then, in a multi-carrier diversity demodulation method of separating and demodulating each subcarrier,
Provided for each of a plurality of antennas, a step of estimating a multipath propagation environment on the time axis and the frequency axis,
The effective symbol length T is provided for each of the plurality of antennas and based on the delay time difference between the delayed waves estimated in the step of estimating the multipath propagation environment so as not to include a portion where the waveform distortion due to the delayed wave occurs. Determining a used symbol portion having a length TM / N (M <N) from the above, and extracting the used symbol portion M points from the complex digital signal orthogonally demodulated at a sampling interval T / N;
A step of separating L subcarriers using a matching filter using the used symbol portion M points, provided for each step of extracting the plurality of used symbol portion M points;
Diversity combining signals of a plurality of sets of L subcarriers which are outputs of the step of separating by the plurality of matched filters;
A small step of tentatively determining the signals of the L subcarriers that have been diversity-combined; and a plurality of L subcarriers provided for each of the plurality of matched filters and separated from the tentatively determined L symbols by the corresponding matched filter. An interference component between the other subcarriers included in each signal of the subcarriers is determined in consideration of the frequency characteristics of the multipath propagation environment, and the L subcarriers separated from the L subcarriers separated by the corresponding matched filter are used. A small step of reducing the interference component between the sub-carriers, and a small step of diversity combining signals of a plurality of sets of L sub-carriers which are outputs of the small step of reducing the plurality of interference components. A multi-carrier diversity demodulation method comprising a plurality of stages of interference component removal steps. The Marciaria modulation is OFDM,
The matched filter is formed using an N-point fast Fourier transform by inputting N complex digital signals of the used symbol portion M points and NM 0s,
In the small step of reducing the interference component, N complex digital signals of the tentatively determined L symbols and NL null symbols are calculated using frequency distortion calculated in a step of estimating a multipath propagation environment. The signal is subjected to N-point inverse fast Fourier transform, M points corresponding to the used symbol portion M points are selected from the result, and the other N−M are replaced with 0, and N-point fast Fourier transform is performed to correspond to subcarriers. 2. The multi-carrier diversity demodulation method according to claim 1, wherein L symbols to be selected are selected and obtained from the difference from the provisionally determined L symbols. A multi-carrier modulation signal in which the effective symbol length is T, the adjacent frequency interval of N subcarriers is 1 / T, and NL (L <N) subcarriers are null carriers is received by a plurality of antennas. Then, in a multi-carrier diversity demodulator that separates and demodulates each subcarrier,
A quadrature demodulation and sampling unit provided for each of the plurality of antennas to obtain N complex digital signals quadrature demodulated at a sampling interval T / N;
Provided for each of the plurality of antennas, a multipath propagation environment estimation unit that estimates the delay time difference of the delayed wave and the frequency distortion of the received wave,
Based on the delay time difference estimated by the multipath propagation environment estimation unit provided for each of the plurality of antennas, the N complex digital signals are used so as not to include a portion in which waveform distortion due to a delayed wave occurs. A symbol extracting unit for extracting M (M <N) complex digital signals as symbols;
A matched filter unit provided for each of the plurality of antennas and separating L subcarriers by a matched filter using the M complex digital signals output from the used symbol extracting unit;
A signal combining unit that diversity-combines a plurality of sets of L subcarrier signals output from the plurality of matched filters;
A provisional decision unit for provisionally determining L symbols from the signals of L subcarriers output from the signal combining unit, and a provisional decision unit corresponding to the provisionally determined L symbols provided for each of the plurality of matched filters. Interference component estimation for obtaining an interference component between each of the L subcarriers separated by the matched filter unit and other subcarriers in consideration of the frequency distortion estimated by the multipath propagation environment estimation unit An interference component subtractor provided for each of the plurality of matched filters, for subtracting the interference component between the L subcarriers from the L subcarrier signals output from the corresponding matched filter unit; A plurality of sets of L subcarriers output from the plurality of interference component subtracters, and a signal combining unit for diversity combining the signals, and one or more interference component removing units. Multicarrier demodulation apparatus according to. The Marciaria modulation is OFDM,
The matched filter section is formed by inputting the M complex digital signals output from the used symbol extracting section and the N complex digital signals of NM 0 and N, and using an N-point fast Fourier transform,
The interference component estimator,
Using the frequency distortion calculated by the multipath propagation environment estimation unit,
A null carrier inserter for adding NL null symbols to the temporarily determined L symbols;
An inverse fast Fourier transformer for N-point inverse fast Fourier transform of the N complex digital signals output from the null carrier inserter;
An effective symbol section extraction window unit that selects M points corresponding to the M complex digital signals output from the used symbol extraction unit from the output of the inverse fast Fourier transformer, and replaces the other N−M with 0;
A fast Fourier transformer for N-point fast Fourier transform of the N complex digital signals output from the effective symbol section extraction window device;
A carrier selector for selecting L subcarriers from the output of the fast Fourier transformer;
4. The multi-carrier diversity demodulator according to claim 3, further comprising a subtractor for calculating a difference between the L complex digital signals output from the carrier selector and the provisionally determined L symbols.

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