JP2003312517A - Electric power steering device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide an electric power steering device for preventing the degradation of a steering feeling caused by switching steering, and for suppressing heat generation in a booster circuit because the steering is performed by PWM drive by non-synchronous rectification in a straight travel mode which is generally most frequent in the travel modes. <P>SOLUTION: The booster circuit 100 of the electric power steering comprises a coil L, a second transistor Q2, a first transistor Q1, and a capacitor C2. The booster circuit 100 controls a current supplied from a battery to a boosting coil by turning on/off the first transistor Q1 by a PWM drive signal, and charges a boosted voltage to the capacitor C2. The CPU 21 determines the load condition of a motor, and performs the synchronous rectification of the first transistor Q1 and the second transistor Q2 when the motor is in a high load condition. When the motor is in a light load condition, only the first transistor Q1 is subjected to the non-synchronous rectification by PWM control when the light load condition continues for a predetermined time. <P>COPYRIGHT: (C)2004,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an electric power steering apparatus for applying an assisting force by a motor to a steering system of an automobile or a vehicle, and more specifically, it can adjust a current supplied from a vehicle battery to the motor. The present invention relates to an electric power steering device including a booster circuit.

[0002]

2. Description of the Related Art Conventionally, the rotational force of a motor is used to
An electric power steering device that assists the operation of a steering wheel is used.

In such an electric power steering apparatus, when the driver rotates the steering wheel to perform steering, an assist command value corresponding to the steering torque by the assist control is calculated, and based on the assist command value. The steering assist force is applied from the motor to the steering mechanism.

By the way, the electric power steering apparatus as described above is a system which requires a large current in order to obtain a large torque. Conventionally, an in-vehicle battery (DC12V) is directly applied, and a motor having a DC12V specification is also used. In order to supply a large current to the motor, the motor is upsized and the wiring used is upsized. (Thick line) cannot be avoided.

In order to solve this problem, an electric power steering device (Japanese Patent Laid-Open No. 8-127350) capable of adjusting the current supplied from the vehicle battery has been proposed.

In this electric power steering system, a booster circuit 300 and a booster circuit controller 301 as shown in FIG. 11 are provided in a circuit for supplying a current to a motor. The booster circuit 300 is provided between the application point P1 of the battery voltage VPIG (DC12V) from the vehicle-mounted battery and the voltage application point P2 of the motor. Boost circuit 30
0 is capacitors C1 and C2, coil L, diode D,
A first transistor Q1 for switching is provided.

The booster circuit controller 301 includes a booster circuit 30.
PW for boosting the first transistor Q1 of 0
The duty ratio as the control amount is calculated by the M calculation. Then, the booster circuit control device 301 outputs a duty ratio drive signal (PWM drive signal) based on this duty ratio, and duty-controls the first transistor Q1 by this duty ratio drive signal. By this duty control, the first transistor Q1 performs a switching operation as shown in FIG. 12, and as a result, the coil L
At this point, energy accumulation and discharge are repeated, and a high voltage at the time of discharge appears on the cathode side of the diode D. In the present specification, as shown in FIG. 12, Tα is the on-time, T is the pulse period, and α is the duty ratio (on-duty). When the first transistor Q1 is turned on, the coil L
When a current flows through the first transistor Q1 and the first transistor Q1 is turned off, the current flowing through the coil L is cut off.

When the current flowing through the coil L is cut off, a high voltage is generated on the cathode side of the diode D so as to prevent a change in magnetic flux due to the cutoff of the current. By repeating this, a high voltage is repeatedly generated on the cathode side of the diode D, smoothed (charged) by the capacitor C2, and generated at the voltage application point P2 as the output voltage VBPIG.

At this time, the voltage boosted by the booster circuit 300 is related to the duty ratio of the duty ratio drive signal output from the booster circuit controller 301. If the duty ratio is high, the output voltage VBPIG will be high, and if the duty ratio is low, the output voltage VBPIG will be low.

Since the diode D is used in the booster circuit 300 described above, when the motor enters the regenerative state, a current can flow from the voltage application point P2 side to the battery B due to the diode D. Output voltage VBP
IG rises. Due to this rise in voltage, the booster circuit 300
Could be damaged. For example, in the above example, the capacitor C2 forming the booster circuit 300 may be destroyed.

Therefore, the present applicant has proposed a booster circuit in which the second transistor Q2 is connected instead of the diode D as shown in FIG. That is, the second transistor Q2 has a source connected to the coil L and a drain connected to the voltage application point P2. In this configuration, the first transistor Q1 and the second transistor Q2 are alternately turned on / off, that is, controlled by the synchronous rectification method.

Therefore, when the motor is in the regenerative state, the second transistor Q2 is turned on by duty control, and the regenerative current is absorbed by flowing to the battery B via the second transistor Q2. As a result, the capacitor C2
Can be prevented from being destroyed.

In the proposed construction, the first transistor Q1 and the second transistor Q2 are driven by the synchronous rectification method in case of high load according to the load state of the motor during power running, and in case of low load, First transistor Q
An asynchronous rectification method is conceivable in which only 1 is PWM-driven and the second transistor Q2 is all off.

This is because when the motor has a low load, the second
This is preferable because the switching loss of the transistor Q2 is eliminated. For example, in a region where the motor rotation speed is high, synchronous rectification is performed as a high load, and in a region where the motor rotation speed is low, asynchronous rectification is performed as a low load.

[0015]

However, in such a case, when the steering wheel (steering wheel) is repeatedly steered to the left and right, the motor stops for a moment at the moment when the steering wheel is turned back. , When the steering wheel starts moving, it shifts to synchronous rectification. Therefore, the asynchronous rectification / synchronous rectification is repeated.

Since the boosted voltage is disturbed at the moment when the asynchronous rectification shifts to the synchronous rectification, the steering feeling may be deteriorated as a result. The object of the present invention is to focus on the above problems and prevent the deterioration of the steering feeling due to the steering back steering. Further, in the straight running, which is the most frequent in normal running, the PWM drive by the asynchronous rectification is used. Therefore, it is an object of the present invention to provide an electric power steering device that can suppress heat generation of a booster circuit.

[0017]

In order to solve the above-mentioned problems, the invention according to claim 1 relates to an electric motor drive means for driving an electric motor based on an electric motor control value, between a DC power supply and the electric motor drive means. And boosting means for boosting the power supply voltage, and boosting control means for generating and outputting a PWM drive signal, wherein the boosting means includes a boosting coil connected to the output terminal of the DC power supply, and the boosting coil. A first switching element and a second switching element both connected to the output terminal of the second switching element, and a boosting capacitor connected to the output terminal of the second switching element, and at least the first switching element of the both switching elements. The PWM drive of the switching element controls the current supplied from the DC power supply to the boost coil, and charges the boost capacitor with the boost voltage. In the electric power steering apparatus, the load state determination means for determining the load state of the electric motor is provided, and the boost control means synchronizes the two switching elements when the electric motor has a high load based on the determination result of the load state determination means. When rectified and the load is low, only the first switching element is switched to P when the low load state continues for a predetermined time.
An electric power steering device characterized by performing WM control and performing asynchronous rectification.

According to a second aspect of the present invention, in the first aspect, the step-up control means sets the carrier cycle of the PWM drive signal to a high cycle to synchronously rectify both of the switching elements when the load is high, thereby reducing the load. In this case, when the low load state continues for a predetermined time, only the first switching element of the two switching elements is asynchronously rectified by setting the carrier cycle of the PWM drive signal to a low cycle.

The invention of claim 3 is the same as claim 1 and claim 2.
In, a steering torque detecting means for detecting a steering torque is provided, and the load state determining means determines that the load state of the electric motor is low when the steering torque detected by the steering torque detecting means is small, It is characterized in that when the steering torque is large, it is determined that the load state of the electric motor is high.

The invention of claim 4 relates to claim 1 and claim 2.
In the above, the motor rotation speed estimation means for estimating the rotation speed of the electric motor is provided, and the load state determination means, when the rotation speed estimated by the motor rotation speed estimation means is small, the load state of the motor is low. It is determined that there is, and when the rotation speed is high, it is determined that the load state of the electric motor is high load.

[0021] The invention of claim 5 relates to claim 1 and claim 2.
In the above, the load state determination means determines the load state of the electric motor based on the electric motor control value or the detected value of the actual current flowing in the electric motor.

[0022]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS (First Embodiment) An embodiment of an electric power steering apparatus embodying the present invention will be described below with reference to FIGS.

FIG. 1 schematically shows a control device 20 of the electric power steering system. A torsion bar 3 is provided on a steering shaft 2 connected to a steering wheel 1 (handle). This torsion bar 3
A torque sensor 4 is attached to the. Then, when the steering shaft 2 rotates and a force is applied to the torsion bar 3, the torsion bar 3 is twisted according to the applied force, and the torque sensor 4 detects the twist, that is, the steering torque τ applied to the steering wheel 1. There is.

The torque sensor 4 constitutes steering torque detecting means. In addition, a reduction gear 5 is attached to the steering shaft 2.
Is stuck. A gear 7 attached to a rotating shaft of an electric motor (hereinafter referred to as a motor 6) as an electric motor is meshed with the speed reducer 5. The motor 6 is a brushless motor composed of a three-phase synchronous permanent magnet motor.

A rotation angle sensor 30 composed of a rotary encoder for detecting the rotation angle of the motor 6 is attached to the motor 6 (see FIG. 2). The rotation angle sensor 30 is π depending on the rotation of the rotor of the motor 6.
It outputs a two-phase pulse train signal having a phase difference of / 2 and a zero-phase pulse train signal indicating the reference rotation position.

Further, a pinion shaft 8 is fixed to the speed reducer 5. A pinion 9 is fixed to the tip of the pinion shaft 8, and the pinion 9 meshes with the rack 10. Tie rods 12 are fixedly provided at both ends of the rack 10, and a knuckle 13 is rotatably connected to a tip portion of the tie rod 12. A front wheel 14 as a tire is fixed to the knuckle 13. Further, one end of the knuckle 13 is rotatably connected to the cross member 15.

Therefore, when the motor 6 rotates, the number of rotations is reduced by the speed reducer 5 and transmitted to the pinion shaft 8 and then to the rack 10 via the pinion and rack mechanism 11. The rack 10 can change the traveling direction of the vehicle by changing the direction of the front wheels 14 provided on the knuckle 13 via the tie rods 12.

A vehicle speed sensor 16 is provided on the front wheel 14. Next, an electric configuration of a control device (hereinafter, referred to as a control device 20) of this electric power steering device will be shown.

The torque sensor 4 outputs a voltage according to the steering torque τ of the steering wheel 1. The vehicle speed sensor 16 outputs the vehicle speed at that time as a pulse signal having a cycle corresponding to the rotation speed of the front wheels 14.

The control unit 20 includes a central processing unit (CPU2).
1), read only memory (ROM 22) and read and write only memory (RAM) for temporarily storing data
23). This ROM 22 has a CPU 21
A control program for performing the arithmetic processing by is stored. The RAM 23 temporarily stores the calculation processing result and the like when the CPU 21 performs the calculation processing.

The ROM 22 stores a basic assist map (not shown). The basic assist map is for obtaining the basic assist current corresponding to the steering torque τ (turning torque) and corresponding to the vehicle speed.
The basic assist current for is stored.

The function of the control device 20 to drive and control the three-phase synchronous permanent magnet motor is a known structure and will be briefly described. FIG. 3 shows the inside of the CPU 21.
It is a control block diagram which shows the function performed by a program. Each unit illustrated in the control block diagram does not represent independent hardware but represents functions executed by the CPU 21.

The CPU 21 is internally provided with a basic assist force calculator 51, a returning force calculator 52 and an adder 53 for calculating the command torque τ *. The basic assist force calculation unit 51 inputs the steering torque τ from the torque sensor 4 and the vehicle speed V detected by the vehicle speed sensor 16, and the assist torque increases as the steering torque τ increases and decreases as the vehicle speed V increases. To calculate.

The return force calculation unit 52 inputs the vehicle speed V as well as the electrical angle θ (corresponding to the rotation angle) of the rotor of the motor 6 and the angular velocity ω, and based on these input values, the steering shaft 2 is moved to the basic position. Return force and steering shaft 2
The return torque corresponding to the resistance to rotation of is calculated. The addition unit 53 calculates the command torque τ * by adding the assist torque and the return torque, and outputs the command torque τ * to the command current setting unit 54.

The command current setting unit 54, based on the command torque τ *, has a two-phase d-axis command current Id * and a q-axis command current Iq *.
To calculate. Both command currents respectively correspond to the d axis in the same direction as the direction of the magnetic flux of the permanent magnet and the q axis orthogonal thereto in the rotating coordinate system synchronized with the rotating magnetic flux generated by the permanent magnet on the rotor of the motor 6.

The d-axis command current Id * and the q-axis command current Iq * are supplied to the subtracters 55 and 56. The subtracters 55 and 56 are d
Axis command current Id *, q axis command current Iq *, and d axis detection current I
Difference values ΔId and Δ from the d- and q-axis detection currents Iq
Iq is calculated and the result is supplied to PI control units (proportional and integral control units) 57 and 58.

The PI control units 57 and 58 are provided with the difference value ΔId,
Based on ΔIq, the d-axis command voltage Vd * and the q-axis command voltage Vq * are calculated so that the d-axis detection current Id and the q-axis detection current Iq follow the d-axis command current Id * and the q-axis command current Iq *, respectively. .

D-axis command voltage Vd * and q-axis command voltage Vq *
Is a non-interference control correction value calculation unit 63 and subtractors 59 and 60.
Thus, the d-axis correction command voltage Vd ** and the q-axis correction command voltage Vq ** are corrected and supplied to the two-phase / 3-phase coordinate conversion unit 61.

The non-interference control correction value calculation unit 63 calculates the d-axis command voltage Vd * and the q-axis command voltage Vq * based on the d-axis detection current Id and the q-axis detection current Iq and the angular velocity ω of the rotor of the motor 6. Non-interference control correction value for ω ・ La ・ Iq, −ω ・
Calculate (φa + La · Id). Note that the inductance L
a and the magnetic flux φa are predetermined constants.

Subtractors 59 and 60 subtract the non-interference control correction values from the d-axis command voltage Vd * and the q-axis command voltage Vq *, respectively, to obtain the d-axis correction command voltages Vd ** and q.
The axis correction command voltage Vq ** is calculated and output to the 2-phase / 3-phase coordinate conversion unit 61. The two-phase / three-phase coordinate conversion unit 61 converts the d-axis correction command voltage Vd ** and the q-axis correction command voltage Vq ** into three-phase command voltages Vu *, Vv *, Vw *, and the same conversion is performed. The phase command voltages Vu *, Vv *, Vw * are output to the PWM control unit 62.

The PWM control section 62 uses the three-phase command voltage V
PWM control signals UU, VU, WU corresponding to u *, Vv *, Vw *
(Including a PWM wave signal and a signal indicating the rotation direction of the motor 6), and the motor drive device 3 that is an inverter circuit
Output to 5.

The q-axis command current Iq * corresponds to the motor control value. The motor drive device 35 corresponds to an electric motor drive means. As shown in FIG. 2, the motor driving device 35 is an FET.
(Field-Effect Transistor) 81U, 82U series circuit, FET 81V, 82V series circuit, FET 81
It is configured by connecting a series circuit of W and 82W in parallel. A voltage boosted higher than the voltage of the battery mounted on the vehicle is applied to each series circuit. And FE
The connection point 83U between T81U and 82U is connected to the U-phase winding of the motor 6, and the connection point 83V between the FETs 81V and 82V.
Is connected to the V phase winding of the motor 6, and FETs 81W and 82
The connection point 83W between W is connected to the W-phase winding of the motor 6.

FET 81U, 82U, FET 81V, 8
The PWM control signals UU, VU, WU (PWM of each phase) are supplied to the 2V and the FETs 81W, 82W from the PWM control unit 62, respectively.
A PWM wave signal and a signal indicating the rotation direction of the motor 6 are input to the control signal.

The motor drive unit 35 uses the PWM control signal U
Three-phase exciting currents corresponding to U, VU, and WU are generated and supplied to the motor 6 via the three-phase exciting current paths. Three
Current sensors 71, 72 are provided in two of the phase excitation current paths.
The respective current sensors 71, 72 detect two exciting currents Iu, Iv of the three-phase exciting currents Iu, Iv, Iw for the motor 6 to convert the three-phase / two-phase coordinates shown in FIG. It is output to the unit 73.

The three-phase / two-phase coordinate converter 73 receives the exciting current Iw calculated by the calculator 74 based on the exciting currents Iu and Iv. 3-phase / 2-phase coordinate conversion unit 73
Converts these exciting currents Iu, Iv, Iw into a two-phase d-axis detection current Id and a q-axis detection current Iq, and subtracters 55, 5
6, input to the non-interference control correction value calculation unit 63.

The two-phase pulse train signal and the zero-phase pulse train signal from the rotation angle sensor 30 are continuously supplied to the electrical angle converter 64 at a predetermined sampling cycle. The electrical angle conversion unit 64 uses the motor 6 based on the pulse train signals.
The electrical angle θ of the rotor with respect to the stator is calculated, and the calculated electrical angle θ is input to the angular velocity conversion unit 65. The angular velocity converter 65 differentiates the electrical angle θ to calculate the angular velocity ω of the rotor with respect to the stator. The positive angular velocity ω represents rotation of the rotor in the positive direction, and the negative angular velocity ω represents rotation of the rotor in the negative direction.

Next, a booster circuit 1 for boosting the battery voltage
00 and the booster circuit control device for controlling the booster circuit 100 will be described. In this embodiment, the booster circuit control device is also used by the CPU 21. The booster circuit 100 corresponds to booster means.

The booster circuit 100 is provided in a current supply circuit between an on-vehicle battery (hereinafter referred to as battery B) as a DC power source and the motor drive device 35. In the booster circuit 100 of the present embodiment, the application point P1 and the voltage application point P
A boost coil (hereinafter simply referred to as coil L) between the two
And the second transistor Q2 is connected. The second
The source of the transistor Q2 is connected to the output terminal of the coil L, and the drain is connected to the voltage application point P2.
The gate of the second transistor Q2 is C of the control device 20.
It is connected to the PU 21. D2 is the second transistor Q
2 parasitic diode.

The application point P1 is a rectifying capacitor C1.
Grounded through. The application point P1 corresponds to the output terminal of the DC power supply. The voltage application point P2 is grounded via a boosting capacitor C2.

The capacitor C2 is the second transistor Q.
It is connected to the drain that serves as the output terminal of No.2. The capacitor C2 corresponds to a boosting capacitor that charges the boosted voltage generated by the boosting coil.

The drain of the first transistor Q1 is connected to the connection point between the output terminal of the coil L and the second transistor Q2, and the source is grounded. The gate of the first transistor Q1 is connected to the CPU 21 of the booster circuit control device 101. D1 is a parasitic diode of the first transistor Q1. To detect the voltage at the voltage application point P2,
The voltage application point P2 is connected to a voltage input port (not shown) of the CPU 21 of the control device 20 so that the output voltage VBPIG can be detected as a measured value.

The first transistor Q1 and the second transistor Q2 are n-channel MOSFETs.
The first transistor Q1 constitutes a first switching element, and the second transistor Q2 corresponds to a second switching element.

Next, CP for controlling both the transistors
U21 will be described. FIG. 5 shows a functional block diagram of the CPU 21. That is, it is a control block diagram showing functions executed by a program inside the CPU 21.

The respective parts shown in the control block diagram do not represent independent hardware but a CPU.
21 shows the functions performed. The CPU 21 constitutes a boost control means and a load state determination means.

The CPU 21 includes a calculator 110, a PID controller 120, a PWM calculator 130, and an A / D converter 150. The arithmetic unit 110 uses the target output voltage VBPIG * (in the present embodiment, 20 V) stored in advance in the ROM 22.
V) and VBPIG input via the A / D converter 150 are calculated, and the deviation is supplied to the PID controller 120.

The PID control unit 120 performs proportional (P) / integral (I) / derivative (D) processing to reduce the deviation, that is, in order to perform feedback control, and the first
This is a circuit for calculating the control amount of the transistor Q1 and the second transistor Q2. The control amount calculated by the PID control unit 120 is further converted into a duty ratio drive signal (PWM drive signal) by calculating a duty ratio α corresponding to the control amount by the PWM calculation unit 130, and the converted duty ratio. The drive signal is applied to each transistor of the booster circuit 100.

In the present embodiment, the calculated duty ratio drive signal is applied to the first transistor Q1 and the second transistor Q2 to perform on / off control alternately (see FIG. 6A). ), Or an asynchronous rectification method in which only the first transistor Q1 is applied and PWM driving is performed (see FIG. 6B).

The synchronous rectification method is performed when the motor 6 is under high load during power running and during regeneration, and the asynchronous rectification method is performed when the motor 6 is under low load during power running. Note that the low load is meant to include the case of no load to which no load is applied in the present specification.

FIG. 6A shows a pulse signal (duty ratio drive signal) applied to the first transistor Q1, where Tα is the on-time, T is the pulse period, and α is the duty ratio of the first transistor Q1 ( On-duty). The duty ratio of the second transistor Q2 is (1- | α |).

When the duty ratio α is "+", it is in the power running state, and when it is "-", it is in the regenerative state. In the first embodiment, the duty ratio α in the power running state is 0 ≦ α ≦ α0 <
1 is set. α0 is a limit value, and the PWM calculation unit 13
If the result of calculating the duty ratio α at 0 exceeds α0, α0 is determined as the duty ratio α.

The duty ratio α in the regenerative state is 0 ≦ | α
| ≦ 1. In addition, starting from the first embodiment,
In another embodiment, when the second transistor Q2 and the first transistor Q1 are alternately turned on and off, the duty ratio of the second transistor Q2 can be calculated by (1− | α |). Omit it.

Further, to the second transistor Q2, a pulse signal (duty ratio drive signal) is applied which turns off when the first transistor Q1 is on and turns on when the first transistor Q1 is off. .

(Operation of First Embodiment) FIG. 7 shows C
7 is a flowchart of a control program executed by the PU 21 at the time of power running, and is executed at a predetermined control cycle when the duty ratio α is “+”.

At S10, the steering torque τ and the threshold value τ0 are read. The threshold value τ0 is stored in the ROM 22 in advance. In S20, the magnitude relationship between the steering torque τ and the threshold value τ0 is determined. That is, it is determined whether the motor 6 is in a low load state or a high load state. Steering torque τ
However, when the steering torque τ exceeds the threshold value τ0, it is determined that the motor 6 has a high load.
Move to 30.

In step S30, the CPU 21 uses the synchronous rectification method.
PW the first transistor Q1 and the second transistor Q2
M drive. That is, according to the duty ratio drive signal of the drive pattern shown in FIG.
The first and second transistors Q2 are alternately turned on and off.

More specifically, at the time of high load during power running, in the booster circuit 100, the first transistor Q1 performs the switching operation by the duty control by the duty ratio drive signal. As a result, energy accumulation and discharge are repeated in the coil L, and the second transistor Q2
A high voltage appears on the drain side of the. That is,
The first transistor Q1 is turned on and the second transistor Q1
When 2 is turned off, a current flows to the ground side through the first transistor Q1. Next, when the first transistor Q1 is turned off, the current flowing through the coil L is cut off. Coil L1
When the current flowing through is cut off, a high voltage is generated on the drain side of the second transistor Q2 which is on so as to prevent the change of the magnetic flux due to the cutoff of the current. By repeating this, a high voltage is repeatedly generated on the drain side of the second transistor Q2, and the capacitor C2 smoothes (charges)
Is generated at the voltage application point P2 as the output voltage VBPIG.

At this time, the voltage boosted by the booster circuit 100 is related to the duty ratio α of the duty ratio drive signal output from the CPU 21. When the duty ratio α is large, the output voltage VBPIG is high, and when the duty ratio α is small, the output voltage VBPIG is low.

After the processing of S30, this flow chart is once ended. In S40, the counter C is incremented by 1, and it is determined in S50 whether or not the value of the counter C has passed the predetermined time C0. The value of the counter C is C0 for a predetermined time.
If has not passed, synchronous rectification is performed in S30.

If the value of the counter C has passed the predetermined time C0 in S50, the process proceeds to S60. After shifting to S60, the CPU 21 PWM-drives only the first transistor Q1 by asynchronous rectification, resets the value of the counter C to 0 in S70, and then ends this flowchart.

The asynchronous rectification will be described. The asynchronous rectification corresponds to the heat generation suppression mode of the booster circuit 100. In asynchronous rectification, the second transistor Q2 is always off,
Only the first transistor Q1 is PWM driven by the duty ratio drive signal. In the case of this asynchronous rectification, when the capacitor C2 is charged and the voltage at the voltage application point P2 reaches the target output voltage VBPIG *, the duty ratio (on-duty) of the first transistor Q1 is actually close to 0. Will be things.

The reason for this is that since the motor 6 has a low load, the discharge current supplied from the capacitor C2 to the motor 6 is small, and particularly when there is no load, the discharge current does not flow.

In such a state, as shown in FIG. 5, since feedback control is performed, once the voltage at the voltage application point P2 reaches the target output voltage VBPIG *, boosting is performed. This is because the duty ratio (on-duty) for is close to zero. The duty ratio (on-duty) approaching 0 means that the capacitor C
This is because a leakage current is generated in 2 and charges are actually discharged little by little, and feedback control corresponding to the leakage current is actually performed so that the duty ratio does not become 0 completely.

If the first transistor Q1 and the second transistor Q2 are driven on and off only by the synchronous rectification method during the power running of the motor 6, when the motor 6 has a low load, the motor 6 is not driven and the capacitor C2 is removed. No discharge current is consumed. That is, in this state, the capacitor C
The charge charged in 2 will be returned to the battery B via the coil L when the second transistor Q2 is turned on. At this time, switching loss due to on / off of the second transistor Q2 and heat generation of the coil L occur.
As described above, due to the occurrence of the switching loss of the second transistor Q2 and the heat generation of the coil L, the booster circuit 100 heats up, resulting in a great decrease in efficiency.

However, as in the present embodiment, when the motor 6 is under a low load and the second transistor Q2 is completely turned off by the asynchronous rectification and the first transistor Q1 is PWM driven, the second transistor Q2 is turned on / off. Switching loss is eliminated, and electric charges do not flow from the capacitor C2 to the coil L. Therefore, the switching loss of the second transistor Q2 and the heat generation of the coil L are eliminated, and the temperature rise of the booster circuit 100 can be suppressed.

When only the first transistor Q1 is turned on and off by PWM driving by only the asynchronous rectification during power running of the motor 6, a current is applied to the voltage application point P2 side via the parasitic diode D2 of the second transistor Q2. It will be in the form of supply. In this case, when the motor 6 has a high load, the current value to the voltage application point P2 side increases, and the loss (heat generation) in the second transistor Q2 (parasitic diode D2) increases, which is not preferable.

When the motor 6 enters the regenerative state, the CPU 21 uses the synchronous rectification method to make the first transistor Q
The first and second transistors Q2 are PWM-driven. At this time, the output voltage VBPIG increases due to the regenerative current from the motor 6, but the second transistor Q2 is turned on by the duty control. Therefore, the second transistor Q2
A regenerative current flows to the battery B via the and is absorbed.

The first embodiment has the following features. (1) The electric power steering apparatus according to the first embodiment includes a motor drive device 35 (electric motor drive means) that drives the motor 6 (electric motor) based on the q-axis command current Iq * (electric motor control value).

The electric power steering device is provided between the battery B (DC power supply) and the motor drive device 35,
Booster circuit 100 for boosting battery voltage (power supply voltage)
(Boosting means) and CPU 21 (Boosting control means) for generating and outputting a duty ratio drive signal (PWM drive signal).

The booster circuit 100 also includes a coil L (a boosting coil), a second transistor Q2 (a second switching element), a first transistor Q1 (a first switching element), and a capacitor C2 (a boosting capacitor). There is. Then, the booster circuit 100 controls the current supplied from the battery B to the boosting coil by turning on and off the first transistor Q1 by the PWM drive signal, and charges the capacitor C2 with the boosted voltage.

Further, the CPU 21 (load state determination means, boost control means) determines the load state of the motor 6, and the motor 6
However, when the load is high, the first transistor Q1 and the second transistor Q2 are synchronously rectified. Further, when the motor 6 has a low load, when the low load state continues for a predetermined time C0,
Only the first transistor Q1 is PWM-controlled to perform asynchronous rectification.

As a result, when the steering wheel is repeatedly steered left and right, the motor 6 is momentarily stopped even when the steering wheel 1 is turned back, and the CPU 21 determines that the load is low. However, if the low load condition is C
If 0 is not continued, the asynchronous rectification is not performed, that is, the synchronous rectification is performed within the predetermined time C0, and thus the asynchronous rectification / synchronous rectification is not repeated. Therefore, the steering feeling does not deteriorate.

Further, when the motor 6 has a low load, for example, in straight running, which is the most frequent running, PWM by asynchronous rectification is used.
Since driving is performed, heat generation of the booster circuit 100 can be suppressed.

As a result, when the motor 6 has a low load, the heat generation of the booster circuit 100 can be suppressed more than when the motor 6 has a high load. (2) In the first embodiment, the torque sensor 4 (steering torque detecting means) for detecting the steering torque τ is provided. When the steering torque τ detected by the torque sensor 4 is less than or equal to the threshold value τ0, the CPU 21 (load state determination means) determines that the load state of the motor 6 is low, and the steering torque τ exceeds the threshold value τ0. In this case, the load condition of the motor 6 is determined to be high.

As a result, the steering torque τ and the threshold value τ0 make it easy to determine whether the motor 6 has a low load or a high load. (Second Embodiment) Next, a second embodiment will be described with reference to FIG.

In the first embodiment, the steering torque τ is used as the parameter for determining the load state of the motor 6, but in the second embodiment, the motor rotation speed n of the motor 6 is used as the parameter for determining the load state of the motor 6. Is different.

That is, in this embodiment, the rotation angle sensor 30 also serves as a rotation position sensor for detecting the rotation position of the motor 6, and the CPU 21 functions as a boost control means, a load state determination means, and an electric motor rotation speed estimation means. Equivalent to.

FIG. 8 is a flow chart of a control program executed by the CPU 21 at the time of power running, and is executed at a predetermined control cycle when the duty ratio α is “+”. In the second embodiment, S10 and S20 of the first embodiment.
Instead of, S10A and S20A are executed respectively.

In S10A, the CPU 21 calculates the motor rotation speed n using a known arithmetic expression based on the detection signal from the rotation angle sensor 30. In S20A, it is determined whether the motor rotation speed n is larger than the rotation speed threshold value n0 stored in the ROM 22 in advance, that is, whether the motor 6 is in a low load state or a high load state. When the motor rotation speed n is equal to or lower than the rotation speed threshold value n0, it is determined that the motor 6 has a low load, and the process proceeds to S40 where the motor rotation speed n is the rotation speed threshold value n0.
If it is larger than the above, it is determined that the motor 6 has a high load, and the process proceeds to S30.

The second embodiment has the following features in addition to (1) of the first embodiment. (1) In the second embodiment, the CPU 21 serves as an electric motor rotation speed estimation unit that estimates the rotation speed of the motor 6. When the estimated rotation speed of the motor 6 (motor rotation speed n) is equal to or lower than the rotation speed threshold value n0, the CPU 21 (load state determination means) determines that the load state of the motor 6 is low, and the motor rotation speed is low. When the number n exceeds the rotation speed threshold value n0,
The load state of the motor 6 is determined to be high.

As a result, the motor speed n and the speed threshold n
With 0, it is possible to easily determine whether the motor 6 has a low load or a high load. You may change the structure of 2nd Embodiment as follows.

(A) In the second embodiment, a brushless motor is used, but a brushed motor is used instead of the brushless motor (hereinafter, simply referred to as a motor in this section).
You may change to. Even in this case, the flowchart of FIG. 8 is executed.

In S10A, as follows,
The motor speed n is calculated (estimated). In order to detect the motor current of the motor, the motor is provided with a motor current detection circuit (not shown) and a motor terminal voltage detection circuit (not shown) for detecting the voltage between the motor terminals.

In order to calculate the motor rotation speed n of the motor, the CPU 21 first calculates the average value of the motor current (motor current average value Ia) detected by the motor current detection circuit (not shown) and the motor terminal voltage. The average value of the inter-terminal voltage detected by the detection circuit (inter-terminal voltage average value Va) is obtained. From the obtained motor current average value Ia and inter-terminal voltage average value Va, the instantaneous value of the motor internal resistance (motor internal resistance instantaneous value R) is calculated according to the equation (1).

R = Va / Ia (1) Next, the motor internal resistance instantaneous value R is integrated over time to obtain the motor internal resistance value Ri, and the internal resistance value Ri, the motor current average value Ia and the inter-terminal voltage are obtained. The back electromotive force Vc of the motor is obtained by using the equation (2) based on the average value Va.

Vc = Va-IaRi (2) Then, using the equation (3), the counter electromotive voltage Vc is multiplied by the motor power generation constant K which is the ratio of the rotation speed to the counter electromotive voltage Vc. , The motor rotation speed n is calculated.

The motor rotation speed n has a sign corresponding to the sign of the back electromotive voltage Vc of the motor. The motor rotation speed n has a positive value for the right rotation of the motor and a negative value for the left rotation of the motor. That is, the motor rotation speed n is the rotation speed including the rotation direction component of the motor.

N = K · Vc (3) Therefore, in this modification, in S20A, | n |> n0
Then, the size relation is judged.

The other structure is the same as that of the second embodiment.
The CPU 21 corresponds to a boost control unit, a load state determination unit, and a motor rotation speed estimation unit.

(B) Further, a steering wheel rotation speed sensor for detecting the rotation speed of the steering wheel 1 (handle) is provided, and the CPU 21 calculates the motor rotation speed n based on the steering wheel rotation speed detected by this steering wheel rotation speed sensor. (Estimation) may be performed. This is also acceptable because the handle rotation speed and the motor rotation speed n are in a proportional relationship.

Also in this case, the CPU 21 corresponds to the boost control means, the load state determination means, and the electric motor rotation speed estimation means. (Third Embodiment) Next, a third embodiment will be described with reference to FIG.

In the first embodiment, the steering torque τ is used as the parameter for determining the load state of the motor 6, but in the third embodiment, the assist command current, that is, the q-axis, is used as the parameter for determining the load state of the motor 6. Command current I
The difference is that q * (motor control value) is used.

That is, also in this embodiment, the CPU
Reference numeral 21 corresponds to boost control means and load state determination means.
FIG. 9 is a flowchart of a control program executed by the CPU 21 at the time of power running, and is executed at a predetermined control cycle when the duty ratio α is “+”.

In the third embodiment, S1 of the first embodiment is used.
S10B and S20B are executed instead of 0 and S20, respectively. In S10B, the CPU 21 reads the q-axis command current Iq * (motor control value).

In S20B, the magnitude relationship between the q-axis command current Iq * and the command value threshold Iq * s stored in the ROM 22 in advance,
That is, it is determined whether the motor 6 is in a low load state or a high load state.

When the q-axis command current Iq * is less than or equal to the command value threshold Iq * s, it is determined that the motor 6 has a low load, and the process proceeds to S40. If it is larger than Iq * s, it is determined that the motor 6 has a high load, and the process proceeds to S30.

In addition to (1) of the first embodiment, the third embodiment has the following features. (1) In the third embodiment, the CPU 21 (load state determination means) determines that the load state of the motor 6 is low load or high load based on the q-axis command current Iq * (motor control value). I chose

That is, the q-axis command current Iq * and RO
It is determined whether the motor 6 is in a low load state or a high load state based on the magnitude relationship with the command value threshold value Iq * s stored in M22.

As a result, the q-axis command current Iq * and RO
With the command value threshold Iq * s stored in M22, the motor 6
Can easily determine whether the load is low or high. The configuration of the third embodiment may be modified as follows. In the third embodiment, the load state of the motor 6 is determined to be low load or high load based on the q-axis command current Iq * (motor control value). Instead of this, the load of the motor 6 may be determined based on the magnitude relationship between the q-axis detection current Iq and the threshold value stored in the ROM 22 in advance.

That is, if the q-axis detection current Iq exceeds the threshold value, the CPU 21 determines that the motor 6 is in a high load state, and if the q-axis detection current Iq is less than or equal to the threshold value,
It is determined that the motor 6 is in the low load state.

The current sensors 71 and 72 correspond to actual current detecting means for detecting the actual current flowing through the motor 6. The q-axis detection current Iq corresponds to the detection value of the actual current flowing through the motor 6.

In this case, the CPU 21 corresponds to the boost control means and the load state determination means. (Fourth Embodiment) Next, a fourth embodiment will be described with reference to FIG.

In the fourth embodiment, in the constitution of the first embodiment,
In the flowchart, instead of S30 and S60,
The difference is that S30A and S60A are executed, and the other configurations are the same.

In S30A, when the steering torque τ is larger than the threshold value τ0, it is determined that the motor 6 is under a heavy load, and therefore the first transistor Q1 and the second transistor Q2.
Is synchronously rectified with a high carrier cycle of the PWM drive signal.

Further, in S60A, the steering torque τ is the threshold τ.
When it is 0 or less, it is determined that the motor 6 has a low load, so that the carrier cycle of the PWM drive signal is set to a low cycle and only the first transistor Q1 is asynchronously rectified.

As a result, in the fourth embodiment, when the motor 6 has a low load, the carrier cycle of the PWM drive signal is set to a low cycle, so that the switching loss is reduced and the heat generation of the transistor due to the switching loss is reduced.
The heat generation of the booster circuit 100 can be suppressed.

On the other hand, in S30A, when the motor 6 has a high load, the first transistor Q1 and the second transistor Q2 are synchronously rectified at a high cycle, so that the ripple voltage when each transistor is turned on and off can be reduced.

Therefore, when both transistors are synchronously rectified at a high cycle, the motor 6 is in a high load state, that is, the steering torque τ is output. Therefore, the more the ripple voltage is suppressed, the worse the steering feeling. Can be prevented.

If the motor 6 has a low load,
When there is no load, the motor 6 does not consume power, so PW
The carrier cycle of the M drive signal is lengthened (low cycle) to synchronously rectify both transistors. That is, in this case
Steering torque τ is not output (state not being steered)
Therefore, the steering feeling is not affected.

The CPU 21 corresponds to boost control means and load state determination means. Therefore, in the fourth embodiment,
In addition to the effect (1) of the first embodiment, there are the following features.

(1) In the fourth embodiment, the CPU 21
When the motor 6 has a high load, the (step-up control means) controls the PW.
The carrier cycle of the M drive signal is set to a high cycle so that both the first transistor Q1 and the second transistor Q2 (both switching elements) are synchronously rectified. Also, C
When the motor 6 has a low load, the PU 21 sets the carrier cycle of the PWM drive signal to a low cycle and synchronously rectifies both the first transistor Q1 and the second transistor Q2 when the low load state continues for a predetermined time. I did it.

As a result, when the asynchronous rectification is performed at a low cycle, the heat generation of the transistor due to the switching loss is reduced, and the heat generation of the booster circuit 100 can be suppressed. Also, the motor 6
In the high load state, since both transistors are synchronously rectified at a high carrier cycle of the PWM drive signal, the ripple voltage is suppressed, and the deterioration of steering feeling can be prevented.

The fourth embodiment may be modified as follows. (A) In the flowchart of the fourth embodiment, S1
0 and S20 should be changed to S10A and S20A of the second embodiment, respectively. As a result, the effect (1) of the second embodiment is achieved.

(B) In the flowchart of the fourth embodiment, S10 and S20 are replaced with S10B and S2 of the third embodiment.
Change to 0B respectively. As a result, the effect (1) of the third embodiment is obtained.

The embodiment of the present invention may be modified as follows. In each of the above embodiments, instead of the embodiment using the steering torque τ and the vehicle speed V, the electric motor control value may be determined only by the steering torque τ.

[0125]

As described in detail above, the inventions of claims 1 to 5 can prevent the deterioration of the steering feeling due to the turning steering of the steering wheel, and are the most straight line running in normal running. Then, since the PWM driving is performed by the asynchronous rectification, it is possible to suppress the heat generation of the booster circuit.

[Brief description of drawings]

FIG. 1 is a schematic view of an electric power steering device embodied in a first embodiment of the present invention.

FIG. 2 is a control block diagram of the electric power steering device.

FIG. 3 is a control block diagram of CPU 21.

FIG. 4 is an electric circuit diagram of the booster circuit.

FIG. 5 is a control block diagram of the control device for boosting the same.

6A is a waveform diagram of PWM drive signals of both transistors in the case of a synchronous rectification method, and FIG. 6B is a waveform diagram of PWM drive signals of both transistors in the case of an asynchronous rectification method.

FIG. 7 is a flowchart of a control program executed by the CPU 21 of the first embodiment.

FIG. 8 is a flowchart of a control program executed by the CPU 21 of the second embodiment.

FIG. 9 is a flowchart of a control program executed by the CPU 21 of the third embodiment.

FIG. 10 is a flowchart of a control program executed by the CPU 21 of the fourth embodiment.

FIG. 11 is an electric circuit diagram of a booster circuit of a conventional electric power steering device.

FIG. 12 is a waveform diagram of a PWM drive signal for the transistor.

[Explanation of symbols]

4 ... Torque sensor (steering torque detection means) 6 ... Motor (electric motor) 20 ... Control device 21 ... CPU (boost control means, load state determination means, and electric motor rotation speed estimation means) 35 ... Motor drive device (electric motor drive means) 100 ... Booster circuit (Boosting means) B ... Battery (DC power supply) L ... Coil (Boosting coil) C2 ... Capacitor (Boosting capacitor) Q1 ... First transistor (first switching element) Q2 ... Second transistor (second) Switching element)

   ─────────────────────────────────────────────────── ─── Continued front page    F term (reference) 3D032 DA15 DA23 DA65 DC01 DC31                       DD10 EC24 GG01                 3D033 CA03 CA13 CA16 CA20                 5H576 AA15 BB05 CC02 DD02 DD07                       EE01 EE10 EE11 GG02 GG04                       HA03 HB02 JJ03 JJ22 JJ23                       JJ24 KK06 LL07 LL22 LL38                       LL41

Claims (5)

[Claims] 1. A motor driving means for driving a motor based on a motor control value, a boosting means provided between a DC power source and the motor driving means for boosting a power supply voltage, and a boosting control for generating and outputting a PWM drive signal. And a first switching element and a second switching element both connected to an output terminal of the boosting coil, the boosting coil being connected to an output terminal of the DC power supply, A boosting capacitor connected to the output terminal of the second switching element, wherein at least the first switching element of the two switching elements is PWM-driven, whereby a current supplied from the DC power supply to the boosting coil. In the electric power steering device that controls the voltage boosting voltage to charge the boosting capacitor, and determines the load state of the electric motor. A load state determination means is provided, and the boost control means synchronously rectifies both of the switching elements when the motor has a high load based on the determination result of the load state determination means, and when the load is low, a low load state is predetermined. An electric power steering device characterized by performing PWM control of only the first switching element to perform asynchronous rectification when the time is continued. 2. The boost control means, when the load is high,
The carrier cycle of the PWM drive signal is set to a high cycle to synchronously rectify both of the switching elements, and when the load is low, only the first switching element among the switching elements when the low load state continues for a predetermined time, 2. The electric power steering apparatus according to claim 1, wherein the PWM drive signal has a low carrier cycle to perform asynchronous rectification. 3. A steering torque detecting means for detecting a steering torque is provided, wherein the load state determining means determines that the load state of the electric motor is low when the steering torque detected by the steering torque detecting means is small. The electric power steering apparatus according to claim 1 or 2, wherein when the steering torque is large, it is determined that the load state of the electric motor is high. 4. An electric motor rotation speed estimating means for estimating the rotation speed of the electric motor is provided, wherein the load state determining means determines that the load status of the electric motor is low when the rotation speed estimated by the electric motor rotation speed estimating means is small. 3. The electric power steering apparatus according to claim 1, wherein it is determined that the load is low, and when the rotational speed is high, the load state of the electric motor is determined to be high. 5. The load state determination means determines the load state of the electric motor based on the electric motor control value or a detected value of an actual current flowing through the electric motor.
And the electric power steering apparatus according to claim 2.