JP2002272118A - High-frequency heating power supply - Google Patents

High-frequency heating power supply

Info

Publication number
JP2002272118A
JP2002272118A JP2001069964A JP2001069964A JP2002272118A JP 2002272118 A JP2002272118 A JP 2002272118A JP 2001069964 A JP2001069964 A JP 2001069964A JP 2001069964 A JP2001069964 A JP 2001069964A JP 2002272118 A JP2002272118 A JP 2002272118A
Authority
JP
Japan
Prior art keywords
power supply
voltage
semiconductor switch
commercial power
magnetron
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Granted
Application number
JP2001069964A
Other languages
Japanese (ja)
Other versions
JP3501133B2 (en
Inventor
Hideaki Moriya
英明 守屋
Kenji Yasui
健治 安井
Yoshiaki Ishio
嘉朗 石尾
Takeshi Kitaizumi
武 北泉
Haruo Suenaga
治雄 末永
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Panasonic Holdings Corp
Original Assignee
Matsushita Electric Industrial Co Ltd
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority to JP2001069964A priority Critical patent/JP3501133B2/en
Application filed by Matsushita Electric Industrial Co Ltd filed Critical Matsushita Electric Industrial Co Ltd
Priority to US10/130,222 priority patent/US6624579B2/en
Priority to PCT/JP2001/008392 priority patent/WO2002028149A2/en
Priority to CNB018029264A priority patent/CN1171505C/en
Priority to EP01972509A priority patent/EP1254590B8/en
Priority to DE60104981T priority patent/DE60104981T2/en
Priority to KR1020027006673A priority patent/KR100766534B1/en
Priority to CN01269659U priority patent/CN2562101Y/en
Publication of JP2002272118A publication Critical patent/JP2002272118A/en
Application granted granted Critical
Publication of JP3501133B2 publication Critical patent/JP3501133B2/en
Anticipated expiration legal-status Critical
Expired - Fee Related legal-status Critical Current

Links

Classifications

    • YGENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02BCLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B40/00Technologies aiming at improving the efficiency of home appliances, e.g. induction cooking or efficient technologies for refrigerators, freezers or dish washers

Landscapes

  • Control Of High-Frequency Heating Circuits (AREA)
  • Rectifiers (AREA)

Abstract

PROBLEM TO BE SOLVED: To provide a control method for reducing the distortion of an input current to stabilize this input current during oscillation of a magnetron in the high-frequency heating power supply for driving the magnetron. SOLUTION: This high-frequency heating power supply has a structure such that the on/off duty ratio of each semiconductor switch element is set to 50% and the role of these elements are interchanged smoothly at an area near the polarity changing point, when one semiconductor switching element 12, which executes the voltage charge-up function and inverter function in the switching operations of the first inversely conductible semiconductor switching element 12 and the second inversely conductible semiconductor switching element 13 interchanges it functions to the other semiconductor switching element 13, which executes only the inverter function at the polarity changing point, where the polarity of the commercial power supply 1 changes. With this structure, since the semiconductor switch elements 12, 13 interchange the roles with the timings, where charge or discharge process of the charge-up capacitor 16 or 17 is completed, a stable input current in which the harmonic elements are controlled can be obtained in a high power-factor with less distortion.

Description

【発明の詳細な説明】DETAILED DESCRIPTION OF THE INVENTION

【0001】[0001]

【発明の属する技術分野】本発明は電子レンジのように
マグネトロンを用いて誘電加熱を行う高周波加熱装置の
分野で、特にマグネトロンを駆動する電源装置の入力電
流の歪を少なくし、安定させる制御に関するものであ
る。
BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to the field of high-frequency heating devices for performing dielectric heating using a magnetron, such as a microwave oven, and more particularly to control for reducing and stabilizing input current of a power supply device for driving a magnetron. Things.

【0002】[0002]

【従来の技術】従来この種のマグネトロンを駆動するた
めの高周波加熱電源装置は図14に示すような回路構成
で、動作を説明すると、交流である商用電源1を一旦ダ
イオードブリッジ2で直流電圧に変換し、この直流電圧
を半導体スイッチ素子3、4のオンオフによってインバ
ータ回路5を動作させ、高圧トランス6の一次巻線に高
周波電圧を発生し、高圧トランス6は二次巻線に高周波
高電圧を励起する。この高周波高電圧は高圧整流回路7
によって直流高電圧に整流され、マグネトロン8に印加
される。マグネトロン8はこの直流高電圧で駆動され、
2.45GHzの電波を発生する。ここでインダクタ9
とコンデンサ10は平滑回路を構成しているが、コンデ
ンサ10の容量はインバータ回路5の小型化が進む現状
として、動作周波数(20KHz〜50KHz)に対して
直流電圧を保持できる程度の容量としており、商用電源
1の周波数に対しては平滑する能力を有していない。こ
のため図15に示す通りコンデンサ10の電圧V10は
商用電源1を単に全波整流した波形を示し、ほぼ0電圧
から商用電源1の最大電圧まで変動する脈動波形を示
す。インバータ回路5はこの脈動するコンデンサ10の
電圧V10によって動作するので、高圧トランス6の一
次巻線に発生する高周波電圧の包絡線波形はV6(Lp)
に示すような波形となりコンデンサ10の電圧V10が
低い期間では同様に低い電圧しか発生し得ない。すなわ
ち、非線形な特性を持つマグネトロン8において発振す
る閾値VAK(TH)に達しなくなる期間が発生する。
この期間はマグネトロン8が発振を停止しているため負
荷であるマグネトロン8で電力消費されないため商用電
源1の電流I1は流れなくなり、電流が0となる期間を
有する非常に歪を持った波形となり、力率の低下と入力
電流に高調波電流を発生する結果となっていた。
2. Description of the Related Art A conventional high frequency heating power supply for driving a magnetron of this type has a circuit configuration as shown in FIG. The DC voltage is converted, and the inverter circuit 5 is operated by turning on and off the semiconductor switching elements 3 and 4 to generate a high-frequency voltage in the primary winding of the high-voltage transformer 6. The high-voltage transformer 6 applies the high-frequency high voltage to the secondary winding. To excite. This high frequency high voltage is supplied to the high voltage rectifier circuit 7
Is rectified to a high DC voltage and applied to the magnetron 8. The magnetron 8 is driven by this DC high voltage,
Generates a 2.45 GHz radio wave. Where inductor 9
And the capacitor 10 constitute a smoothing circuit, but the capacity of the capacitor 10 is such that the DC voltage can be held at the operating frequency (20 KHz to 50 KHz) as the miniaturization of the inverter circuit 5 progresses. It does not have the ability to smooth the frequency of the commercial power supply 1. Therefore, as shown in FIG. 15, the voltage V10 of the capacitor 10 shows a waveform obtained by simply rectifying the commercial power supply 1 in full wave, and shows a pulsating waveform that varies from almost zero voltage to the maximum voltage of the commercial power supply 1. Since the inverter circuit 5 is operated by the pulsating voltage V10 of the capacitor 10, the envelope waveform of the high-frequency voltage generated in the primary winding of the high-voltage transformer 6 is V6 (Lp).
In the period where the voltage V10 of the capacitor 10 is low, only a low voltage can be generated. That is, a period occurs in which the oscillation does not reach the threshold value VAK (TH) in the magnetron 8 having the non-linear characteristic.
During this period, since the magnetron 8 stops oscillating, power is not consumed by the magnetron 8 as a load, and thus the current I1 of the commercial power supply 1 does not flow, resulting in a very distorted waveform having a period in which the current becomes zero. This resulted in a reduction in power factor and generation of harmonic current in the input current.

【0003】このため、商用電源からの脈動波形の谷間
付近で電圧を補完するための回路構成として昇圧チョパ
回路を利用した方法で部品点数の減数や小型化という面
から構成部品、回路構成の共用化を図り、この昇圧機能
と先のインバータ機能の動作を一度に行う方法が多数提
案されており、代表的なものが特開平10−27184
6にて公開されている。図16はこの発明の回路構成を
示した回路図である。しかしながらこの特許における負
荷回路11は放電灯のような消費電力の小さなものであ
り、電子レンジのように大きな電力を扱う電源装置にお
いては昇圧動作とインバータ動作を司る半導体スイッチ
素子Q1,Q2のオンオフのドライブ信号は昇圧用のコ
ンデンサの充放電をするための期間、いわゆるデッドダ
イムが必要ない。さらにまた、電子レンジのように加熱
設定における強・中・弱といった加熱電力(消費電力)の
調整が必要ないため、商用電源1の極性が変わる瞬間や
商用電源1の0電圧部と最大電圧部での半導体スイッチ
素子Q1,Q2のドライブ信号の制御も特に注意する必
要がなかった。
For this reason, a method using a boosting chopper circuit as a circuit configuration for complementing a voltage near a valley of a pulsating waveform from a commercial power supply uses a common component and a circuit configuration in terms of reduction in the number of parts and miniaturization. A number of methods have been proposed for implementing the boosting function and the inverter function at the same time, and a typical method is disclosed in Japanese Patent Application Laid-Open No. 10-27184.
6 has been published. FIG. 16 is a circuit diagram showing a circuit configuration of the present invention. However, the load circuit 11 in this patent consumes a small amount of power, such as a discharge lamp. In a power supply device that handles a large amount of power, such as a microwave oven, it turns on and off the semiconductor switching elements Q1 and Q2 that control the boosting operation and the inverter operation. The drive signal does not require a so-called dead dime during a period for charging and discharging the boosting capacitor. Furthermore, since there is no need to adjust the heating power (power consumption) such as strong, medium, and weak in the heating setting as in a microwave oven, the moment the polarity of the commercial power supply 1 changes, the 0 voltage section and the maximum voltage section of the commercial power supply 1 There is no need to pay special attention to the control of the drive signals of the semiconductor switching elements Q1 and Q2 in the above.

【0004】[0004]

【発明が解決しようとする課題】しかしながら、上記の
ような構成では下記の課題があった。すなわち、実際に
電子レンジのような大電力を扱う機器の制御においては
電源電圧の極性により半導体スイッチ素子のオンオフの
タイミングを切り替える必要のある回路構成を使用した
場合、その極性が変わる変極点でのドライブ信号の制御
は非常に重要となる。なぜならば昇圧チャージアップ機
能とインバータ機能とを司る一方の半導体スイッチ素子
とインバータ機能のみを司るもう一方の半導体スイッチ
素子が変極点で入れ替わる際のデューティ比や入れ替え
タイミングによりチャージアップコンデンサの充放電が
うまく切り替わらなければ入力電流には変極点付近で針
状の歪が発生してしまう。従来この手の回路構成は負荷
回路が放電灯のような消費電力が少なく電流値が微小な
ため、またチャージアップ用コンデンサの容量も小さい
ため入力電流歪はほとんど確認されなかったが、電子レ
ンジのような消費電力の大きな負荷回路の場合、入力電
流波形が大きく歪み、力率低下と高調波成分の拡大が懸
念される。
However, the above configuration has the following problems. In other words, when actually controlling a device that handles large power such as a microwave oven, when using a circuit configuration that needs to switch the on / off timing of the semiconductor switch element according to the polarity of the power supply voltage, at the inflection point where the polarity changes. Control of the drive signal is very important. This is because the charge-discharge of the charge-up capacitor depends on the duty ratio and the switching timing when one semiconductor switch element that controls the boost charge-up function and the inverter function and the other semiconductor switch element that controls the inverter function only at the inflection point. If the input current is not switched, a needle-like distortion occurs in the input current near the inflection point. Conventionally, with this kind of circuit configuration, input current distortion was hardly observed because the load circuit had low power consumption like a discharge lamp and the current value was minute, and the capacity of the charge-up capacitor was also small. In the case of such a load circuit having large power consumption, the input current waveform is greatly distorted, and there is a concern that the power factor is reduced and harmonic components are expanded.

【0005】さらに、特開平10−271846号公報
で開示されている従来例で示した構成は照明機器を対象
としたものであり、これらの機器の変換電力は最大でも
100Wから200W程度である。したがって回路を流
れる電流もおのずと数A程度の微小な電流しか流れない
のでダイオードをスイッチングスピード重視の設計とし
て順方向オン電圧VFが高くなるようにしてもダイオー
ドの損失はそれほど増加することなく設計することが可
能である。
Furthermore, the configuration shown in the conventional example disclosed in Japanese Patent Application Laid-Open No. 10-271846 is for lighting equipment, and the conversion power of these equipment is at most about 100 W to 200 W. Therefore, only a very small current of about several A flows through the circuit. Therefore, even when the forward ON voltage VF is increased, the diode loss should not be increased even if the forward ON voltage VF is increased. Is possible.

【0006】一方、電子レンジ等に用いられるマグネト
ロン駆動用電源は変換電力が1000Wから1500W
程度の大電力を扱うので、回路を流れる電流は最大で4
0Aから50Aの大電流が流れる。このためスイッチン
グスピードを重視してダイオードを設計すると順方向オ
ン電圧VFが高くなるのでダイオードが導通していると
きの損失(導通損失)が大きくなり、スイッチングスピー
ドを速くする事によって損失を低減しようとする効果が
薄れてしまう。また家庭用電子レンジの冷却能力は電子
レンジのサイズやコストの関係からおのずと限られたも
のとなるのでスイッチングスピードを速め、順方向オン
電圧VFの上昇を抑えるためにはダイオードの大型化あ
るいは限られた冷却条件下で放熱するための大型の放熱
フィンなどが必要となってくる。このためマグネトロン
駆動用電源においては変換効率を高め、回路の各部品で
の発生損失低減が必須条件となる。したがって従来例で
示した構成をマグネトロン駆動用電源に適用することは
損失低減の観点からすると非常に困難を伴ってしまう。
そのためマグネトロン駆動用電源に適用する場合にはダ
イオードのスイッチング損失もオン損失も増加させない
ような回路構成とすることが必要となる。また、変換電
力の大きさゆえマグネトロン駆動用電源に電解コンデン
サを用いると電解コンデンサの脈動電流を押さえるため
に高容量でかつ耐電圧の高いスペックの電解コンデンサ
を必要としてしまう。その結果、電源自体の大型化を招
くことからマグネトロン駆動用電源を搭載する電子レン
ジのサイズアップを誘発し、高周波スイッチング動作に
よってマグネトロン駆動用電源を小型軽量化する効果が
損なわれてしまうという課題も有している。
On the other hand, a power supply for driving a magnetron used for a microwave oven or the like has a converted power of 1000 W to 1500 W.
Since it handles large amounts of power, the current flowing through the circuit is at most 4
A large current of 0 A to 50 A flows. For this reason, when a diode is designed with a focus on switching speed, the forward ON voltage VF becomes high, so that the loss (conduction loss) when the diode is conducting increases, and an attempt is made to reduce the loss by increasing the switching speed. The effect of doing is diminished. In addition, the cooling capacity of a home microwave oven is naturally limited due to the size and cost of the microwave oven. Therefore, in order to increase the switching speed and suppress the rise of the forward ON voltage VF, the size or the size of the diode is limited. Large radiating fins and the like for radiating heat under cooling conditions are required. For this reason, in the power supply for driving the magnetron, it is indispensable to increase the conversion efficiency and reduce the loss generated in each component of the circuit. Therefore, it is very difficult to apply the configuration shown in the conventional example to a magnetron driving power supply from the viewpoint of loss reduction.
Therefore, when applied to a magnetron drive power supply, it is necessary to provide a circuit configuration that does not increase the switching loss and the on-loss of the diode. Further, if an electrolytic capacitor is used as a power supply for driving the magnetron because of the large amount of converted power, an electrolytic capacitor having a high capacity and a high withstand voltage specification is required to suppress the pulsating current of the electrolytic capacitor. As a result, the size of the power supply itself is increased, which causes an increase in the size of the microwave oven equipped with the power supply for driving the magnetron, and the high-frequency switching operation impairs the effect of reducing the size and weight of the power supply for driving the magnetron. Have.

【0007】[0007]

【課題を解決するための手段】本発明は上記課題を解決
するために、まず大電力を扱えるような電源装置として
第1および第2の逆導通可能な半導体スイッチ素子の直
列接続体と、第1および第2のダイオードの直列接続体
を並列接続し、前記第1および第2のダイオードにそれ
ぞれ第1と第2のコンデンサを並列接続し、前記第1お
よび第2のダイオードの接続点と前記第1および第2の
逆導通可能な半導体スイッチ素子の接続点との間に商用
電源と高圧トランスの直列回路を接続するとともに、前
記高圧トランスの高圧出力は高圧整流回路を介してマグ
ネトロンに電力を供給する構成としている。
SUMMARY OF THE INVENTION In order to solve the above-mentioned problems, the present invention provides a power supply device capable of handling a large amount of power, a first and a second series-connected semiconductor switch element capable of reverse conduction, and A first and a second diode are connected in parallel, a first and a second capacitor are connected in parallel to the first and the second diode, respectively, and a connection point of the first and the second diode is connected to the A series circuit of a commercial power supply and a high-voltage transformer is connected between the connection point of the first and second semiconductor switches capable of reverse conduction, and the high-voltage output of the high-voltage transformer supplies power to the magnetron via a high-voltage rectifier circuit. It is configured to supply.

【0008】これにより、前記第1および第2の半導体
スイッチ素子が相補的にオンオフすることによって前期
商用電源の電圧極性が正の場合は前記第2のコンデンサ
に商用電源の電圧を昇圧した電圧が加えられ、これとは
逆の電圧極性(負極性)の場合は前記第1のコンデンサに
前記商用電源を昇圧した電圧が加えられる。前記高圧ト
ランスの1次巻線に印加する電圧はこの昇圧電圧に依存
するので前記商用電源の電圧が低い期間であってもマグ
ネトロンが発振するために必要な電圧以上を常に前記高
圧トランスの1次巻線に印加することができ前記商用電
源のほぼ全域に渡り入力電流を流すことができ、歪の少
ない入力電流とすることができる。また、前記第1およ
び第2の半導体スイッチ素子は前記高圧トランスの1次
巻線に高周波電流を流すインバータ動作と前記第1およ
び第2のコンデンサに昇圧電圧を印加する動作を一度に
行うことができるのでインバータ構成部品を最小で構成
することができインバータ回路を小型化できる。また、
回路動作において前記第1および第2のダイオードのタ
ーンオフは半導体スイッチ素子が行って回路モードが切
り替わるのでこれらのダイオードはスイッチングスピー
ドに対する制約を受けることなく順方向オン電圧VFを
重視した設計とすることができ、このダイオードの損失
をきわめて少なくしてインバータ回路を高効率化するこ
とができる。
When the voltage polarity of the commercial power supply is positive as a result of the first and second semiconductor switch elements being turned on and off complementarily, the voltage obtained by boosting the voltage of the commercial power supply is applied to the second capacitor. In the case of the opposite voltage polarity (negative polarity), a voltage obtained by boosting the commercial power supply is applied to the first capacitor. Since the voltage applied to the primary winding of the high-voltage transformer depends on the boosted voltage, the voltage required for the magnetron to oscillate always exceeds the voltage required for the magnetron to oscillate even during the period when the voltage of the commercial power supply is low. The input current can be applied to the winding, and the input current can flow over almost the entire area of the commercial power supply, so that the input current can be reduced in distortion. Further, the first and second semiconductor switch elements may perform an inverter operation for supplying a high-frequency current to a primary winding of the high-voltage transformer and an operation for applying a boosted voltage to the first and second capacitors at a time. Therefore, the number of inverter components can be minimized, and the inverter circuit can be downsized. Also,
In the circuit operation, the first and second diodes are turned off by the semiconductor switching element and the circuit mode is switched. Therefore, these diodes should be designed with emphasis on the forward ON voltage VF without being restricted by the switching speed. Therefore, the efficiency of the inverter circuit can be improved by minimizing the loss of the diode.

【0009】上記のような構成において本発明は、極性
の変わる変極点付近において第1および第2の半導体ス
イッチ素子のオンオフデューティー比を各々50%と
し、変極点付近で昇圧チャージアップ機能とインバータ
機能を司る一方の半導体スイッチ素子とインバータ機能
のみを司るもう一方の半導体スイッチ素子を切り替え
る。このような手段により、極性の変わる変極点付近に
おいて昇圧チャージアップ機能とインバータ機能とを司
る一方のスイッチング素子をチャージアップコンデンサ
の充放電が完了するタイミングで切り替えることができ
るので、安定した高力率で高調波成分のカットされた入
力電流が得られる。
In the above configuration, the present invention provides an on-off duty ratio of each of the first and second semiconductor switching elements of 50% near the inflection point where the polarity changes, and a boost charge-up function and an inverter function near the inflection point. And the other semiconductor switch element that controls only the inverter function. By such means, one of the switching elements, which controls the boosting charge-up function and the inverter function, can be switched at the timing when the charging and discharging of the charge-up capacitor is completed near the inflection point where the polarity changes, so that a stable high power factor As a result, an input current from which harmonic components have been cut is obtained.

【0010】[0010]

【発明の実施の形態】請求項1記載の発明は、第1およ
び第2の逆導通可能な半導体スイッチ素子の直列接続体
と、第1および第2のダイオードの直列接続体を並列接
続し、前記第1および第2のダイオードに各々並列に第
1と第2のコンデンサを接続し、前記第1および第2の
逆導通可能な半導体スイッチ素子の接続点と、前記第1
および第2のダイオードの接続点間に商用電源と高圧ト
ランスの1次巻線の直列回路とを接続して前記高圧トラ
ンスの2次巻線の出力は高圧整流回路を介してマグネト
ロンを駆動する構成としたもののうち、前記第1および
第2の逆導通可能な半導体スイッチ素子のオンオフデュ
ーティ比を前記商用電源の極性が変わる変極点付近では
各々50%となるように構成した。
According to the first aspect of the present invention, a series connection of first and second semiconductor switches capable of reverse conduction and a series connection of first and second diodes are connected in parallel. First and second capacitors are connected in parallel to the first and second diodes, respectively, and a connection point of the first and second reverse-conducting semiconductor switch elements is connected to the first and second diodes.
A configuration in which a commercial power supply and a series circuit of a primary winding of a high-voltage transformer are connected between a connection point of a second diode and an output of a secondary winding of the high-voltage transformer drives a magnetron via a high-voltage rectifier circuit. The on-off duty ratios of the first and second reverse-conducting semiconductor switch elements are each set to 50% near the inflection point at which the polarity of the commercial power supply changes.

【0011】請求項2に記載の発明は、第1および第2
の逆導通可能な半導体スイッチ素子の直列接続体と、第
1および第2のダイオードの直列接続体を並列接続し、
前記第1および第2のダイオードに各々並列に第1と第
2のコンデンサを接続し、前記第1および第2の逆導通
可能な半導体スイッチ素子の接続点と、前記第1および
第2のダイオードの接続点間に商用電源と高圧トランス
の1次巻線の直列回路とを接続して前記高圧トランスの
2次巻線の出力は高圧整流回路を介してマグネトロンを
駆動する構成としたもののうち、前記商用電源の変極点
付近での制御において極性判別手段を備えつつその変極
点を検知することにより昇圧チャージアッブ機能とイン
バータ機能の両方とインバータ機能のみの役割を相補的
に同時に行う前記第1および第2の逆導通可能な半導体
スイッチ素子の役割を入れ替える構成とした。
[0011] The invention as defined in claim 2 is the first and second inventions.
A series connection of a semiconductor switch element capable of reverse conduction and a series connection of a first and a second diode are connected in parallel;
First and second capacitors connected in parallel to the first and second diodes, respectively, and a connection point of the first and second reverse-conducting semiconductor switch elements; and a first and second diode. And a power supply and a series circuit of a primary winding of a high-voltage transformer are connected between the connection points, and an output of a secondary winding of the high-voltage transformer is configured to drive a magnetron through a high-voltage rectifier circuit. In the control near the inflection point of the commercial power supply, by detecting the inflection point while providing the polarity discriminating means, both the step-up charge-up function and the inverter function and the roles of the inverter function and the inverter function alone are simultaneously performed in a complementary manner. The configuration is such that the role of the second reverse-conducting semiconductor switch element is switched.

【0012】上記構成により、電子レンジのような消費
電力の大きな負荷回路においても商用電源の極性が変わ
る変極点における針状の入力電流波形歪みを抑え、さら
に力率低下と高調波成分の拡大も抑えることを可能とし
た。
With the above configuration, even in a load circuit such as a microwave oven which consumes a large amount of power, a needle-shaped input current waveform distortion at an inflection point at which the polarity of the commercial power supply changes is suppressed, and a power factor is reduced and harmonic components are expanded. It is possible to suppress.

【0013】[0013]

【実施例】以下、本発明の実施例について図面を用いて
説明する。
Embodiments of the present invention will be described below with reference to the drawings.

【0014】(実施例1)本発明の第1の実施例に用い
る回路構成を図1に示す。第1および第2の半導体スイ
ッチ素子12、13の直列接続体と第1および第2のダ
イオード14、15の直列接続体を並列接続し、第1お
よび第2のダイオード14、15に各々並列に第1およ
び第2のコンデンサ16、17を接続するとともに半導
体スイッチ素子12、13の接続点とダイオード14、
15の接続点間に商用電源1と高圧トランス18の直列
回路を接続する構成となっている。高圧トランス18の
2次巻線出力は高圧整流回路7に接続されマグネトロン
8に直流高電圧を印加する。マグネトロン8はこの直流
高電圧によって駆動され、2.45GHzの電波を発生
する。なお本実施例では第1、第2の半導体スイッチ素
子は順方向に導通するIGBT(絶縁ゲートバイポーラ
トランジスタ)とこれと逆並列に接続したダイオードに
て記載しているが、MOSFETのように素子内部にダ
イオードを構成したような素子を用いても適用可能であ
ることは言うまでもない。
(Embodiment 1) FIG. 1 shows a circuit configuration used in a first embodiment of the present invention. A series connection of the first and second semiconductor switch elements 12 and 13 and a series connection of the first and second diodes 14 and 15 are connected in parallel, and are connected in parallel to the first and second diodes 14 and 15, respectively. The first and second capacitors 16 and 17 are connected, and the connection point between the semiconductor switch elements 12 and 13 and the diode 14 are connected.
In this configuration, a series circuit of the commercial power supply 1 and the high-voltage transformer 18 is connected between the 15 connection points. The output of the secondary winding of the high voltage transformer 18 is connected to the high voltage rectifier circuit 7 and applies a high DC voltage to the magnetron 8. The magnetron 8 is driven by the high DC voltage and generates a radio wave of 2.45 GHz. In the present embodiment, the first and second semiconductor switch elements are described as IGBTs (insulated gate bipolar transistors) that conduct in the forward direction and diodes connected in anti-parallel to the IGBTs. It is needless to say that the present invention can be applied even if an element such as a diode is used.

【0015】図2はインバータ回路の各期間における電
流が流れる経路を示した図であり、図3はそれに対応し
た動作波形図である。商用電源1の極性が図示の状態で
半導体スイッチ素子13がオンの状態から説明をはじめ
る。この状態では図2(a)に示すように商用電源1→
高圧トランス18の1次巻線→半導体スイッチ素子13
→ダイオード15の経路で電流が流れ、図3(a)の期
間のI13に示す電流が半導体スイッチ素子13および
高圧トランス18の1次巻線に電流が流れることによっ
て高圧トランス18の1次巻線にエネルギーを蓄積す
る。半導体スイッチ素子13を所定の時間でオフすると
高圧トランス18の1次巻線電流は同じ方向に流れ続け
ようとするので今度は図2(b)に示すごとく商用電源
1→高圧トランス18の1次巻線→半導体スイッチ素子
12の並列ダイオード→コンデンサ16の経路で高圧ト
ランス18の1次巻線に蓄えたエネルギーをコンデンサ
16に充電する。この動作によってコンデンサ16には
商用電源1の電圧を昇圧した電圧が蓄えられる。高圧ト
ランス18の1次巻線に蓄えたエネルギーをすべて放出
すると図2(c)の経路が形成され今度はコンデンサ1
6に充電したエネルギーをコンデンサ16→半導体スイ
ッチ素子12→高圧トランス18の1次巻線→商用電源
1の経路で取り出す。そして半導体スイッチ素子12を
所定の時間でオフすると高圧トランス18の1次巻線は
同じ方向に電流を流し続けようとするので図2(d)の
ように高圧トランス18の1次巻線→商用電源1→コン
デンサ17→半導体スイッチ素子13の並列ダイオード
の経路で電流が流れる。商用電源1の電圧極性が図示と
逆極性の場合は半導体スイッチ素子12、13とダイオ
ード14、15とコンデンサ16、17の動作がそれぞ
れ入れ替わるだけで同様の動作となる。
FIG. 2 is a diagram showing paths through which current flows in each period of the inverter circuit, and FIG. 3 is an operation waveform diagram corresponding thereto. The description starts from the state where the polarity of the commercial power supply 1 is shown and the semiconductor switch element 13 is on. In this state, as shown in FIG.
Primary winding of high-voltage transformer 18 → Semiconductor switch element 13
→ A current flows through the path of the diode 15, and the current indicated by I13 in the period of FIG. 3A flows through the semiconductor switch element 13 and the primary winding of the high-voltage transformer 18, so that the primary winding of the high-voltage transformer 18 To store energy. When the semiconductor switch element 13 is turned off for a predetermined time, the primary winding current of the high voltage transformer 18 tends to continue to flow in the same direction. Therefore, this time, as shown in FIG. The energy stored in the primary winding of the high-voltage transformer 18 is charged to the capacitor 16 through the path of the winding → the parallel diode of the semiconductor switching element 12 → the capacitor 16. By this operation, a voltage obtained by boosting the voltage of the commercial power supply 1 is stored in the capacitor 16. When all the energy stored in the primary winding of the high-voltage transformer 18 is released, a path shown in FIG.
The energy charged in 6 is taken out through the path of the capacitor 16 → the semiconductor switching element 12 → the primary winding of the high voltage transformer 18 → the commercial power supply 1. Then, when the semiconductor switch element 12 is turned off for a predetermined time, the primary winding of the high voltage transformer 18 tries to keep the current flowing in the same direction. Therefore, as shown in FIG. A current flows through a path from the power supply 1 → the capacitor 17 → the parallel diode of the semiconductor switch element 13. When the voltage polarity of the commercial power supply 1 is opposite to that shown in the figure, the same operation is performed except that the operations of the semiconductor switch elements 12 and 13, the diodes 14 and 15, and the capacitors 16 and 17 are switched.

【0016】上記の動作においてコンデンサ16、17
は半導体スイッチ素子12、13のオンオフによって高
圧トランス18の1次巻線に高周波電流を発生させるイ
ンバータ動作と商用電源1の電圧に対して昇圧した電圧
をコンデンサ16、17に発生させる動作を兼用できる
ような容量に設計され、コンデンサ16、17の容量は
相等しい容量で構成されている。この結果商用電源1の
電圧極性が図示の場合はコンデンサ16に商用電源1の
電圧を昇圧した電圧を蓄え、反対に商用電源1の電圧極
性が図示とは逆極性の場合はコンデンサ17に商用電源
1の電圧を昇圧した電圧を蓄える動作をする。したがっ
て商用電源1の電圧極性によらずコンデンサ16、17
に発生する電圧を等しくすることができるので商用電源
1の電流は正負対称な波形とすることができる。そし
て、このような動作を継続することで図4に示すように
商用電源1の周期に対してコンデンサ16、17の電圧
波形は商用電源1の電圧極性に応じて昇圧した電圧を発
生する。このため高圧トランス18の1次巻線に流れる
電流の包絡線波形はV18(Lp)に示すような波形と
なる。この電圧を高圧トランス18は昇圧してマグネト
ロン8に印加するのでマグネトロン8に印加する電圧は
V8のような波形を示し、常に発振電圧VAK(TH)以
上の電圧を維持することが可能となる。この結果入力電
流I1は商用電源1のいずれの期間においても電流を流
すことができ、力率の改善、高調波の抑制を実現するこ
とができる。
In the above operation, the capacitors 16, 17
Can perform both the inverter operation of generating a high-frequency current in the primary winding of the high-voltage transformer 18 by turning on and off the semiconductor switch elements 12 and 13 and the operation of generating the voltage boosted with respect to the voltage of the commercial power supply 1 in the capacitors 16 and 17. The capacitors are designed to have such a capacity, and the capacitors 16 and 17 have the same capacity. As a result, when the voltage polarity of the commercial power supply 1 is as shown, the boosted voltage of the voltage of the commercial power supply 1 is stored in the capacitor 16, and when the voltage polarity of the commercial power supply 1 is opposite to that shown in the figure, the capacitor 17 is connected to the capacitor 17. An operation of storing a voltage obtained by boosting the voltage of 1 is performed. Therefore, regardless of the voltage polarity of the commercial power supply 1, the capacitors 16, 17
Can be made equal, the current of the commercial power supply 1 can have a positive and negative symmetric waveform. By continuing such an operation, the voltage waveform of the capacitors 16 and 17 with respect to the cycle of the commercial power supply 1 generates a voltage boosted according to the voltage polarity of the commercial power supply 1 as shown in FIG. For this reason, the envelope waveform of the current flowing through the primary winding of the high-voltage transformer 18 has a waveform as indicated by V18 (Lp). Since this voltage is boosted by the high-voltage transformer 18 and applied to the magnetron 8, the voltage applied to the magnetron 8 has a waveform like V8, and it is possible to always maintain the voltage equal to or higher than the oscillation voltage VAK (TH). As a result, the input current I1 can flow during any period of the commercial power supply 1, thereby improving the power factor and suppressing harmonics.

【0017】また、図3において期間(a)から(b)
へ移行する際ダイオード15をカットオフする動作にな
るが電流経路として半導体スイッチ素子13が直列に接
続されているので電流の遮断は半導体スイッチ素子13
が行うことになりダイオード15のスイッチングスピー
ドは要求されない。また、オフ時にダイオード15に印
加する電圧は零であるのでターンオフ時のスイッチング
損失はまったく生じない。したがってダイオード14、
15の設計としては順方向オン電圧VFを重視した設計
で導通時の損失を重点的に抑制するように設計すること
が可能となり、ダイオード14、15の小型化と同時に
ダイオード14、15を冷却する構成の簡素化を図るこ
とが容易となる。特に電子レンジで用いるようなマグネ
トロン駆動用電源は1000W以上の高電力を扱うので
インバータ回路の電流は40Aから50A程度の非常に
大きな電流レベルとなりダイオード14、15の設計に
おいて順方向オン電圧VFを重視して導通損失を低減す
ることはインバータ回路の効率向上に有益である。この
ためインバータ回路のトータルの電力損失をきわめて低
く抑えることができ、効率の高いマグネトロン駆動用電
源を実現することができる。
Also, in FIG. 3, the periods (a) to (b)
The operation is to cut off the diode 15 at the time of shifting to the operation mode. However, since the semiconductor switching element 13 is connected in series as a current path, the current is cut off by the semiconductor switching element 13.
Therefore, the switching speed of the diode 15 is not required. Further, since the voltage applied to the diode 15 at the time of turning off is zero, there is no switching loss at the time of turning off. Therefore diode 14,
As for the design of 15, the forward ON voltage VF is emphasized, and it is possible to design so as to mainly suppress the loss at the time of conduction, and to cool down the diodes 14 and 15 at the same time as the diodes 14 and 15 are downsized. It is easy to simplify the configuration. In particular, a magnetron driving power supply used in a microwave oven handles a high power of 1000 W or more, so the current of the inverter circuit becomes a very large current level of about 40 A to 50 A, and the forward ON voltage VF is emphasized in designing the diodes 14 and 15. Reducing the conduction loss by doing so is beneficial for improving the efficiency of the inverter circuit. For this reason, the total power loss of the inverter circuit can be suppressed extremely low, and a highly efficient magnetron drive power supply can be realized.

【0018】このように本実施例で使用するマグネトロ
ン駆動用電源においては従来例で示した回路とは全く異
なった回路動作によってダイオード14、15の設計を
順方向オン電圧VF重視の設計とすることが可能となり
ダイオード14、15の損失を極小化しマグネトロン駆
動用電源全体の電力変換効率を向上している。この効果
はコンデンサ16、17がインバータ動作と商用電源1
の電圧を昇圧した電圧を加える動作を兼用することによ
って発揮される特有の効果であり、従来例にて挙げた特
開平10−271846の構成とは異なったコンデンサ
の回路機能と回路動作によって実現されるものである。
As described above, in the magnetron driving power supply used in this embodiment, the design of the diodes 14 and 15 is designed to emphasize the forward ON voltage VF by a circuit operation completely different from the circuit shown in the conventional example. And the loss of the diodes 14 and 15 is minimized, and the power conversion efficiency of the entire magnetron drive power supply is improved. The effect is that the capacitors 16 and 17 operate with inverter and commercial power 1
This is a unique effect exhibited by also using the operation of applying the boosted voltage of the above-mentioned voltage, and is realized by the circuit function and circuit operation of the capacitor different from the configuration of JP-A-10-271846 mentioned in the conventional example. Things.

【0019】図5は本実施例のマグネトロン駆動用電源
のより実際的な回路構成を示したものであり、商用電源
1の出力にインダクタ19とコンデンサ20からなるロ
ーパスフィルタ21を設けることによってインバータ回
路の高周波電流が商用電源へ流れないように構成したも
のである。このように商用電源1とインバータ回路の間
にローパスフィルタ21を挿入する構成とすることによ
りインバータ回路の高周波電流あるいは電圧が商用電源
側へ回り込まないようにすることによって端子雑音の低
減を図ることが可能となる。なお、本構成によっても上
述の動作は何ら変わらない。以下半導体スイッチ素子1
2、13のドライブ信号の制御を行う駆動回路22を中
心に説明していく。
FIG. 5 shows a more practical circuit configuration of the magnetron driving power supply according to the present embodiment. An inverter circuit is provided by providing a low-pass filter 21 composed of an inductor 19 and a capacitor 20 at the output of the commercial power supply 1. In such a manner that the high-frequency current does not flow to the commercial power supply. As described above, by inserting the low-pass filter 21 between the commercial power supply 1 and the inverter circuit, terminal noise can be reduced by preventing high-frequency current or voltage of the inverter circuit from flowing to the commercial power supply side. It becomes possible. The above operation does not change at all with this configuration. The following semiconductor switch element 1
The description will focus on the drive circuit 22 that controls the drive signals 2 and 13.

【0020】駆動回路22は半導体スイッチ素子12、
13を駆動してインバータ回路を動作させる。駆動回路
22が半導体スイッチ素子12、13へ送る駆動信号V
g12、Vg13は図6(a)に示すようにデッドタイ
ムを有し相補的にオンオフするような波形である。この
ように半導体スイッチ素子12、13を相補的にオンオ
フすることによってインバータ回路はマグネトロン8へ
電力を伝送している。
The drive circuit 22 includes the semiconductor switch element 12,
13 to drive the inverter circuit. The drive signal V that the drive circuit 22 sends to the semiconductor switch elements 12 and 13
As shown in FIG. 6A, g12 and Vg13 have waveforms that have a dead time and turn on and off complementarily. The inverter circuit transmits power to the magnetron 8 by turning on and off the semiconductor switch elements 12 and 13 in a complementary manner.

【0021】ここで半導体スイッチ素子13のオン時間
比率Don13とインバータ回路の変換電力Pとの関係
を図7に示す。図中で実線にて示した曲線は商用電源1
の電圧極性が図5に示す極性の場合での変換電力Pの変
化を示したものであり、破線で示した曲線はこれとは逆
に商用電源1の電圧極性が図5とは逆の電圧極性の場合
での変換電力Pの変化を示している。このように商用電
源1の電圧極性によって半導体スイッチ素子13のオン
時間比率Don13とインバータ回路の変換電力Pの関
係は異なったものとなる。したがって、半導体スイッチ
素子13のオン時間比率Don13が略50%の状態で
は商用電源1の電圧極性によらずいずれの電圧極性にお
いても同じ電力変換が可能なので商用電源1の電流は図
8(b)に示すように正負対称の波形とすることができ
る。しかしながら商用電源1の電流を正負対称な正弦波
にしようとすると変換電力はこの一点のみで限定されて
しまう。家庭用の電子レンジなどでは電子レンジで加熱
する際に食品に応じてさまざまな加熱電力が選択され
る。たとえばレンジ“強”、“中”、“弱”などの設定
によって加熱電力の調整が必要とされる。これに対応す
るためには半導体スイッチ素子13のオン時間比率Do
n13を所望の出力電力に応じて変化させることが必要
となるが、商用電源1の電圧極性に関係なく一定のオン
時間比Don13で所望の出力電力に調整しようとする
と図7に示した半導体スイッチ素子13のオン時間比率
Don13と変換電力Pの関係からオン時間比Don1
3が50%から外れることとなり、商用電源1の電圧が
正の期間と負の期間で異なった電流波形を示すことにな
る。そして制御方法を誤れば図8(a)に示すような正
負がアンバランスな電流波形となってしまう。この場
合、電流波形が対称波形とならないので偶数次の高調波
が発生することになり結局のところ力率を向上すること
はできなくなってしまう。
FIG. 7 shows the relationship between the ON time ratio Don13 of the semiconductor switch element 13 and the converted power P of the inverter circuit. The curve shown by the solid line in the figure is the commercial power supply 1
5 shows the change of the converted power P when the voltage polarity is the polarity shown in FIG. 5, and the curve shown by the broken line is opposite to this. The change of the conversion power P in the case of the polarity is shown. As described above, the relationship between the on-time ratio Don13 of the semiconductor switch element 13 and the converted power P of the inverter circuit differs depending on the voltage polarity of the commercial power supply 1. Therefore, when the ON time ratio Don13 of the semiconductor switch element 13 is approximately 50%, the same power conversion can be performed at any voltage polarity regardless of the voltage polarity of the commercial power supply 1, so that the current of the commercial power supply 1 is as shown in FIG. As shown in FIG. However, if the current of the commercial power supply 1 is converted into a symmetrical sine wave, the converted power is limited only at this one point. In a microwave oven for home use, various heating powers are selected according to foods when heating with a microwave oven. For example, it is necessary to adjust the heating power by setting the ranges “strong”, “medium”, “weak”, and the like. In order to cope with this, the ON time ratio Do of the semiconductor switch element 13 is
It is necessary to change n13 in accordance with the desired output power. However, if an attempt is made to adjust the output power to a desired value with a constant on-time ratio Don13 regardless of the voltage polarity of the commercial power supply 1, the semiconductor switch shown in FIG. From the relationship between the on-time ratio Don13 of the element 13 and the conversion power P, the on-time ratio Don1
3 deviates from 50%, and the voltage of the commercial power supply 1 shows different current waveforms in the positive period and the negative period. If the control method is erroneous, a positive / negative current waveform as shown in FIG. In this case, since the current waveform does not become a symmetrical waveform, even-order harmonics are generated, so that the power factor cannot be improved after all.

【0022】そこで本実施例においては商用電源1の電
圧極性に応じて半導体スイッチ素子12、13の駆動信
号を入れ替えるように駆動回路22が動作するようにし
ている。すなわち図9に示すように商用電源1の電圧波
形V1(点線)の電圧極性が正の場合には昇圧チャージア
ップとインバータ動作を司る半導体スイッチ素子13の
オン時間比率(実線)Don13を高め、その逆極性時に
は半導体スイッチ素子13のオン時間比率Don13を
低める。また、商用電源電圧V1の正極時において、0
電圧から最大電圧に達するまでの谷間部では最も昇圧チ
ャージアップができるオン時間比率Don13とし、最
大電圧付近(ピーク部)では逆にオン時間比率Don13
を少し下げている。このように半導体スイッチ素子13
のオン時間比率Don13を変えることにより、歪みの
少ない入力電流を得ることが可能となり、容易にレンジ
“強”、“中”、“弱”などの加熱電力の調整が行え
る。ここで、負極性時に昇圧チャージアップ機能とイン
バータ機能を司ることとなるもう一方の半導体スイッチ
素子12は上記説明した半導体スイッチ素子13を追従
して相補的な動作となることは言うまでもない。
Therefore, in this embodiment, the drive circuit 22 operates so that the drive signals of the semiconductor switch elements 12 and 13 are switched according to the voltage polarity of the commercial power supply 1. That is, as shown in FIG. 9, when the voltage polarity of the voltage waveform V1 (dotted line) of the commercial power supply 1 is positive, the on-time ratio (solid line) Don13 of the semiconductor switch element 13 that controls the boosting charge-up and the inverter operation is increased. At the time of reverse polarity, the on-time ratio Don13 of the semiconductor switch element 13 is reduced. When the commercial power supply voltage V1 is positive,
In the valley from the voltage to the maximum voltage, the on-time ratio Don13 at which the boosting charge-up can be performed most is set. On the other hand, the on-time ratio Don13 near the maximum voltage (peak portion).
A little lower. Thus, the semiconductor switch element 13
By changing the on-time ratio Don13, an input current with less distortion can be obtained, and the heating power in the range “strong”, “medium”, “weak” can be easily adjusted. Here, it is needless to say that the other semiconductor switch element 12 which performs the boosting charge-up function and the inverter function at the time of the negative polarity follows the above-described semiconductor switch element 13 and performs a complementary operation.

【0023】この際、図9の商用電源電圧V1の極性が
入れ替わる変極点においてオン時間比率Don13は5
0%となっており、変極点部を詳細に示した図10のよ
うにVg13とVg12のオン時間T1とT2は等しく
なっている。この制御により、変極点において昇圧チャ
ージアップ機能とインバータ機能の両方を司る一方の半
導体スイッチ素子とインバータ機能のみを司るもう一方
の半導体スイッチ素子と役割交代をスムーズに行うこと
ができ、その結果、入力電流における変極点付近で発生
していた針状の歪みを抑えることができ、安定した入力
電流が得られる。
At this time, the on-time ratio Don13 is 5 at the inflection point where the polarity of the commercial power supply voltage V1 in FIG.
0%, and the on-time T1 and T2 of Vg13 and Vg12 are equal as shown in FIG. 10 showing the inflection point in detail. With this control, at the inflection point, the role can be smoothly switched between one semiconductor switch element that controls both the boost charge-up function and the inverter function and the other semiconductor switch element that controls only the inverter function. Needle-like distortion generated near the inflection point in the current can be suppressed, and a stable input current can be obtained.

【0024】また、全体としてもマグネトロン駆動用電
源の回路での発生損失を低減しつつマグネトロン駆動電
源の変換電力が変化して、半導体スイッチ素子13のオ
ン時間比率Don13が略50%の状態からいずれかの
方向へずれて変換電力が増減しても常に商用電源1の電
流波形は正負対称な正弦波状の波形を維持することが可
能となる。このため変換電力を変化させても常に高い電
力変換効率を維持したまま高力率で電流の歪の少ない動
作を実現することが可能となる。
In addition, the conversion power of the magnetron drive power supply changes while reducing the loss generated in the magnetron drive power supply circuit as a whole, and the on-time ratio Don13 of the semiconductor switch element 13 changes from approximately 50%. Even if the converted power increases or decreases in such a direction, the current waveform of the commercial power supply 1 can always maintain a sine-wave shape symmetrical in positive and negative directions. For this reason, even if the conversion power is changed, it is possible to realize an operation with a high power factor and little current distortion while always maintaining a high power conversion efficiency.

【0025】(実施例2)本発明の第2の実施例に用い
る回路図を図11に示す。図11に示す電源極性判定手
段23は商用電源1の電圧極性を判定し駆動回路22に
商用電源1が正の電圧極性か負の電圧極性かの信号を伝
達する。その一例として図12に示す通り、商用電源V
1が正極性のときは伝達信号V23をLOWレベル、負
極性の時はV23をHIGHレベルという具合にして判
別を行う。駆動回路22はこの判定信号に基づき半導体
スイッチ素子12、13のオン時間比率を入れ替えるよ
うに動作すると同時に図9で示した様に商用電源1の極
性に応じて各々のオン時間比率を変化させ、商用電源の
谷間部では昇圧チャージアップ機能を司る側の半導体ス
イッチ素子のオン時間比率を高め、逆にピーク部では低
める制御を行い、歪みの少ない入力電流波形I1を得
る。
(Embodiment 2) FIG. 11 shows a circuit diagram used in a second embodiment of the present invention. The power polarity determining means 23 shown in FIG. 11 determines the voltage polarity of the commercial power supply 1 and transmits a signal indicating whether the commercial power supply 1 has a positive voltage polarity or a negative voltage polarity to the drive circuit 22. As an example, as shown in FIG.
When 1 has a positive polarity, the transmission signal V23 is at a LOW level, and when it has a negative polarity, V23 is at a HIGH level. The drive circuit 22 operates to switch the on-time ratios of the semiconductor switch elements 12 and 13 based on the determination signal, and at the same time, changes each on-time ratio according to the polarity of the commercial power supply 1 as shown in FIG. In the valley portion of the commercial power supply, the on-time ratio of the semiconductor switch element that performs the boost charge-up function is increased, and conversely, control is performed to reduce the peak time in the peak portion, thereby obtaining an input current waveform I1 with less distortion.

【0026】この際、商用電源の極性は極性判定手段2
3により判別できるため、図13に示す通り、変極点と
なるZVPを検知した後、インバータ動作の一周期分だ
け同時オフの休止期間を設けてコンデンサの放電を十分
行い、各々の半導体スイッチ素子の役割を入れ替えるこ
とができる。この構成では変極点での各々の半導体スイ
ッチ素子のオンオフデューティ比は図13で示すように
Vg13とVg12のオン時間をT1≠T2としても変
極点における針状の入力電流歪みを抑制することが可能
となる。
At this time, the polarity of the commercial power supply is
Therefore, as shown in FIG. 13, after detecting ZVP as an inflection point, a pause period of simultaneous OFF for one cycle of the inverter operation is provided to sufficiently discharge the capacitors, and as shown in FIG. Roles can be swapped. In this configuration, the on / off duty ratio of each semiconductor switch element at the inflection point can suppress needle-like input current distortion at the inflection point even when the on time of Vg13 and Vg12 is T1 ≠ T2 as shown in FIG. Becomes

【0027】[0027]

【発明の効果】以上のように本発明の高周波加熱電源装
置によれば、昇圧チャージアップ機能とインバータ機能
とを司る一方の半導体スイッチ素子とインバータ機能の
みを司るもう一方の半導体スイッチ素子との役割交代が
商用交流電源の極性が入れ替わる度に必要となる回路構
成において発生していた変極点部の針状の入力電流歪み
を抑えることができ、安定した入力電流を得ることがで
きる。
As described above, according to the high-frequency heating power supply device of the present invention, the role of one semiconductor switch element that controls the boost charge-up function and the inverter function and the other semiconductor switch element that controls only the inverter function. Needle-like input current distortion at the inflection point, which has occurred in the circuit configuration required every time the polarity of the commercial AC power source is switched, can be suppressed, and a stable input current can be obtained.

【図面の簡単な説明】[Brief description of the drawings]

【図1】本発明の実施例1におけるマグネトロン駆動用
電源の動作説明用回路図
FIG. 1 is a circuit diagram illustrating the operation of a power supply for driving a magnetron according to a first embodiment of the present invention.

【図2】同マグネトロン駆動用電源の各動作モードでの
電流経路図
FIG. 2 is a current path diagram in each operation mode of the magnetron driving power supply.

【図3】同マグネトロン駆動用電源のインバータ回路の
動作波形図
FIG. 3 is an operation waveform diagram of the inverter circuit of the power supply for driving the magnetron.

【図4】同マグネトロン駆動用電源の動作波形図FIG. 4 is an operation waveform diagram of the magnetron driving power supply.

【図5】同マグネトロン駆動用電源の構成を示す回路図FIG. 5 is a circuit diagram showing a configuration of the magnetron driving power supply.

【図6】同マグネトロン駆動用電源の半導体スイッチ素
子の駆動信号波形図
FIG. 6 is a drive signal waveform diagram of a semiconductor switch element of the magnetron drive power supply.

【図7】同マグネトロン駆動用電源の半導体スイッチ素
子のオン時間比率Don13と変換電力Pのグラフ
FIG. 7 is a graph of the on-time ratio Don13 and the converted power P of the semiconductor switch element of the power supply for driving the magnetron.

【図8】同マグネトロン駆動用電源の商用電源電流波形
の一例を示す図
FIG. 8 is a diagram showing an example of a commercial power supply current waveform of the magnetron drive power supply.

【図9】同マグネトロン駆動用電源における商用電源と
半導体スイッチ素子オン時間比率の関係を示す図
FIG. 9 is a diagram showing a relationship between a commercial power supply and a semiconductor switch element on-time ratio in the magnetron driving power supply.

【図10】同マグネトロン駆動用電源における変極点付
近の半導体スイッチ素子の駆動信号波形図
FIG. 10 is a drive signal waveform diagram of a semiconductor switch element near an inflection point in the magnetron drive power supply.

【図11】本発明の実施例2におけるマグネトロン駆動
用電源の回路図
FIG. 11 is a circuit diagram of a magnetron driving power supply according to a second embodiment of the present invention.

【図12】同マグネトロン駆動用電源における電源極性
判定手段の出力波形図
FIG. 12 is an output waveform diagram of a power supply polarity determining means in the magnetron driving power supply.

【図13】同マグネトロン駆動用電源における変極点付
近の半導体スイッチ素子の駆動信号波形図
FIG. 13 is a drive signal waveform diagram of a semiconductor switch element near an inflection point in the magnetron drive power supply.

【図14】従来のマグネトロン駆動用電源を示す回路図FIG. 14 is a circuit diagram showing a conventional magnetron driving power supply.

【図15】同マグネトロン駆動用電源の動作波形図FIG. 15 is an operation waveform diagram of the magnetron driving power supply.

【図16】特開平10−271846号公報に公開され
た電源装置の回路図
FIG. 16 is a circuit diagram of a power supply disclosed in Japanese Patent Application Laid-Open No. 10-271846.

【符号の説明】[Explanation of symbols]

1 商用電源 7 高圧整流回路 8 マグネトロン 12 第1の半導体スイッチ素子 13 第2の半導体スイッチ素子 14 第1のダイオード 15 第2のダイオード 16 第1のコンデンサ 17 第2のコンデンサ 18 高圧トランス 22 駆動回路 23 電源極性判定手段 DESCRIPTION OF SYMBOLS 1 Commercial power supply 7 High voltage rectifier circuit 8 Magnetron 12 First semiconductor switch element 13 Second semiconductor switch element 14 First diode 15 Second diode 16 First capacitor 17 Second capacitor 18 High voltage transformer 22 Drive circuit 23 Power supply polarity determination means

───────────────────────────────────────────────────── フロントページの続き (72)発明者 石尾 嘉朗 大阪府門真市大字門真1006番地 松下電器 産業株式会社内 (72)発明者 北泉 武 大阪府門真市大字門真1006番地 松下電器 産業株式会社内 (72)発明者 末永 治雄 大阪府門真市大字門真1006番地 松下電器 産業株式会社内 Fターム(参考) 3K086 AA09 BA08 DB03 DB05 DB11 DB21 FA02 FA03 FA04 FA05 5H006 AA01 AA02 BB06 CA01 CB08 DA04 DB01  ──────────────────────────────────────────────────続 き Continued on front page (72) Inventor Yoshiro Ishio 1006 Kazuma Kadoma, Osaka Prefecture Matsushita Electric Industrial Co., Ltd. 72) Inventor Haruo Suenaga 1006 Kazuma Kadoma, Kazuma, Osaka Prefecture F-term in Matsushita Electric Industrial Co., Ltd.

Claims (2)

【特許請求の範囲】[Claims] 【請求項1】 第1および第2の逆導通可能な半導体ス
イッチ素子の直列接続体と、第1および第2のダイオー
ドの直列接続体を並列接続し、前記第1および第2のダ
イオードに各々並列に第1と第2のコンデンサを接続
し、前記第1および第2の逆導通可能な半導体スイッチ
素子の接続点と前記第1および第2のダイオードの接続
点との間に商用電源と高圧トランスの1次巻線の直列回
路とを接続し、前記高圧トランスの2次巻線の出力は高
圧整流回路を介してマグネトロンを駆動する構成とし、
前記第1および第2の逆導通可能な半導体スイッチ素子
のオンオフデューティ比を前記商用電源の極性が変わる
変極点付近で各々約50%とした高周波加熱電源装置。
1. A series connection of first and second semiconductor switches capable of reverse conduction and a series connection of first and second diodes are connected in parallel, and the first and second diodes are connected in series. A first and a second capacitor are connected in parallel, and a commercial power supply and a high voltage are connected between a connection point of the first and second semiconductor switches capable of reverse conduction and a connection point of the first and second diodes. A series circuit of a primary winding of a transformer is connected, and an output of a secondary winding of the high-voltage transformer is configured to drive a magnetron via a high-voltage rectifier circuit;
A high-frequency heating power supply device wherein each of the first and second reverse-conducting semiconductor switch elements has an on / off duty ratio of about 50% near an inflection point at which the polarity of the commercial power supply changes.
【請求項2】 第1および第2の逆導通可能な半導体ス
イッチ素子の直列接続体と、第1および第2のダイオー
ドの直列接続体を並列接続し、前記第1および第2のダ
イオードに各々並列に第1と第2のコンデンサを接続
し、前記第1および第2の逆導通可能な半導体スイッチ
素子の接続点と前記第1および第2のダイオードの接続
点との間に商用電源と高圧トランスの1次巻線の直列回
路とを接続し、前記高圧トランスの2次巻線の出力は高
圧整流回路を介してマグネトロンを駆動する構成とし、
前記商用電源の変極点付近の制御において極性判別手段
を持ちつつその変極点を検知することにより、昇圧チャ
ージアッブ機能とインバータ機能の両方の役割とインバ
ータ機能のみの役割とを相補的に同時に行う前記第1お
よび第2の逆導通可能な半導体スイッチ素子の役割を入
れ替える構成とした高周波加熱電源装置。
2. A series connection of a first and a second semiconductor switch element capable of reverse conduction and a series connection of a first and a second diode are connected in parallel, and the first and the second diodes are respectively connected to the first and the second diodes. A first and a second capacitor are connected in parallel, and a commercial power supply and a high voltage are connected between a connection point of the first and second semiconductor switches capable of reverse conduction and a connection point of the first and second diodes. A series circuit of a primary winding of a transformer is connected, and an output of a secondary winding of the high-voltage transformer is configured to drive a magnetron via a high-voltage rectifier circuit;
In the control near the inflection point of the commercial power supply, by detecting the inflection point while having the polarity discriminating means, the roles of both the boost charge-up function and the inverter function and the role of only the inverter function are simultaneously performed complementarily. A high-frequency heating power supply device having a configuration in which the roles of the first and second reversely conductive semiconductor switch elements are switched.
JP2001069964A 2000-09-27 2001-03-13 High frequency heating power supply Expired - Fee Related JP3501133B2 (en)

Priority Applications (8)

Application Number Priority Date Filing Date Title
JP2001069964A JP3501133B2 (en) 2001-03-13 2001-03-13 High frequency heating power supply
PCT/JP2001/008392 WO2002028149A2 (en) 2000-09-27 2001-09-26 Magnetron drive power supply
CNB018029264A CN1171505C (en) 2000-09-27 2001-09-26 Magnetron drive power supply
EP01972509A EP1254590B8 (en) 2000-09-27 2001-09-26 Magnetron drive power supply
US10/130,222 US6624579B2 (en) 2000-09-27 2001-09-26 Magnetron drive power supply
DE60104981T DE60104981T2 (en) 2000-09-27 2001-09-26 MAGNETRONSVERSORGUNGSSTEUERMITTEL
KR1020027006673A KR100766534B1 (en) 2000-09-27 2001-09-26 Magnetron drive power supply
CN01269659U CN2562101Y (en) 2000-09-27 2001-09-26 Magnetron driving power supply

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
JP2001069964A JP3501133B2 (en) 2001-03-13 2001-03-13 High frequency heating power supply

Publications (2)

Publication Number Publication Date
JP2002272118A true JP2002272118A (en) 2002-09-20
JP3501133B2 JP3501133B2 (en) 2004-03-02

Family

ID=18927918

Family Applications (1)

Application Number Title Priority Date Filing Date
JP2001069964A Expired - Fee Related JP3501133B2 (en) 2000-09-27 2001-03-13 High frequency heating power supply

Country Status (1)

Country Link
JP (1) JP3501133B2 (en)

Also Published As

Publication number Publication date
JP3501133B2 (en) 2004-03-02

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