JP2002208802A - Bandwidth variable filter circuit and high-frequency front-end circuit using the same - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a bandwidth variable filter for realizing the suppression of image signals, when received and the suppression of spurious when the image signal is transmitted, and a high-frequency front-end circuit that is small and operates with small power consumption. SOLUTION: This bandwidth variable filter circuit is for high frequency, including a plurality of switchable lines as a part of a transmission line, has a by-pass where each of the plurality of the switchable lines includes capacitors and/or dielectric resonators (DRO(dielectric resonator oscillator)), has a by-pass where the rest part of the transmission line includes DROs and/or capacitors, switches the characteristic values of the DROs and the capacitance of the capacitors, in parallel with the DROs in the by-pass of the transmission line by selecting one of the plurality of the switchable lines to construct the transmission line, and changes filter characteristics; and the front-end uses the bandwidth variable filter circuit.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a band variable filter circuit and a high frequency front end circuit using the same. More specifically, the present invention includes a band-variable BPF circuit capable of realizing a large band shift in a high-frequency region and a front-end for a portable terminal using the same.

[0002]

2. Description of the Related Art Communication in portable terminals such as portable telephones is performed by:
This is performed using a band of several hundred MHz to several GHz. In order to use frequency resources effectively, a multiplexing method of time division multiple access (TDMA), frequency division multiple access (FDMA) or code division multiple access (CDMA) is adopted, each of which includes various systems. In any case, the transmitting circuit converts the audio signal or the like into the intermediate frequency signal (I
In the receiving circuit, the signal is converted into a high-frequency signal (RF signal) via the F signal, and the portion from the antenna to the RF-IF converter is generally called a high-frequency front-end circuit or simply a front-end.

As an example of the front end, FIG. 7A schematically shows a front end configuration of a narrow band CDMA system currently used in Japan. As shown in the drawing, the signal wave is split into a reception wave (Rx) and a transmission wave (Tx) by a duplexer that functions as an antenna switch. On the reception side, a high-frequency signal (RF signal) is filtered by a reception filter ( SAW1), the signal is amplified by a low noise amplifier (LNA), the image signal is suppressed by an inter-stage receiving filter (SAW2), and then mixed with a local oscillation signal (LOC signal) by a frequency mixer (MIX). It is extracted as a signal (IF signal). On the transmission side, on the other hand, the modulated signal is a power amplifier (PA) and a directional coupler (CUP).
And antenna (ANT) via isolator (ISO)
Sent from. Transmission interstage filter (SAW3) is L
Removes harmonic components of OC and only PA
Filter to send to

[0004] The duplexer includes a band-pass filter (BPF) on the receiving side (Rx) and a band-stop filter (BEP) on the transmitting side (Tx). The duplexer also includes a low-pass filter (LPF) for suppressing harmonics (spurious) generated in the power amplifier (PA).

The narrow band CDMA system (cdm) used in Japan
aone), Narrowband CDMA system (US-PCS) operated in the United States
In, the Tx band and the Rx band are divided into a plurality of frequency bands. For this reason, a variable-bandwidth type duplexer can be applied to the terminal, and the size can be reduced as compared with a fixed-bandwidth type duplexer. Conventionally, a variable-bandwidth BPF used for this purpose
As a circuit, as shown in FIG. 26, a plurality of BPF circuits including a coaxial dielectric resonator (DRO) are provided on a bypass of the transmission line, and switches (SWa and SWb) on the transmission line and switches on the bypass are provided. There is known a configuration in which switches (SWc, SWd, SWe, SWf) are opened and closed to switch between BPFs (BPF-a and BPF-b in the figure). In the figure, BPF-b is separated and BP
Fa is selected. However, in the variable-bandwidth BPF circuit having this configuration, it is necessary to provide a large number of switches that are equal to or larger than the band to be switched. However, a low-loss switch (for example, a switch using a GaAs semiconductor) is expensive, and is generally sold in a resin-molded state instead of a bare chip. Therefore, if a BPF circuit as shown in the figure is to be constructed using a switch (GaAsSW) using a GaAs semiconductor, a large number of
GaAsSW must be mounted, which increases the manufacturing cost and makes it difficult to reduce the size of the device.

FIG. 27 shows a specific configuration of a conventional example different from that of FIG. The BPF shown in FIG. 27A is composed of a coaxial dielectric resonator DROa and DROb connected to a coupling capacitor C0.
, And the inner conductor of DROa is connected to the transmission line via a capacitor C1 and a PIN diode D1.
A path is provided to connect the inner conductor of Ob to GND via the capacitor C2 and the PIN diode D2 (the outer conductor of DRO is also connected to GND).
D1 and D2 function as switches by applying a DC voltage to their terminals. Turning on each diode (FIG. 27 (b)) adds a capacitance in parallel with DRO and shifts the frequency to a lower frequency range (for example, Japanese Patent Application Laid-Open No. 7-58597 discloses a structure similar to this). , JP-A-8-172
333, JP-A-11-46102, JP-A-11-243304, etc.). However, in this variable-type BPF circuit, as the center frequency of the BPF departs from the resonance frequency of the DRO, deterioration in characteristics is inevitable.

Japanese Unexamined Patent Publication No. 2000-68703 discloses a band-variable BPF corresponding to the above-mentioned problem of the conventional example as shown in FIG.
Has been proposed. This is because the metallized portion (outer conductor) on the outer surface of the coaxial dielectric resonator (DRO) is separated from the others by a small area near the opening, and the main area M
And a small area m are connected by a PIN diode (PIN). When the PIN diode is on, that is, when there is no separation between M and m, the electromagnetic field distribution inside the DRO is as shown in FIG. 29 (a), but when the PIN diode is off, that is, when M and m Are separated as shown in FIG. 29B, and the center frequency of the BPF shifts to the higher frequency side. However, BP by this method
F has a complicated manufacturing process. Further, as is clear from FIG. 29B, when the small area portion m is widened to increase the shift amount, the distortion of the electromagnetic field distribution inside the DRO increases, and deterioration of the characteristics is inevitable.

As described above, in any of the conventional examples, when the amount of band shift is large, the characteristic is significantly deteriorated. For this reason, it is not possible to obtain a product in which the desired shift amount is realized and the bandwidth, the out-of-band attenuation, and other high-frequency characteristics are maintained at a satisfactory level. Is about several tens of MHz.

[0009] Further, regardless of CDMA, TDMA, or FDMA, in any of the conventional front-end configurations, it is necessary to provide an LPF between the antenna and the front end in order to suppress spurious noise, and the insertion loss thereof. In consideration of the above, there is a limit in reducing power consumption during transmission. Further, in FIG. 7A, in addition to SAW1, an interstage filter S
As in the case of using AW2, it is necessary to use a plurality of filters for suppressing the image signal. For this reason, in the configuration of the conventional front end, there is a limit in miniaturization and power saving of a product, and a high-frequency front end capable of miniaturization, power saving, and cost reduction is desired. Filter circuit technology that can accurately control a larger frequency shift rate and bandwidth, out-of-band attenuation, and other high-frequency characteristics is indispensable.

[0010]

SUMMARY OF THE INVENTION An object of the present invention is to solve the above-mentioned problems in the prior art. Even if the center frequency is largely shifted, the out-of-band attenuation and other high-frequency characteristics do not deteriorate much. It is another object of the present invention to provide a high-frequency band variable filter circuit which is inexpensive, easily miniaturized, and excellent in practical use. In addition, the present invention realizes downsizing of a product and improvement of characteristics by simplification of a configuration, and reduction of manufacturing cost and power consumption by applying the above-described high-frequency band variable filter circuit to various mobile phone systems. Provide a front-end circuit.

[0011]

As described above, in the variable-bandwidth BPF shown in FIG. 27, when the switch provided in parallel with the DRO is on, the center frequency of the BPF deviates from the resonance frequency of the DRO. Functions as a BPF. However, when the center frequency is shifted greatly (for example, about several hundred MHz or more), that is, when the difference between the resonance frequency of the DRO and the center frequency of the BPF is increased, the BPF
The characteristics deteriorate significantly. For this reason, the band shift amount of the band variable BPF shown in FIG. 27 is limited to several tens of MHz, and the band variable BPF that switches the parallel capacitance between a plurality of values to realize band switching is practical. It was not thought to be something.

However, according to the study of the present inventors, B
The PF characteristic is strongly affected by the inductance of the DRO terminal, and it has been found that the effect is particularly large when the difference between the resonance frequency of the DRO and the center frequency of the BPF is large. That is, D having the conventional terminal structure
In the RO, for example, as shown in FIG. 28, a terminal is drawn out from an internal conductor in an L-shape and connected to a transmission line. This is the same in the prior art described with reference to FIGS. Such a terminal structure provides an inductance on the bypass.

By adopting a configuration in which the terminal inductance of the DRO is shifted from the side path to the transmission path, the present inventors have realized that even if the center frequency shift amount is increased, the insertion loss is small and the out-of-band attenuation amount is further reduced. Without deteriorating high-frequency characteristics such as
Further, it is possible to accurately design the pass band and the harmonic characteristics of the BPF after the band shift, and as a result, the DRO
High-performance band-variable BPF by controlling the capacitance of each stage of the filter circuit by using a variable capacitor.
Was found to be realizable.

Further, this band variable type BPF is provided with (a) GH
It is possible to shift the band up to the z-order, and (b) it is possible to suppress the image signal in one stage due to its high steepness.
(c) Since harmonic-free characteristics can be obtained by adjusting the capacitance of each stage, spurious (unwanted waves) from the transmitting power amplifier can be suppressed without providing an LPF. Therefore, the BPF having this feature
By using for mobile terminals, the front end configuration can be greatly simplified and power consumption can be reduced.

The present invention is based on these findings and provides the following band variable filter circuit and a front end for a portable terminal using the same.

(1) A high-frequency band variable filter including a plurality of switchable lines as a part of a transmission line, wherein (a) each of the plurality of switchable lines includes a bypass including a capacitor. The remaining portion of the transmission line has a bypass including a dielectric resonator (DRO);
(B) each of the plurality of switchable lines has a bypass including a dielectric resonator (DRO), and the remaining part of the transmission line has a bypass including a capacitor; or (c).
Each of the plurality of switchable lines has a bypass including a dielectric resonator (DRO) and a bypass including a capacitor, and one of the plurality of switchable lines is selected to form a transmission line. A variable band filter circuit characterized in that by changing the filter characteristic, the characteristic value of DRO in the bypass of the transmission line and / or the capacitance of a capacitor in parallel with the DRO are switched.

(2) In any one of (a) and (b), at least one of the plurality of switchable lines includes a side path including DRO and a capacitor in addition to the side path including the capacitor or DRO. 2. The variable-bandwidth filter circuit according to 1 above, wherein the band variable filter circuit has one or more sets of side paths including: (3) The terminal described in the above item 1 or 2, wherein the terminal of the DRO has a two-branch structure from the end of the DRO, and the two ends generated by the branch are inserted into the transmission line to directly connect the DRO to the transmission line. Variable bandwidth filter circuit.

(4) The capacitor as described in (1), wherein the capacitor means connected in parallel with the DRO has a capacity for shifting the center frequency of the filter to a low frequency side of 40 MHz or more.
4. The variable-bandwidth filter circuit according to any one of claims 1 to 3. (5) The non-harmonic band variable filter circuit according to any one of 1 to 4, wherein the input admittance characteristics of the DRO and the bypass capacitor in each bypass are selected so as to cancel out the harmonic characteristics.

(6) A high-frequency front-end circuit characterized in that the band limiting means near the input / output section to the antenna includes any of the band variable filters (1) to (5). (7) The high-frequency wave according to the above (6), wherein the band limiting means near the input / output unit to the antenna includes a duplexer having an image signal suppressing function and a spurious suppressing function constituted by any of the band variable BPFs of the above 1 to 5. Front end circuit.

(8) The band limiting means in the vicinity of the input / output portion to / from the antenna is constituted by any of the band variable BPFs 1 to 5, whereby a plurality of transmission frequency bands in the same signal multiplexing system are provided. 7. The high-frequency front-end circuit according to 6, wherein the switching is performed in each of the transmission frequency bands in a plurality of signal multiplexing systems, and the spurious during transmission is substantially suppressed in any of the bands. .

(9) The band limiting means in the vicinity of the input / output portion to / from the antenna is constituted by any one of the band variable BPFs 1 to 5 described above. 7. The high-frequency front end according to 6, wherein switching is performed between the respective reception frequency bands in a plurality of signal multiplexing systems, and the image signal at the time of reception is substantially suppressed in any band. circuit. (10) The high-frequency front-end circuit according to any one of the above items 6 to 9, which is used for a multi-band mobile phone that switches and uses a plurality of bands or a mobile phone that includes a plurality of signal multiplexing systems.

[0022]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS (A) Band variable filter circuit The configuration of the band variable filter circuit of the present invention will be described in detail below. In the following description, a variable-bandwidth BPF circuit will be described as an example. However, a variable-bandwidth BEF circuit may be used by changing the circuit configuration of the BPF to, for example, a BEF circuit configuration.

FIGS. 1 to 3 show the basic structure of a band variable BPF circuit according to the present invention. As shown in the figure, this BPF circuit is a filter circuit having, on a transmission line, one or more sets of a bypass including a dielectric resonator (DRO) and a bypass including a capacitor, and as a part of the transmission line. It is a circuit having a plurality of switchable lines. Any one of these lines is selected and becomes a path for transmission.

In the example shown in FIGS. 1 to 3, lines A to P are provided. At the branch points of these lines, a path switching means SWF is provided on the input side, and a path switching means SWL is provided on the output side, and a line from the input end to the SWF and a line from the output end to the SWL (hereinafter referred to as these). Collectively referred to as a common line) and any one of the switching lines.
And SWL, a path is formed. For example, when the transmission lines at the input and output terminals are connected to the line A by SWF and SWL, a path passing through the line A is formed. The same applies to the line B and the subsequent lines. By making the filter characteristics different in each path, the filter characteristics can be switched by path switching. The number of lines is not limited as long as it is two or more, and depends on the required number of switching stages.

The common line and the plurality of switchable lines must satisfy one of the following conditions. (A) Each of the plurality of switchable lines has a bypass including a capacitor, and the common line has a bypass including a dielectric resonator (DRO). (B) Each of the plurality of switchable lines has a bypass including a dielectric resonator (DRO), and the common line has a bypass including a capacitor. (C) Each of the plurality of switchable lines has a bypass including a dielectric resonator (DRO) and a bypass including a capacitor.

FIG. 1 is a diagram showing the above mode (a). Each line is a capacitor [for example, line A has C
A0 or a capacitor represented by CA (α + 1)], and has a DROF at the input end and a DROL at the output end of the common line.

When the line A is selected and becomes the path of the transmission line, when the transmission line is traced from the input terminal to the output terminal, the high frequency passing through the CCF is formed by first connecting DROF and CA0 in parallel. Via the first-stage BPF. Thereafter, according to the value (integer value) of the subscript α in the figure, if α = 0, the final stage BPF and CCL formed by connecting CA1 and DROL in parallel via the coupling capacitor CCA1 Through the output end. If α> 0, an intermediate-stage BPF composed of a parallel capacitor and a DRO surrounded by a broken line in the figure passes between the first-stage BPF and the last-stage BPF.

The number of BPF stages in the path A is (α + 2)
In the figure, each time the value of α increases by 1, B surrounded by a broken line
The PF unit increases by one. The same applies to the case of passing through the lines B to P, and β,..., Π DRO +
By connecting a capacitor, (β + 2) stages,
, A path including (π + 2) stages of BPFs. α, β,
.., Π and the like are integers of 0 or more, which are determined for each line according to the purpose, and are preferably 1 or more. The values may be different from each other, or any combination of two or more may be equal. All may have the same value. Increasing the number of filter stages per line improves the steepness of the BPF.

FIGS. 2 and 3 show the above (b),
It is the figure which showed the aspect of (c). D for DRO-BPF
Since there is a limit in reducing the size of the RO, usually, the mode (a) in which the number of applied DROs is the smallest is preferable. In the above description, the example of the band variable BPF circuit in which the capacitance is provided corresponding to all the bypasses including the DRO in any of the paths has been described. May be provided with a parallel capacitor.

Although the parallel capacitor depends on the use of the filter, it is generally 0.5 pF or more, preferably 1 pF or more.
More preferably, it is about 3 pF or more. Below the lower limit,
The shift amount of the center frequency of the BPF is small (GHz
(At most about ten and several MHz in the band), the meaning of lower frequency is poor. Further, the effect of improving the out-of-band attenuation described below cannot be sufficiently obtained. The upper limit is not particularly limited. Basically 4
0 MHz or more, preferably 100 MHz or more, more preferably 200 MHz or more, still more preferably 500 MH
The effect of the present invention is sufficiently exhibited by the capacitance value that realizes the shift amount of z or more. Although the individual parallel capacitors may be formed in any manner, in order to more effectively obtain the effects of the present invention,
Provided as a separate element outside the DRO. For example, DR
The capacitor is mounted on the O-mounted substrate as a capacitor, or the DRO-mounted substrate is applied with a multilayer substrate for a multilayer capacitor to make the capacitor an inner layer.

The switch includes any means capable of short-circuiting / opening the path in a direct current manner.
Although an IN diode or the like may be used, a low-loss GaAs switch is preferable in order to minimize the insertion loss of the filter. Although not shown in the figure, wiring and control means necessary for driving the switch can use a conventional method. The control means is, for example, a control IC for a mobile phone.
It may be incorporated in In addition, SWL is omitted and lines A to P
May be connected to the output side. Even in such a configuration, when the line A is selected by the SWF, the branch point SWL
, And line P, the input impedance becomes infinite because SWF is OFF for these lines. Therefore, at the branch point SWL, the current does not flow through the lines B,... However, at high frequencies, the current may flow around the lines B,... This effect can be eliminated by providing the switch SWL.

When performing a band shift of several hundred MHz or more using the circuit of FIG. 1, it is necessary to transfer the inductance of the terminal of the DRO to the transmission line. To move the terminal inductance of the DRO from the bypass to the transmission line, as shown schematically in FIG.
A branch (split) structure is used, and the DRO is directly connected to the transmission line by inserting the branch end into the transmission line.

The influence of the inductance of the terminal portion is very sharp, and even a minute amount of about 0.1 [nH] affects the characteristics. Even a gold wire for wire bonding having a diameter of several tens of μm has an inductance of about 1 [nH / mm]. By designing the circuit at the position of the bypass, the filter characteristics, especially the insertion loss, increase significantly. In the present invention, by moving the DRO terminal position from the side path to the transmission line, the problem of characteristic deterioration is solved even when the difference between the center frequency of the BPF and the resonance frequency of the DRO is large. Therefore, a variable-bandwidth BPF capable of shifting the band to the GHz order is realized.

In addition, since it is difficult to evaluate the influence of the terminal inductance of the DRO, in the conventional band variable BPF using the conventional terminal DRO, when a plurality of parallel capacitors are combined, the characteristic after the center frequency shift is reduced. Although it is extremely difficult to predict and its optimization and precise control are practically impossible, the present invention virtually eliminates the influence of the terminal inductance L, and as a result, corrects circuit design and band shift etc. Adjustments are possible.

For this reason, the variable-bandwidth BPF of the present invention
It is possible to perform elaborate band switching control in combination with appropriate control means. More specifically, by using a DRO and a parallel capacitor having a large out-of-band attenuation and different specifications in each stage, it is possible to obtain characteristics that cannot be realized by a conventional band-variable BPF using DROs having the same specifications. For example, suppression of spurious is realized by increasing the harmonics or adjusting the admittance of each stage of the BPF.

The higher frequency of the harmonic can be obtained by the BP used in the present invention.
This is a phenomenon peculiar to the F circuit, and the ratio of the harmonic frequency F0 'to the center frequency F0 of the BPF is increased by increasing the capacity of the parallel capacitor. That is, by using a large-capacity parallel capacitor, the harmonics are made higher in frequency. The frequency of the spurious to be suppressed by the power amplifier is F
s2, Fs3, Fs4,..., usually Fsx (x
= 3) is sufficient, so that Fsx <F0 '
What is necessary is just to satisfy the conditions which satisfy.

Suppression of spurious can also be performed by adjusting admittance of each stage of the BPF. That is, the DRO is set so that harmonics cancel each other at each stage of the filter.
In addition, by using the parallel capacitor, if a harmonic characteristic at a frequency equal to or lower than Fsx is canceled, an antenna filter having substantially no harmonics can be obtained. Adjustment of the out-of-band attenuation and improvement of the steepness of the filter can be similarly realized.

Although the capacitances of the DRO and the parallel capacitor are determined by the above conditions, it is preferable to further add a condition for imparting symmetry to the circuit. Specifically, in the circuit shown in FIG.
= CcL, CcA1 = CcA (α + 1), CcA2 = CcAα, C
cA3 = CcA (α-1) etc. [In the general formula, CcAk = CcA (α + 2-
k)], and for the parallel capacitor, CA0 = CA (α +
1), CA1 = CAα, CA2 = CA (α-1), etc.
k = Cm + 1-k], a symmetrical circuit configuration is obtained (where k is an integer of 1 or more and (α + 1) / 2 or less in an odd-numbered circuit, and 1 or more and α in an even-numbered circuit). / 2 or less). Track B
The same applies to .about.P. The BPF that satisfies this condition has the effect of greatly improving the characteristics. If DROs having different specifications are used for each stage, the pole frequency can be arbitrarily designed. Therefore, the adjustment of the attenuation outside the band and the improvement of the steepness of the filter can be realized.

Further, an additional circuit may be provided for the purpose of improving steepness such as adding a pole to the circuit or for other purposes. Such a circuit is usually provided with a circuit including L or C or L and C in series or in parallel at the end of the filter circuit, or a circuit including a parallel circuit of L and C in the transmission line. It is realized by adding. A corresponding distributed constant type additional circuit may be used. If the filter circuit has a balanced circuit configuration, which will be described later, the additional circuit is also a balanced type.

An example of a typical additional circuit is shown in FIG.
In the drawing, upper and lower horizontal lines represent the transmission lines in FIG. 1 and the like. However, in the unbalanced type, one is generally GND.
For example, (A-1) is an unbalanced and distributed constant type additional circuit with open and short-circuited DROs on the bypass, and (A-2) is an unbalanced type with a capacitor C and inductor L on the bypass. And a lumped constant type additional circuit, (A-3) is a balanced and distributed constant type additional circuit having an open and short parallel plate type DRO on the bypass, (A-3)
Reference numeral -4) denotes a balanced and lumped constant type additional circuit provided with a capacitor C and an inductor L on a bypass. The same applies to (B), (C), and (E), and in (D), a DRO or the like is provided on the transmission line. (F) and (G) are useful for adjusting the characteristic impedance. In the present invention, any of them can be used. Conventionally, these circuits are provided at one end of the filter circuit. However, in the present invention, it is preferable to provide additional circuits having substantially the same configuration at both ends of the filter circuit.

In the present invention, DRO may be square or cylindrical. A parallel plate type dielectric resonator described later may be used. The line ends may be short-circuited or open, and the number of resonant lines per DRO is not limited. It also includes a distributed constant line through which TEM waves propagate. For example, a microstrip line may be used. When a microstrip line is used, electrode pads are provided on both sides of the end of the dielectric line, and these are used as a part of the transmission line, whereby the terminal inductance of the DRO can be transferred from the side line to the transmission line.

In the present invention, a structure in which a dielectric is sandwiched between two parallel conductor plates is useful as a dielectric resonator.
It may be either an open-end type in which the bipolar plates are insulated (in a DC manner) or a short-circuit type in which the plates are connected. Conventionally, such a structure
It is not used as a dielectric resonator. The parallel plate dielectric resonator (referred to as “parallel plate dielectric resonator” in this specification) has the following features.

First, the parallel plate type dielectric resonator can easily obtain a large characteristic impedance and can improve the steepness of the filter. That is, the steepness of the filter circuit can be improved by increasing Z0, but the practical characteristic impedance Z0 of the coaxial type DRO is 10Ω or less, and there is a limit to the characteristic improvement. Even in the coaxial DRO, Z0 can be reduced to 10 by increasing the outer diameter or decreasing the inner diameter.
Although it is not impossible to make it larger than Ω, increasing the outer diameter contradicts the demand for miniaturization, while reducing the inner diameter poses problems in manufacturing technology and cost. Therefore, in reality, D
There is a limit to the improvement in filter steepness due to RO Z0. In contrast, Z0 of 10Ω or more can be easily obtained with the parallel plate type dielectric resonator. If a dielectric having a dielectric constant of about 40 is used, 40Ω or more is possible.

Second, the parallel plate type dielectric resonator can be manufactured simply by cutting a dielectric substrate having electrodes provided on both sides into appropriate dimensions, and thus mass production is easy. Also, no characteristic adjustment is required. Furthermore, terminals can be dispensed with because they can be mounted directly on the substrate.

Third, by using a parallel plate type dielectric resonator, the filter circuit can be of a balanced type. That is, when two high-frequency lines or devices are generally connected, not only the characteristic impedance but also the distribution of the electromagnetic field at the connection point must be the same. At present, a balanced power amplifier, a low noise amplifier (LNA), and a mixer that are driven by a low voltage and realize a wide dynamic range and a high gain are being put into practical use. By balancing high-frequency circuits (transmitter / receiver for mobile phones, etc.), a circuit configuration that does not require an MMIC such as a power switch and a DC / DC converter for generating a negative voltage is possible.
The mobile phone transmitter / receiver can be significantly reduced in size and price. Balanced high-frequency circuits are promising as the next-generation high-frequency circuit technology, and the BPF applied here also needs to be balanced. However, a balanced SAW filter is difficult and expensive to design. It is also possible to convert the electromagnetic field mode of an unbalanced SAW filter to a balanced type by applying a balanced-unbalanced conversion circuit and apply it to a balanced high-frequency circuit. Is also used. Therefore, this method hinders downsizing and cost reduction of the balanced high-frequency circuit, and becomes less practical.

The solution is to balance the entire circuit configuration. FIG. 6 shows an example of a balanced band variable DRO-BPF circuit. In the figure, in the example of two-stage switching between the line A and the line B, an example in which the switch unit is provided only on the input end side is shown. The portion indicated by the broken line in FIG. 6A is the line B in FIG. When the two contacts on the input end side are connected to the terminals 1a, 1'a, 2a, 2'a of the line A, the line A becomes a path and is connected to the terminals 1b, 1'b, 2b, 2'b. In the state, the line B is a path.

In order to change the circuit from the unbalanced type to the balanced type, the unbalanced type DRO is replaced with a DRO having a balanced type structure such as a parallel plate type DRO so that the transmission line is the same in the forward path and the return path. Is provided with a circuit element. In addition, several tens of MHz
In order to set the band shift amount to exceed
It is necessary to transfer the inductance of the terminal of the RO to the transmission line. The method is the same as described above.

Variable band balanced DRO-BP according to the present invention
In the F circuit, it is possible to form band variable filters that can be switched in multiple stages by the number of switching lines. In this case, as understood from FIG. 6, the number of switches is only four regardless of the number of switching stages and the number of BPF stages. On the other hand, FIG.
In the conventional band variable filter as shown in (1), the number of switches increases as the number of stages of the BPF increases. Therefore, the variable band balanced DRO-BPF circuit according to the present invention has an advantage over the prior art in manufacturing cost and ease of miniaturization.

The dimensions of the parallel plate type dielectric resonator are not particularly limited. However, in order to reduce the size of the entire filter, the thickness (distance between electrode plates) is 1 mm or less, the width is 1 to 10 mm, and the length is 1 to 1.
About 10 mm, preferably 0.5 mm or less in thickness and 1 to 1 in width
5 mm and a length of about 1 to 5 mm are preferable. The width of the parallel plate type dielectric resonator may be continuously changed in the length direction. Thereby, a unique DRO in which the characteristic impedance Z0 continuously changes in the width direction is obtained. By using a parallel-plate dielectric resonator of a continuously changing impedance type, it is possible to suppress harmonics.

As the dielectric material and the electrode material, the materials used in the conventional coaxial DRO can be used.
Further, the application of the electrode to the dielectric substrate can be performed by an existing method of applying a conductor layer such as printing a thick film or forming a thin film.

Although the variable band BPF of the present invention has been described in connection with the portable terminal, the variable band BPF of the present invention is effective in the range of several hundreds of megahertz to several tens of GHz. Not only to the front end, but also useful as a high-frequency circuit or an intermediate frequency band generally as a band variable BPF.

(B) High-Frequency Front-End Circuit As described above, a band-variable BPF with no harmonics can be realized by using the band-variable filter circuit of the present invention. For this reason, the present invention provides an unconventional configuration as a high-frequency front end, particularly a front end for a portable terminal.

For example, FIG. 7A schematically shows a conventional front end in a narrow band CDMA system. As described in detail in the related art section of this specification, in such a front end, an LPF is indispensable in front of the antenna as viewed from the transmitting side in order to remove spurious components generated in the power amplifier (PA). For this reason, the insertion loss becomes 3 dB or more as a whole, which is a great limitation in reducing the power consumption during transmission. However, according to the present invention, such a LPF becomes unnecessary by using a band variable BPF having a non-harmonic characteristic. Further, according to the present invention, it is possible to suppress the image signal at the time of reception, and the SAW1 shown in FIG. 7A becomes unnecessary. Further, according to the present invention, it is possible to suppress the image signal by 70 dB or more. In such a configuration, the SA in FIG.
W2 is also unnecessary.

As described above, according to the present invention, the spurious is removed by the band-variable BPF (indicated by V-BPF in the figure), so that the high-frequency front end which eliminates the spurious removing LPF in the conventional configuration is unnecessary. Is provided. An example of such a front end configuration is schematically shown in FIG. In the present invention, a configuration including or not including the SAW2 in FIG. 7B is also included in the present invention (hereinafter, FIGS. 8B to 8B).
Same in FIG. ).

FIG. 7B shows a comparison with FIG. 7A to clarify the difference from the prior art, and the front end according to the present invention is a front end constituted by the blocks shown here. It is not limited to only the end. Similarly to this example, a high-frequency front end in which a spurious component removing LPF was essential in the conventional configuration is a front end that can be configured by removing the spurious component removing LPF and replacing the BEF with the band variable BPF of the present invention. For example, any front end after the change is included in the scope of the present invention.

Examples of such a front end circuit include a CDMA system such as ADC, US-PCS and K-PCS, GSM, EGSM, PCN, AMPS, DCS, P
A front-end of a TDMA type mobile phone such as DC800 or PDC1500 may be used. Also, W-CDMA
In wideband CDMA such as CDMA and cdma2000, the front-end circuit of the present invention is particularly effective.

Further, according to the present invention, the band limiting means in the vicinity of the input / output portion to / from the antenna is provided with the band variable B of the present invention.
There is provided a high-frequency front end including a duplexer having a function of suppressing an image signal and a function of suppressing a spurious component constituted by a PF.

FIG. 8A shows a conventional technique in a single mode TDMA, and FIG. 8B schematically shows an application example of the present invention. In conventional TDMA, transmission and reception are switched by an antenna switch, but it is necessary to remove noise generated at the time of switching to prevent radiation from the antenna. In addition, a SAW filter is generally applied to the receiving filter, but the allowable power is small, and the antenna is unprotected against spike noise such as static electricity flowing from the antenna when the antenna switch is switched to the receiving side. However, according to the present invention, by configuring an antenna filter using the above-mentioned band-variable BPF, it is possible to block spike noise and protect the transceiver. Since it is not necessary to transmit and receive at the same time in the TDMA system, the application of the band variable BPF may be single. In many mobile telephone systems, each of the transmission wave and the reception wave is an independent frequency band, and the frequency bandwidth used for transmission and reception is a wide band of 8% or more in a fractional band. With the filter according to the state of the art, the steepness deteriorates significantly, and practical steepness cannot be obtained.

Further, the present invention is also useful as a front end for a multi-mode system, which is a combination of these systems, and a portable telephone system in which these systems are combined with PHS or DECT. For example, according to the present invention, the band limiting means in the vicinity of the input / output unit to the antenna is constituted by the band variable BPF of the present invention, whereby switching between a plurality of transmission frequency bands in the same signal multiplexing system is performed. In addition, there is provided a high-frequency front end that switches between transmission frequency bands in a plurality of signal multiplexing schemes and that substantially suppresses spurious at the time of transmission in any band.

Further, according to the present invention, the band limiting means in the vicinity of the input / output unit to the antenna is constituted by the band variable BPF of the present invention, whereby a plurality of reception frequency bands in the same signal multiplexing system are provided. A high-frequency front end is provided that performs not only switching at the same time, but also switches between respective reception frequency bands in a plurality of signal multiplexing systems, and substantially suppresses image signals at the time of reception in any band. Is done.

FIG. 9 shows a front-end configuration for a dual-mode mobile phone in which two TDMA systems such as GSM and DCS are combined. As shown in the upper section, a band variable BPF suitable for each type of communication may be provided,
A band variable BPF may be provided on each of the transmission side (Tx) and the reception side (Rx) to perform switching between systems (this is not shown). Further, in the present invention, a large shift width can be realized by the band variable BPF, so that the dual mode can be realized by the front end of the single BPF configuration shown in the lower part. In this case, the band used for transmission and reception of the two systems is switched by the band variable BPF.

FIG. 10 shows an example of a front end configuration for a triple mode. GSM / DCS /
It is a combination of US-PCS. A configuration in which one of these is replaced with W-CDMA is also possible. Since transmission and reception are performed simultaneously in the CDMA system, in the case of a multi-mode portable device including the CDMA system, two or more band variable BPFs are required. In the conventional example, in such a case, at least a duplexer is required for each CDMA system. However, according to the present invention, since two variable BPFs can include all of them, the configuration becomes extremely simple.

Further, as shown in an embodiment described later, the band variable BPF of the present invention can suppress an image signal by 70 dB or more. This eliminates the need for an interstage filter (SAW2 in FIGS. 7 and 8) on the receiving side. The noise figure (NF) of the entire receiver greatly depends on the noise figure of the first-stage LNA. In receivers for mobile phones, the first stage LNA has H
EMT is being adopted. In this case, NF is 1.4d
B can be reduced to 0.7 dB or less. However, a large insertion loss of the receiving filter and the receiving interstage filter impairs the NF improvement effect. In particular, the first stage L
Reducing the loss from the antenna to the receiving filter together with improving the NF of the NA is effective for improving the NF of the entire receiver. The omission of the interstage filter for reception is effective in reducing signal distortion due to the reduction in gain.

FIGS. 7 to 10 show MIX up to the front end configuration, but the present invention is also applicable to direct conversion (and low IF system) that does not require MIX. In the present invention, since the interstage filter for reception can be omitted, improvement in reception sensitivity can be expected. For this reason, direct conversion can be suitably applied.

[0065]

The present invention will be described more specifically with reference to the following examples. In each of the following examples, DR
A circuit in which the terminal inductance of O is transferred to the transmission line is used.

Embodiment 1 Two-stage DRO-BPF to which DRO of the same specification is applied
1 shows a configuration example of DCS Tx / DCS Rx according to the first embodiment. In FIG. 11, the capacitance of the capacitor and the size of the DRO are set to the values shown in the following table. The switch SW is connected to the line A (upper side in the figure) at the time of transmission (Tx), and the line B is connected at the time of reception (Rx).
(Lower side in the figure) to switch the center frequency. Note that CCL = CCF = 1.57 [pF].

[0067]

[Table 1]

The characteristics of this variable band (switching) type DRO-BPF are shown in FIG. 12 [(a) passband characteristics, (b) delay characteristics, (c) standing wave ratio, (d) harmonic characteristics]. . A thick line corresponds to DCS Tx, and a thin line corresponds to DCS Rx. The DCS (see FIG. 8 for the front-end configuration) has a frequency band of the received wave of 1805 to 1880M.
Hz and a lower heterodyne system with an intermediate frequency of 246 MHz, and the image signal appears at 1313 to 1388 MHz.

However, as shown in FIGS.
In the two-stage switching DRO-BPF of the present invention, the attenuation in each band is about 1 dB, the delay in the band is less than 3 to 4 ns, and the attenuation of the image signal is 34.5 d.
B, 3F0 attenuation is 58.7 dB, which is useful as a duplexer. Using this duplexer, FIG. 8 (b)
Of the conventional DCS
It is also possible to integrate the Rx filter (SAW1 filter) and the Tx filter (LPF) used in the front end in the above, and a circuit configuration smaller and consuming less power than conventional products can be realized.

Embodiment 2 DRO-BPF of Two-Stage Configuration Applying DRO with Same Specifications
1 shows a configuration example of GSM Tx / DCS Tx according to the first embodiment. In FIG. 11, the capacitor capacity, the DRO dimension, and the like are set to the values shown in the following table. The switch SW is connected to the line A (upper side in the figure) as GSM Tx, and the switch SW is connected to the line B (upper side in FIG. (Lower side) to switch the center frequency. Note that CCL = CCF =
It was set to 2.5 [pF].

[0071]

[Table 2]

The characteristic characteristics of the variable band (switching) type DRO-BPF are shown in FIG. 13 [(a) pass band characteristic, (b) delay characteristic,
(c) Standing wave ratio, (d) Harmonic characteristics]. Thick line is GSM T
x and a thin line correspond to DCS Tx. In GSM, the frequency band of the transmission wave is 890 to 915 MHz, and the two-stage switching D of the present invention is used.
In the RO-BPF, the band shift is 1747.5 MHz (DCS)
$ 902.5 MHz (GSM). The 3F0 attenuation is
In GSM-Tx, it is 53.1 dB, and in DCS-Tx, it is 55.2 dB. In any case, it is possible to eliminate the need for an LPF for spurious suppression, and it is possible to reduce the size and power consumption of the front end compared to conventional products. Circuit configuration can be realized.

Embodiment 3 DRO-BPF of Three-Stage Configuration Applying DRO of the Same Specification
1 shows a configuration example of DCS Rx / WCDMA according to the first embodiment. FIG.
In DRO, the dimensions are: 2 x φ0.7 x L3 (mm), E
r: 82, Z0: 8.55 [Ω], and the capacitance of the capacitor in the figure is [Line A] CA0 = CA2 = 10.85 [pF], CA1 = 11.06 [pF], CcA1 = CcA2 = 0 .48 [pF]; [Line B] CB0 = CB3 = 7.3 [pF], CB1 = CB2 = 7.708 [pF], CcB1 = CcB3 = 0.312 [pF], CcB2 = 0.238 [pF] [Common line] CcF = CcL = 0.87 [pF]. The DCS Rx connects the switch SW to the line A (upper side in the figure), and the WCDMA connects the switch SW to the line B (lower side in the figure) to switch the center frequency.

The characteristic characteristics of the variable band (switching) type DRO-BPF are shown in FIG. 15 [(a) pass band characteristic, (b) delay characteristic,
(c) Standing wave ratio, (d) Harmonic characteristics]. Bold line is DCS R
The thin line corresponds to WCDMA Rx and x. As mentioned above, D
CS (see FIG. 8 for the front-end configuration) has a frequency band of the received wave of 1805 to 1880 MHz and an intermediate frequency of 246 Mhz.
Hz lower heterodyne system, and the image signal is 1
Appears at 313-1388MHz. In WCDMA, the frequency band of the received wave is 2110 to 2170 MHz, and the intermediate frequency
It is a 190 MHz upper heterodyne system, and the image signal appears at 2490 to 2550 MHz. In the two-stage switching DRO-BPF of the present invention, the band shift is 1747.5 MHz (DC
S) ⇔2140 MHz (WCDMA). Further, the image signal attenuation amount is 7 for both DCS Rx and WCDMA Rx.
0 dB or more, and in any case, it is possible to eliminate the need for an LPF for spurious suppression, thereby realizing a front-end circuit configuration that is smaller and consumes less power than conventional products.

Embodiment 4 DRO-BPF of Three-Stage Configuration Applying DRO of the Same Specification
2 shows a configuration example of a Cdmaone Tx provided with a parallel capacitor type additional circuit. In the circuit shown in FIG. 16, each DRO has dimensions: 2 × φ0.7 × L6 (mm), Er: 82, Z0: 8.
55 [Ω], and the capacitances of the capacitors of the transmission line, the bypass, and the additional circuit have the values shown in the figure, and the center frequency is switched by selecting the low or high capacitor in the figure. . The additional circuit was provided as a bypass capacitor (10 pF) between the input / output terminal on the common line and the coupling capacitor.

The variable bandwidth (switching) type DRO-BP shown in FIG.
The characteristics of F are shown in FIG. 17 ((a) pass band characteristics, (b) delay characteristics,
(c) Standing wave ratio, (d) Harmonic characteristics]. The thin line is Low,
The thick line corresponds to High. FIG. 1 shows a comparison of characteristics between the circuit of FIG. 16 and the same circuit except that no additional circuit is provided.
8 to 25.

FIG. 18 shows the low-pass band characteristic in FIG.
9 shows a low-delay characteristic, FIG. 20 shows a low-standing wave ratio, and FIG. 21 shows a low-harmonic characteristic, a circuit without an additional circuit (a thin line in the figure) and a circuit with an additional circuit (FIG. (Bold line). FIG. 22 shows the High-passband characteristic, FIG. 23 shows the High-delay characteristic, FIG. 24 shows the High-standing wave ratio, FIG.
The circuit without the additional circuit (thin line in the figure) was compared with the circuit with the additional circuit (thick line in the figure).

From the comparison of these graphs, when additional circuits are connected to both sides of the input / output port, the pass band characteristics [Log
Mag | S21 | characteristic (0.78 to 0.98 GHz)], it can be seen that the steepness of the BPF characteristic is improved on the higher frequency side than the pass band. Harmonic characteristics [Log Mag | S21 | characteristics (0.5 to 3.5G
Hz)], it can be seen that the effect of improving steepness increases as the frequency increases.

[0079]

According to the variable band DRO-BPF of the present invention, the attenuation in each band is about 1 dB, and it is easy to adjust the attenuation outside each band and to improve the harmonic characteristics.
Therefore, a transmission filter for realizing suppression of spurious, a reception filter for realizing suppression of an image signal, and a duplexer for realizing suppression of an image signal at the time of reception and suppression of spurious at the time of transmission are provided.

Further, the variable bandwidth DRO-
The BPF can be used in a wide range of various mobile phone systems, and by using this as a component, a transmission filter or a reception filter that can support different multiplexing systems with a single component can be configured.

In the present invention, a switch made of GaAs semiconductor (GaAsSW) can be used. Since a GaAs semiconductor has low loss, a variable-bandwidth BPF (VB
Even if (PF) is formed, the deterioration of the BPF characteristic is small. But,
All standard products are resin mold type and have no bare chip. There is no standard GaAs MMIC in which a plurality of SWs are integrated. On the other hand, since the external dimensions of commercially available GaAsSW are almost the same as those of DRO, it is difficult to reduce the size if a large number of them are provided.

In the present invention, even if standard or expensive GaAsSW is used, only one or two switches are required per BPF, and the number of parts such as DRO is minimized. For this reason, a reduction in the size of the external dimensions and a reduction in the product price can be realized.

Therefore, by applying the variable band DRO-BPF, the problem of insertion loss of the LPF, which has hindered power saving, can be solved, and the image signal suppressing circuit can be simplified. A new front-end configuration with small size and low power consumption is provided by sharing parts in different modes in a mode mobile phone or the like.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a first embodiment of a variable band DRO-BPF according to the present invention.

FIG. 2 is a circuit diagram showing a second embodiment of the variable band DRO-BPF according to the present invention.

FIG. 3 is a circuit diagram showing a third mode of the band variable DRO-BPF according to the present invention.

FIG. 4 is a perspective view showing a terminal structure of a DRO used in the band variable DRO-BPF according to the present invention.

FIG. 5 is a circuit diagram listing additional circuits applicable to the present invention.

FIG. 6 is a circuit diagram schematically showing a balanced type configuration of a band variable DRO-BPF according to the present invention.

FIG. 7 is a block diagram showing a narrow-band CDMA system front-end configuration in comparison with a conventional example (a) and a configuration example (b) of the present invention.

FIG. 8 is a block diagram showing a front end configuration of a single mode TDMA in contrast to a conventional example (a) and a configuration example (b) of the present invention.

FIG. 9 is a block diagram schematically illustrating a configuration example of a front end of a dual mode TDMA according to the present invention.

FIG. 10 is a block diagram schematically illustrating a configuration example of a triple mode front end according to the present invention.

FIG. 11 is a circuit diagram showing a basic configuration of a band variable BPF circuit according to the first and second embodiments.

FIG. 12 shows DCST by the band variable BPF of the first embodiment.
x / DCS Rx circuit (a) passband characteristics, (b) delay characteristics, (c)
The graph which shows a standing wave ratio and (d) harmonic characteristics.

FIG. 13 shows a GSM T by the band variable BPF of the second embodiment.
x / DCS Tx circuit (a) passband characteristic, (b) delay characteristic, (c)
The graph which shows a standing wave ratio and (d) harmonic characteristics.

FIG. 14 is a circuit diagram illustrating a basic configuration of a band variable BPF circuit according to a third embodiment.

FIG. 15 shows DCS R by the variable band BPF of the third embodiment.
x / WCDMA circuit (a) passband characteristic, (b) delay characteristic,
(c) Graph showing the standing wave ratio and (d) harmonic characteristics.

FIG. 16 is a circuit diagram showing a basic configuration of a band variable BPF circuit according to a fourth embodiment.

FIG. 17 is a diagram showing Cdmaone using the band variable BPF according to the fourth embodiment.
Tx for (a) passband characteristics, (b) delay characteristics, (c) standing wave ratio,
(d) Graph showing harmonic characteristics.

FIG. 18 is a graph showing the low-pass band characteristic of the Tx for Cdmaone of the fourth embodiment.

FIG. 19 is a graph showing low-delay characteristics of Tx for Cdmaone of Example 4 in comparison with the characteristics of (a) a circuit provided with an additional circuit and (b) a circuit not provided with an additional circuit.

FIG. 20 is a graph showing the low-standing wave ratio of Tx for Cdmaone of Example 4 by comparing the characteristics of (a) a circuit provided with an additional circuit and (b) a circuit not provided with an additional circuit.

FIG. 21 is a graph showing low-harmonic characteristics of Tx for Cdmaone of Example 4 in comparison with the characteristics of a circuit provided with an additional circuit and a circuit not provided with an additional circuit.

FIG. 22 is a graph showing the high-pass band characteristics of the Tx for Cdmaone of the fourth embodiment.

FIG. 23 is a graph showing high-delay characteristics of Tx for Cdmaone of Example 4 in comparison with the characteristics of (a) a circuit provided with an additional circuit and (b) a circuit not provided with an additional circuit.

FIG. 24 is a graph showing the characteristics of (a) a circuit provided with an additional circuit and (b) a circuit not provided with an additional circuit with respect to the High-standing wave ratio of Tx for Cdmaone of Example 4;

FIG. 25 is a graph showing the high-harmonic characteristics of the Tx for Cdmaone of Example 4 in comparison with the characteristics of a circuit provided with an additional circuit and a circuit not provided with an additional circuit.

FIG. 26 is a circuit diagram showing a band variable DRO-BPF according to the related art.

27A and 27B are a schematic diagram (a) and a circuit diagram (b) of a band variable DRO-BPF different from that of FIG. 26 according to the related art.

FIG. 28 is a perspective view showing a band variable DRO-BPF different from that of FIG. 26 according to the related art.

FIG. 29 is an explanatory view schematically showing the operation principle of the band variable DRO-BPF in FIG. 28;

Claims (10)

[Claims] 1. A high-frequency band variable filter including a plurality of switchable lines as a part of a transmission line,
(A) each of the plurality of switchable lines has a bypass including a capacitor, and the remaining portion of the transmission line has a bypass including a dielectric resonator (DRO);
Wherein each of the plurality of switchable lines has a bypass including a dielectric resonator (DRO), and the remaining portion of the transmission line has a bypass including a capacitor; Each of the plurality of lines is a dielectric resonator (D
RO) and a bypass including a capacitor, and by selecting one of the plurality of switchable lines to form a transmission line, the characteristic value of the DRO in the bypass of the transmission line and And / or switching the capacitance of a capacitor in parallel therewith to change the filter characteristics. 2. In either (a) or (b), at least one of the plurality of switchable lines is provided in addition to the bypass including the capacitor or DRO,
The variable-bandwidth filter circuit according to claim 1, further comprising at least one set of a bypass including a DRO and a bypass including a capacitor. 3. A structure in which the terminal of the DRO has a two-branch structure from the end of the DRO, and the two ends generated by the branch are inserted into the transmission line to directly connect the DRO to the transmission line. 3. The band variable filter circuit according to 2. 4. The band variable filter according to claim 1, wherein the capacitor means connected in parallel with the DRO has a capacity for shifting the center frequency of the filter to a low frequency side by 40 MHz or more. circuit. 5. The non-harmonic band variable filter circuit according to claim 1, wherein the input admittance characteristics of the DRO and the bypass capacitor in each bypass are selected so as to cancel the harmonic characteristics. . 6. A high-frequency front-end circuit, wherein a band limiting unit near an input / output unit to an antenna includes the band variable filter according to any one of claims 1 to 5. 7. The band variable BP according to claim 1, wherein the band limiting means in the vicinity of the input / output section to the antenna is provided.
7. The high-frequency front-end circuit according to claim 6, further comprising a duplexer having an image signal suppressing function and a spurious suppressing function constituted by F. 8. The band variable BP according to claim 1, wherein the band limiting means in the vicinity of the input / output section to the antenna is provided.
F, thereby performing switching between a plurality of transmission frequency bands in the same signal multiplexing scheme, or switching between respective transmission frequency bands in a plurality of signal multiplexing schemes, and 7. The high-frequency front-end circuit according to claim 6, wherein spurious noise during transmission is substantially suppressed. 9. The band variable BP according to claim 1, wherein the band limiting means in the vicinity of the input / output section to the antenna is provided.
F, thereby switching between a plurality of reception frequency bands in the same signal multiplexing scheme or switching between respective reception frequency bands in a plurality of signal multiplexing schemes, and 7. The high-frequency front-end circuit according to claim 6, wherein the image signal at the time of reception is substantially suppressed. 10. The high-frequency front-end circuit according to claim 6, wherein the high-frequency front-end circuit is used for a multi-band mobile phone that switches between a plurality of bands and uses a plurality of signal multiplexing systems.