JP2002100906A - Microwave circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a microwave circuit which is high in flexibity of design and smaller in both the size and power loss as compared with. SOLUTION: A high-frequency transmission line is composed of a signal line 67, a first ground pattern 72 and a second ground pattern 74 arranged on the sides of the signal line 67, and four bridge conductors 161, 162, 163 and 164 provided at a regular pitch, so as to short-circuit the ground patterns 72 and 74. A first insulating film 261 is sandwiched between the bridge conductor 161 and the signal line 67 to form a first reactance factor, formed of an MIM capacitor. Likewise, a second to a fourth insulating films, 262 to 264, are sandwiched between the second to fourth bridge conductor, 162 to 164, and the signal line 67 to form a second to a fourth reactance factor, with each being formed of a MIM capacitor.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a microwave circuit using a transmission line. Especially microwave / monolithic IC (MMIC) and microwave / hybrid IC
The present invention relates to an impedance matching circuit in a high-frequency semiconductor device for microwave band or millimeter wave band circuits such as (MHIC).

[0002]

2. Description of the Related Art Due to the rapid growth of demand in the IT technology field mainly for information communication in recent years, it is urgent to increase the number of communication lines. For this reason, practical use of a system using a microwave / millimeter wave band, which has not been widely used, has been rapidly progressing.

[0003] The RF section of a high-frequency band wireless communication device generally includes an oscillator, a synthesizer, a modulator, a power amplifier, a low-noise amplifier, a demodulator, and an antenna. In the communication device,
It is desirable that the electrical characteristics be excellent and that they be compact. Especially for portable communication equipment, in addition to the demand for small size and light weight,
Low loss and long battery life are also required. When considering the miniaturization of such a high-frequency circuit section, the MHIC integrates necessary circuits as much as possible on one circuit board.
Also, an MMIC integrated in a semiconductor chip is effective.

[0004] With the rapid development of semiconductor integration technology, integration of MMICs is progressing. That is, circuits formed in one semiconductor chip are developing in a direction of higher integration density. For this reason, the integration density has been improved from a conventional semiconductor chip on which a single high-frequency active element is mounted to a semiconductor chip on which a functional circuit block that performs one circuit function of a device is mounted. Furthermore, the integration density has been increasing so that a plurality of functional circuit blocks can be mounted on the same semiconductor chip. M
The MIC has a high electron mobility transistor (HEMT),
A high-frequency active element such as a heterojunction bipolar transistor (HBT) and a Schottky gate type FET (MESFET), a passive element such as a capacitor, an inductor, and a resistor, and a transmission line are formed. Transmission line types used in high-frequency bands are microstrip lines,
A coplanar line or the like is common. A planar circuit such as a microstrip line or a coplanar line is often used for a microwave circuit because it has higher mass productivity and can be constructed at a lower cost than a three-dimensional circuit such as a waveguide.

However, as is well known, the characteristic impedance of the microstrip line is determined by the relative permittivity ε _{r of} the dielectric substrate to be used, the substrate thickness, and the wiring width of the strip conductor (signal line). That is, in order to the characteristic impedance lower, the dielectric constant epsilon _r is high, the substrate thickness is thin, the wiring width of the signal line may be wider. In order to increase the characteristic impedance, the opposite may be performed.
Similarly, in the case of a coplanar line, the relative permittivity ε _{r of} the substrate,
The characteristic impedance Z depends on the thickness of the substrate, the width of the central signal line, and the distance between the signal line and the ground pattern on both sides of the signal line.
₀ is determined.

[0006]

For this reason, if the thickness of the same substrate is reduced, the width of the signal line of the planar circuit must be reduced in order to realize a desired characteristic impedance. This leads to an increase in conductor loss of the planar circuit. For example, when used in a feed line, a filter, an impedance matching circuit, or the like to an antenna used before and after the final amplification stage of a wireless device, the power use efficiency of the system is deteriorated. That is, there has been a limit in reducing the thickness of the conventional microwave circuit. In particular, in the case of MHIC using a ceramic substrate such as alumina (Al ₂ O ₃ ) for high frequency applications, conductor loss is more dominant as a system loss than dielectric loss, so the conductor loss has an effect on the performance of the microwave circuit. It depends greatly. When the thickness of the dielectric substrate is increased, an unnecessary mode is excited depending on the transmission line, and there is a risk that the loss is further deteriorated.

On the other hand, when a low characteristic impedance is required in the impedance matching circuit, the wiring width of the signal line of the planar circuit becomes large, and it may not be possible to reduce the size. MHI
In microwave circuits such as C, the thickness and material of the dielectric substrate are determined not only by electrical characteristics but also by mechanical strength, compatibility with the mounting process, mass production, etc., so transmission lines with specific impedance There is a restriction on the method of realizing the above, and there is a case where the transmission loss has a great effect on miniaturization.

As described above, in the conventional microwave circuit,
There are limits to reducing transmission line loss, miniaturization, and high functionality.
New techniques have been desired to improve the performance of high frequency systems in the microwave and millimeter wave bands. For example, to realize a circuit element structure such as a filter, a diplexer, an impedance converter, and an impedance matching circuit using a transmission line, a finite length transmission line having a certain characteristic impedance is required. The use of the technique described above has limited the characteristics of the transmission line.

The present invention has been made in view of the above circumstances, and its purpose is to increase the degree of freedom in design.
Another object of the present invention is to provide a microwave circuit that is smaller and has lower loss than before.

Another object of the present invention is to provide a microwave circuit having a novel structure capable of expanding the adjustable range of the characteristic impedance of a high-frequency transmission line.

It is still another object of the present invention to provide a microwave circuit having a novel structure capable of reducing crosstalk with low loss.

Still another object of the present invention is to provide a microwave circuit capable of reducing an occupied area required for wiring as a whole.

Still another object of the present invention is to provide a microwave circuit having a high-frequency transmission line in which electric characteristics at a high frequency can be easily adjusted.

[0014]

In order to achieve the above object, a first feature of the present invention is to provide a basic transmission line designed by low frequency limit approximation and a constant pitch d with respect to the basic transmission line. And a microwave circuit comprising a plurality of reactance components having the same reactance value and loaded periodically. Furthermore, in the microwave circuit having this periodic structure, by selecting the pitch d and the reactance value, a passive element structure in which the characteristic impedance is set to a Bloch impedance different from a value specific to the basic transmission line is partially used. I have it. "Basic transmission line designed by low frequency limit approximation" refers to a transmission line in which the wave number k of the microwave propagating through the basic transmission line is small and the effect of "Bloch scattering" due to the periodic boundary conditions of the transmission line can be ignored. This is a conventional transmission line having no periodic structure. As basic transmission lines used in the microwave circuit according to the first feature of the present invention, conventionally known microstrip lines, coplanar lines, grounded coplanar lines, slot lines, triplate strip lines,
Various high-frequency transmission lines such as waveguides can be used.
The “passive element structure” of the present invention corresponds to a physical structure for realizing a high-frequency circuit such as a filter, a diplexer, an impedance converter, and an impedance matching circuit.

In the microwave circuit according to the first aspect of the present invention, the electrical characteristics of the high-frequency transmission line are periodically changed by a plurality of reactance components periodically loaded at a constant pitch d. Various dispersion relations are realized by boundary conditions. In addition, various Bloch impedances are developed using “Bloch scattering” based on a periodic boundary condition. That is, by realizing a microwave circuit using the various dispersion relations and Bloch impedance, the degree of freedom for determining the characteristic impedance of the high-frequency transmission line can be increased.

As described at the beginning, for example, the characteristic impedance of the conventional microstrip line which is the “basic transmission line” of the present invention is the relative dielectric constant ε _{r of} the substrate, the thickness of the substrate, the signal line (strip wiring). ). In the case of the conventional coplanar line which is the “basic transmission line” of the present invention, the relative permittivity ε _{r of} the substrate, the thickness of the substrate, the width of the central signal line, the signal line and the ground pattern on both sides of the signal line the distance, the characteristic impedance Z ₀ is determined, the variable range is constrained. There are only three independent variables that determine the characteristic impedance of the microstrip line, and there are only four independent variables that determine the characteristic impedance of the coplanar line. For this reason, in order to increase the characteristic impedance of the high-frequency transmission line, it is necessary to reduce the width of the signal line, and there is a problem that the conductor loss increases. On the other hand, when trying to lower the characteristic impedance, the wiring width of the signal line becomes large, which hinders miniaturization.

In the microwave circuit having the periodic structure according to the first feature of the present invention, a reactance component jb and a pitch d are added as independent variables for determining the characteristic impedance of the microwave circuit. That is, by using the Bloch impedance of the microwave circuit having the periodic structure according to the first feature of the present invention, the independent variable for determining the characteristic impedance of the microwave circuit increases, and the constraint is greatly eased. . For this reason, a more diverse and high-performance microwave circuit than before can be easily realized. For example, without narrowing the wiring width of the signal line, it is easy to increase the characteristic impedance Z _0, without thicker line width of the signal line, the characteristic impedance can also be reduced. For this reason, it is possible to obtain a desired characteristic impedance without increasing conductor loss and hindering downsizing of the circuit. As a result, the size and the loss of the microwave circuit can be reduced.

In the first aspect of the present invention, each reactance value of a plurality of reactance components may be made variable by a control signal. For example, if a variable reactor such as a varactor diode whose reactance component is variable by a control signal is used, the Bloch impedance can be dynamically varied. Alternatively, a capacitor is formed by a three-dimensional microstructure called a microsystem, micromechatronics, or MEMS (microelectromechanical system), and the distance between the electrodes of the capacitor is changed by a microactuator or the like. However, a variable reactor can be configured. For example, MEMS
When a capacitor having a structure is used, a metal substrate (bridge conductor) can be formed above a signal line (strip conductor). When a potential difference between the bridge conductor and the strip conductor is changed, the distance between the bridge conductor and the strip conductor is increased by Coulomb force. Is changed, and as a result, the magnitude of the loaded reactance component is changed, so that a function equivalent to the case of using the capacitor having the MEMS structure can be realized. As long as the reactance component can be changed by the control signal, it goes without saying that the loaded element may have a structure other than the MEMS capacitor or the varactor diode.

In the first aspect of the present invention, the wave number k
When the product y = kd of pitch and the pitch d is defined, the y value dependency of the Bloch impedance has a plurality of pass bands whose order increases sequentially from the first pass band as the y value increases.
Here, the wave number k is the wave number of the microwave propagating through the basic transmission line when there is no reactance component loaded periodically. Forbidden bands exist between adjacent order pass bands. That is, the first pass band, the first forbidden band, the second pass band,
A plurality of pass bands of which the order increases as the y value increases in the order of the second forbidden band, the third pass band, the third forbidden band, the fourth pass band, the fourth forbidden band, ... And forbidden zones are arranged. Detail is,
As will be described later, the characteristic impedance of odd-numbered (first, third,...) Pass bands decreases as the order increases. On the other hand, even order (second, fourth,...)
In the passband of ()), the characteristic impedance increases as the order becomes higher. If the characteristic of the odd-order pass band is used, the Bloch impedance in the used frequency band can be easily made lower than the characteristic impedance of the transmission line approximated by the low frequency limit. On the other hand, if the characteristic of the even-order pass band is used, the Bloch impedance in the used frequency band can be easily made higher than the characteristic impedance of the transmission line approximated by the low frequency limit.

Considering the y value dependence of the Bloch impedance, the “passive element structure” constituting the microwave circuit according to the first feature of the present invention has a high-order passband higher than the second passband. It is preferable that the pitch d is selected so as to be the impedance of the band. This is because the characteristic of the higher-order pass band equal to or higher than the second pass band is more remarkably different from the characteristic impedance of the transmission line in the low frequency limit approximation. Therefore, a desired characteristic impedance can be easily obtained while maintaining the basic form of the conventional transmission line by actively utilizing the characteristics of the higher-order passbands equal to or higher than the second passband. In addition, at this time, there is no need to sacrifice low loss or miniaturization.

The following is a description of a coplanar line. That is, as a “basic transmission line”, a dielectric substrate, a signal line disposed on the surface of the dielectric substrate, and first side lines disposed on both sides of the signal line and separated from the signal line by a predetermined distance.
And a second ground pattern, a coplanar line is selected, and each of the reactance components is formed as a bridge conductor connecting the first and second ground patterns to each other, and a capacitor is formed between the bridge conductor and the signal line. What is necessary is just to comprise from the insulating film arrange | positioned as much as possible. The coplanar line having this periodic structure has a reactance component jb determined by a capacitor value formed between a bridge conductor and a signal line as an independent variable for determining its characteristic impedance.
And the pitch d of the arrangement of the plurality of bridge conductors is added to the factor that determines the Bloch impedance. That is, the width of the bridge conductor, the relative permittivity of the insulating film disposed between the bridge conductor and the signal line, the thickness of the insulating film, and the pitch d of the arrangement of the plurality of bridge conductors determine the Bloch impedance of the coplanar line. Independent variables increase dramatically as they are added to the deciding factors. For this reason, a coplanar line having a wider characteristic impedance than the conventional one can be easily realized. In the coplanar line having the periodic structure, the characteristic impedance can be increased or decreased while the wiring width of the signal line is maintained at an optimum value that does not increase conduction loss. That is, it is possible to realize a coplanar line having a desired characteristic impedance without increasing conductor loss and hindering downsizing of the circuit. Therefore, the coplanar line having this periodic structure is suitable for an impedance converter and an impedance matching circuit.

A second feature of the present invention is that (a) a dielectric substrate, (b) first and second ground patterns arranged opposite to each other on the dielectric substrate, and (c) a dielectric substrate. At
An active element having first and second main electrodes and a control electrode disposed between the first and second ground patterns;
(D) On the dielectric substrate, the input-side signal line sandwiched between the first and second ground patterns and connected to the control electrode, and (e) the first and second signal lines on the dielectric substrate. And (f) pass over the input signal line, and periodically at a constant pitch d, between the first and the second signal lines. A plurality of bridge conductors connecting the second ground pattern to each other;
(G) A microwave circuit comprising a plurality of insulating films respectively arranged to form a capacitor between the plurality of bridge conductors and the input side signal line. Furthermore,
In this microwave circuit, the input impedance of the active element is adjusted to a desired value by selecting the pitch and the reactance value exhibited by the capacitor.

In the microwave circuit according to the second feature of the present invention, the first and second ground patterns and the first and second ground patterns are provided.
The input side coplanar line is formed by the input side signal line and the input side signal line sandwiched between the second ground patterns. A plurality of bridge conductors which pass over the input signal line and connect the first and second ground patterns to each other; and an insulation disposed between the plurality of bridge conductors and the input signal line, respectively. A plurality of capacitors serving as reactance components are formed from the film. Therefore, the input-side coplanar line according to the second feature of the present invention has a reactance determined by a capacitor value formed between each of the plurality of bridge conductors and the input-side signal line as an independent variable for determining its characteristic impedance. The component jb and the pitch d of the arrangement of the plurality of bridge conductors are added to the factors that determine the Bloch impedance. That is, the width of the bridge conductor, the relative permittivity of the insulating film disposed between the bridge conductor and the input-side signal line, the thickness of the insulating film, and the pitch d of the arrangement of the plurality of bridge conductors are determined by the input-side coplanar line. In addition, the independent variable for determining the characteristic impedance of the input-side coplanar line is dramatically increased.

As a result, in the microwave circuit according to the second aspect of the present invention, an input impedance matching circuit having a wider adjustment range and a smaller occupation area than the conventional one is inserted on the input side of the active element. You can do it. In addition, since the input impedance matching circuit can adjust the impedance while maintaining the wiring width of the input signal line at an optimum value, the high frequency characteristics of the microwave circuit can be improved.

In the microwave circuit according to the second feature of the present invention, the "active element" includes various high-frequency devices such as a bipolar transistor (BJT) such as a HEMT and an HBT, a MESFET, and an electrostatic induction transistor (SIT). Semiconductor elements can be used. The “first main electrode” of the active element refers to one of the emitter region and the collector region in the BJT, HEMT, MESFE.
In T and SIT, it means either the source region or the drain region. The “second main electrode” of the active element is one of the emitter region and the collector region which does not become the first main electrode in the BJT, and the HEMT,
In MESFETs and SITs, it means either the source region or the drain region that does not become the first main electrode. That is, if the first main electrode is the emitter region,
The second main electrode is a collector region, and if the first main electrode is a source region, the second main electrode is a drain region. or,
The “control electrode” of the active element is the base electrode of BJT and H
Needless to say, it means the gate electrodes of EMT, MESFET, and SIT.

The third feature of the present invention is that (a) a dielectric substrate, (b) first and second ground patterns arranged opposite to each other on the dielectric substrate, and (c) a dielectric substrate. At
An active element having first and second main electrodes and a control electrode disposed between the first and second ground patterns;
(D) On the dielectric substrate, the input-side signal line sandwiched between the first and second ground patterns and connected to the control electrode, and (e) the first and second signal lines on the dielectric substrate. And (f) passing over the output-side signal line and periodically at a constant pitch d, between the first and the second signal lines. A plurality of bridge conductors connecting the second ground pattern to each other;
(G) A microwave circuit comprising a plurality of insulating films arranged to form a capacitor between the plurality of bridge conductors and the output signal line. Furthermore,
In this microwave circuit, the output impedance of the active element is adjusted to a desired value by selecting the pitch and the reactance value exhibited by the capacitor.

In the microwave circuit according to the third feature of the present invention, the first and second ground patterns and the first and second ground patterns are provided.
And an output-side signal line sandwiched between the second ground patterns forms an output-side coplanar line. A plurality of bridge conductors which pass over the output signal line and connect the first and second ground patterns to each other; and insulation provided between the plurality of bridge conductors and the output signal line, respectively. A plurality of capacitors serving as reactance components are formed from the film. For this reason, the output-side coplanar line according to the third feature of the present invention has, as independent variables for determining its characteristic impedance, a reactance component jb and a reactance component jb determined by a capacitor value formed between the bridge conductor and the output-side signal line. Pitch d of array of multiple bridge conductors
Is added to the factors that determine Bloch impedance. That is, the width of the bridge conductor, the relative permittivity of the insulating film disposed between the bridge conductor and the output signal line, the thickness of the insulating film, and the pitch d of the arrangement of the plurality of bridge conductors
Is added to the factors that determine the Bloch impedance of the output-side coplanar line, so that the independent variable that determines the characteristic impedance of the output-side coplanar line greatly increases.

As a result, in the microwave circuit according to the third feature of the present invention, an output impedance matching circuit having a wider adjustment range and a smaller occupation area than the conventional one is inserted on the output side of the active element. You can do it. In addition, the output impedance matching circuit can adjust the impedance while maintaining the wiring width of the output-side signal line at an optimum value, thereby improving the high-frequency characteristics of the microwave circuit.

[0029]

DESCRIPTION OF THE PREFERRED EMBODIMENTS The present invention will be described below in detail.

(Principle of the Present Invention) First, the principle of the present invention will be described using a simple model. FIG. 2 is a schematic diagram of a transmission line in which a reactance component jb is periodically loaded at a pitch d. A signal propagating in a transmission line having a periodic structure as shown in FIG. 2 is delayed as the signal frequency increases, and the speed becomes zero (standing wave) at a frequency at which the period becomes a half wavelength. As is well known. Although several topologies are conceivable in terms of an equivalent circuit, as shown in FIG. 2, a dispersion relationship is derived using a model in which a reactance component jb is periodically inserted into a shunt in a transmission line as an example.

First, k is a propagation constant (wave number) of a line in the absence of a periodic structure, and ABC of a unit of a transmission line periodically loaded with a reactance component jb at a pitch d.
Find the D matrix. Unit unit, the ABC since shunt reactance, and is sandwiched from both sides in a characteristic impedance Z ₀ by the length d / 2 structure
The D matrix is given by kd = y

(Equation 1) Becomes On the other hand, periodic boundary conditions;

(Equation 2) From this, AD = ^e2γd− (A + D) ^eγd− BC = 0 (3) is derived. γ is the propagation constant of the line. AD-BC =
Consider 1 and the above equation, cos (βd) = (A + D) / 2 = cos (y) - a _{(bZ 0/2) sin (} y) ... (4). However, the line was assumed to have no loss (γ = jβ).
Here, β is a propagation constant of a transmission line having a periodic structure.

Assuming that the reactance component is constituted by the capacitance value C ₀ , the reactance factor ξ is defined by the following equation (5): ξ = C ₀ Z ₀ c / 2d (5) Here, Z ₀ is the characteristic impedance of the transmission line in a no-load state, and c is the propagation speed (light speed) in a no-load state.
It is. The reactance factor ξ defined by equation (5)
Is used, equation (4) can be rewritten as equation (6).

Cos (βd) = cos (y) −ξ · (y) · sin (y) (6) Here, the reactance factor 定義 defined by equation (5) is equal to 0.
When the k-β dispersion relation of the equation (6) is calculated for the cases 1 and 10, the k-β diagram shown in FIG. 3 is obtained.

The k-β diagram shown in FIG. 3 shows two bands, the lowest order band and the second band from the bottom. When the reactance factor ０ is 0, the transmission line is a no-load transmission line, and therefore, for TEM waves, k and β coincide. That is, the k-β diagram is a straight line having a slope of 1. In the k-β diagram shown in FIG. 3, since the higher-order band outside the Brillouin zone is convolved in the zone, the slope of the second band is 1.

In the case where the reactance factor ａ shown in FIG. 3A is 0.1, the reactance factor 小 さ い is small. It can be seen from FIG. 7 that the band gap between the second bands is small. It can also be seen that the higher the order of the band, the narrower the pass band (pass band). From this fact, it can be seen that when a band-pass filter having a narrow band is to be formed, it is preferable to increase the reactance factor ξ and use a higher-order pass band.

When a band of interest is to be realized in a low-pass filter to achieve a forbidden band (stop band), if the dimension of the periodic structure increases, the reactance component is increased to increase the early frequency. Can realize the forbidden zone. In FIG. 3, when the reactance factor １０ is 10, the wave number k that reaches the band stop is 0.5, which is 1 at no load (π = 3.14).
/ 6 or less. In other words, if band stop is to be realized at the same frequency, the size can be reduced to 1/6.

An important point in application of the microwave circuit is the dependency of the k-β dispersion relation on the reactance factor ξ shown in FIG. That is, it is necessary to determine what the capacitance value C ₀ should be when a cut-off at a certain frequency is desired.

(Relationship Between Band and Reactance) In order to obtain a condition for starting a forbidden band (stop band), βd = π in Expression (6).

−1 = cosy−ξ · y · siny (7) Here, using the well-known relationship of cosy = ² cos ² (y / 2) -1 (8) siny = 2 sin (y / 2) cos (y / 2) (9), cos (y / 2) = Ξ · y · sin (y / 2) (10) When the equation (10) is solved for the reactance factor ξ, the reactance factor ξ becomes a function ｙ of y = kd.
(Y), and becomes the following equation (11).

Ξ (y) = 1 / (y · tan (y / 2)) (11) The horizontal axis is the reactance factor ξ (y), and the vertical axis is y (= kd)
And the relationship of equation (11) is shown in the graph of FIG.

FIG. 4A shows the first band, and FIG.
Shows the second band. FIG. 4 shows how much the pitch d can be reduced by a periodically arranged reactance component when it is desired to provide a forbidden band at a certain wave number k. Naturally, the limit value when no reactance is loaded is y = kd = π = 3.14. This means that adjacent bands are connected (there is no forbidden band). As the reactance load is increased, the band gap rapidly widens at first, and gradually approaches the value of each band. FIG. 4 shows only the second band, but as described above, the asymptotic value becomes smaller for higher-order bands. Here, the reactance factor ξ is as defined by equation (5). Therefore, for example, the pitch d = 1 mm = 1 × 10
^{When a} capacitance value C ₀ is calculated for a microstrip line having a characteristic impedance Z ₀ = 50Ω on a dielectric substrate having a relative permittivity of ^-3 m and a relative permittivity of ε _r = 10, the light propagation speed becomes _In an alumina substrate with _r = 10,
Considering that 3.0 × 10 ⁸ / ε _r ^1/2 , C ₀ = (ξ · ε _r ^1/2 · 2 × 10 ⁻³ ) / (50 × 3.0 × 10 ⁸ ) = 0.133 · ξ · ε _r ^1/2 (pF) (12)

Specific permittivity ε_r= 10 alumina substrate
Impedance Z₀= Effectiveness of coplanar line 7 of 50Ω
Relative permittivity ε_effIs about 6.7. For this reason, prohibited
In order to make the band start 1 / 3.14 earlier, from FIG.
Since ξ = 2, the capacitance value C ₀From equation (12), 7p
It can be seen that F should be set. Therefore, the previous 60 GHz
Considering the application to the belt, the length is 6mm to 2mm or less
If it is formed by meander line,
It is a sufficiently realistic solution even on an MMIC chip. At that time,
To place capacitors equivalent to 7 pF at 0.4 mm intervals
Become.

[0043] When using a transmission line with real periodic structure (the characteristic impedance of the line) to the impedance matching circuit, not only the band structure, the characteristic impedance Z ₀ of the transmission _line, calculating the like input impedance Z _in There is a need. Here, first, we calculate the characteristic impedance Z ₀ of the transmission line having a periodic structure, after having repeated a finite, obtaining the input impedance Z _in when terminated by the terminating impedance Z _L. The Bloch impedance Z _B (y) of the transmission line is as follows: Z _B ^± (y) = ± B / (A ² −1) ^1/2 (13) when the ABCD matrix of the expression (1) is used. 5, the real part R of the Bloch impedance _{Z B}
showing the e _(Z B _(y)) and the imaginary part _{Im (Z B (y))} .

In calculating FIG. 5, the reactance factor ξ was set to 0.1. Therefore, FIG. 3A corresponds to the band diagram. In the first pass band (pass band) P _1, the wave number k is small, when the influence of Bloch scattering can be ignored (when low-frequency limit approximation possible) is,
The characteristic impedance determined by the Bloch impedance Z _B (y) is a real number, and is 45.6 Ω, which is slightly lower than the characteristic impedance Z ₀ = 50 Ω of the transmission line under no load (average characteristic because the capacitance is arranged in the shunt). The impedance is lower than that at no load). In the first pass band P _1, than in the case of the low-frequency limit approximation, the wave number k
Becomes larger and the effect of Bloch scattering becomes more pronounced, the value of the characteristic impedance becomes smaller.

The characteristic impedance is zero at the first of the forbidden band (stop band) S ₁ begins. Then, in the first forbidden band S _1, the characteristic impedance becomes purely imaginary. The fact that the characteristic impedance is a pure imaginary number corresponds to a state in which no energy is transmitted, that is, an evanescent mode, because the phase of the voltage and the current is shifted by 90 degrees and the power factor of the pointing vector becomes zero. In Figure 5, the first forbidden band S ₁ is appears at, y = kd is approximately 2.6 to 3.14
And coincides with the band diagram of FIG.

In the second pass band P ₂ , the characteristic impedance is a pure real number, but has a negative value. As can be seen from equation (13), a term having a square root is included, so that a positive or negative solution comes out. Physically, it means a traveling wave and a backward wave, so it is sufficient to take a positive value. In a second pass band P _2, when the wave number k increases, the second
Th of the forbidden band S ₂ begins.

The third pass band P ₃ is 6.28 (= 2
π) -8.1. In a third pass band P _3, when the wave number k increases, begins the third forbidden band S _3. The fourth pass band P ₄ starts near 9.42 (= 3π).

In FIG. 5, the first to fourth pass bands P
Looking at the values of the characteristic impedance (Bloch impedance) at ₁ , P ₂ , P ₃ , and P ₄ , the characteristic impedance of the odd-order (first and third) decreases as the order increases, and the even-order decreases. In (second, the illustration is omitted, the same applies to the fourth), the characteristic impedance increases as the order increases. This phenomenon is also very interesting in application. For example, in order to obtain a desired characteristic impedance using a conventional transmission line, if the present invention is used even when it is necessary to sacrifice low loss or downsizing, the number of design parameters increases. A desired characteristic impedance can be realized while maintaining low loss and miniaturization.

(Microstrip Line) A microstrip line will be described below as an example. The microstrip line is formed by forming a ground pattern on the back surface of a dielectric substrate and forming a strip-shaped wiring (signal line) on the upper surface with metal. Since it is inexpensive and can be mass-produced by printing technology, which has been making remarkable progress in recent years, it is often used for microwave circuits. In this case, the characteristic impedance of the transmission line is determined by the relative permittivity ε _{r of} the dielectric substrate, the thickness of the substrate, and the width of the signal line (strip line). Relative permittivity ε _r
In the case where alumina of = 10 is used, a characteristic impedance of approximately 50Ω can be realized when the ratio of the wiring width of the signal line to the height of the substrate is 1: 1. Therefore, as the size is reduced, the width of the signal line becomes narrower, and the conductor loss increases. On the other hand, if a line having a low characteristic impedance is used to constitute a filter or an impedance matching circuit in general, the wiring width of the signal line becomes large, which is one factor that hinders miniaturization. Such a problem occurs because there are only three independent variables that determine the characteristic impedance: the relative permittivity ε _{r of} the substrate, the thickness of the substrate, and the width of the signal line of the strip line. By using the present invention, a variety of circuits can be realized more than ever because the number of independent variables increases. In the case of the transmission line having the periodic structure shown in FIG. 2, the reactance component jb and the pitch d are added to the independent variables. When this value is determined, the characteristic impedance shown in FIG. 5 is obtained by calculation. From an actual design standpoint, once the frequency band to be used and the characteristic impedance are determined, the characteristic impedance of the transmission line before loading the reactance component jb, the pitch d, and the reactance component is determined so as to satisfy the specifications. Become.

Although FIG. 2 is used for describing the present invention, the present invention is characterized in that a microwave is constituted by a transmission line having a periodic structure in which Bloch impedance appears, so the circuit topology is not limited to FIG. . For example, a series may be loaded with a reactance component periodically, or both a series and a shunt may be loaded. The reactance component to be loaded may be a lumped constant type inductor or capacitor, or transmission lines having different characteristic impedances may be alternately arranged. Further, although a microstrip line is used in the description, the present invention is not limited to the type of transmission line used. Other transmission lines, namely coplanar lines,
It may be used for a grounded coplanar line, slot line, triplate strip line, waveguide, or the like.

Although FIG. 2 shows an equivalent circuit of a transmission line having a single periodic structure for simplicity, a transmission line having a multiple periodic structure may be used.

The behavior of the Bloch impedance Z _B (y) of the transmission line shown in FIG. 5 suggests numerous possible applications. Actually, when a circuit is configured using a transmission line having a periodic structure, a transmission line having an infinite number of repetitions n cannot be realized. Then, n
After periodic transmission, to be seen when it is terminated with the terminating impedance Z _L, the behavior of the input impedance Z _in. Wavenumber k, the pitch d, the number of repetitions n of the Bloch impedance _{Z B} of the transmission line termination impedance _{Z L} input impedance wished terminating the transmission line when it is terminated by _Z in (y) is, _Z in (y) = a _{Z B (Z L + jZ B} tan (y · n)) / (Z B + jZ L tan (y · n)) ... (14). Wavenumber zero termination impedance _{Z L,} i.e. low input impedance when the Bloch impedance 45.6Ω the frequency extremes _Z in (y) is as shown in FIG. 6.

Similarly, the pass characteristic S of the S parameter
₂₁ (y) is

(Equation 3) From the relational expression: S ₂₁ (y) = 2 / (A + (B / Z ₀ ) + CZ ₀ + D) = 2 / (2 cos (βd · n) + j ((Z _B / 50) + (50 / Z _B )) sin (βd · n)) (16) From Equations (16) and (13), the y = kd dependence characteristic of the pass characteristic S ₂₁ (y) shown in FIG.

In FIG. 7, the number of repetitions n is 4, 10, 20
Is calculated. Naturally, the band edge becomes sharper as the number of repetitions n increases.

When the number of repetitions n is increased, the microwave circuit generally becomes large, but the amount of out-of-band attenuation can be increased. Therefore, it is necessary to determine the number of repetitions n based on the electrical specifications of the circuit and constraints on the outer shape. . In FIG. 7, pass bands P ₁ , P
Ripple is seen in ₂ , P ₃ and P ₄ because the Bloch impedance differs from the system impedance in that frequency band.

In order to eliminate ripples in desired pass bands P ₁ , P ₂ , P ₃ , P ₄ ,..., Frequency bands P ₁ , P ₂ , P ₃ , P ₄ ,
The characteristic impedance of the transmission line before the loading of the reactance component, the unit amount of the reactance to be loaded, the loading cycle, and the number of repetitions n may be determined so that the Bloch impedance of...

(Application to Antenna Feeding Line) In a wireless device, the reduction in loss from the power amplifier at the last stage to the antenna feeding port greatly affects the characteristics of the wireless device. In other words, if the loss is reduced, there is no need to generate useless power in the power amplifier, which leads to an increase in battery life in the terminal, and it is possible to improve the linearity of the base station by the loss. , Which can lead to further improvement in frequency use efficiency. Therefore, if the microwave circuit of the present invention is used for a feed line from the output of the power amplifier to the antenna feed, the antenna feed can be performed with a desired characteristic impedance and low loss. In addition, by actively using various dispersion relations of the characteristics of the periodic transmission line, physically separated transmission lines can be used.
In addition to the connection between points, it is possible to hold the functions of a filter, a diplexer, a coupler, and the like, which are often required for a wireless device, so that a high-performance front-end unit can be configured.

(Application to Impedance Converter) The present invention can also be applied as an impedance converter.
An input impedance Z _in is converted to an output impedance Z.
If you want to convert to _out , the wavelength is λ / 4
+ N · λ / 2 in length, may be inserted a characteristic impedance of _{Z B} to satisfy (17).

Z_in= Z_B ²/ Z_out (17) As described above, the conventional microstrip line
Characteristic impedance Z ₀Is the relative permittivity of the substrate
ε_rDepending on the thickness of the board and the width of the signal line (strip line).
Is restricted. In the case of coplanar lines, the board
Electric power ε_r, Board thickness, center signal line width, signal line and signal
The characteristic impedance depends on the distance from the ground pattern on both sides of the line.
-Dance Z₀Is constrained. However, the blower of the present invention
Loch impedance Z_BIf you use
Be relaxed.

FIG. 1 shows a structure of an impedance converter according to an embodiment of the present invention. This impedance converter is realized by a coplanar line having a periodic structure. That is, as shown in FIG. 1, the coplanar line in which the signal line 67 and the first ground pattern 72 and the second ground pattern 74 are arranged at a certain distance on both sides of the signal line 67 is referred to as “the present invention”. It is used as a "basic transmission line." The impedance converter further comprises four bridge conductors 161, 162, 16 which are periodically arranged at a constant pitch.
3, 164 short-circuits between the first ground pattern 72 and the second ground pattern 74. The first insulating film 261 is provided between the first bridge conductor 161 and the signal line 67.
Are interposed to constitute a first reactance factor composed of an MIM capacitor. A second insulating film 262 is interposed between the second bridge conductor 162 and the signal line 67,
It constitutes a second reactance factor composed of an IM capacitor. A third insulating film 263 is interposed between the third bridge conductor 163 and the signal line 67, and forms a third reactance factor composed of an MIM capacitor. The fourth insulating film 264 is interposed between the fourth bridge conductor 164 and the signal line 67, and constitutes a fourth reactance factor composed of an MIM capacitor.

FIG. 2 is an equivalent circuit diagram of the transmission line having a periodic structure according to the embodiment of the present invention shown in FIG. At a constant pitch d, a reactance factor jb is periodically loaded on the transmission line in a shunt. The characteristic impedance of the microstrip line wiring width of the signal line, the thickness of the dielectric substrate, the characteristic impedance is determined by the relative dielectric constant epsilon _r of the dielectric substrate. When it is desired to reduce the thickness of the dielectric substrate, there is a case where the wiring width of the signal line must be reduced in order to realize a desired characteristic impedance. in this case,
Conductor loss increases. By using the impedance converter according to the embodiment of the present invention, it is possible to increase the characteristic impedance while keeping the signal line width wide.
That is, in FIG. 5, if the frequency band of the _second pass band P2 is designed to be the applicable frequency, a high characteristic impedance can be realized using a wide wiring. Conversely, when trying to realize a low impedance, if the Bloch impedance of the higher-order pass bands P ₃ , P ₄ ,... Shown in FIG. Desired characteristics can be realized without widening.

In FIG. 2, the reactance component jb is loaded in the shunt with respect to the transmission line. However, the present invention may be applied to any transmission line having a periodic structure. good. Also, the characteristic impedance of the transmission line may be periodically changed using a distributed constant reactance component instead of the lumped constant.

In the description, only a single periodicity is shown, but a transmission line having multiple periods may of course be used. For example, a transmission line having a double period can be realized by alternately loading different reactance components on the transmission line. In this case, a sub-band occurs, and the width of the band gap changes depending on the difference between the values of the different reactance components.

Furthermore, the Bloch impedance can be dynamically adjusted if the reactance component is constituted by a capacitor having a three-dimensional microstructure such as MEMS or a varactor diode, and the reactance component is made variable by a control signal.

For example, when a capacitor having a MEMS structure is used, a bridge conductor can be formed above the strip conductor. When the potential difference between the bridge conductor and the strip conductor is changed, the interval between the bridge conductor and the strip conductor changes due to Coulomb force. As a result, the magnitude of the loaded reactance component changes, so that a function equivalent to the case of using a capacitor having a MEMS structure can be realized. Since the reactance component only needs to be variable by the control signal, the loaded element may have a structure other than the MEMS capacitor or the varactor diode.

(Application to MMIC) The microwave circuit according to the embodiment of the present invention shown in FIG. 8 includes a first transistor (first active element) Tr1 and a second transistor (second active element) This is an MMIC on which a two-stage high-frequency amplifier having Tr2 is mounted. Specifically, from the input terminal 81 (RFin) to the output terminal 86 (RFout)
Between the coupling capacitor C1 and the first transistor Tr
1, coupling capacitor C4, second transistor Tr2,
A high-frequency transmission line is formed by the path of the coupling capacitor C7. Then, an RF signal is input from the input terminal 81, propagates through the high-frequency transmission line, and is output from the output terminal 86. Between the coupling capacitor C1 and the input terminal 81, the open stub impedance Z _s for adjusting the impedance of the RF transmission line is provided.
The source of the first transistor Tr1 is grounded, and the gate receives a gate voltage Vg1 from a DC bias electrode 82 via a bypass capacitor (decoupling capacitor) C2 and an impedance Z _g1 for separating DC and high frequency. It is configured to be able to supply. The drain of the first transistor Tr1 includes a bypass capacitor C3 for separating direct current and high frequency and an impedance Z.
It is configured such that the drain voltage Vd1 can be supplied from the DC bias electrode 84 via _d1 . Similarly, the gate of the second transistor Tr2 is supplied with the gate voltage Vg2 from the DC bias electrode 83 via the bypass capacitor C5 and the impedance _Zg2, and the drain of the second transistor Tr2 is connected to the bypass capacitor C6. and via an impedance Z _d2, the drain voltage Vd2 is configured to be supplied from the DC bias electrode 85. The source of the second transistor Tr2 is grounded.

Thus, the high-frequency signal input from the input terminal 81 is input to the first transistor Tr1 through the coupling capacitor C1, and is first amplified. The amplified high-frequency signal is input to the second transistor Tr2 through the coupling capacitor C4, is further amplified, passes through the coupling capacitor C7, and is output from the output terminal 86 to the outside.
In FIG. 8, Z _B11 , Z _B12 , Z _B21 , Z _B22 , Z _B31 , Z
_B32 , for the basic transmission line as shown in FIG.
This is an impedance matching circuit having a structure in which reactance components are periodically loaded at a constant pitch d, and each has a predetermined Bloch impedance component.

These first transistor (first active element) Tr1 and second transistor (second active element)
Tr2, impedance matching circuit, bias circuit, etc.
Dielectric substrate (semiconductor substrate) such as GaAs or InP
FIG. 9 is a schematic plan view in the case where an MMIC is configured by being integrated on the MMIC 1. In FIG. 9, for the first transistor (first active element) Tr1 and the second transistor (second active element) Tr2, for example, HEMT can be used. The first transistor Tr1 has a first main electrode (source ohmic electrode), a second main electrode (drain ohmic electrode), and a control electrode (gate electrode) disposed on the dielectric substrate 1. Similarly, the second transistor Tr2 has a first main electrode (source ohmic electrode), a second main electrode (drain ohmic electrode), and a control electrode (gate electrode) disposed on the dielectric substrate 1.

Further, the microwave circuit according to the embodiment of the present invention comprises a first ground pattern 72 and a second ground pattern 7 which are opposed to each other on a dielectric substrate (semiconductor substrate) 1.
Four. The input side signal line 68 connected to the control electrode of the first transistor Tr1 is sandwiched between the first ground pattern 72 and the second ground pattern 74. The source ohmic electrode of the first transistor Tr1 is divided into two regions with the gate electrode lead-out portion of the first transistor Tr1 having a T shape as a plane pattern interposed therebetween. The two source ohmic electrodes are connected to the first ground pattern 72, respectively.
And the second ground pattern 74 and are grounded. Similarly, the output signal line 69 connected to the second main electrode of the first transistor Tr1 is connected to the dielectric substrate 1
Above, it is sandwiched between the first ground pattern 72 and the second ground pattern 74. The first ground pattern 72 and the second ground pattern 74 and the input signal line 68 of the first transistor Tr1 sandwiched between the first ground pattern 72 and the second ground pattern 74 form the first An input side coplanar line of the transistor Tr1 is formed. Similarly, a first ground pattern 72 and a second ground pattern 74 and a first transistor T sandwiched between the first ground pattern 72 and the second ground pattern 74
The output side signal line 69 of r1 forms an output side coplanar line of the first transistor Tr1.

Further, the input signal line 70 connected to the control electrode of the second transistor Tr2 is also sandwiched between the first ground pattern 72 and the second ground pattern 74. Similarly to the first transistor Tr1, the source ohmic electrode of the second transistor Tr2 is divided into two regions with the gate electrode lead-out portion of the second transistor Tr2 having a T-shaped planar pattern interposed therebetween. ing. The two source ohmic electrodes are connected to the first ground pattern 72 and the second ground pattern 74, respectively, and are grounded. Similarly, the output signal line 73 connected to the second main electrode of the second transistor Tr2 is sandwiched between the first ground pattern 72 and the second ground pattern 74 on the dielectric substrate 1. ing. The first ground pattern 72 and the second ground pattern 74, and the first ground pattern 72 and the second ground pattern
Transistor Tr sandwiched by the ground pattern 74 of FIG.
2 input-side signal line 70, the second transistor T
An input side coplanar line of r2 is formed. Similarly,
The first ground pattern 72 and the second ground pattern 74, and the output signal line 73 of the second transistor Tr2 sandwiched between the first ground pattern 72 and the second ground pattern 74, An output-side coplanar line of the transistor Tr2 is formed.

The coupling capacitors C1, C4 and C7 shown in FIG. 9 are constituted by MIM capacitors. Similarly,
The bypass capacitors C2, C3, C5 and C6 are also constituted by MIM capacitors. Coupling capacitors C1, C
4, C7 also functions as an element of the high-frequency transmission line at the same time. The intermediate signal line 6 is connected to the input side signal line 68 of the first transistor Tr1 via the MIM capacitor C1.
7 is connected, and an input terminal 81 is connected to the intermediate signal line 67. A first ground pattern 72 and a second ground pattern 7 are placed at a certain distance on both sides of the intermediate signal line 67.
4, the input-side coplanar line of the first transistor Tr1 is formed to be extended. First transistor T
r1 output-side coplanar line and second transistor Tr
A connection coplanar line is configured by the two input-side coplanar lines. An MIM capacitor C4 is inserted between the output signal line 69 of the first transistor Tr1 and the input signal line 70 of the second transistor Tr2. Further, an intermediate signal line 76 is connected to the output-side signal line 73 connected to the drain of the second transistor Tr2 via the MIM capacitor C7. The output terminal 86 is connected to the intermediate signal line 76. Intermediate signal line 7
The first ground pattern 7 is also provided at a certain distance on both sides of the first ground pattern 7.
The second and second ground patterns 74 are arranged to form a coplanar line.

In the impedance matching circuit Z _{B11 according} to the embodiment of the present invention shown in FIG.
Four bridge conductors 161, 16 arranged periodically
2,163,164 pass over the intermediate signal line 67,
The first ground pattern 72 and the second ground pattern 74 are short-circuited. As shown in FIG. 1, the first insulating film 261 is interposed between the first bridge conductor 161 and the intermediate signal line 67, and constitutes a first reactance factor composed of an MIM capacitor. Similarly, a second insulating film 262 is interposed between the second bridge conductor 162 and the intermediate signal line 67 to form a second reactance factor composed of an MIM capacitor. Third bridge conductor 1
A third insulating film 263 is provided between the intermediate signal line 63 and the intermediate signal line 67.
Are interposed to constitute a third reactance factor composed of an MIM capacitor. And the fourth bridge conductor 1
A fourth insulating film 264 is provided between the intermediate signal line 64 and the intermediate signal line 67.
Are interposed to constitute a fourth reactance factor composed of an MIM capacitor. Thus, the input-side coplanar line of the first transistor Tr1 and the first coplanar line that is periodically arranged at a constant pitch with respect to the input-side coplanar line.
To an impedance matching circuit Z _B11 composed of a predetermined Bloch impedance component by the fourth reactance factor
Is configured. That is, the impedance matching circuit Z
_{In B11} , the Bloch impedance is adjusted to a desired value by selecting the pitch of the arrangement of the first to fourth reactance factors and the reactance value exhibited by the capacitor. Furthermore, four bridge conductors 161, 162,
The 163 and 164 also have a function of making the first metal layers 72 and 74, which serve as ground patterns on both sides of the coplanar line, electrically equal in potential to each other.

Similarly, impedance matching circuit Z
_B12 consists of five bridge conductors 171, 172, 173, 174, 175 periodically arranged at a constant pitch,
The first ground pattern 7 passes over the input side signal line 68 and
The second and the second ground patterns 74 are short-circuited, and a reactance factor including an MIM capacitor is formed between the second and the ground patterns 74 and the input-side signal line 68, respectively. That is, the input-side coplanar line of the first transistor Tr1 and the five bridge conductors 171, 172, 173, and 17 periodically arranged at a constant pitch with respect to the input-side coplanar line.
The first to fifth reactance factor corresponding to 4,175, the impedance matching circuit Z _B12 having a predetermined Bloch impedance component is configured. That is, in the impedance matching circuit _ZB12 , the Bloch impedance is adjusted to a desired value by selecting the pitch of the arrangement of the reactance factors and the reactance value exhibited by the capacitor. Furthermore, five bridge conductors 17
1, 172, 173, 174, and 17 also have the function of electrically setting the first metal layers 72 and 74, which are the ground patterns on both sides of the coplanar line, to the same electric potential as described above.

Further, the impedance matching circuits Z _B21 , Z _B
Similarly, in _B22 , _ZB31 , and _ZB32 , the first ground pattern 72 and the second ground pattern 74 are short-circuited by bridge conductors periodically arranged at a constant pitch. That is, the output signal line 69, the input signal line 70,
Between the output side signal line 73 and the intermediate signal line 76, a plurality of reactance factors, each of which is periodically loaded at a constant pitch and composed of an MIM capacitor, are formed. The Bloch scattering is caused by the plurality of reactance factors loaded periodically and the output-side coplanar line of the first transistor Tr1, the input-side coplanar line of the second transistor Tr2, and the output-side coplanar line of the second transistor Tr2. Are given, and impedance matching circuits Z _B21 , Z _B22 , Z _B31 , and Z _{B32 each} having a predetermined Bloch impedance component are configured. That is, in Z _B21 , Z _B22 , Z _B31 , and Z _B32 , each Bloch impedance is adjusted to a desired value by selecting the pitch of the arrangement of each reactance factor and the reactance value exhibited by each capacitor.

[0075] Impedance according to the embodiment of the present invention
Matching circuit Z_B11, Z_B12, Z_B21, Z_{B twenty two}, Z_B31, Z
_B32Is the independent variable that determines its characteristic impedance
And a plurality of bridge conductors and the intermediate signal line 67,
Signal line 68, output side signal line 69, input side signal line 70, output
Formed between the side signal line 73 and the intermediate signal line 76, respectively.
Reactance component determined by the capacitor value
The pitch d of the bridge conductor arrangement is
Added to the factors that determine dance. In other words, yellowtail
Width of bridge conductor, between bridge conductor and each signal line
Dielectric constant, thickness of insulating film,
The pitch of the array of bridge conductors
Circuit Z_B11, Z_B12, Z_B21, Z_B22, Z_B31, Z_B32Bu
In addition to the factors that determine Loch impedance
And the impedance matching circuit Z_B11, Z _B12, Z_B21, Z
_B22, Z_B31, Z_B32Independent variable that determines the impedance of
The number is dramatically increased, and the adjustment range is large. For example,
The bridge conductor is 3μm thick, gold (Au) metal pattern
Can be used, and the width of the bridge conductor is
What is necessary is just to select according to an actance value. For example,
Select a conductor width within the range of 10 to 50 μm.
It is possible.

If the MMIC has an operating frequency of about 60 GHz, the signal lines 68 to 70, 7 constituting the coplanar line are used.
The width of 3,76 may be selected to be about 20 μm. And
Approximately 15 on each side of these signal lines 68-70, 73, 76.
The first of about 250 to 500 μm width at a distance of μm
The ground pattern 72 and the second ground pattern 74 may be arranged. The signal lines 68 to 70, 73, and 76, the first ground pattern 72, and the second ground pattern 74 are made of a gold (Au) thin film having a thickness of 0.1 to 3 μm. If the dielectric substrate (semiconductor substrate) 1 is a semi-insulating substrate, the gold (Au) thin film may be directly deposited on the semi-insulating substrate. If the dielectric substrate (semiconductor substrate) 1 is a conductive substrate, a silicon oxide film (Si) is formed on the conductive substrate.
An insulating film such as an O ₂ film) or a silicon nitride film (Si ₃ N ₄ film) is deposited, and the signal lines 68 to 70, 73,
It is sufficient to deposit a gold (Au) thin film constituting the first ground pattern 76 and the first ground pattern 72 and the second ground pattern 74.

As shown in FIG. 9, the second transistor T
The output side DC bias stub wiring Z _d2 connected to the drain of r2 is connected to the DC bias electrode 85 for short-circuiting the high frequency by the MIM capacitor C6 and supplying the drain voltage Vd2. Although not shown, the output-side DC bias stub wiring Z _d2 includes, for example, a meander line (signal line) formed on the surface of the thin film dielectric layer 302, the thin film dielectric layer 302, and the second ground pattern. 74 and a thin-film microstrip line. The input-side DC bias stub wiring Z _g2 connected to the gate of the second transistor Tr2 is short-circuited to a high frequency by the MIM capacitor C5 and connected to the DC bias electrode 83 for supplying the gate voltage Vg2. I have. The input-side DC bias stub wiring Z _g2 is, like the output-side DC bias stub wiring Z _d2 , a thin-film dielectric layer 3.
01, a thin film microstrip line composed of a meander line (signal line), a thin film dielectric layer 301, and a first ground pattern 72. The output side DC bias stub wiring Z _d1 connected to the drain of the first transistor Tr1 is connected to the DC bias electrode 84 for short-circuiting the high frequency by the MIM capacitor C3 and supplying the drain voltage Vd1. . The output side DC bias stub wiring Z _d1 also includes a meander line (signal line) formed on the surface of the thin film dielectric layer 302 and the thin film dielectric layer 3.
02 and the second ground pattern 74. The input side DC bias stub wiring Z _g1 connected to the gate of the first transistor Tr1 is connected to the DC bias electrode 82 for short-circuiting the high frequency by the MIM capacitor C2 and supplying the gate voltage Vg1. . The input side DC bias stub wiring Z _g1 is formed by a meander line (signal line) formed on the surface of the thin film dielectric layer 301, the thin film dielectric layer 301, and the first thin film dielectric layer 301.
And a thin film microstrip line composed of the ground pattern 72 described above.

[0078] The intermediate signal line 67 is further connected to the input terminal 81, the open stub line Z _s as an impedance adjustment stub line is connected. Stub wiring for impedance adjustment (open stub wiring) Z
_s also the second metal layer Z _s and the thin-film dielectric layers 301, 302
And a second ground pattern 74 (not shown in cross section).

The MIM capacitor C1 and the open stub wiring _Zs also function as an input impedance matching circuit for the first transistor Tr1. Also, DC bias stub wirings Z _s , Z composed of thin film microstrip lines
_{g1, Z d1, Z g2,} Z d2 plays at the same time, part of the role of the impedance matching circuit. Therefore, the impedance matching circuits Z _B11 , Z _B12 , Z _B21 , Z _B22 , Z _B31 , Z
A part of _B32 may be omitted. Alternatively, the open stub wiring _Zs or the DC bias stub wiring _Zs , Z
_g1, Z _d1, Z _g2, a part of the Z _d2, may be configured by a circuit consisting of the Bloch impedance component having a periodic structure of the present invention. Input terminal 81, DC bias electrode 82 to
Reference numeral 85 and the output terminal 86 are bonding pads, which are connected to corresponding pins of the package via gold (Au) or aluminum (Al) bonding wires, respectively.

As shown in FIG. 9, in the embodiment of the present invention,
Such MMICs have thin film masks according to dimensional accuracy and occupied area.
Use of cross strip track and coplanar track
Impedance, the impedance of Bloch impedance component
Matching circuit Z_B11, Z_B12, Z _B21, Z_B22, Z_B31, Z_B32
Achieved circuit miniaturization and high performance by using
are doing. That is, the high frequency gain is improved,
Performance can be improved. And the implementation of the present invention
Information terminal such as a portable wireless device using the MMIC according to the embodiment
No need to generate unnecessary power
This leads to an increase in battery life. In addition, at the base station
However, it is possible to improve the linearity by the loss.
Frequency utilization efficiency.
You.

(Other Embodiments) Although the present invention has been described with the above embodiments, it should not be understood that the description and drawings constituting a part of this disclosure limit the present invention. From this disclosure, various alternative embodiments, examples, and operation techniques will be apparent to those skilled in the art.

In the description of the embodiment already described,
The high-frequency active element and the passive element are exemplified by the MMIC integrated on the semiconductor substrate 1 such as GaAs or InP. However, the dielectric substrate of the present invention may be a ceramic substrate such as alumina, aluminum nitride (AlN), and beryllia (BeO), or may be an MHIC in which high-frequency active elements and passive elements are integrated on the ceramic substrate.

FIG. 1 shows a structure in which a reactance factor jb is periodically loaded on a coplanar line at a constant pitch d. Similarly, in the case of a microstrip line,
The reactance factor jb can be periodically loaded at a constant pitch d. For example, if openings are provided periodically at a constant pitch d along the direction of the signal line in the metal ground pattern opposed to the signal line via the dielectric substrate, the reactance factor jb can be loaded periodically. . Alternatively, the signal line and the metal ground pattern may be connected by an inductor via a via (through hole) provided in the dielectric substrate. Furthermore,
A fine MIM capacitor or the like may be periodically loaded on the microstrip line at a constant pitch d.

As described above, the present invention naturally includes various embodiments and the like not described herein. Therefore, the technical scope of the present invention is determined only by the invention specifying matters according to the claims that are appropriate from the above description.

[0085]

As described above, according to the present invention, a microwave circuit having a passive element structure having flat characteristics over a wide band can be obtained.

Further, according to the present invention, the same characteristic impedance as that of the conventional transmission line can be realized by making the wiring width of the signal line wider than that of the conventional transmission line. A circuit can be configured.

Further, according to the present invention, it is possible to provide a microwave circuit which has a greater degree of freedom in design, is smaller than conventional ones, and has low loss.

Further, according to the present invention, it is possible to provide a microwave circuit capable of expanding the adjustable range of the characteristic impedance of the high-frequency transmission line.

Further, according to the present invention, it is possible to provide a microwave circuit capable of reducing crosstalk with low loss.

Further, according to the present invention, it is possible to provide a microwave circuit capable of reducing an occupied area required for wiring as a whole.

Further, according to the present invention, it is possible to provide a microwave circuit whose electric characteristics at a high frequency can be easily adjusted.

[Brief description of the drawings]

FIG. 1 is a bird's-eye view of an impedance converter according to an embodiment of the present invention.

FIG. 2 is an equivalent circuit diagram of a transmission line having a periodic structure according to the embodiment of the present invention.

FIG. 3 is a band diagram of a transmission line having a periodic structure according to the embodiment of the present invention.

FIG. 4 is a diagram showing a relationship between a band gap start point and a reactance factor in a transmission line having a periodic structure according to an embodiment of the present invention.

FIG. 5 is a diagram showing y = kd dependence of Bloch impedance of a transmission line having a periodic structure according to the embodiment of the present invention.

FIG. 6 is a diagram showing y = kd dependence of input impedance of a transmission line having a periodic structure according to the embodiment of the present invention.

FIG. 7 is a diagram showing y = kd dependence of a transmission characteristic of a transmission line having a periodic structure according to the embodiment of the present invention.

FIG. 8 is an equivalent circuit diagram of the MMIC according to the embodiment of the present invention.

9 is a plan view of an MMIC in which the circuit configuration shown in FIG. 8 is integrated on a dielectric substrate (semiconductor substrate).

[Explanation of symbols]

Reference Signs List 1 dielectric substrate 67 signal line (intermediate signal line) 68 input-side signal line 69 output-side signal line 70 input-side signal line 72 first ground pattern 73 output-side signal line 74 second ground pattern 76 intermediate signal line 81 input Terminals 82 to 85 DC bias electrode 86 Output terminals 161 to 164, 171 to 175 Bridge conductors 261 to 264 Insulating film 301, 302 Thin film dielectric layer jb Periodic reactance factor C1, C4, C7 Coupling capacitors C2, C3, C5, C6 bypass capacitor Tr1 first transistor (first active element) Tr2 second transistor (second active _{element) Z B11, Z B12, Z} B21, Z B22, Z B31, Z B32 impedance matching circuit Z _g1, Z _g2 input side DC bias stub line Z _d1, Z _d2 output side DC bias stub line Z _s open stub Line

Claims (6)

[Claims] 1. A basic transmission line designed by a low frequency limit approximation, and a plurality of reactance components having the same reactance value, which are periodically loaded at a constant pitch with respect to the basic transmission line, A microwave circuit comprising a passive element structure in which a characteristic impedance is set to a Bloch impedance different from a value specific to the basic transmission line by selecting the pitch and the reactance value. 2. The microwave circuit according to claim 1, wherein said reactance value is variable by a control signal. 3. The pitch is d, and the wave number of a microwave propagating through the basic transmission line when there is no periodically loaded reactance component is k, and y is the product of the wave number k and the pitch. = Kd, the y value dependency of the Bloch impedance has a plurality of pass bands whose order increases sequentially from a first pass band with an increase in the y value, and the Bloch impedance of the passive element structure 3. The microwave circuit according to claim 1, wherein the pitch is selected so that the impedance is an impedance of a higher-order pass band equal to or higher than the second pass band. 4. The basic transmission line includes a dielectric substrate, a signal line disposed on a surface of the dielectric substrate, and first and second lines disposed on both sides of the signal line at a predetermined distance from the signal line. A coplanar line including a second ground pattern, wherein each of the reactance components is the first and second
4. A bridge conductor for connecting said ground pattern to each other, and an insulating film arranged to form a capacitor between said bridge conductor and said signal line. The microwave circuit according to the item. 5. A dielectric substrate, first and second ground patterns disposed on the dielectric substrate to face each other, and sandwiched between the first and second ground patterns on the dielectric substrate. An active element having first and second main electrodes and a control electrode disposed thereon; and an input interposed between the first and second ground patterns on the dielectric substrate and connected to the control electrode. A side signal line, an output side signal line disposed on the dielectric substrate and connected to the second main electrode and sandwiched between the first and second ground patterns, and an input side signal line. A plurality of bridge conductors that pass over and periodically connect the first and second ground patterns to each other at a constant pitch; and a capacitor between the plurality of bridge conductors and the input signal line. Multiple cavities each arranged to make up A microwave circuit comprising an edge film, wherein the input impedance of the active element is adjusted to a desired value by selecting the pitch and a reactance value exhibited by the capacitor. 6. A dielectric substrate, first and second ground patterns opposed to each other on the dielectric substrate, and sandwiched between the first and second ground patterns on the dielectric substrate. An active element having first and second main electrodes and a control electrode disposed thereon; and an input interposed between the first and second ground patterns on the dielectric substrate and connected to the control electrode. A side signal line, an output side signal line disposed between the first and second ground patterns on the dielectric substrate and connected to the second main electrode, and an output side signal line. A plurality of bridge conductors that pass over and periodically connect the first and second ground patterns at a constant pitch; and a capacitor between the plurality of bridge conductors and the output signal line. Multiple cavities each arranged to make up A microwave circuit comprising an edge film, wherein the output impedance of the active element is adjusted to a desired value by selecting the pitch and the reactance value exhibited by the capacitor.