JP2002027747A - Switching power supply - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve a power factor and AC/DC conversion efficiency. SOLUTION: In a switching power supply circuit provided with a power factor improving circuit for a compound resonance converter, voltage resonant pulse voltages generated in a primary side voltage resonance converter are divided by two couples of parallel resonance capacitors, and voltage feedback is performed to a power factor improving rectifier means through capacitance coupling. Consequently, a power factor is improved up to, e.g. about 0.9 and simultaneously improvement of AC/DC conversion efficiency and reduction of ripple component of a DC output voltage are realized. A charging current to a filter means is divided by a first rectifier circuit and a second rectifier circuit, so that the operation range of zero voltage switching is not narrowed when the power factor is improved.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit having a power factor improving function.

[0002]

2. Description of the Related Art The applicant has previously proposed various power supply circuits having a resonance type converter on the primary side. Further, various power supply circuits including a power factor improving circuit for improving the power factor of the resonant converter have been proposed.

FIGS. 8 to 11 are circuit diagrams showing various examples of a power factor improving circuit in a switching power supply circuit constructed based on the invention previously filed by the present applicant.
9, 10, and 11 show only the power factor correction circuit portion, and the other portions are the same as those in FIG. 8. FIG. 8 shows a power factor improving circuit 20 of a capacitive coupling type of a capacitor voltage dividing system.
9 is a circuit example of a magnetic coupling type power factor improvement circuit 20b of a capacitor voltage dividing type, FIG. 10 is a circuit example of a magnetic coupling type power factor improvement circuit 20c of a tertiary winding type, and FIG. Circuit examples of a tertiary winding type diode-coupled power factor correction circuit 20d are shown.

First, an example of the entire switching power supply circuit will be described with reference to FIG. This power supply circuit has a configuration in which a power factor improving circuit 20a for improving a power factor is provided for a self-excited voltage resonance type switching converter.

In the power supply circuit shown in FIG. 1, a line filter transformer LFT and an across capacitor CL are provided for a commercial AC power supply AC. Further, a bridge rectifier circuit Di for performing full-wave rectification of the commercial AC power supply AC is provided. The rectified output rectified by the bridge rectification circuit Di is passed through the power factor correction circuit 20 to the smoothing capacitor Ci.
And a rectified smoothed voltage Ei is obtained across the smoothing capacitor Ci.

The configuration of the power factor improving circuit 20a will be described later, and first, the configuration of the voltage resonance type converter will be described. Here, the voltage resonance type converter employs a self-excited configuration including one switching element Q1.

[0007] The base of the switching element Q1 is connected to the positive electrode of a smoothing capacitor Ci (rectified smoothing voltage Ei) via a starting resistor RS and a base current limiting resistor RB.
The base current at the time of starting is obtained from the rectifying and smoothing line. A self-excited oscillation driving resonance circuit (self-excited oscillation drive circuit) including a series connection circuit of a drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB is provided between the base of the switching element Q1 and the primary side ground. Connected.
The clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is off. The collector of the switching element Q1 is connected to the primary winding N1.
A smoothing capacitor Ci through a series connection of the detection windings ND
Is connected to the positive electrode terminal. The emitter is grounded to the primary side ground.

A parallel resonance capacitor Cr is connected to the collector of the switching element Q1. The parallel resonance capacitor Cr forms a primary parallel resonance circuit of the voltage resonance type converter by its own capacitance and a leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT described later. When the switching element Q1 is turned off, the operation of the parallel resonance circuit causes the voltage across the resonance capacitor Cr to be actually a sinusoidal pulse waveform, thereby obtaining a voltage resonance type operation.

The orthogonal control transformer PRT has a detection winding N
D is a saturable reactor on which a drive winding NB and a control winding NC are wound. The orthogonal transformer PRT drives the switching element Q1 and is provided for constant voltage control. As the structure of the orthogonal control transformer PRT, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. Then, the detection winding ND and the drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is further connected to the detection winding ND, It is configured by being wound in a direction orthogonal to the drive winding NB.

In this case, the detection winding ND of the orthogonal control transformer PRT (frequency variable means) is connected in series with a primary winding N1 of an insulation converter transformer PIT, which will be described later, so that the switching of the switching element Q1 is performed. The output is transmitted to the detection winding ND via the primary winding N1.
In the orthogonal control transformer PRT, the switching output obtained on the detection winding ND is excited by the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is generated on the drive winding NB. . This drive voltage is applied to a series resonance circuit (NB,
CB) is output to the base of the switching element Q1 as a drive current via the base current limiting resistor RB.
As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit (NB, CB).

One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, and the other end is connected to the positive electrode (rectified smoothing voltage Ei) of the smoothing capacitor Ci.

On the secondary side of the insulated converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage excited by the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is also provided with a parallel resonance circuit for obtaining the voltage resonance operation. . In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

In this case, a half-wave rectification circuit is provided for the secondary parallel resonance circuit formed as described above by connecting a rectifier diode DO1 and a smoothing capacitor CO1 as shown in the figure. A DC output voltage EO1 is generated. The DC output voltage EO1 is also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1
Is used as a detection voltage to control the resonance frequency for switching of the switching element Q1, thereby performing constant voltage control.

That is, the control circuit 1 varies the level of the control current (DC current) flowing through the control winding NC according to the change in the secondary-side DC output voltage level (EO1), thereby providing the orthogonal control transformer PRT with a control signal. The inductance LB of the wound drive winding NB is variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes. This is the switching element Q1
This operation has the effect of stabilizing the secondary-side DC output voltage.

In the power factor correction circuit 20a, a choke coil Ls and a high-speed recovery type diode D1 are inserted in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The filter capacitor CN is provided in parallel with the series connection circuit of the choke coil Ls and the high-speed recovery type diode D1, thereby forming a normal mode low-pass filter together with the choke coil Ls.

The parallel resonance capacitor C10 is provided in parallel with the high speed recovery type diode D1. Although a detailed description is omitted here, for example, the parallel resonance capacitor C10 forms a series resonance circuit together with, for example, the choke coil Ls. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced. In the power factor improving circuit 20a, the above-described parallel resonance capacitor Cr is connected to a connection point between the choke coil Ls, the anode of the high-speed recovery type diode D1, and the parallel resonance capacitor C10, so that the primary The switching output (voltage resonance pulse voltage) obtained in the side parallel resonance circuit is fed back.
That is, in the configuration of the power factor improvement circuit 20a shown in this figure, the switching output obtained in the primary side parallel resonance circuit is fed back to the rectification current path via the capacitive coupling of the parallel resonance capacitor Cr. As described above, the parallel resonance capacitor Cr forming the primary side parallel resonance circuit of the primary side voltage resonance type converter is connected to the anode of the high-speed recovery type diode D1 of the power factor correction circuit 20a. Capacitor Cr and parallel resonance capacitor C10
Are connected in series, that is, the voltage resonance pulse voltage that appears as the voltage across the parallel resonance capacitor Cr is changed between the parallel resonance capacitor Cr and the parallel resonance capacitor C10.
Is divided by the capacitance ratio of. Then, a circuit system is formed as a voltage feedback system in which a voltage is fed back to the smoothing capacitor Ci via a parallel resonance capacitor C10 connected in parallel with the high-speed recovery type diode D1.

With this circuit configuration, for example, the primary-side voltage resonance pulse voltage Vcp = 600V is divided into about 3: 1 by the primary-side parallel resonance capacitors Cr and C10, and a 150V high-frequency sinusoidal pulse voltage is fed back. ing. At a time near the positive and negative peaks of the AC input voltage VAC, the fast recovery type diode D1 conducts, and a steep pulse charging current charges the smoothing capacitor Ci from the AC input power supply AC. At times other than around the positive and negative peaks of the AC input voltage VAC, the high speed recovery type diode D1 repeats the switching operation by the pulse voltage which is fed back, and when the high speed recovery type diode D1 is off, the parallel resonance capacitor Cr, the inductance LS and the capacitor When a parallel resonance current flows by C N and the fast recovery type diode D1 is on,
A high-frequency charging current flows from the AC input power supply AC to the smoothing capacitor Ci via the inductance LS. With this operation, the conduction angle of the AC input current IAC is increased, and the power factor can be improved.

FIG. 9 shows a magnetic coupling type power factor improving circuit 20b which employs the capacitor voltage dividing system as in FIG. In the power factor correction circuit 20b, a high-speed recovery type diode D1-choke coil Ls is inserted in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The filter capacitor CN is provided in parallel with the series connection circuit of the high-speed recovery type diode D1 and the choke coil Ls, thereby forming a normal mode low-pass filter together with the choke coil Ls.

The parallel resonance capacitor C10 is provided in parallel with the choke coil Ls, and forms a parallel resonance circuit with the choke coil Ls. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced. Also, for the power factor improvement circuit 20b,
The parallel resonance capacitor Cr is connected to the connection point of the cathode of the high-speed recovery type diode D1, the choke coil Ls, and the parallel resonance capacitor C10, and the switching output (voltage resonance pulse voltage) obtained in the primary side parallel resonance circuit is obtained. ) To be returned.

That is, in the configuration of the power factor improvement circuit 20b shown in FIG. 2, the switching output obtained in the primary side parallel resonance circuit is supplied to the rectified current via the inductive reactance (magnetic coupling) assumed to be included in the choke coil Ls itself. Returning to the route. 8, the parallel resonance capacitor Cr and the parallel resonance capacitor C10 are connected in series, so that the voltage resonance pulse voltage is reduced by the parallel resonance capacitor C10.
A circuit system is formed as a voltage feedback system in which the voltage is divided and fed back by the capacitance ratio of r and C10.

In this case, the fast recovery type diode D1 conducts during the time near the positive and negative peaks of the AC input voltage VAC, and a steep pulse charging current charges the smoothing capacitor Ci from the AC input power supply AC. When the AC input voltage VAC becomes low, the high speed recovery type diode D1 is turned off, and the primary side voltage resonance pulse voltage divided by the parallel resonance capacitors Cr and C10 is subjected to voltage resonance by the parallel resonance circuit of the choke coil Ls and the parallel resonance capacitor C10. Occurs.
Due to this voltage resonance, a sine-wave pulse voltage is superimposed on the potential V2 of the cathode of the high-speed recovery diode D1, and the anode potential V1 of the high-speed recovery diode D1 is superimposed.
, The high-speed recovery type diode D1 repeats the switching operation. When the high-speed recovery type diode D1 is turned on, a charging current flows from the capacitor CN to the smoothing capacitor Ci. With this operation, the conduction angle of the AC input current IAC is increased, and the power factor is improved.

FIG. 10 is a circuit example of a tertiary winding type magnetic coupling type power factor improving circuit 20c. In the power factor correction circuit 20c, a high-speed recovery type diode D1 and a choke coil LS are connected in series and inserted between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The filter capacitor CN is inserted between the anode side of the high-speed recovery type diode D1 and the positive terminal of the smoothing capacitor Ci, so that the choke coil L
A normal mode low-pass filter is formed together with s.

For the power factor correction circuit 20c, the insulation converter transformer PI is connected to the connection point between the cathode of the high-speed recovery type diode D1 and the choke coil LS.
The tertiary winding N3 of T is connected via the series resonance capacitor C3, so that the switching output voltage (voltage resonance pulse voltage) obtained in the primary side parallel resonance circuit is fed back. That is, a tertiary winding N3 is formed by winding up the primary winding N1 of the insulating converter transformer PIT to generate a negative-pulse voltage resonance pulse voltage, and a magnetic coupling type power factor improvement circuit is connected via the series resonance capacitor C3. The voltage is fed back to 20c.

In this method, the pulse voltage generated in the tertiary winding N3 is determined by the series resonance capacitor C3 and the choke coil L
Voltage resonance occurs due to the series resonance circuit of s. Due to this voltage resonance, when the AC input voltage VAC is high, the high-speed recovery type diode D1 performs a switching operation. When the high-speed recovery type diode D1 is on, a charging current flows from the capacitor CN to the smoothing capacitor Ci, and the AC input current IAC
And the power factor is improved. When the AC input voltage VAC is low, the voltage V1 <V2, and the fast recovery type diode D1 is turned off, and the smoothing capacitor C
No charging current flows to i.

FIG. 11 shows an example of a tertiary winding type diode-coupled power factor correction circuit 20d. Power factor improvement circuit 2
At 0d, the choke coil LS and the Schottky diode D1s are inserted in series between the positive output terminal of the bridge rectifier circuit Di and the positive terminal of the smoothing capacitor Ci. The filter capacitor CN is inserted in parallel with the series connection of the choke coil LS and the Schottky diode D1s, so that the choke coil L
A normal mode low-pass filter is formed together with s.

For the power factor improving circuit 20d, an insulation converter transformer PI is connected to a connection point between the anode of the Schottky diode D1s and the choke coil LS.
The tertiary winding N3 of T is connected via the series resonance capacitor C3, so that the switching output voltage (voltage resonance pulse voltage) obtained in the primary side parallel resonance circuit is fed back.

In this case, when the absolute value of the AC input voltage VAC is near the peak, the Schottky diode D1s conducts,
The charging flow I1 from the AC input power supply AC to the smoothing capacitor Ci is a choke coil Ls and a Schottky diode D
1s, but at the same time, the voltage resonance pulse voltage of the tertiary winding N3 is fed back to the series circuit of the series resonance capacitor C3 and the Schottky diode D1s, and by switching the Schottky diode D1s, the AC input current IAC is reduced. The conduction angle is increased and the power factor improving function is realized. When the absolute value of the AC input voltage VAC decreases, the Schottky diode D1s becomes non-conductive, and the voltage resonance pulse voltage of the tertiary winding N3 forms a series resonance circuit by the series circuit of the series resonance capacitor C3, the choke coil Ls, and the filter capacitor CN. I do.

As described above, four circuit examples are shown. Among them, the configuration having the highest AC / DC power conversion efficiency (η _{AC / DC} ) is shown in FIG. The characteristics of power conversion efficiency (η _{AC / DC} ) and power factor PF are shown in FIG.
As shown in FIG. FIG. 12 shows load power Po = 200 W to 40
FIG. 13 shows a change characteristic of the power factor PF and the AC / DC power conversion efficiency (η _{AC / DC} ) at the time of W. FIG. 13 shows the power factor PF and the AC / DC power conversion efficiency at the time of AC input voltage VAC = 80 V to 260 V. η _{AC / DC} ).

As can be seen from these figures, the power factor PF can be maintained at 0.7 or more, and an AC / DC power conversion efficiency (η _{AC / DC} ) of 90% or more can be realized in a wide range.

[0031]

However, the switching power supply circuit provided with the power factor correction circuits 20a to 20d as the prior art as described above has the following problems.

First, when the amount of voltage feedback to the power factor improvement circuit 20 is increased to increase the power factor to 0.8 or more in the state of the maximum load power, a stable operation condition of the primary side voltage resonance converter is obtained. Since the region where the zero voltage switching operation is not performed is expanded, it is impossible to improve the power factor to 0.8 or more.

For example, FIG. 14 shows the operation waveforms of the respective parts in the case of the circuit example shown in FIG. 11. In the vicinity of the positive and negative peak values of the AC input voltage VAC, the inductance Ls, the current from the Schottky diode D1s, Since the series resonance current of the winding N3 and the series resonance capacitor C3 is superimposed and flows, an excessive charging current flows as the current I1 to the smoothing capacitor Ci. That is, the currents ID1, ILS, IC3 and I1 have operation waveforms as shown in FIGS. 14 (d) (f) (h) (i). Therefore, if the number of turns of the tertiary winding N3 is increased to increase the amount of voltage feedback and improve the power factor PF,
The operating range of zero voltage switching, which is a stable operation condition of the switching element Q1 of the primary side voltage resonance converter, becomes narrow, and the operation mode becomes unstable with respect to fluctuations of the AC input voltage VAC and the additional power Po. Therefore, the power factor cannot be improved to 0.8 or more.

In the case of the power factor improvement circuits 20a, 20b, and 20c, the voltage factor is increased by changing the voltage division ratio of the parallel resonance capacitors Cr and C10 and the number of turns of the tertiary winding to improve the power factor. Then, an excessive current flows through the high-speed recovery type diode D1 in the vicinity of the positive and negative peak values of the AC input voltage VAC, and similarly, the operating range of the zero voltage switching is narrowed, and therefore, the power factor can be improved to 0.8 or more. could not.

Further, comparing the AC / DC power conversion efficiency before the power factor is improved and the AC / DC power conversion efficiency when the power factor is increased to about 0.8, for example, in the example of FIG. Although it is improved by about 3%, an expensive large-capacity Schottky diode D1s is required. When the maximum load power is 200 W or more and the AC input voltage VAC is 100 V,
According to the full-wave rectification method using a bridge rectifier diode, A
The limit of the C / DC power conversion efficiency was about 92%.

When the maximum load power is 200 W or more and the AC input voltage VAC is 100 V, according to the full-wave rectification method using the bridge rectifier diode, the high-speed recovery type diode D1 (or Schottky diode D1s) and the switching element ( Since the current flowing through the transistor (Q1) is large, the loss of the semiconductor is large and the heat generation is large. Therefore, the fast recovery type diode D1 (or Schottky diode D1s) and the switching element Q
A heat sink is required for the transistor 1. Further, since the ripple voltage of the DC output voltage Ei in the commercial power supply cycle is almost equal to that before the power factor improvement, the electrostatic capacitance of the smoothing capacitor Ci for smoothing the DC input voltage or the smoothing capacitor C01 for smoothing the DC output voltage. The capacity cannot be reduced, and the size cannot be reduced. For these reasons, it is disadvantageous for cost reduction and miniaturization.

[0037]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, a gap is formed between the smoothing means for smoothing the rectified current by two sets of smoothing capacitors connected in series and outputting a doubled DC input voltage, and a gap is formed so as to obtain a required coupling coefficient which is loosely coupled. An insulating converter transformer provided for transmitting the side output to the secondary side, and switching means adapted to intermittently output the doubled voltage DC input voltage by a switching element and to output to the primary winding of the insulating converter transformer, A primary-side resonance circuit formed by at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of a primary-side parallel resonance capacitor and making the operation of the switching means a voltage resonance type, and rectifying an AC power supply. A switch that supplies a rectified current to the smoothing means and is obtained by the primary-side resonance circuit. Power-factor improvement rectifier for improving the power factor by intermittently commutating a rectified current based on the returned switching output voltage, and a secondary winding of the insulating converter transformer. A secondary resonance circuit formed on the secondary side by the leakage inductance component of the wire and the capacitance of the secondary side resonance capacitor; and a secondary winding of the insulating converter transformer formed including the secondary side resonance circuit. DC voltage generating means configured to input the alternating voltage obtained in step (1) and perform a rectification operation to generate a secondary DC output voltage, and a secondary DC output voltage according to the level of the secondary DC output voltage. And a constant voltage control means configured to perform a constant voltage control on the secondary side DC output voltage to constitute a switching power supply circuit.

Here, the power factor improving rectifier includes a first rectifier circuit composed of two sets of high-speed recovery type diodes connected in series and a second rectifier circuit composed of two sets of low-speed recovery type diodes connected in series. A capacitor is connected in parallel to each of the two sets of fast recovery type diodes, and the primary parallel resonance capacitor is connected to a connection point of the two sets of fast recovery type diodes, and the switching output voltage is fed back. Each of the two sets of fast recovery type diodes is configured to improve the power factor by interrupting the rectified current based on the returned switching output voltage.

Alternatively, the power factor improving rectifier includes a rectifier circuit including two sets of high-speed recovery type diodes connected in series, and a capacitor is connected in parallel to each of the two sets of high-speed recovery type diodes. The primary side parallel resonance capacitor is connected to a connection point of the set of fast recovery type diodes, and the switching output voltage is fed back. Each of the two sets of fast recovery type diodes is rectified based on the returned switching output voltage. The power factor is improved by interrupting the current.

According to the above configuration, in a power supply circuit called a switching frequency control type composite resonance type converter, a voltage resonance pulse voltage generated in a primary side voltage resonance converter is divided by two sets of parallel resonance capacitors to improve power factor rectification means. As a result, the power factor can be improved to, for example, about 0.9, and at the same time, the AC / DC conversion efficiency can be improved and the ripple component of the DC output voltage can be reduced. In particular, since the charging current to the smoothing means is divided by the first rectifier circuit and the second rectifier circuit, the operating range of the zero-voltage switching is not narrowed even if the power factor is improved.

[0041]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit according to an embodiment of the present invention.
On the primary side of the power supply circuit shown in this figure, a voltage resonance type switching converter (voltage resonance type converter) is provided. A rectifier circuit having a power factor improving function for this voltage resonance type converter, that is, a power factor improving rectifier circuit 10
Is provided. In this example, the load power Po =
A circuit suitable for a system of 200 W or more and AC input voltage VAC = 100 V system.

In the power supply circuit shown in this figure, a line filter transformer LFT is provided for a commercial AC power supply AC. The AC input current IAC is rectified by the power factor improving rectifier circuit 10 and is smoothed by the two sets of smoothing capacitors Ci1 and Ci2 connected in series. Ei.

The configuration of the power factor improving rectifier circuit 10 will be described later. First, the configuration of the voltage resonance type converter will be described. Here, the voltage resonance type converter employs a self-excited configuration including one switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1. For example, the withstand voltage is about 1500 V.

The base of the switching element Q1 is connected to the positive electrode of a smoothing capacitor Ci1 (rectified smoothing voltage Ei) via a starting resistor RS and a base current limiting resistor RB, so that a base current at the time of starting is obtained from a rectifying smoothing line. I am trying to be. A drive winding NB and a resonance capacitor C are provided between the base of the switching element Q1 and the primary side ground.
A self-excited oscillation drive resonance circuit (self-excited oscillation drive circuit) composed of a series connection circuit of B and a base current limiting resistor RB is connected. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci2 forms a path for a clamp current flowing when the switching element Q1 is turned off. The collector of the switching element Q1 is connected to the positive terminal of the smoothing capacitor Ci1 via a series connection of the primary winding N1 and the detection winding ND. The emitter is grounded to the primary side ground.

A parallel resonance capacitor Cr is connected to the collector of the switching element Q1. The parallel resonance capacitor Cr forms a primary parallel resonance circuit of the voltage resonance type converter by its own capacitance and a leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT described later. When the switching element Q1 is turned off, the voltage across the resonance capacitor Cr becomes
In practice, a sinusoidal pulse waveform is obtained to obtain a voltage resonance type operation.

The orthogonal control transformer PRT shown in FIG.
Is a saturable reactor on which a detection winding ND, a drive winding NB, and a control winding NC are wound. The orthogonal transformer PRT drives the switching element Q1 and is provided for constant voltage control. As the structure of the orthogonal control transformer PRT, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. Then, the detection winding N is applied to the predetermined two magnetic legs of the three-dimensional core in the same winding direction.
D, the drive winding NB is wound, and the control winding NC is wound in a direction orthogonal to the detection winding ND and the drive winding NB.

In this case, the detection winding ND of the orthogonal control transformer PRT (frequency variable means) is connected in series with the primary winding N1 of the insulation converter transformer PIT, which will be described later, so that the switching of the switching element Q1 is performed. The output is transmitted to the detection winding ND via the primary winding N1.
In the orthogonal control transformer PRT, the switching output obtained on the detection winding ND is excited by the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is generated on the drive winding NB. . This drive voltage is applied to a series resonance circuit (NB,
CB) is output to the base of the switching element Q1 as a drive current via the base current limiting resistor RB.
As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit (NB, CB).

Insulated converter transformer P of the present embodiment
As shown in FIG. 2, the IT is provided with an EE-type core in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other. , The primary winding N using the divided bobbin B
1 and the secondary winding N2 are wound separately. As shown in the figure, the gap G
Is formed. As a result, loose coupling with a required coupling coefficient can be obtained. The gap G can be formed by forming the central magnetic legs of the E-shaped cores CR1 and CR2 shorter than the two outer magnetic legs.
The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, and the other end is connected to the positive electrode (rectified smoothing voltage) of the smoothing capacitor Ci1 through a series connection of the detecting winding ND. Ei).

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage excited by the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for performing a voltage resonance type switching operation, and the secondary side is provided with a parallel resonance circuit for obtaining a voltage resonance operation. . That is, it is configured as a complex resonance type switching converter referred to in this specification.

In this case, a half-wave rectification circuit is provided for the secondary parallel resonance circuit formed as described above by connecting a rectifier diode DO1 and a smoothing capacitor CO1 as shown in the figure. , And generates a DC output voltage EO1.
The DC output voltage EO1 is also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage E
The constant voltage control is performed by controlling the resonance frequency for switching of the switching element Q1 using O1 as the detection voltage. That is, the control circuit 1 uses a DC current whose level is varied in accordance with the level of the DC voltage output E01 on the secondary side as a control current, for example, as the drive transformer PRT.
, The constant voltage control is performed as described later.

By the way, the insulation converter transformer PIT
, The inductance L1 of the primary winding N1 and the inductance L1 of the secondary winding N2 depend on the relationship between the polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the connection of the rectifier diode DO (DO1).
In some cases, the mutual inductance M with the inductance L2 is + M or -M. For example, when the connection configuration shown in FIG. 3A is adopted, the mutual inductance is + M (additive polarity: forward method), and when the connection configuration shown in FIG. 3B is employed, the mutual inductance is −M (depolarization: Flyback method). If this is made to correspond to the operation of the secondary side of the power supply circuit shown in FIG. 1, for example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the operation in which the rectified current flows through the rectifier diode DO1 is as follows. It can be regarded as an operation mode (forward mode) of + M.

In the control circuit 1, the level of the control current (DC current) flowing through the control winding NC is varied according to the change in the secondary DC output voltage level (EO1), so that the winding is wound around the orthogonal control transformer PRT. The inductance LB of the mounted drive winding NB is variably controlled. Thereby, the switching element Q1 formed including the inductance LB of the drive winding NB is formed.
The resonance condition of the series resonance circuit in the self-excited oscillation driving circuit for the above changes. This is an operation of varying the switching frequency of the switching element Q1, and this operation has an effect of stabilizing the secondary DC output voltage.

In the circuit shown in this figure, when the switching frequency is varied, the period during which the switching element Q1 is off is fixed, and the period during which the switching element Q1 is on is variably controlled. In other words, in this power supply circuit, as a constant voltage control operation, an operation is performed to variably control the switching frequency, thereby performing resonance impedance control on the switching output, and at the same time, controlling the conduction angle of the switching element in the switching cycle (PWM). Control). This complex control operation is realized by a set of control circuit systems. Here, as the switching frequency control, when the secondary output voltage increases due to, for example, a tendency toward light load, the control is performed so as to suppress the secondary output by increasing the switching frequency. It is supposed to be done.

Next, the configuration of the power factor improving rectifier circuit 10 will be described. The power factor improving rectifier circuit 10 has a function of rectifying the AC input current IAC and also has a function of improving the power factor. Specifically, the capacitor voltage dividing system, the electrostatic coupling type voltage feedback system, and the power factor improving power source are configured by the voltage doubler rectification system.

In the power factor improving rectifying circuit 10, a capacitor CN for suppressing normal mode noise is provided between the AC lines.
Is arranged. Also choke coil (inductance) L
two sets of fast recovery type diodes D1A and D1B
Is provided. The high-speed recovery type diodes D1A and D1B are connected in series and arranged between the positive terminal of the smoothing capacitor Ci1 and the primary side ground. The choke coil Ls is connected to a connection point between the high-speed recovery type diodes D1A and D1B. Parallel resonance capacitors C10 and C11 are connected in parallel to the high-speed recovery type diodes D1A and D1B, respectively.

A series circuit of low-speed recovery type diodes Di1 and Di2 is arranged on the AC line. The series circuit of the slow recovery type diodes Di1 and Di2 is arranged between the positive terminal of the smoothing capacitor Ci1 and the primary side ground.

First, the rectifying function of the power factor improving rectifier circuit 10 will be described. This power factor improving rectifier circuit 1
At 0, the high speed recovery type diodes D1A and D1B function as a first rectifier circuit, and the low speed recovery type diodes Di1 and Di2 function as a second rectifier circuit.

That is, during the positive period of the AC input voltage VAC, the first rectifier circuit is used in a system of AC power supply AC → line filter transformer LFT → choke coil Ls → high speed recovery type diode D1B → smoothing capacitor Ci1 →. A rectified current flows to charge the smoothing capacitor Ci1, and at the same time, the AC power supply AC → the line filter transformer LFT
→ Low speed recovery type diode Di1 → Smoothing capacitor C
In the system of i1..., a rectified current flows through the second rectifier circuit and charges the smoothing capacitor Ci1. In the negative period of the AC input voltage VAC, the system of AC power supply AC → line filter transformer LFT → smoothing capacitor Ci2 → primary side ground → high speed recovery type diode D1A →.
A rectified current by the first rectifier circuit flows to charge the smoothing capacitor Ci2, and at the same time, in a system of AC power supply AC → line filter transformer LFT → smoothing capacitor Ci2 → primary side ground → low speed recovery type diode Di2 →. Then, a rectified current flows from the second rectifier circuit to charge the smoothing capacitor Ci2.

That is, the rectification current is divided into two systems by the first and second rectification circuits, and the smoothed capacitors Ci1 and Ci
2 will be supplied. And the smoothing capacitor Ci
1 and Ci2 are connected in series, and a rectified smoothed voltage Ei is taken out from the positive terminal side of the smoothing capacitor Ci1, whereby a double voltage rectification method is realized.

The power factor improving function of the power factor improving rectifier circuit 10 is as follows. As described above, the parallel resonance capacitors C10 and C11 are connected in parallel to the two sets of the fast recovery type diodes D1A and D1B, respectively. Although detailed description is omitted here, for example, the parallel resonance capacitors C10 and C11 form a series resonance circuit together with, for example, the choke coil Ls. This has the effect of suppressing an increase in the rectified smoothed voltage Ei when the load is reduced. For the power factor improving rectifier circuit 10, the choke coil Ls, the anode of the high-speed recovery type diode D1B, and the high-speed recovery type diode D1B
The above-described parallel resonance capacitor Cr is connected to the connection point between the 1A cathode and the parallel resonance capacitors C10 and C11, and the switching output (voltage resonance pulse voltage) obtained in the primary-side parallel resonance circuit is fed back. Like that. With such a configuration, the switching output obtained in the primary-side parallel resonance circuit is fed back to the rectified current path via the capacitive coupling of the parallel resonance capacitor Cr.

The switching output fed back in this manner causes the alternating voltage of the switching cycle to be superimposed on the rectified current path. The superimposed voltage of the switching cycle causes the high-speed recovery type diode D1A ( In D1B), an operation of interrupting the rectified current in the switching cycle is obtained, and the apparent inductance of the choke coil Ls also increases due to the interrupting action. Further, a voltage is generated at both ends of the parallel resonance capacitor C10 (or C11) when a current of a switching period flows, but the level of the rectified smoothed voltage Ei is only the voltage between both ends of the parallel resonance capacitor C10 (or C11). Will be reduced. Thus, the smoothing capacitor C1 is also maintained during the period in which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci1 (or Ci2).
The charging current to i1 (or Ci2) is caused to flow. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased. As a result, the power factor is improved.

According to the above-described configuration, the parallel resonance capacitor Cr forming the primary side parallel resonance circuit of the primary side voltage resonance type converter is connected in series with the parallel resonance capacitor C10. The capacitor Cr is connected in series with the parallel resonance capacitor C11. That is, the voltage resonance pulse voltage that appears as the voltage across the parallel resonance capacitor Cr is divided by the capacitance ratio of the parallel resonance capacitor Cr and the parallel resonance capacitor C10 (or C11). Then, via a parallel resonance capacitor C10 connected in parallel with the fast recovery type diode D1A (or via a parallel resonance capacitor C11 connected in parallel with the fast recovery type diode D1B),
A circuit is formed as a voltage feedback system in which voltage is fed back to the smoothing capacitor Ci1 (or Ci2).

The capacitance of the parallel resonance capacitors Cr and C10 (C11) is defined as Cr <C10 (C11). In particular, when the capacitance of the capacitor C10 (C11) is increased, the power factor P
F will improve. The switching frequency fs is controlled to be high during the period when the AC input voltage VAC is high, and the switching frequency fs is controlled to be low during the period when the AC input voltage VAC is low.
In the vicinity of the peak value, the voltage resonance pulse voltage is not fed back to the power factor correction rectifier circuit 10, and as the AC input voltage VAC decreases, the voltage resonance pulse voltage
The amount of return to is increased.

In the case of this example, during the positive period of the AC input voltage VAC, the voltage resonance pulse voltage is divided by the parallel resonance capacitors Cr and C11 and fed back, and a rectified current is generated by switching of the high-speed recovery type diode D1B. As a result, the conduction angle of the AC input current is increased and the power factor is improved. Also, during the negative period of the AC input voltage VAC, the voltage resonance pulse voltage
The voltage is divided and fed back by r and C10, the rectified current is intermittently switched by the switching of the high-speed recovery type diode D1A, and the conduction angle of the AC input current is enlarged. As a result, the power factor is improved.

Also, the charging current to the smoothing capacitors Ci1 and Ci2 is shunted by the operation of the first and second rectifier circuits described above. This prevents an excessive charging current from flowing through the inductance Ls and the high-speed recovery type diode D1B or D1A near the positive and negative peak values of the AC input voltage VAC. The current flowing through the parallel resonance capacitors C11 and C10 connected in parallel with the high-speed recovery type diodes D1B and D1A is changed to the AC input voltage V
It is possible to prevent the zero voltage switching operation from being restricted due to the influence near the peak value of AC. That is, even if the voltage feedback amount is increased by increasing the voltage dividing ratio by the parallel resonance capacitor Cr and the parallel resonance capacitors C10 and C11, the stable condition of the zero voltage switching in the entire region with respect to the fluctuation of the AC input voltage VAC and the load power Po. To be satisfied. Therefore, there is no problem in reducing the capacitance of the parallel resonance capacitors C10 and C11 to increase the amount of voltage feedback and improving the power factor to, for example, 0.8 or more.

FIGS. 4, 5 and 6 show experimental results and operation waveforms of the switching power supply circuit of FIG. In the experiment,
Capacitor CN = 1 μF, parallel resonance capacitor Cr = 2
400 pF, parallel resonance capacitor C10 = C11 = 3900
pF, primary winding N1 = 90T, inductance Ls = 7
5 μH was set. In the prior art shown in FIG. 8, for example, the parallel resonance capacitor Cr = 8200 pF, the parallel resonance capacitor C10 = 0.027 μF, and the primary winding N1 = 4
Note that 5T and the inductance Ls were set to 75 μH.

FIG. 4 shows the change characteristics of the power factor PF and the AC / DC power conversion efficiency (η _{AC / DC} ) with respect to the fluctuation of the load power Po = 200 W to 40 W when the AC input voltage VAC = 100 V. Is AC input voltage VAC = 80V ~
Power factor PF at 140 V and AC / DC power conversion efficiency (η
_{AC / DC} ).

As can be seen from FIGS. 5 and 6, the power factor PF can be improved to 0.9 or more, and at the same time, the AC / DC power conversion efficiency (η _{AC / DC} ) can be improved. . As described above, in this example, the load power Po = 200
W or more, is applied when the AC input voltage VAC is a 100 V system, and the rectified smoothed voltage Ei is obtained by doubling the AC input voltage VAC by a voltage doubler rectification method to obtain a voltage twice that of the full-wave rectification method.
The withstand voltage of the switching element Q1 is 1500 V. For this reason, the primary current flowing through the primary winding N1 of the insulating converter transformer PIT is reduced to one half of that of the prior art, so that the AC / DC power conversion efficiency (η _{AC / DC} ) can be improved to about 94%. , And the switching element Q
A heat sink for 1 is also unnecessary. For example, the AC / DC power conversion efficiency (η _{AC / DC} ) is 9 in the circuit example of FIG. 8 (when a full-wave rectification type electrostatic coupling type power factor improvement circuit is provided).
From 0.9%, it is improved to 93.4% in this example. This also reduces the input power by about 5.9W. of course,
The power factor PF and the AC / DC power conversion efficiency (η _{AC / DC} ) can be maintained in a wide range with respect to variations in the load power Po and the AC input voltage VAC.

FIGS. 6A to 6F show load power Po.
= 200 W and the AC input voltage VAC = 100 V shows operation waveforms of each unit. FIGS. 6C and 6D show the current I3 flowing through the second rectifier circuit and the current I3 flowing through the first rectifier circuit.
1 is shown. In the vicinity of the peak value of the AC input voltage VAC, for example, a current of 10 Ap flows. In this example, the current I1 is divided into 4 Ap and the current I3 is divided into 6 Ap. This is the slow recovery diode Di
1 and Di2, and the current flowing through the high-speed recovery type diodes D1A and D1B does not become excessive. Further, as can be seen from the current I3 in FIG. 6D, during the period when the absolute value of the AC input voltage VAC is low, the power factor is improved by the switching operation of the high-speed recovery type diodes D1A and D1B.

In this example, since the current flowing through the high-speed recovery type diodes D1A and D1B does not become excessive,
The zero voltage switching operation is not restricted near the peak value of the AC input voltage VAC. Therefore, there is no problem in increasing the voltage feedback amount by changing the voltage dividing ratio by the parallel resonance capacitors Cr and C10 (and C11) as described above, and improving the power factor to 0.9 or more.

The low-speed recovery type diode Di1,
Both Di2 and the high-speed recovery type diodes D1A and D1B suppress heat generation due to a large current, so that a heat radiating plate becomes unnecessary and a diode having a small current capacity can be selected. Further, the ripple voltage of the DC output voltage Eo is reduced to 30 mV. Therefore, the smoothing capacitor Ci1,
The capacitance of Ci2 and C01 can be reduced. For example, in the case of the prior art, the smoothing capacitor Ci = 1000 μF,
Although C01 = 220 μF, the capacitance of each of the smoothing capacitor Ci1 (Ci2) and the smoothing capacitor C01 in this example can be reduced to about half that of the prior art.
From these facts, it is possible to reduce the size and cost of the circuit.

FIG. 7 shows a circuit according to another embodiment of the present invention. The power supply circuit shown in FIG. 7 also has a configuration in which a voltage resonance type converter is provided on the primary side and a parallel resonance circuit is provided on the secondary side as a composite resonance type converter, similarly to the power supply circuit of FIG. The power factor improving rectifier circuit 10 of the switching power supply circuit of FIG. 7 is obtained by removing the low-speed recovery type diodes Di1 and Di2 from the configuration of the power factor improving rectifier circuit of FIG. Also, an example in which a separately excited oscillation type is used as the primary side switching converter is described. Further, the smoothing capacitors Ci1 and Ci2 are connected in series, and the power factor improving rectifier circuit 10 employs the voltage doubler rectification method as in FIG.

In this example, one MOS-FE is provided on the primary side.
A separately-excited voltage-resonant converter employing T as the switching element Q10 is provided. Switching element Q10
Are driven by the oscillation circuit 2 and the drive circuit 3 for switching. The oscillating circuit 2 generates an oscillating signal of a required frequency based on the control of the control circuit 1 so that the drive circuit 3
Output to The drive circuit 3 generates a drive voltage for driving the switching element Q10 based on the oscillation signal and outputs the drive voltage to the switching element Q10. The control circuit 1 controls the oscillation frequency of the oscillation circuit 2 based on the DC output voltage E01 on the secondary side. As a result, the switching element Q10 performs a separately-excited switching operation and also has a function of stabilizing the DC output voltage. The functions of the damper diode DD and the parallel resonance capacitor Cr are the same as those in the example of FIG.

The secondary side of the insulating converter transformer PIT is provided with a rectifying circuit system formed by connecting a secondary side series resonance capacitor C2, rectifying diodes DO1, DO2, and a smoothing capacitor CO1 as shown in the figure. That is, a voltage doubled half-wave rectifier circuit including a secondary side series resonance circuit composed of the secondary side series resonance capacitor C2 and the secondary winding N2 is formed.

The power factor improving rectifier circuit 10 is the same as that of FIG. 1 except that the low-speed recovery type diodes Di1 and Di2 of FIG. 1, that is, the above-described portion as the second rectifier circuit are not formed. Therefore, the rectification is performed by the high-speed recovery type diodes D1A and D1B forming the first rectifier circuit, and the power factor improving action is realized as described above. In the case of the circuit of FIG. 7, since the second rectifier circuit does not exist, the charging current to the smoothing capacitor Ci is not shunted even near the peak value of the AC input voltage VAC. Improving the power factor is inappropriate in view of the stability of the zero-voltage switching operation. However, when the power factor is 0.8 or less, it can be used as a practical circuit because of the improvement in AC / DC conversion efficiency and the reduction in ripple voltage.

Although the embodiments have been described above, the present invention may have various modifications. For example, the present applicant has already proposed a configuration including a full-wave rectifier circuit using a secondary-side series resonant circuit, a double voltage rectifier circuit, a quadruple voltage rectifier circuit, and the like, as a composite resonant switching converter. Such a configuration can also be realized as a modification of the present embodiment. That is, the present embodiment is not particularly limited as the configuration of the secondary-side resonance circuit and the rectifier circuit.

In addition, the so-called single-end type configuration having one switching element has been described as the primary-side voltage resonance type converter, but the present invention is also applicable to the so-called push-pull type in which two switching elements are alternately switched. The invention can be applied.

[0080]

As can be seen from the above description, according to the present invention, even if the power factor is increased to 0.90 or more by increasing the amount of voltage feedback to the power factor improving circuit, the zero-voltage switching of the primary side voltage resonance converter can be performed. Since the operation area is secured,
Power factor can be improved.

Further, the power conversion efficiency of AC / DC is improved, and energy is saved by reducing the input power. Further, the ripple voltage of the rectified smoothed voltage and the DC output voltage in the commercial power supply cycle can be reduced, the capacitance of each smoothing capacitor (electrolytic capacitor) can be reduced, and the size of each smoothing capacitor can be reduced. In addition, since two charging current paths are formed by the first and second rectifier circuits, current is shunted, thereby reducing heat generation. Therefore, each diode can be selected to have a small current capacity, and a heat sink is not required. As a result, the size and cost of the circuit can be reduced.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a switching power supply circuit according to an embodiment of the present invention.

FIG. 2 is a side sectional view showing a structure of an insulating converter transformer employed in the power supply circuit of the present embodiment.

FIG. 3 is an explanatory diagram showing each operation when the mutual inductance is + M / -M.

FIG. 4 shows the power factor and A of the switching power supply circuit according to the embodiment.
FIG. 4 is an explanatory diagram of characteristics of C / DC conversion efficiency.

FIG. 5 shows the power factor and A of the switching power supply circuit according to the embodiment.
FIG. 4 is an explanatory diagram of characteristics of C / DC conversion efficiency.

FIG. 6 is a waveform chart showing an operation of the switching power supply circuit of the embodiment.

FIG. 7 is a circuit diagram of a switching power supply circuit according to another embodiment of the present invention.

FIG. 8 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 9 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 10 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 11 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 12 shows the power factor and A of the prior art switching power supply circuit.
FIG. 4 is an explanatory diagram of characteristics of C / DC conversion efficiency.

FIG. 13 shows the power factor and A of the prior art switching power supply circuit.
FIG. 4 is an explanatory diagram of characteristics of C / DC conversion efficiency.

FIG. 14 is a waveform chart showing the operation of the switching power supply circuit of the prior art.

[Explanation of symbols]

1 control circuit, 10 power factor improvement rectifier circuit, Ci smoothing capacitor, D1A, D1B high speed recovery type diode, Di
1, Di2 low speed recovery type diode, Cr, C10,
C11 parallel resonance capacitor, C2 secondary parallel resonance capacitor, PRT orthogonal control transformer, PIT isolation converter transformer, Q1, Q10 switching element

Claims (3)

[Claims] 1. A smoothing means for smoothing a rectified current by two sets of smoothing capacitors connected in series and outputting a doubled DC input voltage, and a gap formed so as to obtain a required coupling coefficient which is loosely coupled. An insulating converter transformer provided for transmitting a primary output to a secondary side; and a switching device for intermittently outputting the doubled DC input voltage by a switching element and outputting the doubled DC input voltage to a primary winding of the insulating converter transformer. Means, a primary resonance circuit formed by at least a leakage inductance component including a primary winding of the insulating converter transformer, and a capacitance of a primary side parallel resonance capacitor to make the operation of the switching means a voltage resonance type; The AC power supply is rectified, a rectified current is supplied to the smoothing means, and the AC power is Power factor improving rectifying means for improving a power factor by intermittently commutating a rectified current based on the returned switching output voltage, and a secondary of the insulating converter transformer. A secondary resonance circuit formed on the secondary side by the leakage inductance component of the winding and the capacitance of the secondary resonance capacitor; and a secondary resonance circuit formed including the secondary resonance circuit, DC output voltage generating means configured to input an alternating voltage obtained to the winding and perform a rectification operation to generate a secondary DC output voltage, and according to a level of the secondary DC output voltage. And a constant-voltage control means configured to perform constant-voltage control on the secondary-side DC output voltage. 2. The power factor improving rectifier includes: a first rectifier circuit including two sets of high-speed recovery type diodes connected in series; and a second rectifier circuit including two sets of low-speed recovery type diodes connected in series. A capacitor is connected in parallel to each of the two sets of fast recovery type diodes, and the primary side parallel resonance capacitor is connected to a connection point of the two sets of fast recovery type diodes, and the switching output voltage is fed back. , 2 above
The switching power supply of claim 1, wherein each of the sets of fast-recovery diodes is configured to improve the power factor by interrupting a rectified current based on the returned switching output voltage. circuit. 3. The power factor improving rectifier includes a rectifier circuit including two sets of high-speed recovery type diodes connected in series, and a capacitor is connected in parallel to each of the two sets of high-speed recovery type diodes. The primary side parallel resonance capacitor is connected to the connection point of the two sets of fast recovery type diodes, and the switching output voltage is fed back.
The switching power supply of claim 1, wherein each of the sets of fast-recovery diodes is configured to improve the power factor by interrupting a rectified current based on the returned switching output voltage. circuit.