JP2001136772A - Drive control device for brushless motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To obtain a drive control device which reduces torque ripples caused by a commutation and by which the noise level and the vibration of a motor are lowered. SOLUTION: In order to perform feedback control operation by using a noncommutative current as a target current, a first duty factor as a PWM control output to a corresponding switching element Q is computed. A second duty factor used to PWM-control a terminal voltage at a phase, at which a commutation current becomes 0 as required, is computed and output in such a way that this drive control circuit is operated in a region, in which the first duty factor is at a prescribed value or smaller during a commutation period.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a rectangular wave drive control device for a brushless motor having a trapezoidal magnetic flux distribution, and more particularly to a torque ripple suppression device.

[0002]

2. Description of the Related Art A brushless motor detects the position of a magnetic pole of a rotor composed of a permanent magnet, and switches the energization mode of a stator coil based on a detection signal to rotate the rotor. If the current fluctuates when the current supply mode of the stator coil is switched (at the time of commutation), torque ripple occurs, causing vibration and noise. Therefore, a rectangular wave drive control device for a trapezoidal wave magnetized brushless motor has been proposed in which the torque ripple is reduced to reduce vibration and noise. An example of such a device is EPEA (Europe Power Corporation) by J. Cros et al. Electronics Association) September 1993, Bright
It is disclosed on pages 266 to 271 (hereinafter abbreviated as reference 1) in the newsletter of the on tournament.

According to this, the normal PWM output Mod. 1 for controlling the non-commutation phase current in the non-commutation section and the zero current during commutation so that the non-commutation phase current does not decrease. That controls the terminal voltage of the phase (hereinafter, simply referred to as a commutation phase)
It is disclosed that the M output Mod. FIG.
Is a configuration diagram showing the operation of the device shown in Document 1. In the figure, Mod. 1 is selectively connected to a non-commutation phase switching element, and Mod. 2 is selectively connected to a commutation phase switching element. ing. In the non-commutation mode and the low-speed commutation mode, the changeover switch is selected to be on the lower side, and P or PI control calculation is performed so that the target current Id and the phase current Is of the non-commutation phase match. The duty ratio is output from Mod. 1 and is controlled to be a target value current by changing the duty ratio of the non-commutation phase. On the other hand, in the case of Mod. 2, the switching element connected at the duty ratio of 0% is in the off state.

Next, in the commutation mode, the changeover switch is selected to the upper side, and the duty ratio of Mod. 1 becomes 100%, while the duty ratio of Mod. Is output, and the voltage applied to the commutation phase is reduced by changing the duty ratio of the switching element in the commutation phase, thereby limiting the decrease in the non-commutation phase current. The output for Mod. 1 has an output voltage = 4E which is a theoretical value at the time of commutation at low speed (here,
An upper limit value of the duty ratio that satisfies the relationship of E = generated voltage / phase) is set. For Mod.2, an upper limit value that is a function of E during commutation at high speed is set. During commutation, Mod. 1 during low-speed rotation,
In the case of Mod. 2 at the time of high-speed rotation, the P or PI control calculation is set so that the output duty ratio is saturated to these upper limits.

[0005]

Since the conventional rectangular wave drive control is performed as described above, the open control is substantially determined by the upper limit during commutation, and the upper limit is determined by the resistance of the brushless motor. When the generated voltage E changes during commutation in addition to the change depending on parameters such as the inductance and the inductance, there is a problem that compensation is difficult and not practical.

SUMMARY OF THE INVENTION The present invention has been made to solve the above problems, and has a rectangular trapezoidal wave magnetized brushless motor capable of ensuring the controllability of following a current target value and reducing torque ripple. It is an object to provide a wave drive control device.

[0007]

According to a first aspect of the present invention, there is provided a drive control apparatus for a brushless motor, comprising: a plurality of switching elements connected in series between a positive electrode and a negative electrode of a DC power supply; And a brushless motor, a current detector for detecting a current flowing through the connection path, a position detector for detecting a magnetic pole position of the brushless motor, and an induced voltage per phase generated in a drive coil of the brushless motor , A drive signal generating means for selectively energizing the switching element based on a position detection signal obtained from the position detector, and a feedback current based on a detection current detected by the current detector. A first calculating means for calculating a first duty ratio, which is an output of the driving signal generating means, so that the target current and the target current coincide with each other; A second duty ratio for a switching element connected to a phase where the commutation current becomes zero, based on the output of the motor generation voltage calculation means and the target current so as to reduce the neutral point potential fluctuation of the brushless motor. It has a calculating means and a correcting means for correcting the first duty ratio based on the second duty ratio.

According to a second aspect of the present invention, there is provided a drive control device for a brushless motor, wherein a circuit has n phases, a circuit resistance per phase is R, a target current is Id, a power supply voltage is Vd, and a power generation voltage per phase. Is Es, the second duty ratio is nI
This is a function of d · R + (n + 1) Es−Vd.

According to a third aspect of the present invention, there is provided a drive control device for a brushless motor, wherein a circuit has n phases, a circuit resistance per phase is R, a target current is Id, a power supply voltage is Vd, and a power generation voltage per phase. the Es, when the circuit resistance correction amount deviation is proportional to the value obtained by dividing the target current Id to the target current Id and the feedback current is was Y _5, second energization rate nI
_{d (R + Y 5) +} (n + 1) is obtained as a function of Es-Vd.

According to a fourth aspect of the present invention, there is provided a drive control device for a brushless motor, wherein the first duty ratio is equal to or less than a predetermined maximum value. 2. Calculate the duty ratio.

According to a fifth aspect of the present invention, in the drive control apparatus for a brushless motor, the maximum value of the first duty ratio is a function of the first duty ratio.

According to a sixth aspect of the present invention, there is provided a drive control apparatus for a brushless motor, wherein a circuit has n phases, a circuit resistance per phase is R, a target current is Id, a power supply voltage is Vd, and a power generation voltage per phase. Is Es and the maximum value is D1max,
The second duty ratio is nIdR + (n + 1) Es-Vd + n
This is a function of (1-D1max) Vd.

[0013]

DESCRIPTION OF THE PREFERRED EMBODIMENTS Embodiment 1 Hereinafter, a first embodiment of the present invention will be described with reference to the drawings. As shown in FIG. 1, the brushless motor 1 includes a rotor 2 having a permanent magnet on the surface, a three-phase drive coil 3 (u, v, w phases) and position detectors 4a, 4b, 4c. One end of each drive coil 3 is connected in common, and the other ends are freewheel diodes D _a , D _b , connected in anti-parallel to each of the upper arm switching elements Q ₁ , Q ₂ , Q ₃ and the switching elements.
As well as connected to the positive electrode of the DC power source 5 via a D _c, it is connected the lower arm switching element Q _4, Q _5, Q ₆ and the diode D _d, D _e, via the D _f to the negative electrode of the DC power source 5 .

A phase current detector 6 is connected to two phases of the drive coil 3.
u, 6w, and sends a signal corresponding to the current flowing through the drive coil 3 to the microcomputer 7. The current value of the remaining one phase without a detector is calculated from the relationship that the sum of the phase currents is always zero. Position detector 4a for detecting the position of the rotor 2 of the brushless motor 1, 4b, 4c to have each Hall element, their position signal H _1, H _2, H ₃
Is input to the microcomputer 7. Further, a signal for detecting the power supply voltage Vd supplied to the switching element is also input to the microcomputer 7. The microcomputer 7 outputs an appropriate drive signal and PWM signal to the gate terminal of the switching element based on these input values.

Next, the position signals H ₁ , H ₂ , H ₃ and the steps will be described with reference to FIG. The position signal H ₁ is a generated voltage (emf) that appears in the U phase of the drive coil 3 when the rotor 2 rotates.
The signal is set to a high level from the time when the Eu waveform reaches the maximum value + E to the lowest value -E, and is set to a low level signal in other sections. Similarly, the position signal H ₂ is a signal corresponding to the generated voltage Ev appearing in the V phase, and the position signal H ₃ is a signal corresponding to the generated voltage Ew appearing in the W phase.

Six sections (π / 3 in electrical angle and 4 magnetic poles in mechanical angle) divided by these position signals H ₁ , H ₂ and H _3.
Are assigned to step numbers 1 to 6 and used for control. For example, when the step is 1, a phase current corresponding to the same polarity as the generated voltage is passed. That is, as shown in FIG. 2, the U-phase current Iu in the positive direction, in order to flow the W-phase current Iw in the negative direction, the switching element Q _1, Q ₆ energized. As a result, the motor generates a positive torque of Iu × Eu + Iw × Ew, and rotates in the direction of step 2 if the load torque is small.

FIG. 3 is a table showing the energized state of the switching element in each step. In FIG. 3, although not described the energization ratio of the switching elements of the lower arm, duty factor of the switching elements of the corresponding lower arm is a complement of the upper arm (lower arm if the upper arm D ₁ 1- D ₁ ). Duty ratio D
If ₁ is greater than or equal 0.5, D ₁ phrase is on the upstream side, 1-D ₁
When the duty ratio D ₁ is 0.5 or less, the upstream and downstream are reversed, that is, the direction of the current is reversed, and a negative-direction torque is generated. , And rotate in the opposite direction (steps 1 → 6 → 5 → 4). FIG.
“Off” in the middle indicates that the switching element is in a non-energized state.

When the step is changed, the phase in which the current flows changes, and this is called commutation. Hereinafter, the phase where the current becomes 0 is simply referred to as a commutation phase. Explaining with reference to the waveforms of FIG. 2, from step 6 to step 1, the V-phase current becomes 0, the U-phase current starts flowing, and the W-phase current does not change. In this case, the V phase is called a commutation phase, and the W phase is called a non-commutation phase.

Assuming that there is symmetry with respect to the drive coil 3, the relationship between the drive coil terminal voltages Vu, Vv, Vw and the drive coil phase currents Iu, Iv, Iw is expressed by the following equation (1).

[0020]

(Equation 1)

Here, R is the resistance per phase, L is the equivalent inductance per phase, Vn is the neutral point potential of the driving coil, and Eu, Ev, and Ew are the generated voltages (emf voltages) of each phase. If the PWM cycle is shorter than the time constant L / R of the circuit, a so-called state averaging method can be applied, and the terminal voltage can be expressed by the duty ratio and the power supply voltage Vd in FIG. In the non-commutated state, equation (1)
Is an equation for only the U phase and the W phase, and is as shown in the following equation (2).

[0022]

(Equation 2)

By adding each of the above equations (2), Iu + Iw
By rearranging by applying = 0, Vn becomes as shown in Expression (3). Vn = Vd / 2 (3) In the commutation state of Step 1, Equation (1) becomes as Equation (4) below (however, the forward voltage drop of the diode is ignored). ).

[0024]

(Equation 3)

Equation (4) is added, and Iu + Iv + I
When rearranging by applying w = 0, Vn becomes as in the following equation (5) (when the current in the commutation phase is positive). Vn = Vd / 2− (Vd / 2−Ew) / 3 (5) Similarly, Vn at the time of commutation in the case of step 2 is expressed by the following equation (when the current of the commutation phase is negative): It becomes like (6). Vn = Vd / 2 + (Vd / 2−Ew) / 3 (6)

This relationship is shown by the waveform of the neutral point potential in FIG. As described above, since the neutral point potential Vn changes abruptly in the commutation section, the current in the non-commutation phase is also affected by the relationship of Expression (1) and fluctuates. This becomes the torque ripple of the motor, causing vibration and noise. In response to this voltage fluctuation, the control device must be designed and adjusted so that the current is controlled to be maintained at the target value. In particular, the output of the control device should not be saturated (the duty ratio is fixed to the maximum value) and the control cannot be temporarily disabled.

Next, the operation of the microcomputer 7 is shown in FIG. In FIG. 4, a power supply voltage Vd, currents Iu, I
w, and the signals of the motor rotational positions H ₁ , H ₂ , and H ₃ are internal signals digitized by an interface circuit (not shown). In the motor-rotation-position determining means 8, by the motor rotational position signal H _1, H _2, combinational logic H _3, to determine what number of steps 1-6 as shown in FIG. The motor angular velocity calculating means 9 calculates the angular velocity ω of the motor by measuring the time interval between two successive step change points and dividing the angle (constant value) corresponding to one step by this time interval. The motor generation voltage calculation means 10 calculates an amount Es corresponding to the induced voltage E per phase generated in the drive coil 3 as a product of the emf coefficient Km (constant value) and the angular velocity ω.

The FB current selection means 11 uses the phase current in which the current does not change even in the commutation region = the non-commutation phase current in the feedback control calculation. Therefore, the phase currents Iu, Iv, Iw (however, Iv is Iv = -Iu-I
(calculated from w), and the polarity is changed as necessary. FIG. 3 shows a table of the FB current and the steps in this operation. CW and CC in the figure
W is the rotation direction of the motor, and the feedback current to be selected differs depending on the rotation direction (determined according to the order of step change: the direction of → 1 → 2 → 3 → is CW and the reverse direction is CCW). The current control calculation means 12 performs the control of the duty ratios D ₁ and D ₂ so that the target current Id generated by the target current generation means 13 and the feedback current Is match.
Is calculated. The details of the calculation will be described later.

The control output phase selecting means 14 converts the duty factors D ₁ and D ₂ calculated by the current control calculating means 12 and their complements into the six switching elements Q _{1 in the} next three-phase PWM circuit 15. It determines whether to set to either of the corresponding PWM generating circuit to Q _6. FIG. 3 and FIG. 5 show this setting work in a table of steps and switching elements.
In these tables, only the switching element of the upper arm is described, but the PWM generation circuit for the switching element of the lower arm has a complement of the duty ratio of the switching element of the upper arm in the pair. The duty ratio is set. For example by D ₁ is 35%, if the step is 1, 35% each of the switching elements Q _1, Q ₄ and 65% duty ratio is set.

The notation "off" in the tables of FIGS. 3 and 5 means that both the upper and lower switching elements of the pair are set to the non-energized state. Three-phase PWM circuit 1
5 is on / o by comparing a triangular wave (not shown) (frequency is about 20 KHz in consideration of both noise and switching loss in many cases) with a level corresponding to the duty ratio.
ff signal is generated, in well-known conventional PWM generating circuit for outputting to the next stage of the drive circuit 16 has six circuits corresponding to the switching element Q ₁ to Q _6.

The detailed operation of the current control calculation means 12 will be described with reference to the flowchart shown in FIG. In the figure, S
In 101, the respective functions of the motor rotational position determining means 8, motor angular velocity calculating means 9, motor generated voltage calculating means 10, FB current selecting means 11 and target current generating means 13 which are inputs of the current control calculating means 12 are executed. I have. Next, S
At 102, a so-called feedforward-attached PI (proportional / integral) control operation is performed. That is, hereinafter, the calculations of Expressions (7) to (9) are performed. ε = Id−Is (7) Y ₁ = Ki · ε + Y ₁ (previous value) (8) Y ₂ = Rm × Id + Kp · ε + Y ₁ + Es + Vd / 2 (9) where Id : Target current, Is: actual current (feedback current), ε: deviation, Y ₁ : integral value, Ki: integral gain, Y ₂ : output voltage, Rm: nominal resistance value per phase,
Kp: proportional gain, Es: generated voltage per phase (em
f voltage), Vd: DC power supply voltage.

Next, it is determined in S103 whether or not commutation is in progress. If commutation is in progress, the process proceeds to S104.
Go to 9. As a first example, the method for determining the end of commutation is a method of terminating the commutation when the commutation current is 0 or a value close to 0 in a commutation state. Then, when a predetermined commutation time which can be calculated from the initial commutation current value, the emf voltage, the circuit resistance, and the circuit inductance has elapsed, a method of terminating the commutation can be considered. Whether or not commutation is being performed is determined based on whether or not commutation is started to end of commutation.

In S104, which is the first calculating means in the case of commutation, the difference (Vd / 2 + Es) × sgn (Ic) / 3 of the difference between the neutral point potentials during commutation and non-commutation is calculated. wherein calculating the first duty ratio D ₁ from the output voltage Y ₃ obtained by adding the Y _2. That is,

Y ₃ = Y ₂ + (Vd / 2 + Es) × sgn (Ic) / 3 (10) D ₁ = Y ₃ / Vd (11) where sgn () is a sign function. When the argument is 0, the function is also 0, and when the argument is other than 0, the function is the sign of the argument.
The argument Ic here is a commutation current, which is a current transiently flowing in the off phase in the table of FIG.

S105, S106, S107 and S10
8 is a portion corresponding to the second calculating means in the first embodiment. First, in S105, the compensation voltage Vcmp is calculated by the following equation (12).
Is calculated. Vcmp = 3Rm · Id + 4Es−Vd (12) Assuming that the circuit has n phases, the equation (12) can be expressed as Vcmp = nId · R
+ (N + 1) Es-Vd. This equation is obtained by substituting Vn of equation (6) into the first equation of equation (4), for example, and as shown in the following equation (13). _{Vd · D 1 = R · Iu} + LdIu / dt + E + Vd / 2 + Vd / 6-Ew / 3 ......... (13) where the limit conditions a non-commutated current Iu can be controlled to match the Id: Iu = Id, D ₁ = 1, Ew = −Es,
Substituting dId / dt ≒ 0 into Equation (13) and rearranging (however, the generated voltage E substitutes Es of the same estimated value, and the resistor R substitutes Rm of the same nominal value), the following Equation (14) is considered. Can be Vd> 3Rm · Id + 4Es (14) The expression (12) is obtained by expressing the relationship of the expression (14) in an equal expression using the compensation voltage Vcmp (> 0).

When the terminal voltage of the commutation phase is controlled to Vcmp or Vd-Vcmp (the former when the commutation current is negative, and the latter when the commutation current is positive) during commutation, the equation corresponding to equation (4) becomes The following equation (15) is obtained. In this case, V-phase terminal voltage = Vd−Vcm
p = Vd-D ₂ · Vd

[0037]

(Equation 4)

Assuming that the equation (5) is derived from the equation (4), the neutral point potential Vn at this time is as shown in the equation (16). Vn = Vd / 2− (Vd / 2 + Ev−Vcmp) / 3 (16) Similarly, Vn corresponding to equation (5) when the commutation current is negative.
Becomes the following equation (17). Vn = Vd / 2 + (Vd / 2 + Ev-Vcmp) / 3 (17) In any case, the value V of the neutral point potential during non-commutation during commutation
The change from d / 2 is reduced by Vcmp / 3.

This means that the condition is the same as Expression (15), which is a condition under which the current of the non-commutation current can be controlled. That is, by appropriately controlling the terminal voltage of the commutation phase of the second duty factor D _2, it is possible to reduce the change of the neutral point potential during commutation, non rolling according to the first duty ratio D ₁ A voltage width for current control of the current phase is secured (= the terminal voltage is not saturated), and the current during commutation is controlled to the target value, so that no torque ripple occurs during commutation. Moreover, as can be seen from Equations (16) and (17), the neutral point potential is not related to the first duty ratio D _1, and only the second duty ratio D ₂ (∵
Vcmp = it relates to D ₂ · Vd), no control interference from D ₂ to control the control and the neutral point potential Vn by D ₁ for controlling the non-commutation current, becomes easy conditions construct a control system I have.

If Vcmp is negative or 0, there is no need to control the terminal voltage of the commutation phase, so in S106, processing is performed to set Vcmp to the lower limit of 0. That is, if Vcmp is negative in S106,
In S107, the processing of Vcmp = 0 is performed. In S108, performs calculations = Vcmp / Vd of the second duty ratio D ₂ for controlling the terminal voltage of the commutation phase. In S109, it is shown correcting means for correcting the first duty ratio D ₁ on the basis of the second duty ratio D ₂ is first energized for the calculated non-commutation phase current control in Vcmp = 0 above The correction of Vcmp for the rate D1 is performed by the following equation (18). D ₁ = D ₁ −D ₂ · sgn (Ic) / 3 (18)

The basis of this equation is as follows: Equation (5) and Equation (16)
Or the difference between Equations (6) and (17) is Vcmp / 3
Which is apparent from the relationship of Vcmp / 3 = D ₂ · Vd / 3. Subsequently, the process proceeds to S111. If not commutation in the S103, the first duty ratio D ₁ at S110 calculates D ₁ = Y ₂ / V
Perform d. Second duty ratio D ₂ used only during commutation at the same time
= 0, and the process proceeds to S111. In step S111, the first duty ratio D ₁ and the second duty ratio D ₂ are set in the control output phase selection means 14 and the three-phase PWM circuit 15 shown in FIG.

The rectangular-wave drive control apparatus for a trapezoidal-wave magnetized brushless motor according to the present invention is configured as described above. Therefore, the required range of the neutral point potential fluctuation in the commutation section can be reduced. The output of the controller does not saturate (the duty ratio is fixed to the maximum value) over a wider range, and continuous feedback control in which the non-commutation current is maintained at its target value is possible. Torque ripple can be greatly reduced, and noise and vibration can be reduced.

Embodiment 2 FIG. 7 is a flowchart showing the operation of the current control calculation means 12 according to the second embodiment of the present invention. The difference between Embodiment 2 and Embodiment 1 is that
In order to ensure the dynamic range of non-commutation current control during commutation (so that the saturation of the first duty ratio D ₁ does not occur dynamically)
Advance first advance set the upper limit value D1max duty factor D _1, lies in the so controls the second duty ratio D ₂ so as not to exceed the upper limit value D1max. The condition for that is equation (13)
From the conditions transformed from equation (14) into equation (14), D ₁ = 1 becomes D ₁ = D
When the same transformation is performed by substituting 1max <1, the equation (1
3) can be transformed into the following equation (19). Vd> 3Rm.Id + 4Es + 3 (1-D1max) Vd (19)

Next, similarly to the above-described relationship between Expressions (14) and (12), Expression (19) is expressed by the following expression using Vcmp. Vcmp = 3Rm · Id + 4Es−Vd + 3 (1-D1max) Vd (20) Assuming that the circuit has n phases, the equation (20) is expressed as Vcmp = nRm · I
d + (n + 1) Es-Vd + n (1-D1max) Vd. That is, the difference between the first embodiment and the second embodiment resides in that the equation (19) is applied instead of the equation (12). In the flowchart of FIG. 7, the same processing blocks as those in FIG. D1max good even fixed value (e.g. 0.85) in S201, but is more suitable for the dynamic compensation by setting as a function of D _1. FIG. 8 shows an example of this function. Next, in S202, the equation (20)
Is calculated. In the following, the same procedure as that of the first embodiment is performed, and the description is omitted.

Embodiment 3 FIG. 9 is a flowchart showing the operation of the current control calculation means 12 according to Embodiment 3 of the present invention. The difference between the third embodiment and the first embodiment is as follows.
A function of adjusting the compensation voltage Vcmp is added in order to improve the robustness of non-commutation current control during commutation with respect to variation or change in the characteristics of components. In particular, there is a possibility that the resistance value change (difference between the resistance R and the nominal resistance Rm) due to the variation of the circuit resistance R such as the winding resistance of the brushless motor, the wiring resistance of each part, the internal resistance of the switching element and the temperature change. Therefore, this is mainly compensated for. In FIG. 9, the same processing blocks as those in FIG. 6 are denoted by the same numbers (100s), and the description is omitted because they have the same contents.

In step S301, for the next commutation section, the deviation ε in the current commutation section is divided by the target current Id to normalize the current, and the value multiplied by the evaluation coefficient Kt is integrated. Perform the calculation. The correction integrated value is stored in the Y _4.
In S302, the compensation voltage Vcmp using a correction integral value Y ₅ previous calculated by the following equation (21). _{Vcmp = 3 (Rm + R 5} ) Id + If 4Es-Vd ......... (21) circuit and n-phase, the formula (21) is Vcmp = n (Rm +
Y ₅ ) Id + (n + 1) Es−Vd. That is, at the time of the previous commutation, the first duty ratio D ₁ is saturated, and the actual current Is has not reached the target current Id (since ε> 0, Y ₅ >).
0), this time compensated voltage Vcmp By increasing in proportion to Y _5, is modified in a direction to prevent the first saturation duty factor D _1, would reduce the torque ripple of the brushless motor.

Here, Y ₅ is an integrated value only in the pre-rotational flow section, but may be a value reflecting the integrated value two or three times before or before. In S303 and S304, the Y ₄ is the final integration value in the current commutation next Vcmp calculation (S30
With transferring the Y ₅ to be used for 2), then initialize the Y ₄ for Y ₄ operation during commutation (Y ₄ = 0) is. The following operation is the same as that of the first embodiment, and thus the description is omitted.

[0048]

According to the drive control apparatus for a brushless motor according to the first aspect of the present invention, a plurality of switching elements connected in series between a positive electrode and a negative electrode of a DC power supply, and a connection point between the switching elements and a brushless motor. A current detector for connecting a motor and detecting a current flowing through the connection path;
A position detector for detecting a magnetic pole position of the brushless motor, motor generation voltage calculating means for calculating an induced voltage per phase generated in a drive coil of the brushless motor, and a position detection signal obtained from the position detector. A drive signal generating means for selectively energizing the switching element, and calculating a first duty ratio as an output of the drive signal generating means such that a feedback current based on the detection current detected by the current detector matches a target current. A switching unit connected to a phase where the commutation current becomes zero so as to reduce the neutral point potential fluctuation of the brushless motor based on the first calculation means, the voltage of the DC power supply, the output of the motor generation voltage calculation means and the target current. Since the second calculating means for calculating the second duty ratio for the element and the correcting means for correcting the first duty ratio based on the second duty ratio are provided, The required range of the neutral point potential fluctuation in the section can be reduced, the controller output does not saturate (the duty ratio is fixed to the maximum value) over a wider range, and the non-commutation current reaches its target. Continuous feedback control that is maintained at a value becomes possible, and as a result, torque ripple of the motor can be significantly reduced, and noise and vibration can be reduced.

According to the brushless motor drive control apparatus according to the second aspect of the present invention, the circuit has n phases, the circuit resistance per phase is R, the target current is Id, the power supply voltage is Vd,
Assuming that the generated voltage per phase is Es, the second duty ratio is a function of nId · R + (n + 1) Es−Vd.
Since the neutral point potential fluctuation can be predicted and controlled based on the detection signal, good control responsiveness can be realized.

According to the brushless motor drive control device of the third aspect of the present invention, the circuit has n phases, the circuit resistance per phase is R, the target current is Id, the power supply voltage is Vd,
When the power generation voltage per phase Es, the circuit resistance correction amount deviation is proportional to the value obtained by dividing the target current Id to the target current Id and the feedback current Is was Y _5, second energization rate nId (R + Y ₅ ) + (N + 1) Es−Vd, so that it is possible to correct for variations and changes in components, and to perform more suitable control.

According to the drive control device for a brushless motor according to a fourth aspect of the present invention, the switching element connected to the phase where the commutation current is zero so that the first duty ratio is equal to or less than the predetermined maximum value. , The dynamic control range of the non-commutation current control corresponding to the variation or change of the components can be secured, and suitable control can be performed.

According to the drive control apparatus for a brushless motor according to the fifth aspect of the present invention, the maximum value is a function of the first duty ratio. Thus, a suitable control range can be secured, and suitable control can be performed.

According to the drive control device for a brushless motor according to claim 6 of the present invention, the circuit has n phases, the circuit resistance per phase is R, the target current is Id, the power supply voltage is Vd,
Assuming that the generated voltage per phase is Es and the maximum value is D1max, the second duty ratio is nIdR + (n + 1) Es-Vd
Since the function is set to + n (1-D1max) Vd, a dynamic control range of the non-commutation current control corresponding to the variation or change of the components can be secured, and more suitable control can be performed.

[Brief description of the drawings]

FIG. 1 is a system configuration diagram showing a drive control device for a brushless motor according to a first embodiment of the present invention.

FIG. 2 is an electric signal waveform diagram of each part.

FIG. 3 is a table showing examples of steps, duty ratios, and FB current selection.

FIG. 4 is a control block diagram of a drive control device.

FIG. 5 is a table showing examples of steps, duty ratios, and FB current selection.

FIG. 6 is a control flowchart showing an operation of the drive control device.

FIG. 7 is a control flowchart showing an operation of the drive control device according to the second embodiment of the present invention.

8 is a diagram showing the relationship between D1max and D _1.

FIG. 9 is a control flowchart showing an operation of the drive control device according to the third embodiment of the present invention.

FIG. 10 is a control block diagram showing a conventional brushless motor drive control device.

[Explanation of symbols]

1 brushless motor, 3 drive coils, 4a, 4b,
4c position detector, 5 the DC power supply, 6u, 6w current detector, 10 a motor generated voltage calculating unit, Q ₁ to Q ₆ switching elements.

Continuation of the front page F term (reference) 5H560 BB04 BB12 DA02 DA19 DC12 EB01 GG04 RR01 SS01 UA05 XA02 XA12 5H570 BB07 CC01 DD04 GG01 HA08 HB03 HB16 JJ03 KK06 LL02 LL15

Claims (6)

[Claims] 1. A current detector for connecting a plurality of switching elements connected in series between a positive electrode and a negative electrode of a DC power supply, a connection point of the switching elements and a brushless motor, and detecting a current flowing through the connection path. A position detector for detecting a magnetic pole position of the brushless motor, a motor generation voltage calculating means for calculating an induced voltage per phase generated in a drive coil of the brushless motor, and a position detection obtained from the position detector A drive signal generating means for selectively energizing the switching element based on the signal; and an output of the drive signal generating means such that a target current matches a feedback current based on a detection current detected by the current detector. First calculating means for calculating a first duty ratio; and a voltage of the DC power supply, an output of the motor power generation voltage calculating means, and the target current. A second calculating means for calculating a second duty ratio for the switching element connected to the phase where the commutation current is zero, so as to reduce a neutral point potential fluctuation of the brushless motor; And a correction means for correcting the first duty ratio based on the following. When the circuit has n phases, the circuit resistance per phase is R, the target current is Id, the power supply voltage is Vd, and the power generation voltage per phase is Es, the second duty ratio is nId · R +
2. The brushless motor drive control device according to claim 1, wherein the drive control device is a function of (n + 1) Es-Vd. 3. The circuit has n phases, R is the circuit resistance per phase, Id is the target current, Vd is the power supply voltage, Es is the power generation voltage per phase, and the deviation between the target current Id and the feedback current Is. when was the circuit resistance correction amount Y ₅ in proportion to a value obtained by dividing the target current Id, second energization rate
_{nId (R + Y 5) +} (n + 1) brushless motor drive control device according to claim 1 which is a function of Es-Vd. 4. The method according to claim 1, wherein the second duty ratio for the switching element connected to the phase where the commutation current is zero is calculated so that the first duty ratio is equal to or less than a predetermined maximum value. A drive control device for a brushless motor according to any one of the preceding claims. 5. The drive control device for a brushless motor according to claim 4, wherein the maximum value is a function of the first duty ratio. 6. When the circuit has n phases, the circuit resistance per phase is R, the target current is Id, the power supply voltage is Vd, the power generation voltage per phase is Es, and the maximum value is D1max. The rate is nId · R + (n + 1) Es−Vd + n (1-D
6. The brushless motor drive control device according to claim 4, wherein the drive control device is a function of 1max) Vd.