JP2000235424A - Current mirror circuit and current sensor and switching circuit and switching device equipped with them - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a current sensor with high current detecting precision which can be loaded on a switching device capable of a quick response at the time of the generation of incomplete short-circuit. SOLUTION: This current sensor is provided with a first semiconductor element Q93 having first and second main electrodes and a control electrode, a second semiconductor element Q94 having a first main electrode and a control electrode connected with the first main electrode and control electrode of the first semiconductor element Q93 and a second main electrode, a comparator CMP401 whose first input terminal is connected with the second main electrode of the first semiconductor element Q93, and whose second input terminal is connected with the second main electrode of the second semiconductor element Q94, and a third semiconductor element Q95 whose first main electrode is connected with the second input terminal of the comparator CMP401, whose control electrode is connected with the output terminal of the comparator CMP401, and whose second main electrode is connected with a reference resistance Rr2. Then, currents flowing through the first semiconductor element Q93 can be detected by detecting currents flowing through the reference resistance Rr2.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a current mirror circuit, a current sensor, a switching circuit having the same, and a switching device, and more particularly to a semiconductor switching device suitable for a power supply control device.

[0002]

2. Description of the Related Art Semiconductor switching devices (power semiconductor devices) used in conventional power supply control devices include:
For example, there is one shown in FIG. The power supply control device shown in FIG. 9 is a device for selectively supplying power from a battery to each load in an automobile and controlling power supply to the load by a thermal FET QF. In the power supply control device shown in FIG. 9, one end of a shunt resistor RS is connected to a power supply 101 that supplies an output voltage VB, and the other end is connected to a drain terminal D of a thermal FET QF. Further, a load 10 is connected to the source terminal S of the thermal FET QF.
2 are connected. Here, the load 102 corresponds to a headlight of an automobile, a drive motor of a power window, and the like. The power supply control device shown in FIG.
A driver 901 that detects the current flowing through the shunt resistor RS and controls the drive of the thermal FET QF by a hardware circuit, and an A / D converter that controls the drive signal of the thermal FET QF on / off based on the current value monitored by the driver 901. Device 902 and microcomputer (CPU) 903
And

As shown in FIG. 10, a thermal FET QF operating as a main device of a semiconductor switching device includes a power device (main FET) QM, a resistor RG, a temperature sensor 121, a latch circuit 122, and an overheat shutoff FE.
It has a built-in TQS, and has an overheat cutoff function of forcibly turning off the thermal FET QF by a built-in gate cutoff circuit when the junction temperature of the thermal FET QF rises to a specified temperature or higher. That is, when the temperature sensor 121 detects that the temperature of the power device (main FET) QM has risen to a temperature equal to or higher than the specified value,
The detection information to that effect is held in the latch circuit 122, and the overheat cutoff FET QS as the gate cutoff circuit is turned on, thereby forcibly turning off the power device QM. Here, the temperature sensor 121 is formed by continuously connecting four diodes made of polysilicon or the like, and the temperature sensor 121 is integrated near the power device QM. As the temperature of the power device QM rises, the reverse leakage current of the temperature sensor 121 increases, and when the voltage across the four diodes lowers the gate potential of the FET Q51 to a potential that is set to “L” level, FE
TQ51 transitions from the on state to the off state. As a result, the gate potential of the FET Q54 is pulled up to the potential of the gate control terminal G of the thermal FET QA.
5 changes from the off state to the on state, and the latch circuit 12
"1" is latched at 2. At this time, the output of the latch circuit 122 becomes “H” level, and the overheat cutoff FET QS transitions from the off state to the on state, so that the potential level of the true gate TG of the power device QM becomes “L” level. , The power device QM transitions from the on-state to the off-state, and is overheated.

In FIG. 9, ZD1 is a thermal FET Q
Keeping the voltage between the gate terminal G and the source terminal S of F at 12V,
This is a Zener diode that bypasses an overvoltage applied to the true gate TG of the power device QM. The driver 901 includes a differential amplifier 911, 913 as a current monitor circuit, a differential amplifier 912 as a current control circuit, a charge pump circuit 915,
The drive circuit 914 is configured to drive the true gate G of the thermal FET QF via the internal resistor RG based on the on / off control signal from the microcomputer 903 and the overcurrent determination result from the current limiting circuit. Shunt resistor R
If an overcurrent is detected via the differential amplifier 912 based on the voltage drop of S and the current exceeds the determination value (upper limit), the drive circuit 914 turns off the thermal FET QF, and then the current decreases. Then, when the value falls below the determination value (lower limit), the thermal FET QF is turned on. On the other hand, the microcomputer 903 includes a current monitor circuit (differential amplifier 9).
11, 913), the current is constantly monitored, and if an abnormal current exceeding the normal value is flowing, the thermal FET
By turning off the drive signal of the QF, the thermal FET Q
F is turned off. If the temperature of the thermal FET QF exceeds the specified value before the drive signal for the off control is output from the microcomputer 903, the thermal FET QF is turned off by the overheat cutoff function.

On the other hand, a semiconductor switching device for a power supply control device as shown in FIG. 11 is also known. FIG.
1 has an output voltage VB
, A drain electrode terminal of a power MOSFET and a drain electrode terminal of a current mirror element are connected together. Power MOS
The load 401 is connected to the source terminal of the FET. In the semiconductor switching device shown in FIG. 9, a resistor R411 is further connected between the source electrode terminal of the power MOSFET and the source electrode terminal of the current mirror element, and a potential difference E between both ends of the resistor R411 is compared with a comparator 400.
Are compared. The output of the comparator is input to the control circuit 403, and the control circuit 403 controls the drive circuit 402. The driving circuit 402 is a power MOSF
A gate voltage is supplied to the gate electrode terminals of the ET and the current mirror element to control on / off of the power MOSFET and the current mirror element.

[0006]

However, the above-described conventional power supply control device requires a shunt resistor RS connected in series to a power supply path for current detection. By increasing the load current,
There is a problem that the heat loss of the shunt resistor cannot be ignored.

The above-mentioned overheat cutoff function and overcurrent control circuit function when a large current flows due to the almost complete short-circuit state occurring in the load 102 and the wiring, but the imperfections having a certain degree of short-circuit resistance When a small short-circuit current flows due to a rare short-circuit such as a short-circuit, it does not function, and the microcomputer 903 detects an abnormal current via a current monitor circuit and controls the thermal FET QF to be turned off. In some cases, the responsiveness of microcomputer control to abnormal current is poor.

Further, the shunt resistor RS and the A / D converter 9
02, the microcomputer 903 and the like are required, so that a large mounting space is required, and there is also a problem that these relatively expensive articles increase the apparatus cost.

Further, the current mirror circuit as shown in FIG. 11 is affected by variations in the value of the resistor R411 connected between the source electrode terminal of the power MOSFET and the source electrode terminal of the current mirror element, variations in the shunt ratio, and the like. There was a problem that the system was susceptible to credibility and lacked reliability. In addition, the current mirror circuit should originally operate with the same potential between the source electrode terminal of the power MOSFET and the source electrode terminal of the current mirror element. However, in the current mirror circuit as shown in FIG. There is a problem that it is difficult to make the same potential and accurate measurement is impossible.

An object of the present invention is to solve the above-mentioned conventional problems and circumstances, and to provide a current mirror circuit operable under ideal conditions.

Another object of the present invention is to provide a current sensor capable of accurately detecting a current.

Still another object of the present invention is to eliminate the need for a shunt resistor and to enable a high-speed response to an abnormal current when a rare short-circuit such as an incomplete short-circuit having a certain short-circuit resistance occurs. To provide a semiconductor switching circuit that is easy to use.

Still another object of the present invention is to eliminate the need for a shunt resistor that is directly connected to a power supply path for detecting current, thereby suppressing heat loss of the device and preventing an incomplete short circuit having a certain degree of short circuit resistance. An object of the present invention is to provide an inexpensive semiconductor switching device that enables high-speed response to an abnormal current when a rare short circuit occurs and that is easy to integrate.

[0014]

According to the present invention, there is provided a semiconductor device comprising: a first semiconductor device having first and second main electrodes and a control electrode; A second semiconductor element having a first main electrode, a control electrode, and a second main electrode respectively connected to the first main electrode and the control electrode; and a first main electrode of the first semiconductor element. And a comparator having a second input terminal connected to the second main electrode of the second semiconductor element, a first main electrode connected to the second input terminal of the comparator, and an output of the comparator. A first feature is that the current mirror circuit includes a third semiconductor element having a control electrode connected to a terminal and a reference resistor connected to a second main electrode. Here, as the first to third semiconductor elements, an FET, a static induction transistor (SIT), or a bipolar transistor (BJT) can be used. MOS composite devices such as emitter-switched thyristors (EST) and MOS-controlled thyristors (MCT); IGBTs;
Other insulated gate power devices can be used. These semiconductor elements may be of an n-channel type or a p-channel type. The “first main electrode” is one of an emitter electrode and a collector electrode in a BJT or IGBT, and an IGFE such as a MOSFET or a MOSSIT.
T means either a source electrode or a drain electrode. "Second main electrode" refers to BJT or IGBT
Means one of the emitter electrode and the collector electrode which does not become the first main electrode, and the IGFET means one of the source electrode and the drain electrode which does not become the first main electrode. That is, if the first main electrode is an emitter electrode, the second main electrode is a collector electrode, and if the first main electrode is a source electrode, the second main electrode is a drain electrode. "Control electrode" means BJ
Needless to say, it means the gate electrodes of T, IGBT and IGFET.

In the current mirror circuit according to the first aspect of the present invention, the potential of the second main electrode of the first semiconductor element is made equal to the potential of the second main electrode of the second semiconductor element. ,
Since the operation of this ideal current mirror circuit can be realized, it is suitable for applications such as extremely accurate current measurement.

A second feature of the present invention resides in that a first semiconductor element having first and second main electrodes and a control electrode, and a first semiconductor element connected to the first main electrode and the control electrode of the first semiconductor element, respectively. A second semiconductor element having a first main electrode, a control electrode, and a second main electrode, a driving circuit that supplies a voltage to each control electrode of the first and second semiconductor elements, Connecting a first input terminal to a second main electrode of the first semiconductor element;
A comparator having a second input terminal connected to a second main electrode of the second semiconductor element; a first main electrode connected to a second input terminal of the comparator; a control electrode connected to an output terminal of the comparator; And a third semiconductor element having a reference resistance connected to the main electrode of the first semiconductor element. A current flowing in a load connected to the second main electrode of the first semiconductor element is detected by detecting a current flowing through the reference resistance. Current sensor. Here, the first to third
FET, SIT, or BJ
T is available. In addition, various MOS composite devices and other insulated gate power devices such as IGBTs can be used. These semiconductor elements may be of an n-channel type or a p-channel type. In the case of a BJT or IGBT, the term “first main electrode” refers to one of an emitter electrode and a collector electrode, and a MOSFET or a MOSSIT.
In IGFETs, etc., it means either the source electrode or the drain electrode. "Second main electrode" means BJ
In the case of T or IGBT, one of the emitter electrode and the collector electrode which does not become the first main electrode, IGFE
T means one of a source electrode and a drain electrode that does not become the first main electrode. That is,
If the first main electrode is an emitter electrode, the second main electrode is a collector electrode, and if the first main electrode is a source electrode,
The second main electrode is a drain electrode. In addition, "control electrode"
Of course, BJT means the gate electrodes of BJT, IGBT and IGFET.

A current sensor according to a second aspect of the present invention comprises:
A so-called "current mirror circuit"
The current mirror circuit can be operated by making the potential of the second main electrode of the semiconductor element equal to the potential of the second main electrode of the second semiconductor element, so that extremely accurate current measurement can be performed.

In the second feature of the present invention, the first semiconductor element may include, for example, a plurality of unit cells (unit cells).
It is possible to employ a power device having a multi-channel structure connected in parallel. Then, the shunt ratio may be determined by adjusting the number of unit cells connected in parallel constituting each semiconductor element so that the current capacity of the second semiconductor element is smaller than the current capacity of the first semiconductor element. . For example, by configuring the number of unit cells of the first semiconductor element to be 1000 for the number of unit cells of the second semiconductor element, the channel width of the second semiconductor element and the first semiconductor element can be increased. The split ratio can be determined by setting the ratio of W to 1: 1000.

A third feature of the present invention is that a first semiconductor element having first and second main electrodes and a control electrode, and a first semiconductor element connected to the first main electrode and the control electrode of the first semiconductor element, respectively. A second semiconductor element having a first main electrode, a control electrode, and a second main electrode, and a first main electrode connected to the first main electrode and the control electrode of the first semiconductor element, respectively. A third semiconductor element having an electrode, a control electrode, and a second main electrode;
A comparator having a first input terminal connected to the second main electrode of the first semiconductor element and a second input terminal connected to the second main electrode of the third semiconductor element; 1st input terminal
A fourth semiconductor element in which a control electrode is connected to the output terminal of the comparator and a reference resistor is connected to the second main electrode;
And a control voltage supply unit that supplies a control voltage to each control electrode of the first to third semiconductor elements according to an output of the comparison unit. At least, an abnormal current flowing through the first semiconductor element is detected, and when an abnormal current occurs, the first current is detected.
On / off control of the semiconductor element of
A switching circuit that interrupts the conduction state of the first semiconductor element by this current oscillation and detects the value of the current flowing through the first semiconductor element by detecting the current in the reference resistor.

When, for example, a power MOSFET is used as the first semiconductor element, the terminal voltage (drain-source voltage) of the power MOSFET forming a part of the power supply path changes from the off state to the on state. The voltage characteristics at the time (for example, the fall in the case of an n-channel FET) change according to the state of the power supply path and the load, that is, the time constant based on the wiring inductance and the wiring resistance and the short-circuit resistance of the path. For example,
In a normal operation in which a short circuit has not occurred, the voltage quickly falls below a predetermined voltage. However, in a case where a complete short circuit has occurred, the voltage does not fall below the predetermined voltage. In addition, when an incomplete short circuit having a certain short-circuit resistance occurs, although it converges to a predetermined voltage, it takes a long time to converge.

A third feature of the present invention utilizes a transient voltage characteristic of the semiconductor element when transitioning from an off state to an on state in such a semiconductor element. That is, by detecting the difference between the voltage between the terminals of the first semiconductor element and the voltage between the terminals of the first semiconductor element (reference voltage), the voltage between the terminals of the first semiconductor element forming a part of the power supply path is detected. At the same time as determining the degree to which the voltage (that is, the current in the power supply path) deviates from the normal state, the potential of the second main electrode of the first semiconductor element, which constitutes a so-called current mirror circuit, and the third By equalizing the potential of the second main electrode of the semiconductor element and operating this current mirror circuit,
This enables extremely accurate current measurement.

Therefore, it is possible to eliminate the need for a conventional shunt resistor connected in series to a power supply path for current detection, and to reduce not only an overcurrent due to a complete short circuit but also a certain short-circuit resistance. It is also possible to easily detect an abnormal current when a rare short such as an incomplete short circuit occurs.

A fourth feature of the present invention is that a first semiconductor element having a first main electrode connected to an input terminal, a second main electrode connected to an output terminal, and a control electrode; A second semiconductor element having a first main electrode, a control electrode, and a second main electrode respectively connected to the first main electrode and the control electrode of the first semiconductor element; A third semiconductor element having a first main electrode, a control electrode, and a second main electrode connected to the main electrode and the control electrode, respectively;
A comparator having a first input terminal connected to the second main electrode of the first semiconductor element and a second input terminal connected to the second main electrode of the third semiconductor element; 1st input terminal
A fourth semiconductor element in which a control electrode is connected to the output terminal of the comparator and a reference resistor is connected to the second main electrode;
And a control voltage supply unit that supplies a control voltage to each control electrode of the first to third semiconductor elements according to an output of the comparison unit. At least, an abnormal current flowing to a load connected to the output terminal is detected, and when an abnormal current occurs, the first semiconductor element is turned on / off to generate a current oscillation. A switching device that interrupts a conduction state between terminals and detects a value of a current flowing in a load by detecting a current in a reference resistor.

In the fourth aspect of the present invention, it is preferable that the first to third semiconductor elements, the comparing means, and the control voltage supply means are integrated on the same semiconductor substrate.

When, for example, a power MOSFET is used as a first semiconductor element constituting a semiconductor switching device, a power MOSFET forming a part of a power supply path
The voltage between the terminals of T (drain-source voltage) is the state of the power supply path and the load, that is, the voltage characteristic when transitioning from the off state to the on state (for example, falling in the case of an n-channel FET). And the time constant based on the wiring inductance of the path and the wiring resistance and the short-circuit resistance. For example, in a normal operation in which a short circuit does not occur, the voltage quickly decreases to a predetermined voltage or less, but when a complete short circuit occurs, the voltage does not decrease to a predetermined voltage or less. In addition, when an incomplete short circuit having a certain short-circuit resistance occurs, although it converges to a predetermined voltage, it takes a long time to converge.

A fourth feature of the present invention utilizes a transient voltage characteristic of the semiconductor element when transitioning from an off state to an on state in such a semiconductor element. That is, by detecting the difference between the voltage between the terminals of the first semiconductor element and the voltage between the terminals of the first semiconductor element (reference voltage), the voltage between the terminals of the first semiconductor element forming a part of the power supply path is detected. At the same time as determining the degree to which the voltage (that is, the current in the power supply path) deviates from the normal state, the potential of the second main electrode of the first semiconductor element, which constitutes a so-called current mirror circuit, and the third By equalizing the potential of the second main electrode of the semiconductor element and operating this current mirror circuit,
This enables extremely accurate current measurement.

Therefore, the heat loss of the device can be suppressed by eliminating the need for the conventional shunt resistor connected in series to the power supply path for detecting the current. It is also possible to easily detect an abnormal current when a rare short-circuit such as an incomplete short-circuit having a certain short-circuit resistance occurs. Furthermore, overcurrent can be detected without using a shunt resistor. In particular, when the on / off control of the semiconductor switching device is configured by a hardware circuit, a microcomputer is not required. The unit price can be reduced.

In particular, the ratio of the number of unit cells constituting each semiconductor element may be determined so that the current capacities of the second and third semiconductor elements are smaller than the current capacity of the first semiconductor element. . By selecting the unit cell number and setting the layout of the planar pattern of the power IC, the circuit configuration of the second and third semiconductor elements can be reduced in size, and the area of the semiconductor chip can be reduced. In addition, the equipment cost can be significantly reduced.

[0029]

Referring to the drawings, a current sensor will be described as a first embodiment of the present invention, and a switching device will be described as a second embodiment of the present invention. In the following description of the drawings, the same or similar parts are denoted by the same or similar reference numerals.

(First Embodiment: Current Sensor) A current sensor according to a first embodiment of the present invention, as shown in FIG. 1, has a first and a second main electrode and a control electrode. One semiconductor element Q93, a first main electrode connected to a first main electrode and a control electrode of the first semiconductor element Q93, respectively,
A first input terminal is connected to a second semiconductor element Q94 having a control electrode and a second main electrode, and a second main electrode of the first semiconductor element Q93. 2
CMP40 in which the second input terminal is connected to the main electrode of
1, a first main electrode at a second input terminal of the comparator CMP401, a control electrode at an output terminal of the comparator CMP401, and a second
And a third semiconductor element Q95 in which a reference resistor Rr2 is connected to the main electrode of the third semiconductor element Q95. The current sensor detects a current flowing through the first semiconductor element Q93 by detecting a current flowing through the reference resistor Rr2.

In FIG. 1, n-channel MOSFETs are used as the first and second semiconductor devices Q93, Q94. Further, a pnp type B is used as the third semiconductor element Q95.
JT is used. A resistor R412 is connected between the base and the emitter of the third semiconductor element Q95, and a resistor R412 is connected between the base of the third semiconductor element Q95 and the output terminal of the comparator CMP401. Thus, the third semiconductor element (pnp type BJT) Q95
It is biased to operate in the unsaturated region.

As the n-channel MOSFET as the first semiconductor element Q93, for example, a power device having a multi-channel structure in which a plurality of unit cells (unit cells) are connected in parallel can be used. And the second
A semiconductor device having a similar multi-channel structure can be adopted for the n-channel MOSFET as the semiconductor element Q94. The division ratio may be determined by adjusting the number of unit cells connected in parallel constituting each semiconductor element so that the current capacity of the second semiconductor element Q94 is smaller than the current capacity of the first semiconductor element Q93. . For example, by configuring the number of unit cells of the first semiconductor element Q94 to be 1000 for the number of unit cells of the second semiconductor element Q94, the second semiconductor element Q94 and the first
The ratio of the channel width W of the semiconductor element Q93 to 1: 1000
As a result, the shunt ratio can be determined.

The current sensor according to the first embodiment of the present invention constitutes a so-called "current mirror circuit". The output of the comparator CMP401 becomes valid ("H" level) when the potentials supplied to its "+" and "-" input terminals substantially match, and invalid ("L") when they do not match. Level). For example, when the potential of the node N1 becomes higher than the potential of the node N2, the comparator CMP401 outputs an “L” level.
When the output of the comparator CMP401 becomes “L” level, p
Since the base potential of np-type BJT (third semiconductor element) Q95 decreases, pnp-type BJT (third semiconductor element) Q95
95 tends to increase the current value. Then, the potential drop at both ends of the reference resistor Rr2 increases, so that the node N
2 has a higher potential. That is, the comparator CMP401 keeps “L” until the potentials of the node N1 and the node N2 become equal.
Output level. In this manner, the potential at the node N1 of the second main electrode of the first semiconductor element Q93 is made equal to the potential at the node N2 of the second main electrode of the second semiconductor element Q94, and this current mirror The circuit can be operated as an ideal current mirror circuit. Therefore, extremely accurate current measurement can be performed.

In FIG. 1, MOSFETs have been exemplified as the first and second semiconductor elements.
MOSSIT, various MOS composite devices, IGBT and the like can be used. FIG. 1 shows an n-channel type, but a p-channel type may be used. Similarly, a pnp-type BJT biased to operate in an unsaturated region is illustrated as the third semiconductor element, but an npn-type BJT may be used if the bias relationship is reversed. Also,
In addition, FETs, SITs, various MOS composite devices, and IGBTs can be used as the third semiconductor element.

(Second Embodiment: Switching Device) A switching device with a current oscillation type cutoff function according to a second embodiment of the present invention has a structure as shown in FIG.
A first semiconductor element QA serving as a main device (power device) and an abnormal current of the main device (first semiconductor element) QA are detected, and when an abnormal current occurs, the main device QA is turned on / off to perform current oscillation. And a control circuit for interrupting the main device QA by the current oscillation is integrated on the same substrate. A hybrid IC using an insulating substrate such as ceramic or glass epoxy, an insulating metal substrate, or the like may be used as the substrate, but more preferably a power IC monolithically integrated on the same semiconductor substrate (same chip). Good.

Normally, this power IC operates by being connected between a power supply 101 for supplying an output voltage VB and a load 102. In FIG. 2, a semiconductor switching element QA having a thermal cutoff function is used as a main device (power device) of a power IC. As the semiconductor switching element QA having a thermal cutoff function, for example, FIG.
0 may be used (in the following second embodiment of the present invention, a thermal FET QF is used).
Will be described). In addition, as can be understood from the following description, when the on / off number-of-times integration circuit (number-of-times control means) is provided, the thermal cutoff function is not essential.
The semiconductor switching element (first semiconductor element) QA is
It has first and second main electrodes and control electrodes. This first
As a semiconductor element of, for example, a DMOS structure, a VMO
A power MOSFET having an S structure or a UMOS structure or a MOSSIT having a structure similar to these can be used. Also, MOS composite devices such as EST and MCT, IGB
Other insulated gate power devices such as T can be used. Furthermore, if you always use the gate with reverse bias,
A junction FET, a junction SIT, an SI thyristor, or the like can also be used. The semiconductor switching element QA as a main device (power device) of the power IC may be an n-channel type or a p-channel type. That is, the switching device with the current oscillation type interruption function according to the second embodiment of the present invention includes both an n-channel type and a p-channel type.

Referring to FIG. 2, a description will be given of a switching device having an n-channel current oscillation type interruption function monolithically integrated on the same semiconductor substrate. FIG.
As shown in the figure, the control circuit of the switching device with a current oscillation type interruption function according to the second embodiment of the present invention includes a first reference device (A) connected in parallel with a main device (first semiconductor element) QA. FE as a second semiconductor element)
TQB, FET QC as a second reference device (third semiconductor element), comparison means (CMP1) for comparing the main electrode voltage of main device QA with the main electrode voltage of reference device QB, and this comparison At least a control voltage supply unit 111 that supplies a control voltage to control electrodes of the main device QA and the reference device QB according to the output of the unit (CMP1). Here, each of the first to third semiconductor elements QA, QB, and QC has a pair of main electrodes each including first and second main electrodes. For example, the first and second main electrodes of the main device (first semiconductor element) QA are connected to the first and second main electrode regions of a power device constituting the main device, respectively. The “first main electrode region” means one of an emitter region and a collector region in an IGBT, and one of a source region and a drain region in an IGFET (power IGFET) such as a power MOSFET or a power MOSSIT. The “second main electrode region” is one of an emitter region and a collector region that does not become the first main electrode region in the IGBT, and a source region or a drain region that does not become the first main electrode region in the power IGFET. Means either one of That is, if the first main electrode region is an emitter region, the second main electrode region is a collector region, and if the first main electrode region is a source region, the second main electrode region is a drain region. The “control electrode” means, of course, the gate electrodes of IGBTs and power IGFETs. Second and third semiconductor elements Q having the same current-voltage characteristics as main device QA
Similarly, “main electrode” and “control electrode” are defined for B and QC.

The thermal FET QA as a main device (first semiconductor element) includes, for example, a power device (main FET) QM as shown in FIG. 10, a resistor RG connected to a true gate of the power device QM, and a temperature sensor. 12
1. FET Q5 whose gate is connected to temperature sensor 121
1. A circuit comprising a latch circuit 122 connected to the output side of the FET Q51 and an overheating cutoff FET QS having a gate connected to the output side of the latch circuit 122. The true gate of the power device QM is connected to the output side of the overheating cutoff FET QS. This thermal F
As the main FET QM of the ETQA, for example, a power device having a multi-channel structure in which a plurality of unit cells (unit cells) are connected in parallel may be used. Then, the second and third semiconductor elements Q are connected in parallel with the main FET (power device) QM of the thermal FET QA.
B and QC are arranged at positions adjacent to the thermal FET QA. The second and third semiconductor elements QB, QC
Has a temperature sensor, a latch circuit or an overheat shutoff FE
A circuit for overheating the reference device such as TQS is not essential. The second and third semiconductor elements QB and QC are:
Since the main device (main FET) QM is arranged at an adjacent position in the same process as that of the main device (main FET) QM, it is possible to eliminate (reduce) variations in electrical characteristics due to the influence of temperature drift and non-uniformity between lots. Second and third semiconductor elements QB, QC
The number of unit cells connected in parallel constituting the second and third semiconductor elements QB and QC is adjusted so that the current capacity of the main FET becomes smaller than the current capacity of the main FET. For example, the second
By configuring the number of unit cells of the main device (main FET) QM to be 1000 with respect to the unit cell number of 1 of the third semiconductor elements QB and QC, the second and third semiconductor elements QB, The ratio between QC and the channel width W of the first semiconductor element QM is set to 1: 1000. The temperature sensor 121 includes the second and third semiconductor elements QB and QB.
C and a plurality of diodes composed of a polysilicon thin film or the like deposited on an interlayer insulating film formed on the first semiconductor element QM, which are connected in series. It is integrated at a position near the area. As the temperature of the power device QM rises, the reverse leakage current of the temperature sensor 121 increases, and the voltage drop across the diodes connected in series reduces the gate potential of the FET Q51 to a potential at which the gate potential becomes “L” level. Then, the FET Q51 transitions from the on state to the off state. As a result, the gate potential of the FET Q54 is pulled up to the potential of the gate control terminal G of the thermal FET QA, and the FET Q65 transitions from the off state to the on state, so that "1" is latched by the latch circuit 122. At this time, the output of the latch circuit 122 becomes “H” level, and the overheat cutoff FET QS transitions from the off state to the on state, so that the potential level of the true gate TG of the power device QM becomes “L” level. ,
The power device QM transitions from the on-state to the off-state, and is overheated.

The switching device with a current oscillation type cutoff function according to the second embodiment of the present invention, more specifically, as shown in FIG.
B, QC, resistors R1, R2, R5, R331, a Zener diode ZD1, a diode D1, a comparator CMP1 as comparing means, and a driving circuit 1 as control voltage supplying means
11, the first input terminal is connected to the second main electrode (source electrode) of the first semiconductor element QA, and the third semiconductor element QC
And a comparator CMP401 having a second input terminal connected to a second main electrode (source electrode) of the same semiconductor substrate (semiconductor chip) 1 together with a main device (first semiconductor element) QA.
10 is monolithically mounted. In FIG.
The Zener diode ZD1 has a function of maintaining the voltage between the gate terminal G and the source terminal S of the thermal FET QA at 12 V, and bypassing an overvoltage applied to the true gate TG of the power device QM. Further, outside the semiconductor chip 110, a fourth semiconductor in which a first main electrode is connected to a second input terminal of the comparator CMP401, a control electrode is connected to an output terminal of the comparator CMP, and a reference resistor Rr2 is connected to the second main electrode. It has an element Q95, a resistor R10, and a switch SW1. A resistor R412 is connected between the base and the emitter of the fourth semiconductor element Q95, and a resistor R412 is connected between the base of the fourth semiconductor element Q95 and the output terminal of the comparator CMP401. In this way,
The fourth semiconductor element (pnp-type BJT) Q95 is biased to operate in the unsaturated region.

The switching device with the current oscillation type interruption function according to the second embodiment of the present invention functions when a user or the like turns on the switch SW1.

Driving circuit 111 as control voltage supply means
Has a source transistor Q5 whose collector side is connected to the potential VP, and a sink transistor Q6 whose emitter side is connected to the ground potential (GND), which are connected in series.
On / off control of the source transistor Q5 and the sink transistor Q6 based on a switching signal by the on / off switching of the switch SW1, and a signal for controlling the drive of these to the control electrodes of the main device (thermal FET) QA and the reference device QB. Output. The drive circuit 111 may be constituted by a MOSFET instead of the BJT shown in FIG. For example, the driving circuit 111 can be formed of CMOS. If the drive circuit 111 is constituted by MOSFETs, it becomes possible to manufacture the power IC (switching device with current oscillation type interruption function) according to the second embodiment of the present invention by a simple MOSFET manufacturing process. Also, if the drive circuit 111 is configured by BJT,
The power IC according to the second embodiment of the present invention can be manufactured by the BIMOS manufacturing process. The output voltage VB of the power supply 101 is, for example, 12 V, and the output voltage VP of the charge pump is, for example, VB + 10 V.

The first main electrode (drain electrode) of the main device (first semiconductor element) QA and the first main electrode (drain electrode) of the second and third semiconductor elements QB and QC are connected to each other and have a common potential. Has been maintained. Further, a first reference resistance Rr1 is connected as an external resistance to the second main electrode (source electrode) of the second semiconductor element QB, and an external resistance is connected to the second main electrode (source electrode) of the third semiconductor element QC. A second reference resistor Rr2 is connected as a resistor. The respective resistance values of the first reference resistor Rr1 and the second reference resistor Rr2 depend on the ratio of the channel width W of the second and third semiconductor elements QB, QC to the main device (first semiconductor element) QM. You just need to select it. For example, as described above, the second and third semiconductor elements QB and QC and the main device (main FET) Q
When the ratio of the channel width W of M is 1: 1000, the resistance may be set to be 1/1000 of the resistance value of the load 102. By setting this first reference resistor Rr1 and second reference resistor Rr2, the same drain and when the load current of the normal operation flows to the thermal FET QA - voltage V _DS between the source second and third semiconductor elements QB, QC can be generated.

First main electrode (drain electrode) and second main electrode (source electrode) of main device (first semiconductor element) QA
A series circuit of the resistors R1 and R2 is connected between them. The "+" input terminal of the comparator CMP1 shown in FIG.
The main electrode voltage voltage obtained by dividing the (drain D- source S voltage) V _DS between the resistor R1 and the resistor R2 of the thermal FETQA is supplied via a resistor R5. Also, the comparator C
The source voltage VS of the FET (second semiconductor element) QB is supplied to the “−” input terminal of MP1. That is,
The output becomes valid ("H" level) when the potentials supplied to both the "+" and "-" input terminals substantially match, and becomes invalid ("L" level) when they do not match.
Note that, as described later, the comparator CMP1 has a constant hysteresis.

FIG. 7 is a conceptual equivalent circuit diagram focusing on a main device (first semiconductor element) QA of a switching device having a current oscillation type interruption function according to a second embodiment of the present invention. . The equivalent circuit of the thermal FET QA as the main device (first semiconductor element) is represented by an equivalent current source g
_m · v _i, drain resistance rd, the gate-source capacitance C _GS, is shown in a simplified using the gate-drain capacitance C _GD and the drain-source capacitance C _DS. When the equivalent circuit of the thermal FET QA is used, the power supply 101
The power supply path from the power supply to the load 102 is represented as a circuit as shown in FIG. The load 102 includes a wiring inductance L0 and a wiring resistance R0 of the power supply path.

FIG. 6 shows the drain-source voltage V of the thermal FET QA forming a part of such a power supply path.
FIG. 9 is a transient response curve showing the falling voltage characteristics when the _DS transitions from the OFF state to the ON state when the load 102 is short-circuited, when the load is a reference load (normal operation), and when the load 102 is a resistor 1 KΩ. The fall characteristic is the transient response according to the impedance of the entire power supply path including the switching device with the current oscillation type cutoff function according to the second embodiment of the present invention, for example, the wiring inductance and the wiring resistance of the path. do.

First, the change in the drain-source voltage V _DS when the resistance of the load 102 in FIG. 6 is 1 KΩ can be considered as follows. In other words, due to the characteristics of the thermal FET QA (“HAF2001” manufactured by Hitachi) used in this measurement, the true gate-source voltage VT _GS is substantially equal to the threshold voltage 1.6 V at a drain current _ID = 12 mA.
Is maintained. Then, the charging of the true gate G of the thermal FET QA by the drive circuit 111 is continued,
Anyway go and true gate - source voltage VT _{G S} will go to rise, but the drain - source voltage V _DS is reduced, the true gate - because it increases the capacitance C _GD drain, so that absorbs the charge reaches the source voltage VT _GS - true gate. In other words, the drain-source voltage V _DS generates a capacitance that does not cause the electric charge reaching the true gate-source voltage VT _GS to raise the potential, and the true gate-source voltage VT _GS is about 1.6 V Is maintained. In other words, at each point in time after the thermal FET QA transitions to the ON state, the charge between the drain and the source is absorbed by the drive circuit 111 to absorb the charge transmitted to the gate G and keep the true gate TG voltage VT _GS constant. Voltage V
_DS .

That is, the drain-source voltage V _DS shown in FIG.
The difference from the curve when the load resistance is 1 KΩ is ΔV _DS GAP, and the charge of Q _GD = ΔV _DS GAP × C _GD is taken as the true gate−
Subtracting from the source-to-source voltage VT _GS means that the true gate-to-source voltage VT _GS becomes 1.6V. In other words, the true gate-source voltage VT _GS means that the potential has increased from 1.6 V by this charge Q _GD . This can be expressed by the following equation.

[0048]

VT _GS −1.6 = ΔV _DS GAP × C _GD / (C _GS + C _GD ) (1) That is, ΔV _DS GAP is proportional to (VT _GS −1.6 V). Incidentally, when the drain current I _D is zero only at the drain circuit and a Miller capacitance for charging the true gate - but is determined curve of the source voltage V _DS, flows a drain current I _D,
The inductance L _C of the whole circuit and the resistance R _{C of the} whole circuit
Will be affected. When the drain current I _D as a dead short (dead short) increases, the drain current rising slope of I _D is substantially determined by the overall circuit inductance L _C and the whole circuit of the resistor R _C, rising slope of the drain current I _D is converged to a constant value, thus the true gate - also becomes possible to converge the curve of source voltage VT _GS.

The ratio of the channel width W of the second semiconductor element (FET) QB to the channel width W of the main device (main FET) QM of the switching device with a current oscillation type cutoff function according to the second embodiment of the present invention is 1: If you have a current mirror circuit as 1000, a (drain current I _D B of the second semiconductor element) (drain current I _DA main device) = 1000 ×. Therefore, when I _DA = 5 A flows as the drain current of the thermal FET QA and I _DQB = 5 mA flows as the drain current of the FET QB,
Thermal FETQA and drains of FET QB - source voltage V _DS and the true gate - source voltage V
T _GS matches. In other _{words, V DSA = V DSB, VT} GS A = VT GSB
Becomes Here, V _DSA and V _DSB are respectively the thermal FE
It is the drain-source voltage of TQA and FET QB,
VT _GSA and VT _GSB are thermal FET QA and FET, respectively.
This is the true gate-source voltage of QB.

Therefore, when the FET QB has completely transitioned to the ON state, the power supply voltage VB is substantially applied to both ends of the first reference resistor Rr1.
As a load of the FET QB equivalent to the load of 5A connected to the A, the resistance value of the first reference resistor Rr1 is Rr1 = 12V
/ 5 mA = 1.4 KΩ.

Next, the operation of the switching device with a current oscillation type cutoff function according to the second embodiment of the present invention in the three-pole characteristic region will be described. Thermal FE
When T (first semiconductor element) QA transitions to the ON state,
The drain current _IDQA rises toward the final load current value determined by the circuit resistance. In addition, thermal FET Q
The true gate-source voltage VT _GSA of A takes a value determined by the drain current _IDQA , and the drain-source voltage V _DSA
This also rises while the brake is applied by the Miller effect of the capacitor capacitance C _GD due to the decrease in. Moreover, FET true gate (second semiconductor device) QB - source voltage VT _GSB until the drain current I _DQB = 5 mA (corresponding to the drain current I _DQA = 5A), the true gate -
The source-to-source voltage VT _GSB = VT _GSA increases, but after the pinch-off point is reached, the drain current I _DQB = 5 mA is constant (constant in the five-pole characteristic region), so that the true gate-source is true. The inter-voltage VT _GSB also becomes constant, and the "H"
In the case of "AF2001", the voltage is constant at about 2.7V.

[0052] Furthermore, the true gate of the thermal FET (first semiconductor element) QA - source voltage VT _GSA Since going increases according to the increase of the drain current I _DQA, true gate - source voltage VT _GSB <VT _GSA . In _{addition, V DSA = VT GSB + VT} GD, from the relationship of _{V DS B = VT GSB + VT} GD, the _{V DSA -V DSB = VT GSA -VT} GSB.
Here, the true gate-source voltage difference VT _GSA -VT _GSB
, Since represents a drain current I _DQA -5A, drain - by detecting the difference V _DSA -V _DSB source voltage, it is possible to obtain a drain current I _DQA -5A.

The drain-source voltage V _DSB of the FET (second semiconductor element) QB is directly input to the comparator CMP1, and the drain-source voltage V DS of the thermal FET QA is
_{The value of DSA divided} by R1 and resistor R2 is input to comparator CMP1. That is, if the variable resistor RV is not taken into account, V ₊ = V _DSA × R1 / (R1 + R2) (2) is input to the “+” input terminal of the comparator CMP1. Become. Immediately after the thermal FETQA transitions to the ON state, (2) the potential of "+" input terminal of the comparator CMP1 V ₊ is determined by the equation, the drain of the FET QB - source voltage V _DSB> V _+. But thermal FET
As the drain current I _{DQA of} QA increases, V ₊ given by equation (2) increases, and eventually becomes larger than the drain-source voltage V _DSB of FET QB. At this time, the output of comparator CMP1 becomes “ The thermal FET QA changes from the "H" level to the "L" level to be turned off.

In the comparator CMP1, the diode D
Hysteresis is formed by 1 and the resistor R5. When the thermal FET QA transitions to the off state, the drive circuit 11
1, the gate potential is grounded by the sink transistor Q6, and the potential on the cathode side of the diode D1 is V _DSA −0.
Since the voltage becomes 7 V (the forward voltage of the Zener diode ZD1), the diode D1 conducts. As a result, the resistance R1
→ A current flows through the path of the resistor R5 → the diode D1, and the potential V ₊ of the “+” input terminal of the comparator CMP1 is
11 is lower than when ON control is performed. Therefore, the thermal FET is reduced to a specific drain-source voltage difference V _DSA -V _DSB which is smaller than immediately after the transition to the off state.
QA maintains the off state, but the potential of input terminal V ₊ further decreases, so that the output of comparator CMP1 changes from “L” level to “H” level. Therefore,
After a lapse of a certain time, the thermal FET QA is turned on again. It should be noted that there are various methods for attaching the hysteresis characteristic, but this is one example.

[0055] The drain when the thermal FETQA transition to the OFF state - threshold V _DSAth the voltage V _{DS A-source}
Then, the following equation is established.

V _DSAth −V _DSA = R2 / R1 × V _DSB (3) The overcurrent determination value in the triode characteristic region is determined by equation (3).

Next, the operation in the five-pole characteristic region will be described. When wiring is normal, thermal FET QA
Transitions to the ON state, the thermal FET QA continuously maintains the ON state. Therefore, after the true gate-source voltages VT _GSA and VT _GSB have reached the pinch-off voltage, the thermal FETs QA, FET QB and FET Q
C also operates in the five-pole characteristic region. "HAF2" made by Hitachi
001 ", the on-resistance is the drain-source voltage V
_{When DS} = 10 V, since R _{DS (ON)} = 30 mΩ, the following equation is obtained.

[0058]

V _DSB = 5A × 30 [mΩ] = 0.15 [V] (4) V _DSA = I _DQA × 30 [mΩ] (5) V _DSA −V _DSB = 30 [mΩ] × (I _DQA -5 [A]) (6) Also, when the drain current I _DQA increases due to a short circuit of the wiring or the like, the value of the equation (6) increases, and the overcurrent increases. When the threshold value is exceeded, the thermal FET (first semiconductor element) QA is turned off. In this case, the state transits to the off state via the pinch-off point, the operation state in the three-pole characteristic region, and the above-described state. Then, the potential V ₊ of the “+” input terminal of the comparator CMP1 decreases after a certain period of time due to the hysteresis of the diode D1 and the resistor R5 shown in FIG.
The output of the comparator CMP1 changes from the "L" level to the "H" level, and the thermal FET QA is again turned on. Thus, the thermal FET QA repeats the transition to the ON state and the OFF state, and finally reaches the overheat cutoff. Note that if the wiring returns to normal before the overheat interruption (example of an intermittent short-circuit failure), the thermal FE
The TQA continuously maintains the ON state.

FIG. 8 (a) shows a switching circuit with a current oscillation type cutoff function which is a basis according to a second embodiment of the present invention.
FIG. 8B shows the device drain current _ID and the corresponding drain-source voltage V _DS . In the figure,
Is the case of a complete short circuit (dead short), is the case of normal operation, and is the case of incomplete short circuit. When a complete short circuit (dead short circuit) has occurred (in the figure), when the thermal FET (first semiconductor element) QA transitions from the off state to the on state, the drain current _ID rapidly flows. , The thermal FET QA is kept on, the thermal FET QA is overheated, and the built-in heat shutoff FET
The thermal FET QA is overheated by the transition of the QS to the ON state. Further, when an incomplete short circuit having a certain short-circuit resistance occurs (in the figure), the on / off control of the thermal FET QA is repeated as described above to greatly change the drain current _ID . Thermal F
Thermal FET by periodic heat generation of ETQA
QA is overheating more quickly.

As shown in FIG. 2, the switching device with the current oscillation type cutoff function according to the second embodiment of the present invention
And a current sensor unit for detecting by detecting a current flowing through the reference resistor Rr2. In FIG.
The fourth semiconductor element Q95 is a pnp-type BJT biased to operate in an unsaturated region. This current sensor unit constitutes a so-called “current mirror circuit”. The output of the comparator CMP401 becomes valid ("H" level) when the potentials supplied to its "+" and "-" input terminals substantially match, and invalid ("L") when they do not match. Level).
For example, when the potential of the node N1 becomes higher than the potential of the node N2, the comparator CMP401 outputs an “L” level. When the output of the comparator CMP401 becomes “L” level, the base potential of the pnp-type BJT (fourth semiconductor element) Q95 decreases, so that the pnp-type BJT (fourth semiconductor element) Q95 tends to have a larger current value. become. Then, the potential drop at both ends of the second reference resistor Rr2 increases, so that the potential of the node N2 increases. That is, until the potentials of the node N1 and the node N2 become equal, the comparator CMP4
01 outputs an “L” level. Thus, the first
The potential at the node N1 of the second main electrode of the semiconductor element QA and the potential at the node N2 of the second main electrode of the third semiconductor element QC to operate as an ideal current mirror circuit Becomes possible. Therefore,
Extremely accurate current measurement becomes possible.

(Other Embodiments) As described above, the present invention has been described with reference to the first and second embodiments.
The discussion and drawings that form part of this disclosure should not be understood as limiting the invention. From this disclosure, various alternative embodiments, examples, and operation techniques will be apparent to those skilled in the art.

For example, in the second embodiment, an ON / OFF count integrating circuit 304 as shown in FIG. 3 is connected to the nodes N51, N52 and N53 shown in FIGS. Thermal FE as semiconductor element 1
TQA shutoff can be accelerated. That is, thermal F
When the number of times of on / off control of the ET (first semiconductor element) QA reaches a predetermined number, an operation of turning off the thermal FET QA by the on / off number integrating circuit (number control means) 304 becomes possible.

As shown in FIG. 3, this on / off frequency integrating circuit 304 includes resistors R131 and R132 connected to node N51 shown in FIG. 2 and node N52 shown in FIG.
C131 connected to the node N5 in FIG.
1, a diode D132, a FET Q131,
It includes a backflow prevention diode D131 and a resistor R133.

In the overcurrent control, the thermal FET (first
Each time the gate potential of QA periodically goes to "H" level, capacitor C131 is charged via resistor R131 and backflow preventing diode D131. F
Since the gate potential of the ETQ 131 is initially lower than the threshold value, it is in the off state. However, when the gate potential rises with the charging of the capacitor C131, the FET Q 131 transitions to the on state. When the FET Q131 transitions to the ON state, FIG.
Node N5 on the anode side of the temperature sensor 121 shown in FIG.
Since 1 is lowered, the same condition as in the high temperature state is reached, and the overheat cutoff FET QS transitions to the ON state to cut off the thermal FET QA.

The overheat cutoff promotion circuit 106 shown in FIG. 4 is connected to the nodes N53 and N71 in FIG.
TQA cutoff may be accelerated. That is, in the case of an overcomplete short circuit, the on / off control of the thermal FET QA is repeatedly performed, and when the overheat cutoff is caused by the periodic heating action of the thermal FET QA, the time until the overheat cutoff is relatively large. It is thought that it becomes longer. In such a case, the overheat cutoff promotion circuit (overheat cutoff promotion means)
It is sufficient that the cutoff of the thermal FET QA is accelerated by 106.

As shown in FIG.
Are FET Q221, diode D221, resistor R22
1 to R223 and a capacitor C221. In the overcurrent control, each time the gate potential of the thermal FET QA periodically goes to "H" level, the capacitor C221 is connected to the resistor R221 and the reverse current blocking diode D
221 is charged. Although the gate potential of the FET Q221 is initially at or below the threshold value, the FET Q221 is in an off state. Terminal TG (thermal FE) located at node N62 via resistor R221
A current flows from the true gate of the TQA) to the ground potential (GND), and the amount of charge stored in the terminal TG (node N62) decreases. For this reason, the drain-source voltage V _DSA increases for the same drain current I _D , and the power consumption of the thermal FET QA increases and the overheat cutoff is accelerated. Note that the smaller the resistance R221, the earlier the overheat cutoff. Further, the resistor R223 is a discharge resistor of the capacitor C221, and is desirably set so that R222〓R223.

The inrush current mask circuit 303 shown in FIG. 5 may be connected to nodes N52, N53, N71. The inrush current mask circuit 303 is connected to the FE connected to the node N71.
TQ311, Q312, diode D311 connected to node N53, resistor R31 connected to node N52
3, a capacitor C311 and resistors R311 and R312. This inrush current mask circuit 303
, When the thermal FET QA transitions to the ON state, the gate-source voltage V _GSA is changed to the diode D311.
And a resistor R312 is supplied to the gate of FETQ312 through and also the gate - source voltage V _GSA via the diode D311 and the resistor R311 FETs Q
311 is supplied to the gate. The gate of the FET Q312 is connected to the source SA (node N52) of the thermal FET QA via the capacitor C311.
Since 11 is uncharged, the gate potential of the FET Q312 does not rise sufficiently and the FET Q312 cannot transition to the ON state. The FET Q311 is on while the FET Q312 is off, and couples the voltage dividing point supplied to the + terminal (node N71) of the comparator CMP1 to the source SA (node N52) of the thermal FET QA. Therefore, the output of the comparator CMP1 is maintained at the “H” level, and even if a large inrush current flows, the thermal F
ETQA will not transition to the off state.

With the passage of time, the capacitor C311 is charged via the resistor R312, and finally the FET C311 is charged.
Q312 transitions to the ON state. With this, FETQ
The mask state in which 311 has transitioned to the off state ends, and the overcurrent detection control functions. Note that the resistance R
313, after the thermal FET QA transitions to the off state,
This is a discharge resistor for resetting the capacitor C311. It is desirable to set R312〓R313 so as not to affect the mask time. Since the mask time is determined by the time constant of R312 × C311,
When a single chip is used, the mask time can be adjusted by arbitrarily changing the capacitance value of the external capacitor C311.

When the load 102 of the switching device with a current oscillation type cutoff function according to the second embodiment of the present invention is turned on, an inrush current several times to several tens times that of a stable state flows. The period during which the rush current flows varies depending on the type and capacity (size) of the load 102, and is about 3 msec to 20 msec. If the overcurrent control as described in the second embodiment is performed during the period when the rush current flows,
It may take time for the load 102 to reach a steady state, and the response of the load itself may be poor, such as a delay in lighting of the light. Inrush current mask circuit 303 shown in FIG.
Is added to the configuration of FIG. 2 to solve such a problem.

As described above, the present invention naturally includes various embodiments and the like not described herein. Therefore, the technical scope of the present invention is defined only by the matters specifying the invention according to the claims that are appropriate from the above description.

[0071]

As described above, according to the current mirror circuit of the present invention, the potential of the second main electrode of the first semiconductor element and the potential of the second main electrode of the second semiconductor element are different. , The ideal circuit operation can be realized.

According to the current sensor of the present invention, the first
Since the operation as an ideal current mirror circuit can be realized by making the potential of the second main electrode of the semiconductor device equal to the potential of the second main electrode of the second semiconductor device, extremely accurate current measurement Becomes possible.

Further, according to the switching circuit of the present invention, the conventional shunt resistor is not required, and not only an overcurrent due to a complete short circuit, but also an abnormal state when a rare short circuit such as an incomplete short circuit having a certain short circuit resistance occurs. The current can be detected easily and accurately.

Further, according to the switching device of the present invention, the heat loss of the device is suppressed by eliminating the need for a conventional shunt resistor, and not only an overcurrent due to a complete short circuit but also an incomplete short circuit having a certain degree of short resistance. An abnormal current when a rare short circuit occurs can be detected easily and accurately. In particular, when the control circuit section of the semiconductor switch serving as the main device is monolithically integrated on the same semiconductor substrate, a microcomputer is not required, so that the chip area can be reduced and the device cost can be significantly reduced.

[Brief description of the drawings]

FIG. 1 is a circuit configuration diagram of a current sensor according to a first embodiment of the present invention.

FIG. 2 is a circuit configuration diagram of a switching device with a current oscillation type cutoff function according to a second embodiment of the present invention.

FIG. 3 is a circuit configuration diagram of an ON / OFF count integrating circuit used in a switching device having a current oscillation type cutoff function according to another embodiment of the present invention.

FIG. 4 is a circuit configuration diagram of an overheat cutoff promotion circuit used in a switching device having a current oscillation type cutoff function according to still another embodiment of the present invention.

FIG. 5 is a circuit configuration diagram of an inrush current mask circuit used in a switching device having a current oscillation type cutoff function according to still another embodiment of the present invention.

FIG. 6 is an explanatory diagram for explaining a principle used by a switching device with a current oscillation type cutoff function according to a second embodiment of the present invention, and shows a state between a drain and a source at the time of transition from an off state to an on state. FIG. 4 is an explanatory diagram of a voltage falling characteristic.

FIG. 7 shows a main device (first device) of a switching device with a current oscillation type cutoff function according to a second embodiment of the present invention.
2 is a conceptual equivalent circuit diagram focusing on (a semiconductor element of FIG. 1).

FIG. 8A shows a transient response characteristic of a drain current of a main device (first semiconductor element) in a switching device having a current oscillation type cutoff function according to a second embodiment of the present invention. 8 (b) shows the corresponding drain-
FIG. 4 is an explanatory diagram showing a transient response characteristic of a source-to-source voltage.

FIG. 9 is a circuit configuration diagram of a conventional semiconductor switch.

FIG. 10 is a circuit configuration diagram of a thermal FET.

FIG. 11 is a circuit configuration diagram of a conventional current mirror type power supply control device.

[Explanation of symbols]

Reference Signs List 101 power supply 102 load 106 overheat cutoff promotion circuit (overheat cutoff promotion means) 110 semiconductor chip 111 drive circuit (control means) 301 overcurrent detection section 302 current enable section 303 inrush current mask circuit (prohibition means) 304 on / off number integration circuit (Number control means 9 305 Charge pump section 306 Cut-off latch circuit C131, C221, C311 Capacitor CMP1, CMP401 Comparator D1, D131, D132, D221, D311 Diode QA, QF Thermal FET (first semiconductor element) QB FET (first semiconductor element) 2 semiconductor switch) QC FET (third semiconductor switch) Q93 first semiconductor element Q94 second semiconductor element Q95 pnp type BJT Q131, Q221, Q311, Q312 MOSFE
T RG Internal resistance R1, R2, R5, R131-R133, R221-R
223, R311 to R313, R331, R412, R
413 Resistance Rr1 First reference resistance Rr2 Reference resistance (second reference resistance) T1, T2, T3, T11 to T18 Input / output terminal ZD1 Zener diode

 ──────────────────────────────────────────────────続 き Continued on the front page F term (reference) 5H420 BB12 CC02 DD02 EA14 EA39 EB37 FF04 FF14 FF21 LL05 LL07 5J055 AX02 AX11 AX12 AX37 AX44 AX47 BX16 CX28 DX08 DX09 DX13 DX14 DX22 DX53 DX54 DX73 DX83 EY10 EX11 EY11 EY17 EY21 EZ04 EZ10 EZ31 EZ55 EZ57 EZ61 FX04 FX06 FX32 GX01 GX06

Claims (5)

[Claims] A first semiconductor element having first and second main electrodes and a control electrode; and a first main element connected to the first main electrode and the control electrode of the first semiconductor element, respectively. A second semiconductor element having an electrode, a control electrode, and a second main electrode; connecting a first input terminal to a second main electrode of the first semiconductor element; Second main electrode
A third input terminal connected to a first main electrode, a second output terminal of the comparator connected to a control electrode, and a second main electrode connected to a reference resistor. A current mirror circuit comprising a semiconductor element. 2. A first semiconductor element having first and second main electrodes and a control electrode, and a first main element connected to the first main electrode and the control electrode of the first semiconductor element, respectively. A second semiconductor element having an electrode, a control electrode, and a second main electrode; a driving circuit for supplying a voltage to each control electrode of the first and second semiconductor elements; and the first semiconductor element The first input terminal is connected to the second main electrode of the second semiconductor device, and the second main electrode of the second semiconductor element is connected to the second input terminal.
A third input terminal connected to a first main electrode, a second output terminal of the comparator connected to a control electrode, and a second main electrode connected to a reference resistor. A current sensor comprising a semiconductor element, wherein a current flowing in a load connected to a second main electrode of the first semiconductor element is detected by detecting a current flowing through the reference resistor. 3. A first semiconductor element having first and second main electrodes and a control electrode, and a first main element connected to the first main electrode and the control electrode of the first semiconductor element, respectively. A second semiconductor element having an electrode, a control electrode, and a second main electrode; a first main electrode, a control electrode connected to the first main electrode and the control electrode of the first semiconductor element, respectively. A first input terminal connected to a second main electrode of the first semiconductor element, and a second main electrode of the third semiconductor element. Second
A fourth input terminal having a first main electrode connected to a second input terminal of the comparator, a control electrode connected to an output terminal of the comparator, and a reference resistor connected to a second main electrode. Comparing means for comparing a voltage between main electrodes of the semiconductor element and the first and second semiconductor elements; and a control electrode of each of the first to third semiconductor elements according to an output of the comparing means. A control voltage supply means for supplying a control voltage, detecting an abnormal current flowing through the first semiconductor element, and controlling the first semiconductor element on / off to generate a current oscillation when an abnormal current occurs The current oscillation interrupts a conduction state of the first semiconductor element, and detects a current value flowing through the first semiconductor element by detecting a current flowing through the reference resistor. Switch Grayed circuit. 4. A first semiconductor element having a first main electrode connected to an input terminal, a second main electrode connected to an output terminal, and a control electrode, and a first semiconductor element of the first semiconductor element. A second semiconductor element having a first main electrode, a control electrode, and a second main electrode connected to the main electrode and the control electrode, respectively; a first main electrode of the first semiconductor element; A third semiconductor element having a first main electrode, a control electrode, and a second main electrode respectively connected to the electrodes; and a first input terminal connected to a second main electrode of the first semiconductor element. Connected to the second main electrode of the third semiconductor element.
A fourth input terminal having a first main electrode connected to a second input terminal of the comparator, a control electrode connected to an output terminal of the comparator, and a reference resistor connected to a second main electrode. Comparing means for comparing the voltage between the main electrodes of the semiconductor element and the first and second semiconductor elements; and controlling electrodes of the first to third semiconductor elements according to the output of the comparing means. A control voltage supply means for supplying a control voltage; detecting an abnormal current flowing through a load connected to the output terminal; Is generated, and the current oscillation causes the input terminal
A switching device for interrupting a conduction state between output terminals and detecting a value of a current flowing through the load by detecting a current flowing through the reference resistor. 5. The switching device according to claim 4, wherein said first to third semiconductor elements, said comparing means and control voltage supply means are integrated on the same semiconductor substrate.