JP2000152618A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To miniaturize a switching power supply circuit, reduce its weight, and reduce costs by compositively controlling the conduction angle between the resonance impedance and the switching element of a primary side parallel resonance circuit, and performing the constant-voltage control of a secondary side output voltage. SOLUTION: Since parts with low breakdown voltage can be used for a switching element Q1, a parallel resonance capacitor Cr, and a secondary side rectification diode, elements can be inexpensive. Therefore, those with improved characteristics (those with improved characteristics of saturation voltage, accumulation time, descent time, current amplification rate, and the like for the switching element Q1 and those with improved characteristics of forward voltage drop, inverse recovery time, and the like for the rectification diode), for example, for the switching element Q1 and the bridge rectification circuit can be selected. By this sort of improvement in characteristics, a switching frequency can be set highly, power loss can be reduced, and each kind of parts can be miniaturized and the weight can be reduced.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 10 is a circuit diagram showing an example of a switching power supply circuit that can be constructed based on the invention proposed by the present applicant. In the switching power supply circuit shown in this figure, for example, a commercial AC power supply in Japan or the U.S.
This corresponds to a condition of 0 W or more.

In the switching power supply circuit shown in FIG. 1, a so-called voltage doubler rectification circuit comprising rectification diodes Di1, Di2 and smoothing capacitors Ci1, Ci2 is provided as a rectification / smoothing circuit for rectifying / smoothing the commercial AC power supply AC. Be provided. In this voltage doubling rectifier circuit, for example, if a DC input voltage corresponding to one time the peak value of the AC input voltage VAC is Ei, the DC input voltage
Generate Ei. For example, if the AC input voltage VAC is 144V, the DC input voltage 2Ei is about 400V. As described above, the reason why the voltage doubler rectifier circuit is employed as the rectifier / smoothing circuit is that the AC input voltage is AC1
The reason is that it is compatible with a relatively heavy load condition of a 00V system and a maximum load power of 150 W or more. That is, by increasing the DC input voltage to twice the normal value, the amount of current flowing into the subsequent switching converter is suppressed, and the reliability of the components forming the switching power supply circuit is ensured. . In the voltage doubler rectifier circuit shown in this figure, an inrush current limiting resistor Ri is inserted in the rectified current path, for example, so as to suppress the inrush current flowing into the smoothing capacitor when the power is turned on. I have.

The voltage-resonant type switching converter shown in FIG. 1 has a self-excited type provided with a single switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT;
Junction type transistor). The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci1 (rectified smoothed voltage 2Ei) via a starting resistor RS so that a base current at the time of starting can be obtained from the rectified smoothing line. An inductor LB and a detection drive winding N are provided between the base of the switching element Q1 and the temporary ground.
A self-excited oscillation driving resonance circuit comprising a series connection circuit of B, a resonance capacitor CB and a base current limiting resistor RB is connected. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a damper current flowing when the switching element Q1 is off. , The collector of switching element Q1 is connected to one end of primary winding N1 of insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Is a voltage resonance type converter based on its own capacitance and a combined inductance (L1 + Lc) obtained by series connection of a leakage inductance L1 on the primary winding N1 side of an orthogonal insulation converter transformer PRT described later and an inductor Lc of a choke coil PCC. To form a primary parallel resonance circuit. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vcr across the resonance capacitor Cr actually becomes a sinusoidal pulse waveform due to the action of the parallel resonance circuit, and the voltage resonance type operation is performed. Is obtained.

[0007] The choke coil PCC has an inductor Lc.
And the detection drive winding NB are transformer-coupled. In the detection drive winding NB, an alternating voltage corresponding to the switching period is excited by the switching output transmitted from the primary winding N1 of the orthogonal type insulated converter transformer PRT to the inductor Lc.

The orthogonal type insulation converter transformer PRT is
It has a function of transmitting the switching output of the switching element Q1 to the secondary side and performing constant voltage control of the secondary side output. For example, as shown in FIG. 11, the orthogonal type insulating converter transformer PRT is formed by joining the ends of two magnetic legs of two double U-shaped cores 201 and 202 having four magnetic legs. A three-dimensional core 200 is provided. Then, the primary winding N1 and the secondary winding N2 are wound around predetermined two magnetic legs of the three-dimensional core 200 in the same winding direction, and the control winding NC is connected to the primary winding N1. , And wound around two predetermined magnetic legs so that the winding direction is orthogonal to the secondary winding N <b> 2, thereby constituting a saturable reactor. In this case, the opposing surfaces of the magnetic legs of the double U-shaped cores 201 and 202 are joined, and no gap is formed.

The above-mentioned orthogonal type insulated converter transformer PRT
One end of the primary winding N1 is connected to the collector of the switching element Q1, and the other end is connected to the choke coil P1 as shown in the figure.
It is connected to the positive electrode (rectified and smoothed voltage 2Ei) of the smoothing capacitor Ci via the series connection of the inductor Lc of CC.

On the secondary side of the orthogonal type insulated converter transformer PRT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. The alternating voltage excited by the secondary winding N2 by this parallel resonance circuit becomes a resonance voltage. That is, a voltage resonance operation is obtained on the secondary side. For the secondary parallel resonance circuit formed as described above, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, DO3, DO4 and the smoothing capacitors CO1, CO2 are provided. As shown in the figure, a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and [rectifier diodes DO3, DO4, smoothing capacitor CO1]
2], two sets of full-wave rectifier circuits are provided.
A full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] receives a resonance voltage supplied from the secondary parallel resonance circuit to generate a DC output voltage EO1, and generates a [rectifier diode DO3, DO4, smoother. Similarly, the full-wave rectifier circuit composed of the capacitor CO2] receives the resonance voltage supplied from the secondary parallel resonance circuit and generates the DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies a DC current, the level of which is varied according to the level of the DC voltage output EO1 on the secondary side, to the control winding NC of the quadrature-type insulating converter transformer PRT, for example. The constant voltage control is performed as described.

For example, when the DC output voltage EO1 on the secondary side fluctuates with the fluctuation of the AC input voltage VAC or the load power, the control circuit 1 changes the control current flowing through the control winding NC within a required range. . Since the control winding NC is wound around the orthogonal type insulated converter transformer PRT, the leakage inductance (L1, L2) in the orthogonal type insulated converter transformer PRT which is a saturable reactor.
Is obtained. As described above, the primary winding N
The leakage inductance L1 of 1 forms a parallel resonance circuit on the primary side and the leakage inductance L2 of the secondary winding N2 forms a parallel resonance circuit on the secondary side. , The inductances L1 and L2 are variably controlled. This operation changes the resonance impedance of the primary side and the secondary side, so that the switching output transmitted from the primary side to the secondary side also changes. As a result, the output of the secondary side DC voltage (EO1, EO2) changes. A constant voltage is achieved. Note that such a constant voltage control method is hereinafter referred to as a “parallel resonance frequency control method”.

FIG. 12 is a circuit diagram showing another example of a switching power supply circuit that can be constructed based on the invention proposed by the present applicant. Similarly to the power supply circuit shown in FIG. 10, the power supply circuit shown in this figure is a system in which a commercial AC power supply in Japan or the United States is a so-called AC 100 V system and corresponds to a condition that the maximum load power is 150 W or more. On the primary side, one switching element Q
A self-excited voltage resonant converter according to 1 is provided.
In FIG. 12, the same parts as those in FIG. 10 are denoted by the same reference numerals, and the description of the parts having the same configuration will be omitted.

The switching power supply circuit shown in FIG. 1 includes an orthogonal control transformer PRT. As shown in FIG. 13, this orthogonal control transformer PRT has two double U-shaped cores 201, 20 having four magnetic legs.
There is provided a three-dimensional core 200 formed so as to join the ends of the two magnetic legs. Then, the controlled winding NR is wound around the predetermined two magnetic legs of the three-dimensional core 200 by a predetermined number of turns, and furthermore, a predetermined winding is performed so as to be orthogonal to the winding direction of the controlled winding NR. The control winding NC is wound around the two magnetic legs to form a saturable reactor. This orthogonal control transformer PRT can be viewed as one variable inductance element.
The size is smaller than the orthogonal insulation converter transformer PRT shown in FIG.

In this case, the controlled winding NR is connected between the positive terminal of the smoothing capacitor Ci1 and the insulating converter transformer PIT.
Is inserted in series with the primary winding N1. Therefore, in the power supply circuit shown in FIG.
The primary-side switching operation is performed by the voltage resonance by the combined inductance (L1 + LR) obtained by connecting the leakage inductance L1 on the primary winding N1 side of the T and the inductance LR of the controlled winding NR in series, and the capacitance of the parallel resonance capacitor Cr. Form a parallel resonant circuit to shape.

The insulation converter transformer PI shown in FIG.
T is an EE-type core 1 formed by two E-type cores 101 and 102 made of a ferrite material, for example, as shown in FIG.
00 is formed. At this time, no gap is formed in the center magnetic leg as shown in the figure. Then, the primary winding N1 (and the detection driving winding NB) and the secondary winding N2 are actually wound around the center magnetic leg in a divided state using a divided bobbin. . Thereby, the primary winding N1
And the secondary winding N2 are loosely coupled (for example, a coupling coefficient k ≒ 0.
The state of 9) is obtained.

In the isolated converter transformer PIT, the polarities (winding direction) of the primary winding N1 and the secondary winding N2 are set.
And the connection of the rectifier diodes DO (DO1, DO2), the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 becomes + M (additive polarity mode). M (depolarized mode) in some cases. For example, FIG.
15B, the mutual inductance becomes + M, and in the operation when the connection form shown in FIG. 15B is adopted, the mutual inductance becomes -M. In the circuit shown in this figure, the polarities of the primary winding N1 and the secondary winding N2 are in an additive polarity mode.

In this case, the insulation converter transformer P
A parallel resonance circuit is formed by connecting a secondary parallel resonance capacitor C2 in parallel to the IT secondary winding N2. The alternating voltage excited by the secondary winding N2 is converted into a resonance voltage by the parallel resonance circuit. The resonance voltage is converted into a half-wave rectification circuit including a rectifier diode DO1 and a smoothing capacitor CO2, a rectifier diode DO2 and a smoothing capacitor CO2.
And two sets of half-wave rectifier circuits. Then, the DC output voltages EO1 and EO2 are obtained by these two sets of half-wave rectifier circuits, respectively. The rectifier diodes DO1 and DO2 forming the half-wave rectifier circuit use a high-speed rectifier to rectify the alternating voltage in the switching cycle.

For example, when the DC output voltage EO2 on the secondary side fluctuates with the fluctuation of the AC input voltage VAC or the load power, the control circuit 1 controls the control current flowing through the control winding NC to a range of, for example, 10 mA to 40 mA. To change. Thus, the inductance LR of the controlled winding NR is changed in a range of, for example, 0.1 mH to 0.6 mH.

Since the controlled winding NR forms a parallel resonance circuit for obtaining a voltage-resonant switching operation as described above, this parallel resonance circuit is not controlled for a fixed switching frequency. The resonance condition of the circuit is changed. At both ends of the parallel connection circuit of the switching element Q1 and the parallel resonance capacitor Cr, a sinusoidal resonance pulse is generated by the action of the parallel resonance circuit corresponding to the OFF period of the switching element Q1, but the resonance condition of the parallel resonance circuit , The width of the resonance pulse is variably controlled. That is, PWM (P
(ulse Width Moduration) control operation is obtained. The PWM control of the resonance pulse width is control of the off-period of the switching element Q1, which means that the on-period of the switching element Q1 is variably controlled under a condition of a fixed switching frequency. By variably controlling the ON period of the switching element Q1 in this manner,
The switching output transmitted from the primary winding N1 forming the parallel resonance circuit to the secondary side changes, and the output level of the DC output voltage (EO1, EO2) on the secondary side also changes. As a result, the secondary DC voltage (EO1, EO2) can be made constant. In addition, such a constant voltage control method is herein referred to as a “primary-side voltage resonance pulse width control method”.

FIG. 16 is a circuit diagram showing another example of a switching power supply circuit which can be constructed based on the invention proposed by the present applicant. In FIG. 16,
The same parts as those in FIGS. 10 and 12 are denoted by the same reference numerals, and the description of the parts having the same configuration will be omitted. In the power supply circuit shown in this figure, an example is shown in which the controlled winding of the orthogonal control transformer PRT is provided on the secondary side. In this case, two controlled windings NR and NR1 are wound and provided as controlled windings of the orthogonal control transformer PRT, and the controlled winding NR is connected to an end of the secondary winding N2. The rectifier diode DO1 is connected so as to be inserted in series between the anodes. Also, the controlled winding NR1
Is inserted in series between the tap output of the secondary winding N2 and the anode of the rectifier diode DO2. In such a connection form, the secondary parallel resonance circuit
It is formed to include the inductance components (LR, LR1) of R and NR1.

As described above, in the case where the controlled windings (NR, NR1) of the orthogonal control transformer PRT are provided on the secondary side, the inductance of the controlled windings NR, NR1 is determined as an inductance control method. Is varied, so that the pulse width of the resonance voltage V2 of the secondary parallel resonance capacitor C2, that is, the conduction angle of the secondary rectifier diode is variably controlled. As a result, constant voltage control is achieved by controlling the output level obtained on the secondary side.

The insulating converter transformer PIT provided in the power supply circuit shown in this figure has the same structure as that shown in FIG. Also, an orthogonal control transformer PRT
13 has the same structure as that of FIG. 13 except that the controlled winding NR1 is additionally wound in a winding direction orthogonal to the control winding NC. However, in the case of the orthogonal control transformer PRT of the power supply circuit shown in FIG.
In order to insert R into the primary side, an insulation distance between the secondary side and the control winding NC which is DC-connected must be ensured, and a corresponding size is required. In the orthogonal control transformer PRT of the power supply circuit, the controlled winding NR1
Is provided on the secondary side, so that an insulating distance from the control winding NC is not required, so that the size of the orthogonal control transformer PRT can be made smaller.

[0024]

The above-mentioned FIG.
In the switching power supply circuit having the configuration described with reference to FIG. 16, since the AC input voltage VAC corresponds to the AC 100 V system and the maximum load power is 150 W or more, the DC input voltage of 2Ei level is I'm trying to get. Therefore, actually, a resonance voltage Vcr of 1800 V is generated between both ends of the switching element Q1 and the parallel resonance capacitor Cr when the switching Q1 is off. Therefore, for the switching element Q1 and the parallel resonance capacitor Cr, it is required to select a high withstand voltage product of 1800V. Therefore, the switching element Q
1 and the parallel resonance capacitor Cr are correspondingly large. In particular, when a high withstand voltage product is selected for the switching element Q1, the switching frequency should be set high because the saturation voltage VCE (SAT), the accumulation time tSTG, the fall time tf are large, and the current amplification factor hFE is small. Becomes difficult. When the switching frequency is low, the switching loss and the drive power increase, so that the power loss of the power supply circuit increases accordingly. Further, a transformer provided in a power supply circuit and a capacitor provided in a drive circuit system and the like are large and expensive, which hinders reduction in size and weight of the circuit and cost reduction.

In any of the constant voltage control systems shown in FIGS. 10, 12 and 16, the (orthogonal type) insulated converter transformer PIT (P
RT) has the required degree of coupling obtained by not forming a gap, and then, in series with the primary winding N1 or the secondary winding N2, a winding (inductor) of a choke coil Lc or the controlled winding NR of the orthogonal control transformer PRT is connected. Therefore, the leakage inductance component in the power supply circuit increases. This increase in the leakage inductance component leads to an increase in the leakage magnetic flux, which may affect an electronic circuit or the like on the load side. Therefore, in practice, in order to reduce the influence of the leakage magnetic flux, for example, the entire switching converter circuit is housed in an aluminum shield case with ventilation holes formed, and the input and output are connected by providing connectors. I have. Also in such a structure, reduction in size and weight and cost of the circuit have been hindered, and the manufacturing time has been correspondingly long.

[0026]

SUMMARY OF THE INVENTION Accordingly, the present invention has been made in consideration of the above-described problems, and has been made to promote reduction in size and weight and cost of a resonance type power supply circuit, as well as improvement in manufacturing efficiency and power conversion efficiency. It is an object of the present invention to improve various characteristics as a resonance type power supply circuit.

For this reason, the rectifying / smoothing means which receives a commercial AC power supply, generates a rectified smoothed voltage having a level corresponding to the same level as the commercial AC power supply level, and outputs the same as a DC input voltage is loosely coupled. An insulating converter transformer provided with a gap so as to obtain a required coupling coefficient and transmitting the primary output to the secondary side, and a primary winding of the insulating converter transformer intermittently connecting a DC input voltage by a switching element. And a primary side parallel resonance circuit formed by a leakage inductance component including at least the primary winding of the insulated converter transformer and the capacitance of the parallel resonance capacitor to make the operation of the switching means a voltage resonance type. And a secondary side series resonance capacitor in series with the secondary winding of the isolated converter transformer. A secondary side series resonance circuit that forms a series resonance circuit by connecting the leakage inductance component of the secondary winding of the insulated converter transformer and the capacitance of the secondary side series resonance capacitor, and a secondary side with respect to the rectified current path A secondary side that is formed by inserting a series resonance capacitor and performs alternating voltage full-wave rectification by inputting an alternating voltage obtained to the secondary winding of an insulating converter transformer, and corresponding to almost twice the input voltage level DC output voltage generating means configured to generate a DC output voltage, and by changing the switching frequency of the switching element according to the level of the secondary DC output voltage, the resonance impedance of the primary side parallel resonance circuit and the switching element Constant voltage control means configured to perform a constant voltage control on the secondary side output voltage by controlling the conduction angle of the It was possible to configure the switching power supply circuit comprises and.

According to the above configuration, the voltage-resonant converter is provided for the primary side, the isolation converter transformer is loosely coupled, and the secondary-side series resonance circuit and the voltage doubler full-wave rectifier circuit are provided on the secondary side. As a result, a secondary-side DC output voltage is generated to supply power to the load. In other words, a required load condition is basically met by providing a voltage doubler full-wave rectifier circuit on the secondary side. Accordingly, the primary side is not provided with a voltage doubler rectifier circuit but with a full-wave rectifier circuit that generates a rectified smoothed voltage corresponding to one time of the AC input voltage level. Further, in the configuration of the constant voltage control according to the above configuration, by changing the switching frequency according to the secondary side output voltage level, the resonance impedance of the primary side parallel resonance circuit and the conduction angle of the switching element are simultaneously controlled. As a result, the control sensitivity is improved by the composite control operation.

[0029]

FIG. 1 is a circuit diagram showing a configuration example of a switching power supply circuit according to a first embodiment of the present invention. In the power supply circuit shown in this figure, as in the case of FIGS. 10, 12 and 16 described above, a self-excited voltage resonance type switching by one switching element (bipolar transistor) is applied to the primary side. A converter is provided. In this figure, the same parts as those in FIGS. 10, 12, and 16 are denoted by the same reference numerals, and description thereof will be omitted.

In the power supply circuit according to the present embodiment shown in this figure, as a rectifying / smoothing circuit for receiving an AC input voltage VAC and obtaining a DC input voltage, a full-wave circuit comprising a bridge rectifier circuit Di and a smoothing capacitor Ci. A rectifier circuit is provided to generate a rectified smoothed voltage Ei corresponding to a level that is one-time the AC input voltage VAC. That is, in the present embodiment, a voltage doubler rectifier circuit is not provided as in the related art. In this specification, a rectifier circuit that generates a rectified smoothed voltage Ei corresponding to one time of the level of the AC input voltage VAC as shown in FIG. 1 is also referred to as an “equal voltage rectifier circuit”. .

The orthogonal control transformer PRT shown in FIG.
Is a saturable reactor on which a detection winding ND, a drive winding NB, and a control winding NC are wound. The structure of the orthogonal control transformer PRT is, as shown in FIG.
Two U-shaped cores 201, 2 having two magnetic legs
The two-dimensional core 200 is formed so that the ends of the magnetic legs 02 are joined to each other. Then, the detection winding N is applied to the predetermined two magnetic legs of the three-dimensional core 200 in the same winding direction.
D, the drive winding NB is wound, and the control winding NC is wound in a direction orthogonal to the detection winding ND and the drive winding NB. Here, as the number of turns (number of turns) of the detection winding ND, the drive winding NB, and the control winding NC, for example, the detection winding ND = 1T (turn), the drive winding NB = 3T, and the control winding NC = It is 1000T. The orthogonal control transformer PRT of the present embodiment is reduced in size and weight, for example, to about 7 gr.

In this case, the detection winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching of the switching element Q1 is performed. The output is transmitted to the detection winding ND via the primary winding N1.
In the orthogonal control transformer PRT, the drive winding NB is excited by the switching output obtained from the detection winding ND, and an alternating voltage is generated in the drive winding NB. This alternating voltage becomes a source of a driving voltage in the self-excited oscillation driving circuit.

The control circuit 1 varies the level of the control current (DC current) supplied to the control winding NC in accordance with the change in the secondary-side DC output voltage level, thereby winding the control coil PRT on the orthogonal control transformer PRT. The inductance LB of the mounted drive winding NB is variably controlled. Thereby, the driving winding NB
The resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB changes. This is an operation of changing the switching frequency of the switching element Q1, as will be described later with reference to FIG. 4. This operation has an effect of stabilizing the secondary-side DC output voltage.

As shown in FIG. 3, the insulating converter transformer PIT according to the present embodiment is provided with an EE-type core 100 in which E-type cores 101 and 102 made of, for example, a ferrite material are combined so that their magnetic legs face each other. With respect to the center magnetic leg of the EE type core 100, actually, a primary winding N1 and a secondary winding N2 (using a divided bobbin in which a winding portion is divided into a primary side and a secondary side). And N2A) are wound separately. In the present embodiment, a gap G is formed for the center magnetic leg as shown in the figure. This gap G is the E-shaped core 10
The central magnetic legs 1, 102 can be formed by forming them shorter than the two outer magnetic legs. As a result, for example, a loose coupling is achieved by a coupling coefficient smaller than that of the insulating converter transformer PIT shown in FIG. 14 as a conventional example, so that a saturated state is hardly obtained. As the coupling coefficient k in this case, for example, k ≒ 0.8
5 is assumed.

Insulated converter transformer P of the present embodiment
The secondary winding N2 of the IT is wound with a different number of windings from the conventional one as described later. One end of the secondary winding N2 is connected to the secondary side ground, and the other end is a series resonance capacitor C.
s is connected to the connection point of the anode of the rectifier diode DO1 and the cathode of the rectifier diode DO2 via the series connection of s. The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO1, and the anode of the rectifier diode DO2 is connected to the secondary side ground. Smoothing capacitor C
The negative side of O1 is connected to the secondary side ground. Also,
In this case, the secondary winding N2A is independent of the secondary winding N2.
Is wound. For the secondary winding N2A, the center tap is grounded to ground, and the rectifier diode DO is
The DC output voltage EO2 is generated by connecting a full-wave rectifier circuit composed of 3, DO4 and a smoothing capacitor CO2.

As a result, in such a connection form, a voltage doubler full-wave rectifier circuit composed of a set of [series resonance capacitor Cs, rectifier diodes DO1, DO2, smoothing capacitor CO1] is formed. Here, the series resonance capacitor Cs forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO1 and DO2 by its own capacitance and the leakage inductance component (L2) of the secondary winding N2. That is, the power supply circuit of the present embodiment is provided with a parallel resonance circuit for performing a voltage resonance type switching operation on the primary side and performing a voltage doubler full-wave rectification operation (current resonance operation) on the secondary side. A series resonant circuit is provided for obtaining.
In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

FIG. 4 is a waveform diagram showing the operation of the main part of the power supply circuit having the configuration shown in FIG. 1. FIGS. 4 (a) to 4 (g) show the maximum load power (Pomax) and the minimum guaranteed AC input voltage. (V
FIG. 4 shows operation waveforms of various parts at the time of AC min).
(H) to (n) respectively show the minimum load power (Pomi)
n) and the operation waveforms of the same parts as in FIGS. 4A to 4G at the time of the maximum guaranteed AC input voltage (VAC max).

In this case, a base current (drive current) IB flows through the base of the switching element Q1 as shown in FIGS. 4D and 4K by the self-excited oscillation driving circuit for the switching element Q1. By this driving current IB,
The switching element Q1 performs a switching operation. At this time, the collector current Icp flowing to the collector of the switching element Q1 has the waveforms shown in FIGS. As shown in FIGS. 4A and 4H, a parallel resonance voltage Vcr is generated at both ends of the parallel connection circuit of the switching element Q1 // parallel resonance capacitor Cr as shown in FIGS. This parallel resonance voltage Vcr
As shown in the figure, a period TON in which the switching element Q1 is on is 0 level, and a waveform of a sine wave pulse is obtained in the period TOFF in which the switching element Q1 is off, which corresponds to the operation as a voltage resonance type.

By the switching operation on the primary side as described above, a switching output is obtained on the primary winding N1.
This operation is shown as the switching output current I1 obtained in the primary winding N1 in FIGS. 4B and 4I, and a smooth waveform close to a sine wave is obtained by the voltage resonance type operation.

The switching output obtained on the primary side as described above is excited by the secondary winding N2. Then, on the secondary side, the voltage doubler full-wave rectification operation is obtained as follows by the above-mentioned set of [series resonance capacitor Cs, rectifier diodes DO1, DO2, and smoothing capacitor CO1].

Here, during a period T1 in which the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the rectifier diode DO1 operates in the reduced polarity mode in which the polarity of the primary winding N1 and the secondary winding N2 is -M. The rectified current I3 (FIG. 4 (f) (m)) rectified by the rectifier diode DO2 is charged to the series resonance capacitor Cs by the series resonance effect of the leakage inductance L2 of the secondary winding N2 and the series resonance capacitor Cs. Operation is obtained. FIG. 4 (g) (n)
5 shows a voltage V2 across the rectifier diode DO2.

Then, during the period T2 in which the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on and the rectification operation is performed, the polarity mode of the primary winding N1 and the secondary winding N2 is + M. The smoothing capacitor CO1 is charged in a state where series resonance (current resonance) occurs in which the potential of the series resonance capacitor Cs is added to the voltage induced in the secondary winding N2. At this time, the rectified current I2 charged in the smoothing capacitor CO1 via the rectifier diode DO1 has a waveform shown in FIGS. As can be seen by comparing this waveform with the rectified current I3 rectified by the rectifier diode DO2 shown in FIGS. 4F and 4M, the level of the rectified current I2 is higher than that of the rectified current I3. This is due to the rectification operation in which the potential of the series resonance capacitor Cs is applied as described above.

As described above, the polarity mode (+ M;
By performing the rectification operation using both the forward operation) and the depolarization mode (-M; flyback operation), the secondary winding N2 in the smoothing capacitor CO1 is performed.
, A DC output voltage EO1 corresponding to almost twice the induced voltage is obtained. That is, in the present embodiment, the secondary side DC output voltage is obtained by performing the double voltage full-wave rectification by using the state in which the mutual inductance is in the operation mode of + M and -M, and the load is accordingly increased. The power supplied to the side also increases, thereby increasing the maximum load power.

As described above with reference to FIG. 3, the configuration for obtaining the above-mentioned voltage doubler full-wave rectification operation is to form a gap G with respect to the insulating converter transformer PIT to achieve loose coupling by a required coupling coefficient. This is achieved by obtaining a state that is more unlikely to be saturated.
For example, when the gap G is not provided for the insulating converter transformer PIT as in the related art, there is a high possibility that the insulating converter transformer PIT becomes saturated during flyback operation and the operation becomes abnormal. It is difficult to desirably perform the voltage doubling rectification operation as described in the first embodiment.

Further, as described above, by providing the voltage doubler full-wave rectifier circuit for performing the rectification operation in both the positive and negative periods of the excitation voltage of the secondary winding N2, in the case of the circuit form of this embodiment, As can be seen from the waveforms shown in FIGS. 4 (g) and 4 (n), the voltage applied to the rectifier diodes DO1, DO2 forming the secondary-side voltage doubler full-wave rectifier circuit is the secondary-side output voltage (EO1) when turned off. ) Clamped to the level. Therefore, assuming that the actual secondary-side output voltage EO1 is 135 V, a 150 V withstand voltage product can be used for the rectifier diodes DO1 and DO2.

Further, since the secondary side DC output voltage is obtained by the voltage doubler rectifier circuit, for example, the same level as the secondary side DC output voltage obtained by the equal voltage rectifier circuit (half-wave rectifier circuit) is obtained. In order to obtain, the number of turns of the secondary winding N2 of the present embodiment is １ ／ of that in the related art. This reduction in the number of turns leads to a reduction in the size and weight of the insulating converter transformer PIT and a reduction in cost.

Further, in the power supply circuit of this embodiment, as can be understood from the above description, the primary side is provided with the voltage resonance type converter (parallel resonance circuit), and the secondary side is provided with the series resonance circuit and the voltage doubler. It is a composite resonant switching converter having a full-wave rectifier circuit. Here, FIG. 5 shows the relationship between the switching frequency fs and the secondary side DC output voltage EO (E01, E02) in the present embodiment. In this case, the horizontal axis indicates the switching frequency, and the vertical axis indicates the level of the secondary side DC output voltage EO. A resonance curve indicated by a broken line indicates characteristics in the power supply circuit illustrated in FIG. As can be seen from this figure, for example, in order to make the secondary side DC output voltage EO = 135 V in accordance with the load fluctuation, the switching frequency fs is set to 7
It is necessary to control in a range of Δ150 KHz from 5 KHz to 225 KHz. However, as described above, the switching frequency of the switching element Q1 is limited to about 50 KHz because of the withstand voltage in the power supply circuit shown in FIG. On the other hand, in the circuit shown in FIG. 1, the switching frequency f
s may be controlled in a range of 100 KHz to 175 KHz in the range of 75 KHz, and the control range is suppressed to about 1/2. Further, as described later, in the present embodiment, a higher switching frequency can be easily realized.

The reason why the control range of the switching frequency is reduced in the present embodiment as described above is as follows. In the present embodiment, the control circuit 1
As described above, the switching voltage of the switching element Q1 is variably controlled by the operation of the constant voltage control circuit system including the quadrature control transformer. This operation is also shown in FIG. 4, for example, FIGS. 4 (a), 4 (c), 4 (d) and 4 (h).
(J) As can be seen by comparing the waveforms (Vcr, Icp, IB) of (k), in changing the switching frequency, in this circuit, the period TOFF during which the switching element Q1 is turned off is fixed. In addition, the ON period TON is variably controlled. That is, in the present embodiment, as the constant voltage control operation, the resonance frequency control for the switching output is performed by operating to variably control the switching frequency. At the same time, the conduction angle control of the switching element in the switching cycle ( PWM control). This complex control operation is realized by a set of control circuit systems. Actually, FIGS. 4 (a) and 4 (c)
Pomax, VAC m corresponding to the operation waveform shown in (d)
In the case of Pomin and VAC max corresponding to the operation waveforms shown in FIGS.
ON is reduced to about 1/3. Accordingly, the amount of the current I1 flowing from the smoothing capacitor Ci to the voltage resonance type converter is also limited to approximately 1/3 as shown in the transition from FIG. 4B to FIG. 4I. Therefore, the control sensitivity is improved and the substantial control range is expanded. Thus, as described above, the variable width of the switching frequency can be smaller than in the related art.

In FIG. 5, the switching frequency is changed by the parallel resonance frequency fo of the primary side parallel resonance circuit.
1 and the series resonance frequency fo2 of the secondary-side series resonance circuit are shown. Here, for example, as shown in the figure, the inductance and the capacitance are set so that the parallel resonance frequency fo1 and the series resonance frequency fo2 become equal at 75 KHz. Is selected, the resonance impedance difference is reduced and the transmission efficiency from the primary side to the secondary side is increased, so that the corresponding maximum load power can be improved.

Further, since the maximum load power has been increased as described above, the present embodiment employs a double voltage rectification method as a rectifying and smoothing circuit for generating a DC input voltage. There is no need to cover the load power.
For this reason, as described with reference to FIG. 1, for example, the configuration of a normal equal-voltage rectifier circuit using a bridge rectifier circuit can be employed. Thereby, for example, the AC input voltage VAC = 14
The rectified smoothed voltage Ei at 4V is about 200V. The switching element Q1 // shown in FIGS.
The resonance voltage Vcr obtained at both ends of the parallel connection circuit of the secondary side parallel resonance capacitor Cr is generated when the switching element Q1 is turned off by the action of the primary side parallel resonance circuit on the rectified smoothed voltage Ei. In the present embodiment, as described above, the rectified smoothed voltage Ei is about 1/2
As a result, the resonance voltage Vcr is first set in FIG.
The resonance voltage Vcr (1800 V) generated in the conventional power supply circuit shown in FIGS. 12 and 16 can be suppressed to about 1/2. Also, as described above, the conduction angle of the switching element Q1 is variably controlled (PWM control).
As a result, the peak value of the resonance voltage Vcr is controlled to be substantially constant regardless of the rise in the AC input voltage VAC. As a result, in the power supply circuit of the present embodiment, the resonance voltage Vcr is suppressed to be substantially constant at a level of about 900 V at the peak. Therefore, in this embodiment, a 900 V withstand voltage product may be selected for the switching element Q1 and the parallel resonance capacitor Cr.

In this embodiment, since the switching element Q1, the parallel resonance capacitor Cr, and the rectifier diode on the secondary side can be used with a lower breakdown voltage than those conventionally required, only the elements are required. It will be cheaper. Therefore, for example, the switching element Q1 and the bridge rectifier circuit DO with improved characteristics (the saturation voltage VCE (SAT), the storage time tST
G, falling time tf, current amplification factor hFE, etc., or a rectifier diode with a forward voltage drop V
F, good characteristics such as the reverse recovery time trr) can be selected. Due to such improvement in characteristics, in the present embodiment, the switching frequency can be set higher than in the conventional case, and accordingly, reduction of power loss and reduction in size and weight of various components are promoted. That is, it is possible to improve various characteristics such as power conversion efficiency as compared with the related art, and to promote reduction in size, weight, and cost.

Further, from the viewpoint of reducing the size and weight of the power supply circuit, the conventional configuration including the voltage doubler rectifier circuit for generating the DC input voltage requires two sets of rectifier diodes and a smoothing capacitor. However, in the present embodiment, for example, a full-wave rectifier circuit using a normal bridge rectifier circuit can be used, so that a set of a block-type smoothing capacitor and a bridge rectifier diode can be employed. Therefore, cost reduction and downsizing of parts can be achieved.

Further, the number of turns of the primary winding N1 is reduced as compared with the conventional case by using the equal-size voltage rectifier circuit, but this also contributes to improvement of the constant voltage control sensitivity.

In the power supply circuit shown in FIG. 10, FIG. 12, or FIG. 16, an inductor of a power choke coil for the primary winding N1 or the secondary winding N2, or a controlled winding of the orthogonal control transformer PRT. NR are connected in series, whereas in the power supply circuit of the present embodiment, the primary winding N1
Alternatively, these windings are not connected in series to the secondary winding N2. In FIG. 1, the detection winding ND is connected in series to the primary winding N1.
Is 1T, for example, and the insulation converter transformer P
It has a negligible inductance value in terms of the effect of increasing the leakage inductance in IT. For this reason, the generation source of the leakage magnetic flux as the entire switching converter (power supply circuit) is the insulation converter transformer PI
T, as a countermeasure against leakage magnetic flux, it is sufficient to provide, for example, a short ring made of a copper plate to the transformer main body. For example, aluminum is required in the power supply circuit shown in FIG. 10, FIG. 12, or FIG. No shield case is required.

For example, as a practical example of the power supply circuit shown in FIG.
Maximum load power Pomax = 200W, minimum load power Po
min = 0 W, and the AC input voltage VAC is VAC
= 100V ± 20% fluctuation condition,
The ferrite EE type core of the insulating converter transformer PIT is EE35 type, the width of the gap G is 1 mm, the primary winding N1 = 50T, the secondary winding N2 = 25T, the parallel resonance capacitor Cr = 4700pF, the secondary side series resonance. In the configuration in which the capacitor Cs = 0.1 μF, as described with reference to FIG. 5, the switching frequency fs = 100 KH
The control range of z to 250 KHz makes it possible to stabilize the output of the secondary side, and the maximum load power Pomax = 200 W,
Under the condition of the AC input voltage VAC = 100 V, a high power conversion efficiency of 93% was obtained.

FIG. 6 is a circuit diagram showing a configuration of a switching power supply circuit according to a second embodiment of the present invention. In this figure, the same parts as those in FIG.

In the power supply circuit shown in FIG.
As the switching element Q1, bipolar transistors (BJT) Q11 and Q12, a damper diode (Zener diode) DD1, a damper diode DD2, and a resistor R1
1, a Darlington circuit formed by connecting R12 as shown. In this case, the connection mode of the Darlington circuit is such that the collector of the transistor Q11 is connected to the collector of the transistor Q12,
Is connected to the emitter of the transistor Q12, and the emitter of the transistor Q12 is grounded. Further, the anode of the damper diode DD1 is connected to the emitter of the transistor Q11, and the cathode of the damper diode DD1 is connected to the base of the transistor Q11 via the resistor R11. The anode of the damper diode DD2 is
The emitter is connected to the transistor Q12, and the cathode is connected to the collector of the transistor Q12. Resistance R12
Are connected in parallel between the base and the emitter of the transistor Q12. In the Darlington circuit thus formed, the base of the transistor Q11 is equivalent to the base of the switching element Q1 described in each of the above embodiments, and the collector contacts of the transistors Q11 and Q12 are equivalent to the collector of the switching element Q1. Become. Also,
The emitter of the transistor Q12 is equivalent to the emitter of the switching element Q1.

In this case, a self-excited oscillation circuit for driving the switching element in a self-excited manner is omitted.
Instead, a configuration in which the oscillation / drive circuit 2 is provided and switching driving by separately excited is performed is employed. For this reason, in the present embodiment, the isolated converter transformer PIT
Are provided with windings N4A and N4B. And winding N
A DC voltage of +12 V is generated by a half-wave rectifier circuit including 4A, rectifier diode D1, and capacitor CA.
A DC voltage of -12 V is generated by a half-wave rectifier circuit composed of a winding N4B, a rectifier diode D2, and a capacitor CB. The DC voltage of +12 V and −12 V is supplied to the oscillation / drive circuit 22 as an operation power supply. The structure of the insulating converter transformer PIT according to the present embodiment may be the same as the structure of the core shown in FIG. The difference from the first embodiment is that the windings N4A and N4B are additionally wound on the primary side.

The oscillation / drive circuit 2 is activated by an activation resistor RS, and generates an oscillation signal having a cycle having a required switching frequency fs. And, by using the operating power supply of + 12V / -12V,
The oscillating signal is converted into a positive (on) / negative (off) switching drive current for each switching cycle and output to the base terminal of the switching element Q1. As a result, the switching element Q1 is driven to perform a switching operation at a required switching frequency. When a Darlington circuit is employed for the switching element Q1 as in the present embodiment, higher power conversion efficiency can be obtained than when the switching element Q1 is a single bipolar transistor, for example.

In the control circuit 1 shown in this figure, for example, a DC signal (detection signal) having a level varied according to the level of the secondary DC output voltage EO1 as a detection input is sent to the oscillation / drive circuit 2. I am trying to supply. here,
In the oscillation / drive circuit 2, if the off-period of the switching element Q1 is made constant and the on-period is variably controlled to vary the switching frequency in accordance with the detection signal input from the control circuit 1, An operation similar to that shown in FIG. 4 is obtained, and FIG.
The same effect as that of the power supply circuit described above can be obtained.

In the above description of the embodiments,
The case where a single bipolar transistor (BJT) or a Darlington circuit having two bipolar transistors is employed as the switching element Q1 has been described as an example, but the embodiment of the present invention will be described below. The switching element can be employed instead of the switching element Q1.

In FIG. 7, a MOS-FET (MOS type field effect transistor;
An example using a metal oxide semiconductor is shown. MO
When an S-FET is used, a Zener diode ZD for forming a feedback current path at the time of switching off is connected in parallel in the direction shown in the figure between the drain and the source. That is, the anode is connected to the source of the MOS-FET, and the cathode is connected to the drain of the Zener diode ZD. In this case, the base, the collector, and the emitter of the switching element Q1 shown in the above embodiment are respectively the gate, the drain, and the MOS-FET.
Will be replaced by the source.

FIG. 8 shows an example in which the switching element Q1 is replaced with
An example using an IGBT (insulated gate bipolar transistor) is shown. A diode D for forming a feedback current path at the time of switching off is connected in parallel between the collector and the emitter of the IGBT. Here, the anode and cathode of the diode D are I
It is connected to the collector and emitter of the GBT.
In this circuit, the base, collector and emitter of the switching element Q1 shown in each of the above embodiments are respectively
Replaced by the gate, collector and emitter of the IGBT.

FIG. 9 shows that the switching element Q1 is replaced with
An example using an SIT (static induction thyristor) is shown. Even between the collector and emitter of this SIT,
A diode D for forming a feedback current path at the time of switching off is connected in parallel, and an anode and a cathode of the diode D are connected to a cathode and an anode of the SIT, respectively. In this circuit, the base, collector and emitter of the switching element Q1 shown in each of the above embodiments are replaced with the gate, anode and cathode of the SIT, respectively.

In this embodiment, even if any of the configurations shown in FIGS. 7 to 9 is employed, it is possible to further increase the efficiency. In the case where the configuration shown in FIGS. 7 to 9 is employed, although not shown here, the switching element Q
The configuration of the drive circuit is changed so as to conform to the characteristics of the element to be adopted in place of 1. For example, a separately-excited configuration may be adopted in accordance with the configuration shown in FIG. It is preferable to expect the stable operation. For example, in the case of the MOS-FET shown in FIG.

[0066]

As described above, according to the present invention, as a switching power supply circuit, a voltage resonance type switching converter is provided on the primary side, and an insulating converter transformer is loosely coupled, so that a primary winding and a secondary winding are connected. An operation mode in which the mutual inductances of the windings have opposite polarities (+ M /-
M) is obtained. Further, on the secondary side, a secondary side series resonance capacitor is connected in series to the secondary winding to form a series resonance circuit, and a voltage doubler full-wave rectifier circuit using the series resonance circuit is provided. As a result, a secondary DC output voltage corresponding to twice the alternating voltage (excitation voltage) obtained in the secondary winding is obtained.

As described above, the power is supplied to the load by the voltage doubler full-wave rectifier circuit. As a result, in the present invention, for example, the secondary DC output of the same size is provided by the full-wave rectifier circuit or the half-wave rectifier circuit as in the related art. The maximum load power that can be handled can be improved as compared with the case where a voltage is obtained. Along with this, the primary side is not a voltage doubler rectifier circuit, but a full-wave rectifier circuit is used to input a rectified smoothed voltage at a level corresponding to the AC input voltage level. It will be able to respond.

Further, as a configuration for constant voltage control for stabilizing the secondary output voltage, the switching frequency is changed in accordance with the secondary output voltage level, so that the resonance impedance in the power supply circuit and the switching The conduction angle of the element is controlled in a complex manner.

The following can be said from the above configuration.
For example, conventionally, in order to meet the above condition, it is necessary to obtain a rectified smoothed voltage corresponding to twice the AC input voltage level by a voltage doubler rectifier circuit. As the capacitor, it is necessary to select a withstand voltage product corresponding to the switching voltage generated according to the rectified smoothing voltage level. Conventionally, when a DC output voltage is generated by a half-wave rectifier circuit on the secondary side, a voltage that is approximately 2.5 to 3.5 times the rectified smoothed voltage during the non-conduction period of the rectifier diode. Therefore, a withstand voltage product corresponding to the voltage level is selected.

On the other hand, in the present invention, the switching voltage depending on the rectified smoothing voltage level is reduced to half of the conventional switching voltage. Can be used. Further, on the secondary side, as described above, the voltage doubler full-wave rectifier circuit is provided. Here, the doubler voltage full-wave rectifier circuit performs a full-wave rectification operation during both positive and negative periods of the alternating voltage. As a result of the rectification operation, the voltage applied to the rectification diode is suppressed to be substantially equal to the rectification smoothed voltage level,
As the rectifier diode on the secondary side, a diode having a lower withstand voltage than the conventional one can be selected. As a result, it is possible to reduce costs for the switching element, the primary-side parallel resonance capacitor, the secondary-side rectifier diode, and the like. In addition, it is also possible to easily select a switching element and a secondary rectifier diode having improved characteristics and set a high switching frequency, thereby improving the power conversion efficiency. Also,
This also makes it possible to reduce the size and weight of circuit components around the switching element. Further, as described above, since the circuit for obtaining the rectified smoothed voltage from the commercial AC power supply is a normal equal-voltage rectifier circuit, for example, a normal set of a block-type smoothing capacitor and a bridge rectifier diode are used. Therefore, the cost and the circuit scale can be reduced in this respect as well. Further, the number of windings of the controlled winding is reduced, so that the orthogonal control transformer used for constant voltage control can be reduced in size, weight and cost.

Further, according to the present invention, the rectifier circuit provided on the secondary side employs a voltage doubler full-wave rectifier circuit, so that a DC output voltage of the same level as that obtained, for example, when an equal voltage rectifier circuit is provided is obtained. By doing so, it is possible to reduce the number of turns of the secondary winding to about 1/2 of the conventional case.

Also, by changing the switching frequency, the resonance impedance with respect to the switching output and the conduction angle of the switching element are controlled in a complex manner, and the constant voltage control is performed by this action.
As a result, the control sensitivity is improved and the controllable range is expanded, so that it is possible to stabilize the secondary-side output voltage in a control range with a narrower switching frequency than in the related art. Such a reduction in the control range of the switching frequency can be achieved by reducing the number of windings wound around a transformer forming a power supply circuit,
It also contributes to the miniaturization of various component elements.

Further, the configuration of the constant voltage control circuit of the present invention is such that the circuit system for driving the switching element is of a self-excited type or a separately-excited type. Inductance elements such as choke coils or controlled windings of a quadrature control transformer are not connected in series. For this reason, the generation of leakage magnetic flux as a power supply circuit is limited only to the insulated converter transformer, so there is no need to cover the power supply circuit with a shield case as in the past, eliminating leakage magnetic flux just by applying a copper plate short ring, for example. it can. As a result, the power supply circuit itself can be reduced in size and weight and cost can be reduced, and the manufacturing time can be reduced.

The switching element can be constituted by a Darlington circuit formed with a bipolar transistor, or a MOS field effect transistor, an insulated gate bipolar transistor, or an electrostatic induction thyristor. For example, the power conversion efficiency can be further improved as compared with the case where the switching means is formed by a single bipolar transistor.

As described above, according to the present invention, the power supply circuit provided with the voltage resonance type converter on the primary side can be improved in various characteristics such as low cost, small size and light weight, and power conversion efficiency.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a perspective view showing a structure of an orthogonal control transformer provided in the power supply circuit shown in FIG.

3 is a perspective view showing a structure of an insulating converter transformer provided in the power supply circuit shown in FIG.

FIG. 4 is a waveform chart showing an operation of a main part in the power supply circuit shown in FIG.

FIG. 5 is an explanatory diagram showing a relationship between a switching frequency and a secondary-side DC output voltage.

FIG. 6 is a circuit diagram showing a configuration of a power supply circuit according to a second embodiment of the present invention.

FIG. 7 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 8 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 9 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 10 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

11 is a perspective view showing a structure of an orthogonal type insulated converter transformer provided in the power supply circuit shown in FIG.

FIG. 12 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

13 is a perspective view showing a structure of an orthogonal control transformer provided in the power supply circuit shown in FIG.

FIG. 14 is a perspective view showing a structure of an insulating converter transformer provided in the power supply circuit shown in FIG.

FIG. 15 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 16 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

[Explanation of symbols]

1 control circuit, 2 oscillation / drive circuit, Ci, CO1,
CO2 smoothing capacitor, Cr parallel resonance capacitor, C
s Secondary side series resonance capacitor, Di bridge rectifier circuit, DO1, DO2, DO3, DO4 rectifier diode, PIT
Insulation converter transformer, PRT orthogonal control transformer, NC control winding, ND detection winding, NB drive winding,
Q1 Switching element

Claims (7)

[Claims] A rectifying / smoothing means for receiving a commercial AC power supply, generating a rectified smoothed voltage having a level corresponding to an equal multiple of the commercial AC power supply level, and outputting the rectified smoothed voltage as a DC input voltage; A gap is formed so as to obtain the coupling coefficient of, and an insulating converter transformer provided for transmitting the primary side output to the secondary side; The switching means is configured to output to a line, and is formed by at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of a parallel resonance capacitor, and the operation of the switching means is a voltage resonance type. The primary side parallel resonance circuit and the secondary side series By connecting a capacitor in series, a leakage inductance component of the secondary winding of the insulating converter transformer and a capacitance of the secondary series resonance capacitor form a secondary series resonance circuit that forms a series resonance circuit; The secondary voltage is formed by inserting the secondary side series resonance capacitor into the current path. DC output voltage generation means configured to be able to generate a secondary DC output voltage corresponding to approximately twice as large as the above, and changing a switching frequency of the switching element according to a level of the secondary DC output voltage. In such a way that the resonance impedance of the primary parallel resonance circuit and the conduction angle of the switching element are controlled in a complex manner,
And a constant voltage control means configured to perform constant voltage control on the secondary side output voltage. 2. The switching means is formed to include at least a series resonance circuit formed by a series connection of a driving winding and a resonance capacitor, and the switching element is self-excited based on a resonance output of the series resonance circuit. The constant voltage control means includes: the detection winding and the drive winding connected in series to a primary winding of the insulating converter transformer; and the detection winding. And a quadrature control transformer as a saturable reactor around which a drive winding and a control winding whose winding direction is orthogonal are provided, and a variable control current is provided in accordance with the level of the secondary DC output voltage. 2. The switching frequency can be variably controlled by flowing through the control winding to change the inductance of the drive winding. Switching power supply circuit according. 3. The switching means includes a separately-excited drive circuit for driving the switching element in a separately-excited manner, and the constant-voltage control means includes a switching element in accordance with a level of the secondary-side DC output voltage. The switching power supply circuit according to claim 1, wherein the switching frequency is variably controllable by variably controlling the ON period while keeping the OFF period constant. 4. The switching power supply circuit according to claim 1, wherein said switching means is configured to use a Darlington circuit formed with a bipolar transistor as one switching element. 5. The switching power supply circuit according to claim 1, wherein said switching means includes a MOS field effect transistor as a switching element. 6. The switching power supply circuit according to claim 1, wherein said switching means includes an insulated gate bipolar transistor as a switching element. 7. The switching power supply circuit according to claim 1, wherein said switching means is formed with an electrostatic induction thyristor as a switching element.