JP2000151554A - Method for deciding frequency deviation, method for tuning image receiver to multiplex carrier wave signal and multiplex carrier wave broadcasting signal image receiver - Google Patents
Method for deciding frequency deviation, method for tuning image receiver to multiplex carrier wave signal and multiplex carrier wave broadcasting signal image receiverInfo
- Publication number
- JP2000151554A JP2000151554A JP11296673A JP29667399A JP2000151554A JP 2000151554 A JP2000151554 A JP 2000151554A JP 11296673 A JP11296673 A JP 11296673A JP 29667399 A JP29667399 A JP 29667399A JP 2000151554 A JP2000151554 A JP 2000151554A
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- Japan
- Prior art keywords
- frequency
- tuning
- frequency deviation
- sum
- carrier wave
- Prior art date
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Links
- 238000000034 method Methods 0.000 title claims abstract description 33
- 230000001419 dependent effect Effects 0.000 claims 1
- 238000011156 evaluation Methods 0.000 abstract description 8
- 238000012545 processing Methods 0.000 abstract description 5
- 238000010586 diagram Methods 0.000 description 11
- 238000004891 communication Methods 0.000 description 9
- 239000000969 carrier Substances 0.000 description 4
- 238000003780 insertion Methods 0.000 description 4
- 230000037431 insertion Effects 0.000 description 4
- 238000009966 trimming Methods 0.000 description 4
- 239000002131 composite material Substances 0.000 description 2
- 238000012937 correction Methods 0.000 description 2
- 238000009826 distribution Methods 0.000 description 2
- 238000005562 fading Methods 0.000 description 2
- 238000012546 transfer Methods 0.000 description 2
- 230000003044 adaptive effect Effects 0.000 description 1
- 230000005540 biological transmission Effects 0.000 description 1
- 230000015556 catabolic process Effects 0.000 description 1
- 230000001427 coherent effect Effects 0.000 description 1
- 230000002301 combined effect Effects 0.000 description 1
- 238000006731 degradation reaction Methods 0.000 description 1
- 238000001514 detection method Methods 0.000 description 1
- 230000000694 effects Effects 0.000 description 1
- 230000001771 impaired effect Effects 0.000 description 1
- 238000012986 modification Methods 0.000 description 1
- 230000004048 modification Effects 0.000 description 1
- 238000000926 separation method Methods 0.000 description 1
- 238000001228 spectrum Methods 0.000 description 1
- 230000001360 synchronised effect Effects 0.000 description 1
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2647—Arrangements specific to the receiver only
- H04L27/2655—Synchronisation arrangements
- H04L27/2668—Details of algorithms
- H04L27/2673—Details of algorithms characterised by synchronisation parameters
- H04L27/2675—Pilot or known symbols
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2647—Arrangements specific to the receiver only
- H04L27/2655—Synchronisation arrangements
- H04L27/2657—Carrier synchronisation
- H04L27/2659—Coarse or integer frequency offset determination and synchronisation
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
- H04L2027/0024—Carrier regulation at the receiver end
- H04L2027/0026—Correction of carrier offset
- H04L2027/0028—Correction of carrier offset at passband only
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
- H04L2027/0044—Control loops for carrier regulation
- H04L2027/0053—Closed loops
- H04L2027/0055—Closed loops single phase
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
- H04L2027/0044—Control loops for carrier regulation
- H04L2027/0063—Elements of loops
- H04L2027/0065—Frequency error detectors
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/0014—Carrier regulation
- H04L2027/0083—Signalling arrangements
- H04L2027/0089—In-band signals
- H04L2027/0091—Continuous signals
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2602—Signal structure
- H04L27/261—Details of reference signals
- H04L27/2613—Structure of the reference signals
Landscapes
- Engineering & Computer Science (AREA)
- Computer Networks & Wireless Communication (AREA)
- Signal Processing (AREA)
- Two-Way Televisions, Distribution Of Moving Picture Or The Like (AREA)
- Synchronisation In Digital Transmission Systems (AREA)
- Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
- Noise Elimination (AREA)
Abstract
Description
【0001】[0001]
【発明の属する技術分野】この発明は、例えば同期化の
目的のために出力を増強したパイロット副搬送波を使用
したような多重搬送波通信システムでの周波数同期化の
分野に関する。本発明は特に、例えばデジタル・ビデオ
放送受像機のようなOFDM(OrthogonalFrequency Di
vision Multiplex)信号用の受像機の同調に関するが、
これに限定されるものではない。The present invention relates to the field of frequency synchronization in a multi-carrier communication system, for example, using a pilot sub-carrier with enhanced power for synchronization purposes. The present invention is particularly applicable to OFDM (Orthogonal Frequency Diode) such as a digital video broadcast receiver.
vision Multiplex) Receiver tuning for signals
It is not limited to this.
【0002】[0002]
【従来の技術】多重搬送波通信システムはデータ伝送お
よびその他の補助機能用に等間隔を隔てた多数の副搬送
波を利用している。復調プロセスを適正に行うため、こ
のようなシステム用に開発された受像機は極めて僅かな
残留周波数偏差で周波数ロックされなければならない。
周波数ロックを容易にするため、あるクラスの多重搬送
波通信システムは、増強された出力を有する選択された
副搬送波で送信されるパイロットの集合を利用してい
る。選択されたこれらの副搬送波はサイドローブが低い
ことを特徴とする最適な自己相関を伴うパイロット挿入
パターンを形成する必要がある。BACKGROUND OF THE INVENTION Multi-carrier communication systems utilize a number of equally spaced sub-carriers for data transmission and other auxiliary functions. In order for the demodulation process to work properly, receivers developed for such systems must be frequency locked with very little residual frequency deviation.
To facilitate frequency locking, some classes of multi-carrier communication systems utilize a set of pilots transmitted on selected sub-carriers having enhanced power. These selected sub-carriers need to form a pilot insertion pattern with optimal autocorrelation characterized by low side lobes.
【0003】[0003]
【発明が解決しようとする課題】図1はノイズや干渉が
なく、かつ通信チャネルにひずみがない理想的な場合に
受像機によって再構成されるDFT係数に関連する出力
の例を示している。しかし、実際の適用においては、受
信された信号は広帯域ノイズ並びに様々な干渉源によ
り、ある周波数で発生される強力な狭帯域ノイズによっ
て損なわれる。更に、通信チャネルの伝達関数が未だ修
正されていない場合には、チャネル自体に振幅と位相の
双方のひずみが誘発される。図2はある副搬送波周波数
で出現する周波数選択的なフェージング、ノイズおよび
強い干渉の複合的な作用に起因するひずみを示してい
る。図示のとおり、この場合はパイロット副搬送波とデ
ータ副搬送波との識別はより一層困難であり、その結
果、周波数ロック用に使用されるいずれのシステムも性
能が劣化する。FIG. 1 shows an example of the output associated with the DFT coefficients reconstructed by the receiver in an ideal case without noise and interference and without distortion in the communication channel. However, in practical applications, the received signal is corrupted by broadband noise as well as strong narrowband noise generated at certain frequencies by various sources of interference. Furthermore, if the transfer function of the communication channel has not yet been modified, both amplitude and phase distortions will be induced in the channel itself. FIG. 2 shows the distortion resulting from the combined effects of frequency selective fading, noise and strong interference appearing at certain subcarrier frequencies. As can be seen, in this case, the distinction between pilot and data subcarriers is much more difficult, resulting in degraded performance of any system used for frequency locking.
【0004】図3は2つの成分すなわち分数部分ξ△f
cと粗周波数偏差J△fcとを有する全周波数偏差△f
を示しており、但しJは図示した例では2であり、△f
cは副搬送波分離である。多重搬送波通信システムでは
分数周波数偏差を評価し、かつ修正するために幾つかの
時間領域方法を利用できる。しかし、粗周波数偏差を評
価するために提案されている方法の大多数は、識別され
ようとする信号間の周知の位相関係を利用する可干渉性
の処理に基づいている。このような方法は、チャネル伝
達関数が未だ修正されていない場合には、周波数獲得プ
ロセスを開始するにはそれほど適してはいない。FIG. 3 shows two components, a fractional part ξ △ f
c and the total frequency deviation Δf having the coarse frequency deviation JΔfc
Where J is 2 in the illustrated example and Δf
c is the subcarrier separation. In a multi-carrier communication system, several time-domain methods are available for evaluating and correcting for fractional frequency deviation. However, the majority of proposed methods for estimating the coarse frequency deviation are based on coherent processing, which makes use of the known phase relationship between the signals to be identified. Such a method is not very suitable for starting the frequency acquisition process if the channel transfer function has not yet been modified.
【0005】[0005]
【課題を解決するための手段】本発明の諸側面は添付の
特許請求の範囲に記載されている。Summary of the Invention Aspects of the present invention are set forth in the appended claims.
【0006】本発明の好適な実施例によって、未修正の
通信チャネルによって激しくひずみ、かつ、広帯域ノイ
ズと強い狭帯域干渉によって損なわれることがある受信
信号を適切に処理することによって、粗周波数偏差の評
価が可能になる。[0006] The preferred embodiment of the present invention provides for the proper processing of received signals which can be severely distorted by unmodified communication channels and impaired by wideband noise and strong narrowband interference, to reduce coarse frequency deviation. Evaluation becomes possible.
【0007】好適な実施例に基づいて、信号処理装置は
複数の周波数偏差候補の各々について、副搬送波の集合
の出力の合計を算出することによって、同調目的のため
に周波数偏差を判定するものである。ある条件が満たさ
れると、上記の合計のうちの最大のものに関連する周波
数偏差が同調を調整するために利用される。In accordance with a preferred embodiment, the signal processor determines the frequency deviation for tuning purposes by calculating the sum of the outputs of a set of subcarriers for each of a plurality of frequency deviation candidates. is there. If certain conditions are met, the frequency deviation associated with the largest of the above sums is used to adjust the tuning.
【0008】好適には、各周波数偏差候補について、出
力の合計は副搬送波出力のうち最大であるUを排除す
る。但し、Uは1以上の整数である。それによって強力
な干渉信号の結果としての周波数偏差選択における誤り
が回避される。[0008] Preferably, for each frequency deviation candidate, the sum of the outputs excludes U, which is the largest of the subcarrier outputs. Here, U is an integer of 1 or more. This avoids errors in frequency deviation selection as a result of strong interference signals.
【0009】好適には、関連する出力の合計が、他の周
波数偏差に関連する出力の合計の少なくとも幾つかと所
定の関係にある場合にのみ、同調目的のために周波数偏
差が選択される。好適には、最大の合計は次に最大であ
るLの出力合計の平均値と所定の関係になければなら
ず、但し、Lは1以上の整数である。Preferably, the frequency deviation is selected for tuning purposes only if the sum of the associated outputs is in a predetermined relationship with at least some of the sums of the outputs associated with other frequency deviations. Preferably, the largest sum must be in a predetermined relationship with the average of the next largest L output sum, where L is an integer greater than or equal to one.
【0010】好適には、各周波数偏差に関連する出力の
合計は、目標の周波数偏差をより高い信頼性で判定する
ために複数の記号周期にわたって積分される。好適に
は、この動作は、所定の判定基準が満たされると停止さ
れ、この判定基準は適正な周波数偏差が判定されたこと
を高度の信頼性を以って示すものである。このことは、
信頼できる周波数偏差の評価のために必要な記号周期の
数は予め定まっていないことを意味しており、ひいては
ロック・イン期間が可能な限り短縮される。Preferably, the sum of the outputs associated with each frequency deviation is integrated over a plurality of symbol periods to more reliably determine the target frequency deviation. Preferably, the operation is stopped when a predetermined criterion is met, the criterion indicating with a high degree of reliability that an appropriate frequency deviation has been determined. This means
This means that the number of symbol periods required for reliable frequency deviation evaluation is not predetermined, and thus the lock-in period is reduced as much as possible.
【0011】可変的な観察期間を許容するため、目標周
波数偏差が判定されたことを確証する判定基準は観察の
ために利用される信号周期数の関数である。[0011] To allow for a variable observation period, the criterion for establishing that the target frequency deviation has been determined is a function of the number of signal periods utilized for observation.
【0012】このようにして本発明は、粗周波数偏差の
信頼できる評価を判定するために必要な時間を最小限に
する逐次的な決定手順を利用した信号処理装置を提供す
るものである。Thus, the present invention provides a signal processing apparatus utilizing a sequential decision procedure that minimizes the time required to determine a reliable estimate of a coarse frequency deviation.
【0013】[0013]
【発明の実施の形態】実施の形態1.次に添付図面を参
照して本発明の実施の形態を説明する。DESCRIPTION OF THE PREFERRED EMBODIMENTS Embodiment 1 Next, embodiments of the present invention will be described with reference to the accompanying drawings.
【0014】デジタル放送受像機の形式の本発明の実施
の形態を、デジタル放送受像機のブロック図である図4
を参照して説明する。FIG. 4 is a block diagram of a digital broadcast receiver according to an embodiment of the present invention in the form of a digital broadcast receiver.
This will be described with reference to FIG.
【0015】このデジタル放送受像機の従来形の部分
は、OFDM多重搬送波放送信号を受信するアンテナ1
と、受信した放送信号を増幅する無線周波数増幅器2
と、増幅された信号を中間周波数信号に逓降変換するミ
キサ3と、中間周波数信号を増幅する中間周波数増幅器
4と、増幅された中間周波数信号を復調して同相(I)
および直角位相(Q)ベースバンド信号を生成する直交
復調器5と、これらのベースバンド信号を複素値デジタ
ル信号に変換するアナログ−デジタル変換器(ADC)
6と、前記デジタル信号を離散フーリエ変換して各々の
副搬送波について副記号データを得るための高速フーリ
エ変換(FFT)プロセッサ7と、副記号データ中のエ
ラーを検出し、かつ修正するエラー修正プロセッサ8
と、副記号データが供給される出力端子9と、電圧制御
発振器(VOC)10とを備える。電圧制御発振器10
は、OFDM放送周波数と所定量だけ異なる周波数に同
調された信号をミキサ3に供給する局部発振器として使
用されている。この発振器10は、FFTプロセッサ7
の出力と結合された同調コントローラ101から制御信
号を受信する。The conventional part of this digital broadcast receiver is an antenna 1 for receiving an OFDM multi-carrier broadcast signal.
And a radio frequency amplifier 2 for amplifying a received broadcast signal
A mixer 3 for down-converting the amplified signal to an intermediate frequency signal, an intermediate frequency amplifier 4 for amplifying the intermediate frequency signal, and an in-phase (I) demodulating the amplified intermediate frequency signal.
And a quadrature demodulator 5 for generating quadrature (Q) baseband signals, and an analog-to-digital converter (ADC) for converting these baseband signals into complex-valued digital signals
6, a fast Fourier transform (FFT) processor 7 for performing a discrete Fourier transform on the digital signal to obtain sub-symbol data for each sub-carrier, and an error correction processor for detecting and correcting errors in the sub-symbol data 8
And an output terminal 9 to which sub-symbol data is supplied, and a voltage controlled oscillator (VOC) 10. Voltage controlled oscillator 10
Is used as a local oscillator that supplies a signal tuned to a frequency different from the OFDM broadcast frequency by a predetermined amount to the mixer 3. This oscillator 10 has an FFT processor 7
Receive a control signal from the tuning controller 101 coupled to the output of the tuning controller 101.
【0016】分数周波数偏差は、ADC6の出力を受け
るように結合された周波数微調整回路102を利用し
て、利用できる当業者には公知の時間領域技術の1つを
用いて既に修正されたものと想定する。同様に、記号の
タイミングは既に評価されているので、受信された信号
を最小限のタイミング・エラーで効率良くサンプリング
できるものと想定する。The fractional frequency deviation has already been corrected using one of the time domain techniques known to those skilled in the art available using a frequency fine adjustment circuit 102 coupled to receive the output of the ADC 6. Assume that Similarly, since the timing of the symbols has been evaluated, it is assumed that the received signal can be efficiently sampled with minimal timing errors.
【0017】周波数粗調整回路103は、それぞれの副
搬送波が確実に適正位置にあるようにするために用いら
れる。The coarse frequency adjustment circuit 103 is used to ensure that each sub-carrier is at an appropriate position.
【0018】指標K1,K2,...KPを有する副搬送波で送
信されたP個のパイロットがあるものと想定してみる。
順序付けされたこれらの指標集合をパイロット挿入パタ
ーンと呼ぶことにする。更に、複数の副搬送波間隔内で
測定された可能な粗周波数偏差の範囲が、以下のものと
想定する。Suppose there are P pilots transmitted on subcarriers having indices K 1 , K 2 ,... K P.
These ordered index sets will be referred to as pilot insertion patterns. Further assume that the range of possible coarse frequency deviations measured within a plurality of sub-carrier intervals is as follows.
【0019】(−Jmin、−Jmin+1、...、−1、0
、1、...Jmax−1、Jmax)[0019] (- J min, -J min +1 , ..., - 1,0
, 1,. . . J max -1, J max )
【0020】従って、可能性がある偏差の総数は(Jmin
+Jmax+1)に等しい。Therefore, the total number of possible deviations is (J min
+ J max +1).
【0021】図5は、粗周波数偏差の非干渉性評価のた
めの周波数粗調整回路103のブロック図である。復調
目的のために離散フーリエ変換(DFT)を行うFFT
プロセッサ7の出力は、出力セレクタおよびマルチプレ
クサ(OMS:Output Selector and Multiplexer)2
0の入力に接続される。OMSは(Jmin+Jmax+1)の出
力チャネルを有しており、各チャネルはパイロット挿入
パターンのそれぞれの周波数偏移バージョンに対応する
Pの複合DFT係数をそれぞれ提供する。これらの係数
はDFT係数を処理するための(Jmin+Jmax+1)のトリ
ミングおよび加算ユニット(TSU:Trimming and Sum
ming Unit)22に送られ、これらの係数はオリジナル
のパイロット挿入パターンおよび、可能性がある全ての
粗周波数偏差によって周波数偏移される(Jmin+Jmax)
パージョンとを表すものである。FIG. 5 is a block diagram of the coarse frequency adjustment circuit 103 for evaluating noncoherence of the coarse frequency deviation. FFT with discrete Fourier transform (DFT) for demodulation purposes
The output of the processor 7 is an output selector and multiplexer (OMS) 2
Connected to 0 input. The OMS has (J min + J max +1) output channels, each channel providing a respective P composite DFT coefficient corresponding to a respective frequency shifted version of the pilot insertion pattern. These coefficients are a (J min + J max +1) trimming and sum unit (TSU) for processing the DFT coefficients.
ming unit 22 and these coefficients are frequency shifted by the original pilot insertion pattern and any possible coarse frequency deviations (J min + J max ).
It represents a version.
【0022】各々のTSU22の機能ブロック図を図6
に示す。TSUは下記の演算を行う。FIG. 6 is a functional block diagram of each TSU 22.
Shown in The TSU performs the following operation.
【0023】先ず、ブロック222で、各複合DFT係
数を用いて、 q=x2+y2 が計算され、但し、xとyはDFT係数のそれぞれ実部
と虚部である。全ての出力値がブロック224で累算器
2によって加算される。更に、これらの出力値はブロッ
ク226で分類される。最大値U(Uは1、または好適
には1より大きい整数である)がブロック228で累算
器1によって合計される。次に230で、その結果が2
24で計算された合計値から差し引かれる。このように
して、Uの最大値qを無視し、かつ残りの(P−U)の
値を加算することで“トリミングされた”合計Sが形成
される。トリミングの目的は、異なる等級の信号を表す
配分間での統計的な距離を増大することで、それらを識
別し易くすることにある。その効果は、大きい出力の検
出により、正しい周波数偏差が見出されたという誤った
仮定を回避することにあり、その場合、これらの大きい
出力は実際には、例えば強力な干渉信号の結果なのであ
る。Uの値は経験式に選択することができる。First, at block 222, q = x 2 + y 2 is calculated using each composite DFT coefficient, where x and y are the real and imaginary parts of the DFT coefficient, respectively. All output values are added by accumulator 2 at block 224. Further, these output values are classified at block 226. The maximum value U (where U is 1, or preferably an integer greater than 1) is summed by accumulator 1 at block 228. Then at 230, the result is 2
It is subtracted from the sum calculated at 24. In this way, a "trimmed" sum S is formed by ignoring the maximum value q of U and adding the remaining (PU) values. The purpose of the trimming is to increase the statistical distance between the distributions representing signals of different grades so that they can be easily identified. The effect is to avoid the false assumption that the correct frequency deviation has been found by the detection of large powers, in which case these large powers are actually a result, for example, of a strong interference signal . The value of U can be chosen empirically.
【0024】TSUは並行して動作するので、(Jmin+
Jmax+1)のトリミングされた合計は図5の24に示した
分類および記憶レジスタ(SSR:Sorting and Storin
g Register)の入力に同時に送られる。Since the TSU operates in parallel, (J min +
J max +1) is the sum of the classification and storage registers (SSR: Sorting and Storin) shown at 24 in FIG.
g Register).
【0025】図7はSSR24の機能ブロック図であ
る。SSRによって行われる動作は下記のように要約す
ることができる。ブロック300で、TSUからのトリ
ミングされた合計Sを分類する、ブロック302で、ト
リミングされた合計のうち最大のものSmaxを選択す
る、−ブロック304で、該最大のトリミングされた合
計に相当する偏移の値、すなわちJ*を記憶する、−ブ
ロック306で、“次に”最大であるLのトリミングさ
れた合計についてその平均値AVLを計算する、−上記
の3つの値を図5の26に示した決定ブロック(DB:
Decision Block)に供給する。FIG. 7 is a functional block diagram of the SSR 24. The operations performed by the SSR can be summarized as follows. At block 300, classify the trimmed sum S from the TSU, at block 302, select the largest of the trimmed sums S max- at block 304, corresponding to the largest trimmed sum. the value of the deviation, i.e., stores the J *, - in block 306, "then" for trimmed sum of L is the maximum for the average value calculated AV L, - in FIG. 5 three values of the Decision block (DB: 26)
Decision Block).
【0026】決定ブロック(DB)26のブロック図を
図8に示す。この決定ブロックはブロック400でSS
R24および記号周期カウンタ(SPC:Symbol Perio
d Counter)28(図5)によって与えられた情報を利
用して、下記の公式に従って、決定しきい値を判定す
る。FIG. 8 shows a block diagram of the decision block (DB) 26. The decision block is SS at block 400
R24 and symbol period counter (SPC: Symbol Period)
Using the information provided by (d Counter) 28 (FIG. 5), a decision threshold is determined according to the following formula:
【0027】TH=AVL[1+h/√(LM)]TH = AV L [1 + h / √ (LM)]
【0028】但し、Mは観察のために用いられる記号周
期の数であり、hは、好適には0.6より大きい定数で
ある。hの値は、(誤ったロックにつながる)正しくな
い評価を得る可能性を低めることと、決定を行うのに必
要な総観察時間(ロックまでの時間)を短縮することと
の良好な調整を図るために経験的に選択してもよい。Where M is the number of symbol periods used for observation and h is preferably a constant greater than 0.6. The value of h is a good compromise between reducing the chance of getting an incorrect evaluation (which leads to false locks) and reducing the total observation time (time to lock) required to make a decision. It may be selected empirically to achieve this.
【0029】次に、ブロック402で、DBはSmaxの
値と計算されたしきい値THとを比較する。Smaxがし
きい値THを超えている場合は、ブロック404で副搬
送波間隔の倍数で測定された粗周波数偏差の評価として
偏移値J* を用いる。しかし、しきい値THを超えな
い場合は、得られる情報は偏差の信頼できる評価を判定
するには不十分であり、次の記号周期から得られた追加
の観察を処理しなければならない。このような順次的決
定手順は、Smaxがしきい値THを超えた場合に終了
し、または観察された記号周期の総数Mが所定の最大値
Mmaxに達した場合に打ち切られる。従って、極端な場
合、特にMmaxの値が小さく、スペクトルひずみが激し
い場合は、周波数偏移の評価を得ずに順次手順を終了す
ることが可能である。ちなみに、装置が連続する全ての
記号周期を考慮に入れることが好適ではなるが、これは
必ずしも不可欠ではなく、例えば予備の周期を無視して
もよい。Next, at block 402, the DB compares the value of Smax with the calculated threshold TH. If S max exceeds the threshold TH, block 404 uses the deviation value J * as an estimate of the coarse frequency deviation measured at multiples of the subcarrier spacing. However, if the threshold TH is not exceeded, the information obtained is not sufficient to determine a reliable estimate of the deviation, and additional observations obtained from the next symbol period have to be processed. Such a sequential decision procedure is terminated when S max exceeds the threshold TH or is aborted when the total number M of observed symbol periods reaches a predetermined maximum value M max . Therefore, in an extreme case, particularly when the value of M max is small and the spectrum distortion is severe, the procedure can be sequentially terminated without obtaining the evaluation of the frequency shift. By the way, it is preferred that the device take into account all successive symbol periods, but this is not essential, for example, spare periods may be ignored.
【0030】上述のブロックおよびユニットの動作は、
全ての累積器、レジスタおよびカウンタをリセットする
図5の制御およびタイミング・ユニット(CTU:Cont
roland Timing Unit)30によって開始される。CTU
は更に、順次的決定手順を終了、または打ち切りにする
時点を判定する。The operation of the blocks and units described above
The control and timing unit (CTU: Cont) of FIG. 5 that resets all accumulators, registers and counters
(Roland Timing Unit) 30. CTU
Further determines when to terminate or discontinue the sequential determination procedure.
【0031】記号周期カウンタ(SPC)28は、順次
的評価手順の開始から現段階に至るまでに処理された記
号周期の数Mを判定する。この情報は上述のような適応
する決定しきい値THを判定するために決定ブロック
(DB)によって利用される。Mの現在値は更に、周波
数偏移の評価を行わずに順次的手順を打ち切るか否かを
決定するためにCTU30によっても利用される。A symbol period counter (SPC) 28 determines the number M of symbol periods processed from the start of the sequential evaluation procedure to the current stage. This information is used by the decision block (DB) to determine an adaptive decision threshold TH as described above. The current value of M is also used by the CTU 30 to determine whether to abort the sequential procedure without evaluating the frequency shift.
【0032】2KのOFDM信号(すなわち2000の
副搬送波を有する信号)と共に使用される特定の実施例
では、Jmin=Jmax=20にすることが特に望ましく、
ひいては可能性がある偏差が41あり、P=45、U=
8または好適には4、およびL=12であることが判明
しているが、必要ならば、勿論これらの各々を個々に変
更することもできる。In a particular embodiment used with a 2K OFDM signal (ie, a signal with 2000 subcarriers), it is particularly desirable to have J min = J max = 20,
Thus there are 41 possible deviations, P = 45, U =
It has been found that 8 or preferably 4 and L = 12, but of course each of these can of course also be modified individually, if desired.
【0033】上記の実施の形態は例えば適宜に設計され
た論理ゲートを有するASICを使用して、全体をハー
ドウェアで実現してもよい。あるいは、機能の幾つか、
または全てを単数または複数の適宜にプログラムされた
汎用プロセッサ・ユニットによって実行してもよい。幾
つかの機能を個々のプロセッサ・ユニットで実施するよ
うな場合は、これらの機能を並行的にではなく、逐次的
に実行することが望ましく、または必要である。The above embodiment may be realized entirely by hardware, for example, by using an ASIC having appropriately designed logic gates. Or some of the features,
Or all may be performed by one or more appropriately programmed general purpose processor units. If some functions are to be performed by individual processor units, it may be desirable or necessary to perform these functions sequentially, rather than in parallel.
【0034】これまで本発明を、パイロット副搬送波が
出力を増大した多重搬送波信号の文脈で説明してきた
が、適宜の修正を加えることによって本発明を(計算さ
れた出力の最小限を探索することによって)パイロット
の出力が減少し、または出力がない信号と共に、または
(計算された出力の所定の配分を探索することによっ
て)時間および/または副搬送波指標に基づいて出力が
変化する信号と共に使用することもできよう。Although the invention has been described in the context of a multi-carrier signal in which the pilot sub-carrier has increased power, the invention can be modified (with a search for a minimum of the calculated power by making appropriate modifications). For use with signals with reduced or no power of the pilot, or with signals whose power varies based on time and / or sub-carrier indicators (by searching for a predetermined distribution of calculated power). I could do it.
【0035】[0035]
【発明の効果】多重搬送波通信システムで、周波数偏差
を選択して、この偏差について所定の副搬送波の集合の
出力の合計を判定し、この合計と他の合計の少なくとも
幾つかとに所定の関係がある場合は、前記合計のうちの
選択された1つに関連する周波数偏差を表す信号を供給
することで設定周波数と目標周波数との間の周波数偏差
を判定することによって、ノイズやひずみによる劣化が
なく、受像機を多重搬送波信号と同期化することができ
る。In a multi-carrier communication system, a frequency deviation is selected and the sum of the outputs of a given set of sub-carriers is determined for this deviation, and a predetermined relationship is established between this sum and at least some of the other sums. In some cases, by determining a frequency deviation between the set frequency and the target frequency by providing a signal representing a frequency deviation associated with a selected one of the sums, degradation due to noise and distortion is reduced. Instead, the receiver can be synchronized with the multi-carrier signal.
【図1】 ノイズや干渉がなく、通信チャネルにひずみ
がない理想的な場合に、受像機によって再構成されるD
FT係数に関連する出力の例を示す。FIG. 1 shows a D reconstructed by a receiver in an ideal case with no noise or interference and no distortion in the communication channel.
5 shows an example of an output related to an FT coefficient.
【図2】 受信信号がノイズおよび強力な干渉によって
損なわれ、更に周波数選択的なフェージングに起因して
チャネルによるひずみがある場合に受像機によって再構
成されるDFT係数に関連する出力の代表例を示す。FIG. 2 shows a representative example of the output associated with the DFT coefficients reconstructed by the receiver when the received signal is corrupted by noise and strong interference and there is distortion due to the channel due to frequency selective fading. Show.
【図3】 分数部分と粗周波数偏差からなる全周波数偏
差の2つの成分を示す図である。FIG. 3 is a diagram showing two components of a total frequency deviation including a fractional part and a coarse frequency deviation.
【図4】 本発明に係る受像機のブロック図である。FIG. 4 is a block diagram of a receiver according to the present invention.
【図5】 粗周波数偏差の非干渉性評価のために受像機
で使用される装置(周波数粗調整回路)のブロック図で
ある。FIG. 5 is a block diagram of an apparatus (frequency coarse adjustment circuit) used in the receiver for noncoherence evaluation of the coarse frequency deviation.
【図6】 該装置(周波数粗調整回路)のトリミングお
よび加算ユニット(TSU)のブロック図である。FIG. 6 is a block diagram of a trimming and adding unit (TSU) of the device (frequency coarse adjustment circuit).
【図7】 前記装置(周波数粗調整回路)の分類および
記憶レジスタ(SSR)のブロック図である。FIG. 7 is a block diagram of a classification and storage register (SSR) of the device (coarse frequency adjustment circuit).
【図8】 前記装置(周波数粗調整回路)の決定ブロッ
ク(DB)のブロック図である。FIG. 8 is a block diagram of a decision block (DB) of the apparatus (coarse frequency adjustment circuit).
1 アンテナ、2 無線周波数増幅器、3 ミキサ、4
中間周波数増幅器、5 直交復調器、6 アナログ−
デジタル変換器(ADC)、7 高速フーリエ変換プロ
セッサ、8 エラー修正プロセッサ、9 出力端子、1
0 電圧制御発振器(VCO)、20 出力セレクタお
よびマルチプレクサ(OSM)、22トリミングおよび
加算ユニット(TSU)、24 分類および記憶レジス
タ(SSR)、26 決定ブロック(DB)、28 記
号周期カウンタ(SPC)、30 制御およびタイミン
グ・ユニット(CTU)、101 同調コントローラ、
102 周波数微調整回路、103 周波数粗調整回
路。1 antenna, 2 radio frequency amplifier, 3 mixer, 4
Intermediate frequency amplifier, 5 quadrature demodulator, 6 analog
Digital Converter (ADC), 7 Fast Fourier Transform Processor, 8 Error Correction Processor, 9 Output Terminal, 1
0 voltage controlled oscillator (VCO), 20 output selector and multiplexer (OSM), 22 trimming and adding unit (TSU), 24 classification and storage register (SSR), 26 decision block (DB), 28 symbol period counter (SPC), 30 control and timing unit (CTU), 101 tuning controller,
102 Frequency fine adjustment circuit, 103 Frequency coarse adjustment circuit.
Claims (14)
周波数と目標周波数との間の周波数偏差を判定する方法
であって、 (a)周波数偏差を選択して、該偏差について所定の副
搬送波の集合の出力の合計を判定するステップと、 (b)選択された他の偏差についてステップ(a)を実
施するステップと、 (c)前記合計と他の合計の少なくとも幾つかとに所定
の関係がある場合は、前記合計のうちの選択された1つ
に関連する周波数偏差を表す信号を供給するステップと
を含む周波数偏差を判定する方法。1. A method for determining a frequency deviation between a set frequency and a target frequency in order to synchronize with a multi-carrier signal, comprising the steps of: (a) selecting a frequency deviation; Determining the sum of the outputs of the set; (b) performing step (a) for the selected other deviation; and (c) there is a predetermined relationship between said sum and at least some of the other sums. Providing a signal representative of a frequency deviation associated with a selected one of said sums.
大のものである請求項1に記載の周波数偏差を判定する
方法。2. The method of claim 1, wherein the selected sum is the largest of the sums.
合計が最大出力Uを無視し、但しUは1以上の整数であ
る請求項2に記載の周波数偏差を判定する方法。3. The method of claim 2, wherein the sum of the outputs determined for each frequency deviation ignores the maximum output U, where U is an integer greater than or equal to one.
数偏差を判定する方法。4. The method according to claim 2, wherein U is 4 or more.
最大であるLの平均AVLの関数であり、但しLは1以
上の整数である請求項2、3、または4に記載の周波数
偏差を判定する方法。5. The method according to claim 2, wherein the predetermined relation is a function of an average AV L of L which is the next largest of the sums, where L is an integer of 1 or more. A method for determining the frequency deviation.
わたって積分される請求項1から請求項5までのいずれ
かに記載の周波数偏差を判定する方法。6. The method according to claim 1, wherein the sum of the outputs is integrated over a plurality of M symbol periods.
差判定方法を停止し、従って数Mは変数である請求項5
に記載の周波数偏差を判定する方法。7. The method according to claim 5, wherein the frequency deviation judging method is stopped when the predetermined relation is satisfied, and the number M is a variable.
A method for determining the frequency deviation described in 1.
7に記載の周波数偏差を判定する方法。8. The method of claim 7, wherein said predetermined relationship is a function of M.
+h/√(LM)]とを比較することによって判定さ
れ、但しhは所定の定数である請求項5に従属する場合
の請求項8に記載の方法。9. The method according to claim 1, wherein the predetermined relationship is the maximum sum, and AV L [1]
+ H / √ (LM)], where h is a predetermined constant, when dependent on claim 5.
法において、請求項1から請求項9までのいずれかに記
載の周波数偏差を判定する方法によって判定された周波
数偏差を利用して粗同調を行う受像機を多重搬送波信号
に同調する方法。10. A method for tuning a receiver to a multi-carrier signal, wherein coarse tuning is performed by using the frequency deviation determined by the method for determining a frequency deviation according to any one of claims 1 to 9. A method of tuning a receiver to a multi-carrier signal.
と所定の関係にするために微同調動作を行った後に粗同
調が行われ、次に同調周波数が整合する副搬送波周波数
を変更するために粗同調が行われる請求項10に記載の
受像機を多重搬送波信号に同調する方法。11. A coarse tuning is performed after a fine tuning operation is performed to make the tuning frequency substantially a predetermined relationship with the subcarrier frequency, and then a subcarrier frequency at which the tuning frequency matches is changed. A method for tuning a receiver to a multi-carrier signal according to claim 10, wherein coarse tuning is performed.
項11に記載の受像機を多重搬送波信号に同調する方
法。12. The method of tuning a receiver to a multi-carrier signal according to claim 11, wherein the fine tuning operation is performed in a time domain.
に記載の同調動作を行うように動作可能な同調制御を有
する多重搬送波放送信号受像機。13. A multicarrier broadcast signal receiver having a tuning control operable to perform the tuning operation according to claim 10.
のに適した請求項13に記載の多重搬送波放送信号受像
機。14. The multi-carrier broadcast signal receiver according to claim 13, which is suitable for receiving a digital video broadcast signal.
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
GB9823812.4 | 1998-10-30 | ||
GBGB9823812.4A GB9823812D0 (en) | 1998-10-30 | 1998-10-30 | Multicarrier communications systems |
Publications (2)
Publication Number | Publication Date |
---|---|
JP2000151554A true JP2000151554A (en) | 2000-05-30 |
JP3274113B2 JP3274113B2 (en) | 2002-04-15 |
Family
ID=10841593
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
JP29667399A Expired - Lifetime JP3274113B2 (en) | 1998-10-30 | 1999-10-19 | Method for determining frequency deviation, method for tuning receiver to multi-carrier signal, and multi-carrier broadcast signal receiver |
Country Status (3)
Country | Link |
---|---|
US (1) | US6647066B1 (en) |
JP (1) | JP3274113B2 (en) |
GB (2) | GB9823812D0 (en) |
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- 1998-10-30 GB GBGB9823812.4A patent/GB9823812D0/en not_active Ceased
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1999
- 1999-10-19 JP JP29667399A patent/JP3274113B2/en not_active Expired - Lifetime
- 1999-10-26 GB GB9925326A patent/GB2343311B/en not_active Expired - Lifetime
- 1999-10-28 US US09/429,309 patent/US6647066B1/en not_active Expired - Lifetime
Cited By (7)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
KR101007770B1 (en) | 2002-08-26 | 2011-01-14 | 콸콤 인코포레이티드 | Beacon signaling in a wireless system |
US7995527B2 (en) | 2002-08-26 | 2011-08-09 | Qualcomm Incorporated | Multiple access wireless communications system using a multi-sector configuration |
CN101095293B (en) * | 2004-10-14 | 2012-07-18 | 高通股份有限公司 | Enhanced beacon signaling method and apparatus |
WO2007142194A1 (en) * | 2006-06-09 | 2007-12-13 | Nec Corporation | Communication system, transmission device, reception device, and synchronization method |
US8489133B2 (en) | 2006-06-09 | 2013-07-16 | Nec Corporation | Communication system, transmission device, reception device, and synchronization method |
US8351405B2 (en) | 2006-07-14 | 2013-01-08 | Qualcomm Incorporated | Method and apparatus for signaling beacons in a communication system |
JP2008167115A (en) * | 2006-12-28 | 2008-07-17 | Sony Corp | Detector, detection method, and receiver |
Also Published As
Publication number | Publication date |
---|---|
GB2343311B (en) | 2003-09-24 |
US6647066B1 (en) | 2003-11-11 |
JP3274113B2 (en) | 2002-04-15 |
GB2343311A8 (en) | 2000-07-27 |
GB2343311A (en) | 2000-05-03 |
GB9823812D0 (en) | 1998-12-23 |
GB9925326D0 (en) | 1999-12-29 |
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