JP2000151554A - Method for deciding frequency deviation, method for tuning image receiver to multiplex carrier wave signal and multiplex carrier wave broadcasting signal image receiver - Google Patents

Abstract

PROBLEM TO BE SOLVED: To synchronize an image receiver with a multiplex carrier wave signal by deciding a sum of set output of a specified sub-carrier wave regarding a selected frequency deviation and deciding a frequency deviation between a set frequency and a target frequency when there is a predetermined relationship between this sum and at least some of other sums. SOLUTION: A decision block decides a decision threshold by using information an SSR and a signal frequency counter given at a block 400. At a block 402, the decision block compares a value of an Smax with a calculated threshold TH and, when the Smax exceeds the threshold TH, a block 404 uses a shift value as evaluation of a rough frequency deviation measured by a multiple of a sub-carrier wave interval. Here, when the threshold value TH is not exceeded, obtained information is insufficient to decide the reliable evaluation of the deviation and processes additional observation obtained from a next sign frequency. A successive decision procedure such as this processing finishes when the Smax exceeds the threshold TH.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to the field of frequency synchronization in a multi-carrier communication system, for example, using a pilot sub-carrier with enhanced power for synchronization purposes. The present invention is particularly applicable to OFDM (Orthogonal Frequency Diode) such as a digital video broadcast receiver.
vision Multiplex) Receiver tuning for signals
It is not limited to this.

[0002]

BACKGROUND OF THE INVENTION Multi-carrier communication systems utilize a number of equally spaced sub-carriers for data transmission and other auxiliary functions. In order for the demodulation process to work properly, receivers developed for such systems must be frequency locked with very little residual frequency deviation.
To facilitate frequency locking, some classes of multi-carrier communication systems utilize a set of pilots transmitted on selected sub-carriers having enhanced power. These selected sub-carriers need to form a pilot insertion pattern with optimal autocorrelation characterized by low side lobes.

[0003]

FIG. 1 shows an example of the output associated with the DFT coefficients reconstructed by the receiver in an ideal case without noise and interference and without distortion in the communication channel. However, in practical applications, the received signal is corrupted by broadband noise as well as strong narrowband noise generated at certain frequencies by various sources of interference. Furthermore, if the transfer function of the communication channel has not yet been modified, both amplitude and phase distortions will be induced in the channel itself. FIG. 2 shows the distortion resulting from the combined effects of frequency selective fading, noise and strong interference appearing at certain subcarrier frequencies. As can be seen, in this case, the distinction between pilot and data subcarriers is much more difficult, resulting in degraded performance of any system used for frequency locking.

FIG. 3 shows two components, a fractional part ξ △ f
c and the total frequency deviation Δf having the coarse frequency deviation JΔfc
Where J is 2 in the illustrated example and Δf
c is the subcarrier separation. In a multi-carrier communication system, several time-domain methods are available for evaluating and correcting for fractional frequency deviation. However, the majority of proposed methods for estimating the coarse frequency deviation are based on coherent processing, which makes use of the known phase relationship between the signals to be identified. Such a method is not very suitable for starting the frequency acquisition process if the channel transfer function has not yet been modified.

[0005]

Summary of the Invention Aspects of the present invention are set forth in the appended claims.

[0006] The preferred embodiment of the present invention provides for the proper processing of received signals which can be severely distorted by unmodified communication channels and impaired by wideband noise and strong narrowband interference, to reduce coarse frequency deviation. Evaluation becomes possible.

In accordance with a preferred embodiment, the signal processor determines the frequency deviation for tuning purposes by calculating the sum of the outputs of a set of subcarriers for each of a plurality of frequency deviation candidates. is there. If certain conditions are met, the frequency deviation associated with the largest of the above sums is used to adjust the tuning.

[0008] Preferably, for each frequency deviation candidate, the sum of the outputs excludes U, which is the largest of the subcarrier outputs. Here, U is an integer of 1 or more. This avoids errors in frequency deviation selection as a result of strong interference signals.

Preferably, the frequency deviation is selected for tuning purposes only if the sum of the associated outputs is in a predetermined relationship with at least some of the sums of the outputs associated with other frequency deviations. Preferably, the largest sum must be in a predetermined relationship with the average of the next largest L output sum, where L is an integer greater than or equal to one.

Preferably, the sum of the outputs associated with each frequency deviation is integrated over a plurality of symbol periods to more reliably determine the target frequency deviation. Preferably, the operation is stopped when a predetermined criterion is met, the criterion indicating with a high degree of reliability that an appropriate frequency deviation has been determined. This means
This means that the number of symbol periods required for reliable frequency deviation evaluation is not predetermined, and thus the lock-in period is reduced as much as possible.

[0011] To allow for a variable observation period, the criterion for establishing that the target frequency deviation has been determined is a function of the number of signal periods utilized for observation.

Thus, the present invention provides a signal processing apparatus utilizing a sequential decision procedure that minimizes the time required to determine a reliable estimate of a coarse frequency deviation.

[0013]

DESCRIPTION OF THE PREFERRED EMBODIMENTS Embodiment 1 Next, embodiments of the present invention will be described with reference to the accompanying drawings.

FIG. 4 is a block diagram of a digital broadcast receiver according to an embodiment of the present invention in the form of a digital broadcast receiver.
This will be described with reference to FIG.

The conventional part of this digital broadcast receiver is an antenna 1 for receiving an OFDM multi-carrier broadcast signal.
And a radio frequency amplifier 2 for amplifying a received broadcast signal
A mixer 3 for down-converting the amplified signal to an intermediate frequency signal, an intermediate frequency amplifier 4 for amplifying the intermediate frequency signal, and an in-phase (I) demodulating the amplified intermediate frequency signal.
And a quadrature demodulator 5 for generating quadrature (Q) baseband signals, and an analog-to-digital converter (ADC) for converting these baseband signals into complex-valued digital signals
6, a fast Fourier transform (FFT) processor 7 for performing a discrete Fourier transform on the digital signal to obtain sub-symbol data for each sub-carrier, and an error correction processor for detecting and correcting errors in the sub-symbol data 8
And an output terminal 9 to which sub-symbol data is supplied, and a voltage controlled oscillator (VOC) 10. Voltage controlled oscillator 10
Is used as a local oscillator that supplies a signal tuned to a frequency different from the OFDM broadcast frequency by a predetermined amount to the mixer 3. This oscillator 10 has an FFT processor 7
Receive a control signal from the tuning controller 101 coupled to the output of the tuning controller 101.

The fractional frequency deviation has already been corrected using one of the time domain techniques known to those skilled in the art available using a frequency fine adjustment circuit 102 coupled to receive the output of the ADC 6. Assume that Similarly, since the timing of the symbols has been evaluated, it is assumed that the received signal can be efficiently sampled with minimal timing errors.

The coarse frequency adjustment circuit 103 is used to ensure that each sub-carrier is at an appropriate position.

Suppose there are P pilots transmitted on subcarriers having indices K ₁ , K ₂ ,... K _P.
These ordered index sets will be referred to as pilot insertion patterns. Further assume that the range of possible coarse frequency deviations measured within a plurality of sub-carrier intervals is as follows.

[0019] _{(- J min, -J min +1} , ..., - 1,0
, 1,. . . J _max -1, J _max )

Therefore, the total number of possible deviations is (J _min
+ J _max +1).

FIG. 5 is a block diagram of the coarse frequency adjustment circuit 103 for evaluating noncoherence of the coarse frequency deviation. FFT with discrete Fourier transform (DFT) for demodulation purposes
The output of the processor 7 is an output selector and multiplexer (OMS) 2
Connected to 0 input. The OMS has (J _min + J _max +1) output channels, each channel providing a respective P composite DFT coefficient corresponding to a respective frequency shifted version of the pilot insertion pattern. These coefficients are a (J _min + J _max +1) trimming and sum unit (TSU) for processing the DFT coefficients.
ming unit 22 and these coefficients are frequency shifted by the original pilot insertion pattern and any possible coarse frequency deviations (J _min + J _max ).
It represents a version.

FIG. 6 is a functional block diagram of each TSU 22.
Shown in The TSU performs the following operation.

First, at block 222, q = x ² + y ² is calculated using each composite DFT coefficient, where x and y are the real and imaginary parts of the DFT coefficient, respectively. All output values are added by accumulator 2 at block 224. Further, these output values are classified at block 226. The maximum value U (where U is 1, or preferably an integer greater than 1) is summed by accumulator 1 at block 228. Then at 230, the result is 2
It is subtracted from the sum calculated at 24. In this way, a "trimmed" sum S is formed by ignoring the maximum value q of U and adding the remaining (PU) values. The purpose of the trimming is to increase the statistical distance between the distributions representing signals of different grades so that they can be easily identified. The effect is to avoid the false assumption that the correct frequency deviation has been found by the detection of large powers, in which case these large powers are actually a result, for example, of a strong interference signal . The value of U can be chosen empirically.

Since the TSU operates in parallel, (J _min +
J _max +1) is the sum of the classification and storage registers (SSR: Sorting and Storin) shown at 24 in FIG.
g Register).

FIG. 7 is a functional block diagram of the SSR 24. The operations performed by the SSR can be summarized as follows. At block 300, classify the trimmed sum S from the TSU, at block 302, select the largest of the trimmed sums S _max- at block 304, corresponding to the largest trimmed sum. the value of the deviation, i.e., stores the J *, - in block 306, "then" for trimmed sum of L is the maximum for the average value calculated AV _L, - in FIG. 5 three values of the Decision block (DB: 26)
Decision Block).

FIG. 8 shows a block diagram of the decision block (DB) 26. The decision block is SS at block 400
R24 and symbol period counter (SPC: Symbol Period)
Using the information provided by (d Counter) 28 (FIG. 5), a decision threshold is determined according to the following formula:

TH = AV _L [1 + h / √ (LM)]

Where M is the number of symbol periods used for observation and h is preferably a constant greater than 0.6. The value of h is a good compromise between reducing the chance of getting an incorrect evaluation (which leads to false locks) and reducing the total observation time (time to lock) required to make a decision. It may be selected empirically to achieve this.

Next, at block 402, the DB compares the value of _Smax with the calculated threshold TH. If S _max exceeds the threshold TH, block 404 uses the deviation value J * as an estimate of the coarse frequency deviation measured at multiples of the subcarrier spacing. However, if the threshold TH is not exceeded, the information obtained is not sufficient to determine a reliable estimate of the deviation, and additional observations obtained from the next symbol period have to be processed. Such a sequential decision procedure is terminated when S _max exceeds the threshold TH or is aborted when the total number M of observed symbol periods reaches a predetermined maximum value M _max . Therefore, in an extreme case, particularly when the value of M _max is small and the spectrum distortion is severe, the procedure can be sequentially terminated without obtaining the evaluation of the frequency shift. By the way, it is preferred that the device take into account all successive symbol periods, but this is not essential, for example, spare periods may be ignored.

The operation of the blocks and units described above
The control and timing unit (CTU: Cont) of FIG. 5 that resets all accumulators, registers and counters
(Roland Timing Unit) 30. CTU
Further determines when to terminate or discontinue the sequential determination procedure.

A symbol period counter (SPC) 28 determines the number M of symbol periods processed from the start of the sequential evaluation procedure to the current stage. This information is used by the decision block (DB) to determine an adaptive decision threshold TH as described above. The current value of M is also used by the CTU 30 to determine whether to abort the sequential procedure without evaluating the frequency shift.

In a particular embodiment used with a 2K OFDM signal (ie, a signal with 2000 subcarriers), it is particularly desirable to have J _min = J _max = 20,
Thus there are 41 possible deviations, P = 45, U =
It has been found that 8 or preferably 4 and L = 12, but of course each of these can of course also be modified individually, if desired.

The above embodiment may be realized entirely by hardware, for example, by using an ASIC having appropriately designed logic gates. Or some of the features,
Or all may be performed by one or more appropriately programmed general purpose processor units. If some functions are to be performed by individual processor units, it may be desirable or necessary to perform these functions sequentially, rather than in parallel.

Although the invention has been described in the context of a multi-carrier signal in which the pilot sub-carrier has increased power, the invention can be modified (with a search for a minimum of the calculated power by making appropriate modifications). For use with signals with reduced or no power of the pilot, or with signals whose power varies based on time and / or sub-carrier indicators (by searching for a predetermined distribution of calculated power). I could do it.

[0035]

In a multi-carrier communication system, a frequency deviation is selected and the sum of the outputs of a given set of sub-carriers is determined for this deviation, and a predetermined relationship is established between this sum and at least some of the other sums. In some cases, by determining a frequency deviation between the set frequency and the target frequency by providing a signal representing a frequency deviation associated with a selected one of the sums, degradation due to noise and distortion is reduced. Instead, the receiver can be synchronized with the multi-carrier signal.

[Brief description of the drawings]

FIG. 1 shows a D reconstructed by a receiver in an ideal case with no noise or interference and no distortion in the communication channel.
5 shows an example of an output related to an FT coefficient.

FIG. 2 shows a representative example of the output associated with the DFT coefficients reconstructed by the receiver when the received signal is corrupted by noise and strong interference and there is distortion due to the channel due to frequency selective fading. Show.

FIG. 3 is a diagram showing two components of a total frequency deviation including a fractional part and a coarse frequency deviation.

FIG. 4 is a block diagram of a receiver according to the present invention.

FIG. 5 is a block diagram of an apparatus (frequency coarse adjustment circuit) used in the receiver for noncoherence evaluation of the coarse frequency deviation.

FIG. 6 is a block diagram of a trimming and adding unit (TSU) of the device (frequency coarse adjustment circuit).

FIG. 7 is a block diagram of a classification and storage register (SSR) of the device (coarse frequency adjustment circuit).

FIG. 8 is a block diagram of a decision block (DB) of the apparatus (coarse frequency adjustment circuit).

[Explanation of symbols]

1 antenna, 2 radio frequency amplifier, 3 mixer, 4
Intermediate frequency amplifier, 5 quadrature demodulator, 6 analog
Digital Converter (ADC), 7 Fast Fourier Transform Processor, 8 Error Correction Processor, 9 Output Terminal, 1
0 voltage controlled oscillator (VCO), 20 output selector and multiplexer (OSM), 22 trimming and adding unit (TSU), 24 classification and storage register (SSR), 26 decision block (DB), 28 symbol period counter (SPC), 30 control and timing unit (CTU), 101 tuning controller,
102 Frequency fine adjustment circuit, 103 Frequency coarse adjustment circuit.

Claims (14)

[Claims] 1. A method for determining a frequency deviation between a set frequency and a target frequency in order to synchronize with a multi-carrier signal, comprising the steps of: (a) selecting a frequency deviation; Determining the sum of the outputs of the set; (b) performing step (a) for the selected other deviation; and (c) there is a predetermined relationship between said sum and at least some of the other sums. Providing a signal representative of a frequency deviation associated with a selected one of said sums. 2. The method of claim 1, wherein the selected sum is the largest of the sums. 3. The method of claim 2, wherein the sum of the outputs determined for each frequency deviation ignores the maximum output U, where U is an integer greater than or equal to one. 4. The method according to claim 2, wherein U is 4 or more. 5. The method according to claim 2, wherein the predetermined relation is a function of an average AV _L of L which is the next largest of the sums, where L is an integer of 1 or more. A method for determining the frequency deviation. 6. The method according to claim 1, wherein the sum of the outputs is integrated over a plurality of M symbol periods. 7. The method according to claim 5, wherein the frequency deviation judging method is stopped when the predetermined relation is satisfied, and the number M is a variable.
A method for determining the frequency deviation described in 1. 8. The method of claim 7, wherein said predetermined relationship is a function of M. 9. The method according to claim 1, wherein the predetermined relationship is the maximum sum, and AV _L [1]
+ H / √ (LM)], where h is a predetermined constant, when dependent on claim 5. 10. A method for tuning a receiver to a multi-carrier signal, wherein coarse tuning is performed by using the frequency deviation determined by the method for determining a frequency deviation according to any one of claims 1 to 9. A method of tuning a receiver to a multi-carrier signal. 11. A coarse tuning is performed after a fine tuning operation is performed to make the tuning frequency substantially a predetermined relationship with the subcarrier frequency, and then a subcarrier frequency at which the tuning frequency matches is changed. A method for tuning a receiver to a multi-carrier signal according to claim 10, wherein coarse tuning is performed. 12. The method of tuning a receiver to a multi-carrier signal according to claim 11, wherein the fine tuning operation is performed in a time domain. 13. A multicarrier broadcast signal receiver having a tuning control operable to perform the tuning operation according to claim 10. 14. The multi-carrier broadcast signal receiver according to claim 13, which is suitable for receiving a digital video broadcast signal.