GB2580649A - Improved generator automatic voltage regulator power supply stages - Google Patents

Abstract

An Automatic Voltage Regulator (AVR) for a generator uses a cascode connection Xc of a high voltage, depletion mode JFET or MOSFET and a low voltage, enhancement mode MOSFET, as the main control device. The cascoded connected pair of devices is controlled by comparators Comp1, Comp2 requiring a single positive internal supply. Comparator Comp1 senses when the AC line voltage has dropped to zero by resistor Rs connected to line voltage L, and its output switches to low, pulling its output capacitor to zero volts. Clamp diodes Dc prevent an over voltage being applied to the Comp1 input. The internal supply voltage Vsup is provided by a charge pump/voltage doubler comprising capacitor Ccp and Schottky diodes D3, D4, with storage capacitor Cst to store the Dc value of the doubled AC voltage. The power stage may be half wave (fig. 3a) or full wave (fig. 3b). The generator may be a brushless or slip-ring type.

Description

Improved generator voltage regulator power supply stage Description

Prior Art

There are many types of power output stages employed in voltage regulators for both brushless and directly excited generators. Among them are half wave and full wave types which are shown in Fig. ! a,b,c,d. In fig la the configuration is a half wave thyristor design, lb is a full wave thyristor design, lc is a triac design, ld is a mosfet, transistor or IGBT design.

In cases la,b,c,d initial build up from the residual voltage of the generator can be problematic. The steel used in the laminations of the generators has to have sufficient remanence to provide a certain residual voltage, which needs to be sufficient to provide enough internal supply voltage to bootstrap the generator into full voltage operation. To do this the main power control devices must be turned on during this residual period. The internal power supply must be able to have sufficient voltage and current capability to do this, yet still operate efficiently from the normal operating supply voltage. This is often non-trivial since in the case of thyristors and Triacs, these devices require a trigger current and voltage which maybe in the order of 10's of mA at several volts. Also such devices also have both latching and holding current requirements which have to be met, often leading to addition power dissipating resistances requiring to be added. In the case of a Mosfet power control device the gate threshold voltage also needs to be achieved. Especially when operating at low speed the residual is often very low and it is difficult to obtain sufficient power to bootstrap the system into normal operation. It is also likely that a short circuit on the generator can lose the remanence in the laminations causing the required residual build up voltage to almost disappear. Providing sufficient residual voltage often requires the use less efficient high remanence steel laminations which increase the internal losses of the generator when normal running under load, which in turn reduces the overall efficiency from what it could achieve, or even additional permanent magnets.

Fig.le shows a proposed circuit according to Patent number GB2494715A which uses a depletion mode (normally on) mosfet in various configurations either assisting a separate power device such as a thyristor or as the main control device, but while this will build up from a low residual voltages it has the inherent disadvantage of requiring a negative supply of several volts to be available quickly to ensure the device turns off when required, otherwise the regulator may not control the generator voltage sufficiently fast on build up and potentially severe voltage overshoot will occur and regulator and/or applied loads may be stressed to the point of failure. This negative supply is non-trivial to provide such as those shown in figs. 5a and 5b and often there will still be the requirement for a positive supply for the remainder of the AVR's control circuitry, so duplication of the internal power supplies is often needed at additional expense and complexity.

In addition, due in part to the inherent leakage inductance of the generator's output winding, these thyristor and mosfet devices create substantial waveform distortion and interference on the main output waveform when switching on at high voltage in the normally utilised mode of Phase advance over each half cycle of the generator's output voltage, and to meet EMI specifications will often require complex line filtering to be added, which is both bulky and expensive. The inductive load provided by the excitation filed requires fast, soft recovery diodes as 'normal' standard recovery diodes create a large spike of current, which snaps to zero abruptly on reverse voltage being applied. This creates considerable high frequency interference, but not only that but especially in the case of Thyristors and Triacs the current flowing in the field immediately commutates to the output of the generator, leading to a transient dip in the output voltage due to the leakage inductance of the output windings. Such a dip depends on the leakage inductance of the windings and the magnitude of the current but on directly excited generators can even cause the voltage to drop to zero, causing distortion to far exceed the specification.

One further aspect of thyristor and potentially mosfet switching is the power factor of the load presented to the generator by the phase control action of the switch timing. In general these provide a substantially lagging power factor, which increases the field power demand on the generator.

Proposed art What is proposed in this application is a method to alleviate all of the above problems leading to a substantially more efficient and faster system of both regulator and generator, which builds up from extremely low residual voltages, creates ultra-low interference which requires no additional filtering, and which reduces the excitation demand by drawing a leading power factor load from the generator, helping to excite the generator thus reducing field power demand. Fig.1f is the proposed basic improved version where a cascode connection of a high voltage depletion mode Jfet or Mosfet X7 and a small series low voltage enhancement mode silicon Mosfet X6 are used to remove the necessity for a negative supply such as in Fig.1e yet still allow build-up from an extremely low residual voltage. This combination of devices is hereafter designated as one component 'Xc'. This requires the use of a cascode connection of a low power, low on resistance, low gate threshold silicon mosfet (a recent innovation) connected in the source of a power depletion mode mosfet or junction field effect transistor. Such low voltage mosfets may have an on-state resistance of less than a milli-ohm so contribute little to additional I2R loss.

Fig 2a and 2b show the minimum EMI and distortion filtering required by conventional thyristor and mosfet stages to meet conventional specifications. Fig 3a, 3b and 3c show the implementation of half wave and full wave versions for the cascode mode. The cascode connected devices are turned on at zero voltage on the AC supply and turned off part way through the half cycle. This not only produces very low EMI and low distortion but also provides a leading power factor demand on the generator and can save between 10 and 50% of the load required by the lagging power factor equivalent. A comparison table of the performance of the various options with a nominal field current of 7 Amps is shown in Fig 4. This assumes a typical small series resistor Rload and inductor Lload on the generator with a Power factor of 0.8, which tends to maximise the effects of the distortion and interference created by the regulator's power device switching. A resistive only or shunt resistor inductor combination load tends to damp out some of the ringing and reduces the distortion and interference.

In all these figures to be described, the generator is an AC sine wave source V1, with winding leakage inductance Lleak, and load resistance Rload and Lload. The field winding comprises Inductance Lf with series resistance Rf. The flywheel diode which allows current to continue flowing when the control device is switched off is Df. The cascode device is turned on and off by the application of a positive voltage on its gate; it does not require a negative supply. The silicon mosfet can have a threshold gate voltage as low as 0.4V so can be turned on with voltage supplied by a simple capacitive voltage doubler, so that under residual voltage conditions the AC voltage required to turn on this cascode connected power switch is no more that that required with a simple depletion mode power field effect transistor, and in fact is entirely limited by the forward drop of the rectifying diode(s). A protective fast current limit is also shown in figs 6a and 6b. Figs 7a and 7b show a method of enhancing the efficiency of the power stage using mosfets as synchronous rectifiers.

A description of the figures for each circuit follows:-la. This is a half-wave thyristor controlled output stage. The thyristor X1 is phase controlled by a pulse or train of pulses from Vg and Rg to adjust the average voltage and therefore current to the field. It causes the highest distortion and draws a 2" harmonic (DC) component from the supply windings. It cannot easily be peak current limited. It generally requires a fast flywheel diode to reduce high frequency ringing. It has a power factor ranging from 0.1 lagging at low load up to around 0.5 lagging at full load.

lb. This is a full wave thyristor controlled output stage. The thyristors X1,2 are phase controlled by a pulse or pulse train from Vg and Rg to adjust the average voltage and therefore current to the field. It cannot easily be peak current limited. It requires a fast recovery flywheel diode to reduce high frequency ringing. It has a power factor ranging from 0.1 lagging at low load up to around 0.5 lagging at full load.

lc. This is a full wave rectifier bridge controlled by a triac X3. It is basically similar as a full wave thyristor bridge. There is less need for a flywheel diode as the field current will flow through the bridge diodes (which need to be fast), when the triac is off.

ld. This is a full wave bridge controlled by a mosfet/transistor/IGBT; a mosfet only is shown. These are all 'enhancement mode' devices which are normally off until turned on by a positive voltage or current signal. It requires a fast recovery flywheel diode to limit EMI. It can be run synchronous with the line frequency or at a higher frequency, which creates more interference and distortion. The silicon mosfet has a high positive temperature coefficient leading to the need for over-sizing and large heat-sinks. An IGBT can used but it has an initially high forward voltage but without the high temperature coefficient of the mosfet. Conventional high voltage bipolar transistor can be used but these required a substantial base current to be provided at a low voltage otherwise there is substantial dissipation. In general DC controlled devices such as these transistor types can be peak current limited relatively easily, providing protection for the regulator in the event of over current or a short circuit on the regulator's output.

Fig le is essentially similar to Fig ld but instead of an enhancement mode device a depletion mode (normally on)device is used. This requires a negative voltage to be applied to the gate relative to the source to turn it off, otherwise full excitation will be applied to the field with the possibility of over voltage being created from the generator and the potential for damage.

Fig if is an improved circuit whereby the depletion mode device is controlled by a second cascode connected mosfet or transistor such that the combination is actually acts as an enhancement mode pair but with an extremely low threshold voltage which is positive and therefore no negative supply is required to be created.

Figs 2a and 2b show the type of filtering typically required for the half wave, full wave thyristor and mosfet power stage. The mosfet stage requires similar complexity to that of the full wave thyristor stage and so is not shown. The action of switching in the field current flowing into the field causes an abrupt dip in the terminal voltage of the generator due to its leakage inductance. The current builds up in this then drops abruptly causing a transient overshoot or spike. The various stages of the filter are necessary to help supply this peak current yet absorb the transient spike, but also cause ringing at the resonant frequencies of the LC combinations. The inductors must be rated for the peak overload current of the regulator as otherwise they will saturate causing the filter to be ineffective. The relatively large series mode inductor required will be expensive and its parasitic shunt capacitance will allow the very high frequency (Megahertz) components created by the recovery action of even fast recovery diodes to bypass the filter and this is likely to also require a common mode inductor with balanced inductances and small Y-grade safety capacitors to earth to ensure compliance. The compliance levels vary from country to country but in general an average measured level of 66 decibels above 1uV ('dBuV') from 150kHz to 500kHz dropping to 60 dBuV above that are maximum allowed values for all conditions of field current and load type.

Fig 3a shows the improved half wave power stage using a cascode connected transistor pair. Xc is the compound power device comprising a low threshold mosfet in series with the source of a depletion mode mosfet or jfet. A low power open drain/collector comparator Compl senses when the AC line has dropped to zero by a resistor Rs connected to the line voltage L and its output switches low, pulling its output capacitor to zero volts. Clamp diodes Dc (2) prevents an over-voltage being applied to the Comp1 input. The internal supply voltage Vsup is provided by a simple charge pump/voltage doubler comprising Ccp, and Schottky diodes D3,4 with a storage capacitor Cst to store the DC value of the doubled AC voltage. The gate voltage for the lower transistor of the cascode pair Xc is supplied via resistor Rg from Vsup which sufficiently high to turn it on even at minimal residual voltage, the limiting factor being the voltage drop of the series diode Dl. This voltage Vsup can be used to supply the rest of the AVR, and can be limited by means of a simple shunt regulator.

When the line-neutral voltage L-N is applied to the power stage, immediately it starts to rise above zero, the comparator Comp1 output goes from low to a high state and this capacitor Cramp then recharges through the pull-up resistor Rramp. This creates a positive going exponential ramp, 'Ramp' synchronous with the line frequency. The negative going edge at Ramp can be used as an interrupt signal when a digital regulator control system is employed. In an analogue regulator, the ramp voltage is compared in Comp2 with an error signal from the main regulator which controls the timing of the switch off signal to 01, pulling its gate to zero. (In a purely analogue regulator, this exponential ramp is important as it affects system stability. When the generator is at no-load, or unity power factor load the system loop gain is at its maximum and therefore it is more sensitive to changes in excitation. Making the ramp slope steeper at low excitation dramatically helps the system stability, since a small change in error voltage has a much smaller effect in excitation than at higher load.) When C11 switches off the current stored in the leakage inductance of the generator causes a transient overshoot which is largely absorbed by capacitor Csnub. The choice of capacitance determines how large this overshoot will be, but in general the distortion that results is far less than that caused by thyristors or mosfet switching. A small series resistor Rsnub damps the ringing of Csnub with Lleak, reducing the distortion.

The turn on of Xc at zero volts ensures there is virtually no high frequency noise present, only a slow progressive commutation of field current from the flywheel diode to the generator winding as the voltage rises. The rate of switch on and off of Ca by means of gate resistances is likewise slow with the end result being that no additional EMI/distortion filter stage is necessary. A small Miller capacitor Cm can be added to provide a very soft turn on and off. There is no need for fast recovery diodes, and this is a cost saving as not only do such devices cost more than standard recovery parts, but being ion-implanted they have a substantially higher forward voltage leading to larger heat-sink requirements. And build up voltage is improved since at this point 01 is fully on and the only forward drop present between line output and the field is one diode, leading to a potential generator turn on from around 390mV AC. This obviates the need for high remanence lossy steel laminations or magnets and can even allow for some degree of permanently connected load. The power factor of the load applied by the regulator and field to the generator has a power factor ranging from 0.1 leading at low load up to around 0.5 leading at full load.

Fig 3b shows the improved full wave version. D1,2,3,4 comprise a full-wave rectifier bridge, and Xc is the cascode connected control pair of transistors as before. The L-N voltage is sensed by Rs1,2, clamped by protection diodes Dc (4). In this case the Compl comparator is supplemented by a second comparator Comp2 each of which when switching low at zero volts produces a positive-going exponential ramp 'Ramp' on positive and negative half cycles, which are summed together with Rsum at the input of Comp3 and compared with the error signal from the control section of the AVR. The power supply and gate drive is provided by either a single charge pump Ccpl and Schottky diodes D6,7 or may have additional Ccp2 and diodes D8,9 to minimise ripple on storage capacitor Cst. Again this voltage is fed to the Xc control gate by Rg. The operation of the rest of the circuit is the same as for the half wave version except that the voltage available to the field is twice that of the half wave. In this case the minimum build up voltage will be two diode forward drops or around 780mV AC. As in 3a there is no need for fast recovery diodes either in the bridge or flywheel paths. The power factor of the load applied by the regulator and field to the generator has a power factor ranging from 0.1 leading at low load up to around 0.5 leading at full load.

It should be noted that a cascode mode device is not essential for this version to provide the leading power factor, but overall the very low RdsOn and low temperature coefficient of Gallium Nitride mosor j-fet reduces the heating of the Xc device dramatically, saving on heat-sinks size (if even required) and increasing the reliability and lifetime of the regulator.

Fig 3c uses a half wave mode employing 2 out of 3 phases which gives more field voltage while maintaining the very low build up voltage of around 390mV AC. In this case however the power factor varies between the two supply phases as the load changes, although it is still predominately leading. Otherwise it gives the same advantages as the two previous versions in that there is negligible EMI, although it does require an extra absorption capacitor Csnubc together with Rsnubc on the second phase to suppress the commutation transient between phases.

Fig 4 summarises the results from the above circuits. To compare performance it uses a nominal field current of 7Adc and field resistance of 7 ohms for the half wave and 14 ohms for the full wave, and a nominal L-N supply voltage of 240Vac. It can be seen that the proposed circuits vastly improve on the prior art versions in terms of build up voltage, EMI, Distortion, and overall efficiency for both generator and regulator.

Fig 5a and 5b show possible methods of obtaining a negative supply for the control circuitry if a depletion mode device only is used, as in fig. le. It is important that this is quickly available to turn off the 01 device as it starts to build up to avoid any tendency to overshoot the target voltage, should the main control regulator be slow to start. When very low power comparators are employed, the overall current requirement is very low and the use of small depletion mode transistors allows sufficient supply voltage to turn off the 01 device with only around 4 volts AC available on the line. In the half wave mode fig 5a, the negative cycle of the AC line voltage is peak rectified by Dr and stored on Cs then passes through the depletion mode transistor Q3 to charge filter capacitor Cf negatively. The current through Q3 is limited by the source resistance Rs which pulls the gate negative when a certain source current is reached. The voltage across Cf is limited by a zener diode to that required for the comparators. It is obvious that such a power supply arrangement is much more complex and expensive than the simple charge pump/voltage doublers shown is Figs 3a.

In fig5b this same approach using a depletion mode mosfet is used via a peak sensing diode from the positive output of the bridge D1,2,3,4, but in this case only positive voltage is available so an inverting charge pump using a single push-pull comparator configured as an oscillator then converts this to a negative voltage sufficient to power the control comparators. Again it is obvious that such a power supply arrangement is much more complex and expensive than the simple charge pump/voltage doublers shown is Figs 3a,b,c.

Fig 6a and 6b show additional peak current limiting applied to the basic regulator. This is often considered desirable where conditions could lead to severe overloads or even a direct accidental short circuit across the field. This uses am additional comparator Comp3 (or 4 for the full-wave version which senses the positive voltage across Rcl compared against a low reference voltage Vcs which controls the peak current level, and by turning on Qc1 switches off Xc via Comp2 (or 3)within a few hundred nanoseconds to ensure its protection. However in the event of a direct short circuit the voltage across the resistor Rcl is high enough to turn off the depletion mode device and this instantaneously limits the current until the gate of Xc is pulled to zero by Comp2/3. A monostable action in the comparator Comp3 or 4 feedback loop comprising Ct1 and Ct2 and the two diodes Dt1,2 prevents Xc turn on until much later in the half cycle, or until the zero voltage trigger reoccurs. This circuit is the same for both half, full and 3 phase wave designs.

Synchronous rectification.

In many applications rectifier diodes are used in half and full wave bridge configuration. Even standard diodes have a forward voltage Vf from 0.7 to 1.5Volts or more, which increases progressively with current. High speed diodes are even worse. This affects the losses in the regulator or other device using these. By providing a floating supply for each gate drive is possible to use mosfets in synchronous rectification mode in all positions where a diode would normally be used. This lowers the running temperature of the regulator components and therefore increases the reliability and efficiency hugely to the point that often a heat-sink(s) is unnecessary. For example a standard diode may have a Vf of 0.9V at 7 amps causing it to dissipate 9.63 watts. Thus a full wave bridge at this current will dissipate 25.2W. A suitable inexpensive high voltage mosfet might have an RdsOn of 57millohm, leading to a dissipation of just 0.5W in each synchronous rectifier, or 2W for the entire bridge at 7A 180degree conduction. Lower RdsOn devices are available. Using this technique applies to all the rectifier diodes in the propose designs including the flywheel diode which will dissipate 0.15W, and would result in much improved efficiency and much smaller heat-sinks (if any), making the design extremely compact. The cost of the heat sinks usually outweighs the extra cost of the mosfets.

A circuit for generating multiple floating but tracking supplies inexpensively is described in co-authored patent number GB2533965. Other inexpensive less complex methods are available, for example the clock output of the voltage inverter similar to that described in fig 5b is used as the clock for a high frequency multi-rail charge pump supply as shown in Figs 7a and b and the current required is very low. Each capacitor-diode combination creates a low power floating supply of each of the synchronous mosfet gates via a comparator. The mosfet gates of the synchronous rectifiers at line frequencies do not have to be switched fast so require virtually zero power. The only standing current is a few micro amps for the ultra low power comparator driving each gate and so there is the ability to use small charge pumps or very small and inexpensive pulse transformers to create the floating gate supplies. The clock frequency is chosen to be less that the 150kHz lower frequency EMI specifications bandwidth so that there is no appreciable interference added to the very low level already present.

Fig 7a and 7b show two typical configurations of synchronous rectifiers driven by comparators CMP, each supplied by a charge pump from a clock generated from a low voltage supply Vcp as described above. Fig7a Is half wave and 7b is full wave, both with polarity sensed by a small resistance Rps in the source of each synchronous rectifier mosfet Xsr, where a low power push-pull comparator CMP is used to detect when current is flowing in the reverse direction of the mosfet and turns it on, leading to very low loss. The charge pump consists of a small capacitor Ccp and two diodes Dcp, charging a small storage capacitor Cs for each stage. Each of the synchronous rectifier stages is identical, equating to a rectifier diode with extremely low forward drop. There is a drain sense resistor Rds from each drain to the non-inverting input of each comparator, the signal being attenuated by Ra so that should the voltage across the mosfet be positive on the drain the comparator holds the gate low, preventing a short circuit current flow. The signal across Rps is clamped by the Rc/Dc combination to -0.W to ensure the input to the CMP does not exceed a safe level or reverse the phase of the comparator. The losses in the half wave circuit are higher for the same 7A current (0.72W and 1.32W) as the flywheel diode can conduct for much longer during the negative half cycle. In this case a lower RdsOn mosfet will reduce the dissipation to below that where a heat-sink is required. This circuit is especially useful for input bridge rectifier configurations fin many different applications, for example high power ultra-high efficiency convection cooled server supplies.

For the ultimate in efficiency each Xsr can in all cases be replaced by the Xc cascode pair, with a further lowering of the build up voltage when a charge pump/voltage doubler is used to supply the gate voltage to reduce the diode forward voltage drop to extremely low levels. The limiting factor in this configuration is the forward drop of the Schottky diodes used in the charge pump supply section. Whether this is worth the extra complexity is a choice to be made by the user, as the values already provided by the above proposed circuitry will be adequate for most applications.

Speed of response to load changes The co-authored Patent GB2533965 teaches that an extremely fast response to voltage changes occurring due to load may be achieved by using a sequence of a suitable number (in this case 4) of all-pass filters generated by 0A1-4 (with positive and negative supplies p -n), unity gain resistors R1 and phase shift components Rps and Cps applied to an attenuated sensed AC line to neutral voltage via Rsensel,2, each all-pass filter output being rectified by the same number of full-wave rectifiers FW and summed via D1-4, and the remaining small high frequency ripple filtered with a much higher 2/3pole Low pas speed filter to provide a DC voltage term which responds to sensed voltage changes in the order of 1-2 milliseconds. The necessary amount of amplification can be a applied via Xamp and the derivative of the transient summed at the error input of the gate drive comparator.

When a conventional single phase full-wave rectifier and filtering is used, then response time to eliminate the voltage ripple at 100/120Hz can be of the order of 50-100 times slower, especially when computational circuitry to calculate True RMS is used. If this high speed sensing method is employed with this regulator, the derivative term of the above fast sensed DC voltage can be summed resistively (i.e. capacitively coupled by Cder and summed by Rsum1,2 with the error signal into the same input in Comp 2 (1/2 wave) or Comp3 (full-wave) designs to greatly improved the response speed of the system. See Fig 8. Once the derivative term thus established has caused the regulator to respond, the slower computed RMS value is used to correct the regulator to the exact value. In some applications ultra fast response is required and it is not unusual to have to over-specify the size of the generator to achieve this. By employing this fast sensing it may be possible to specify a smaller generator and still meet the response time specifications.

Claims (1)

1. Improved generator voltage regulator power supply stage -Claims 1 / An arrangement of power control circuitry for an automatic voltage regulator for a generator using a normally off cascode connected pair of switching devices controlled by simple comparators requiring only a single positive internal supply which improves on existing designs of generator voltage regulators by reducing the required residual voltage to cause the regulator to build up to its required voltage from very low levels whilst increasing the efficiency and response speed of the regulating function and generator and removing the need for electromagnetic interference and distortion suppression and the need for extra permanent magnets.

   2/ An arrangement of power control circuitry as in claim 1 which draws power from the generator at a leading rather than lagging power factor from the supply winding thus reducing the current required and improving the efficiency of the generator.

   3/ An arrangement of power control circuitry as in claim 1 which reduces the generator waveform distortion caused by the switching action of the power devices in such a regulator thus minimising the circuitry required to achieve this.

   4/ An arrangement of power control circuitry as in claim 1 which reduces the electromagnetic interference caused by the switching action of the power devices to levels well below the statutory requirements without the need for additional suppression circuitry.

   5/ An arrangement of power control circuitry as in claim 1 which incorporates extremely fast current limiting to protect the regulator from short circuit or adjustable over-current on its output.

   6/ An arrangement of power control circuitry as in claim 1 which improves the efficiency of the regulator itself by replacing rectifier diodes by mosfet transistors or cascode connected transistors thus reducing the heat loss of said diodes and thus the requirement for heat-sinking.

   7/ An arrangement of power control circuitry as in claim 1 which can speed up the response time of the generator to load changes thus minimising the size of the generator required.

   8/ An arrangement of power control circuitry as in claim 1 which incorporates a non-linear ramp such that stability of the system is optimised at all generator load conditions from no-load through unity power factor loads and on to full load at zero power factor