GB2433381A

GB2433381A - Resonant circuits

Info

Publication number: GB2433381A
Application number: GB0525624A
Authority: GB
Inventors: Nicholas Patrick Roland Hill
Original assignee: Individual
Current assignee: Individual
Priority date: 2005-12-16
Filing date: 2005-12-16
Publication date: 2007-06-20
Anticipated expiration: 2025-12-16
Also published as: GB2433381B; CN101375497A; CN101375497B; GB0525624D0

Abstract

A controllable resonant circuit e.g. for an RFID or a tag reader, comprises an inductor L1 and capacitors C1, C2, C3. A stimulus square wave generates an oscillation in the resonant circuit. A non-linear element FET1 is able to change the amount of capacitance according to whether it is conducting or not. The switching point of FET1 is controlled by its gate voltage, such that the resonant frequency changes according to the mark/space ratio of the oscillation. The change in resonant frequency also affects the amplitude of the relevant portions of the oscillation, such that the overall amplitude of the oscillation is thus controlled by the mark/space ratio. The tag may be an implanted tag for a cat, and detection of the tag by the reader leads to operation of a cat flap.

Description

I

Resonant Circuits

Field of Invention

Embodiments of the invention relate to resonant circuits; particularly but not exclusively the embodiments relate to resonant circuits in RFID (radio frequency identification) responsive to a wide frequency range.

Background of invention

In an RFID system resonant circuits are generally used in both the reader and the transponder. Their use increases the efficiency of energy transfer between the two circuits, which would otherwise be much lower, severely limiting the range of operation. Optimal read range may be achieved when the reader is stimulated at its resonance frequency, and this also matches the resonant frequency of the transponder.

The task of operating both these units at their resonant frequencies is complicated by the following factors: 1) The LC manufactured components have tolerances on their values, and the resonant frequency will vary between different examples of nominally identical resonant systems.

2) Temperature can change the value of the components, making the resonance frequency drift over time and environment.

3) Metallic or magnetic objects placed in the vicinity of either antenna can change their inductance and therefore change the resonant frequency.

4) The desired resonant frequency can change, for example the regulatory frequency band across national boundaries can change.

The impact of these complications increases with the Q of the two resonances. As the Q increases then the resonance bandwidth drops proportionally and a closer match between the two frequencies is required for efficient power transfer. Such constraints may limit the Q of the two circuits to relatively low levels, which may in turn limit the read range and/or lead to excessive power requirements.

In RFID systems one approach to mitigate some of these complications is to use a tuning circuit. A large number of different tuning variants have been disclosed in the prior art, however they have some common features. In order to tune the resonance frequency then either an electrically tuneable component, such as a varactor or electrically variable inductor, or more commonly a combination of discrete capacitors or inductors are coupled into the resonance. An exemplary reference is US63 17027, where a set of tuning capacitors in binary weighting are selectively coupled into the reader resonance with corresponding variations in the resonance frequency.

The adjustment may be carried out in a tuning cycle separate from the normal read and/or write operations. US63 17027 is an example of such an approach, where a sweep of the tuning capacitance is carried out and the resonance amplitude profile determined; the chosen tuning capacitance achieves the maximum amplitude in the reader antenna.

Alternatively 1JS5491715 discloses a method based on the phase difference between the stimulus and the resonance to determine how far from resonance the reader circuit is, These prior art methods to tune a system to resonance at a desired frequency have some significant drawbacks, which are now outlined.

The tuning circuit can require many components, comprising the set of capacitors and the switches to couple them into the resonance; this adds to the system complexity and cost. If fine control over the frequency is required then the selectable capacitors are required at high accuracy in order to achieve a monotonic setting of the total tuning capacitance. Such constraints can limit the feasible tuning accuracy and resolution.

The tuning is not generally real-time. If it is made real-time, such as the method described in US5491715, then this adds complexity and cost. Real-time control is beneficial to cope with transient dc-tuning effects such as temperature drift and metallic obj ects coming into range.

These prior art tuning methods are generally applied to the reader antenna only. A transponder is required to have its natural resonance close to the radio frequency energising field in order to obtain its power. If significantly dc-tuned, then the transponder would not be able to charge up enough to run a tuning circuit, As such, the prior art tuning methods described are usually unsuitable for a passive transponder.

Because of tight cost and power constraints it is more common that transponders are set up at manufacture with a fixed tuning to their target operating. This fixed tuning step adds to the cost of manufacture and cannot adjust to environmental changes that might affect the transponder resonant frequency.

A further transponder consideration is that it may be advantageous for a single transponder to be able to respond to a range of frequencies. One application example is the use of' the same transponder across borders with different regulatory operating bands. If the transponders may respond to the different frequencies permitted in each region without a re-tuning step then this facilitates international travel, a key requirement in the common RFID application of asset tracking. Prior art methods provide no facility for this.

Summary of the invention

We will describe an LC circuit that is responsive to a range of stimulus frequencies without the requirement for control of a tuning circuit. In embodiments the circuit is able to adjust naturally to external environmental influences without additional control mechanisms. In embodiments the circuit requires only a small number of low tolerance electrical components.

The conventional tuning method outlined in the prior art description uses a range of capacitors that are fully coupled into the resonance i.e. 100% duty cycle. In embodiments an alternative approach is taken where the LC resonance comprises a primary capacitive path and one or more secondary capacitive paths that are coupled into the resonance with a variable duty cycle. Depending on the duty cycle, the resonance matches a different stimulus frequency.

The duty cycle is set by a FET with a source potential that varies with the resonance amplitude. The charge required to turn the FET on or off is supplied by the resonance current in the inductor; it is therefore efficient and does not require an external power source; as such the method may have application both in a reader and also a transponder. If implemented in a transponder then the transponder will be responsive to a range of energising frequencies, potentially allowing operation across locations with different regulatory frequencies.

The duty cycle is determined both by the resonance amplitude and also the FET gate voltage. The resonance amplitude may be conveniently controlled with the FET gate voltage, which through the relative phase of the resonance and stimulus sets the steady state amplitude. The amplitude may be straightforwardly controlled up to the maximum level corresponding to a conventional antenna circuit matched to the stimulus frequency with the same Q. At this maximum level the resonance is in phase with the stimulus.

In high coupling situations a conventional transponder includes a regulator to limit the pickup voltage and avoid damage to the rest of the circuitry. The transponder can pickup a significant proportion of the energising field, which is subsequently dissipated as heat in the regulator. This embodiment however limits the pickup voltage through the relative phase between the pickup voltage and the energising field. The level of pickup is limited in the high coupling case, which can avoid issues with the transponder shading the energising field. This may be beneficial when reading multiple transponders.

In embodiments the reader and/or the transponder to be made tolerant to a degree of detuning due to the environment. For example a metallic element placed close to the antenna will modify its inductance, However, provided the desired operating frequency is still encompassed in the frequency range over which the resonance is responsive then the system will continue to operate normally. This offers an improvement relative to the prior art where such environmental changes are generally only corrected in the reader and require a separate control/measurement process.

Further Aspects of the Embodiments Below is a list of points detailing further aspects of the embodiments.

1) A LC resonant circuit comprising an inductor, a primary capacitive path, and at least one other secondary capacitive paths that are coupled into the resonance with a variable duty cycle. Depending on the duty cycle the circuit matches a different stimulus frequency.

2) Point I where the variable duty cycle is controlled by a FET 3) Point 2 where the FET gate voltage is kept constant and the FET source varies with the amplitude of the resonance, turning the FET on and off.

4) Point 2 where the FET gate voltage controlled with an external voltage source 5) A reader according to any one of the preceding points where the stimulus period is a sub-harmonic of the desired energising field frequency.

6) A reader based on any one of the preceding points where a high stability crystal oscillator sets the operating frequency of the energising field.

7) A transponder based on any one of the preceding points.

8) A reader antenna tuning method where the stimulus frequency is varied and the chosen frequency maximises the energy input into the antenna.

9) A reader antenna tuning method where the stimulus frequency is varied and the chosen frequency minimises the threshold amplitude for transponder modulation.

10) A reader antenna tuning method where the stimulus frequency is varied and the chosen frequency makes the modulation transformed transponder impedance resistive.

1l)A reader according to any one of the preceding points where the tuning frequency to match to the transponder is taken from a low power proximity detector using a chirp decay to measure both the transponder proximity and also its resonant frequency.

12)A RFID reader based on any of the preceding points such that the transponder is responsive to a range of frequencies.

13) A RFID transponder based on any of the preceding points such that the transponder is responsive to a range of frequencies.

14) A transmitter based on any of the preceding points 15) A receiver based on any of the preceding points 16) A near-field communication device based on any of the preceding points 17) An inductive charger based on any of the preceding points 18) A device that is inductively charged based on any of the preceding points 19) An oscillator based on any of the preceding points 20) A voltage converter based on any of the preceding points 21)A capacitive energy/information transfer system based on any of the preceding points 22) An energy harvesting device based on any of the preceding points.

Brief Description of the Drawings

Figure 1 is a schematic of the first embodiment of a reader.

Figure 2A is a plot of the voltage waveforms of the first embodiment of the reader with zero volts on the FET gate. Figure 2B is the corresponding plot of the antenna current.

Figure 3A is a plot of the FET gate voltage reducing to negative potential when the circuit is resonating. Figures 3B and 3C are the corresponding voltage and antenna current waveforms, respectively.

Figure 4 is a schematic of the first embodiment of a reader with added sections to control the FET gate voltage and supply the stimulus signals.

Figure 5 is a schematic of a second embodiment of a reader.

Figure 6 is a schematic of a third embodiment of a transponder.

Figure 7A and 7B are plots of the antenna current and voltage respectively for the transponder of figure 6, excited by an external field.

Figure 8 is a schematic of the third embodiment of a transponder with added circuit elements to build up the amplitude of the resonance through the FET gate voltage.

Figure 9A is a plot of the antenna current for the transponder of figure 8, excited by an external field. Figures 9B and 9C are the corresponding plots of the FET gate voltage and antenna voltage, respectively.

Detailed Description of the Embodiments

The following description of a resonant circuit responsive to a wide frequency range is merely exemplary in nature and is in no way intended to limit the invention or its applications or uses. Those skilled in the art would readily recognise that in addition to the field of RF1D it may equally well be applied in alternative fields benefiting from the properties of such a resonant circuit. These may include the following: I) radio transmitters and receivers, including mobile telephony

2) near field communications

3) inductive charging of devices 4) oscillators 5) voltage converters 6) capacitive energy/information transfer between systems 7) energy harvesting e.g. wirelessly from an electromagnetic source or within a circuit such as a mechanical pickup.

Figure 1 shows the first embodiment of the invention. A high Q antenna comprises 32 turns of 660-strand 46AW0 Litz wire, with overall diameter approximately 20cm.

Around the target operating frequency of 125kHz the antenna has inductance 300tH (Li) and effective series resistance 0.7Q (RI), giving a Q of 340. The antenna is placed in series with the capacitor network Cl, C2, C3 and an n-type FET. The FET gate voltage is controlled by the voltage source Vgate and the system is excited by the voltage source Vstimulus. The operating principle is now described with reference to the current and voltage waveforms plotted in figure 2.

Figure 2 shows the stimulus voltage, which has been set to a 125kHz square wave of amplitude 5V. Also shown are the voltages of the FET drain (also the bottom of the inductor) and the FET source. The FET gate voltage (Vgate) is set to OV. There are two clear states of the circuit as follows: 1) The FET is turned on when the source voltage is below the gate voltage (OV) by at least the 2V threshold voltage (Vth) of the FET. In this state the source-drain capacitor C2 is shorted out and the source and drain voltages are very close to each other.

2) The FET is turned off when the source voltage is greater than -Yth, in which case the source and drain voltages diverge and the source-drain capacitor charges up.

Each of these two states results in a different effective capacitance in series with the inductor. The first state gives a higher capacitance with 2.2nF (Cl) in parallel with lOnF (C3), giving a total of 12.2nF. The second state gives a lower capacitance due to the extra 2. 2riF (C2) in series with lOnF when the FET is off. This gives a total capacitance of 4.OnF.

With the gate voltage fixed (at OV), the oscillation of the source voltage causes a transition between these two states of FET on and FET off. The duty cycle (fraction of the cycle that the FET is on) is controlled by the amplitude of the oscillation. A high level of oscillation gives a near 50% duty cycle, whereas an amplitude less than Vth gives a 0% duty cycle. These two extremes of duty cycle correspond to two extremes of frequency, given by the following equations: f 1 7r(JL. (ci + (c2' + C3))+ .JL. (Cl + C3)) fO% 22rJL. (ci + (c2 + C3)I) For the capacitance values of this example these two frequencies are f5o0 = I 06kHz, f0% = 145kHz. Therefore, depending on the amplitude of oscillation, this circuit is able to respond at any frequency within this range. The frequency chosen in this example leads to an amplitude of oscillation that peaks at approximately +26, which together with the 2V Vth of the FET gives the appropriate duty cycle for 125kHz. The asymmetry of the waveforms is a natural consequence of the change in effective capacitance in series with the inductor.

Figure 2B shows the inductor current, which has a more symmetric shape than the voltage. The phase difference between the current and the stimulus voltage is approximately 90 degrees and the net power taken from the stimulus voltage by the resonance is low. In fact it is the slight different from 90 degrees that leads to some net input of energy that maintains the required amplitude for matching to 125kHz. When the amplitude is not matched to the stimulus frequency (e.g. after a turn on transient, or the stimulus frequency changes, etc) then the relative phase between the stimulus and the current adjusts, resulting in a different power being drained from the stimulus to maintain a new fixed amplitude of resonance. This fixed amplitude of resonance sets the duty cycle to match the new stimulus frequency. As such the circuit naturally adjusts to changes in frequency. At this point it is worth noting that a change in any of the components through temperature or metallic detuning of the inductor would cause a similar response of the circuit, re-tuning to give the correct duty cycle for the stimulus frequency.

The amplitude of the resonance may be further controlled through the gate voltage Vg.

Figure 3A shows the profile of Vg, which is initially set to OV for 5ms to allow turn-on transients to settle. Subsequently it is reduced from OV to -25V over a 5ms window, and a further 5ms at -25V. The effect on the drain voltage and the inductor current is shown in figures 3B and 3C, respectively. The lower value of gate voltage means that a larger amplitude is required in order to turn the FET off, since the source needs to be at least Vth below the gate voltage. In order to set the same duty cycle to match 125kHz the amplitude of the resonance increases. In this manner, the resonance amplitude may be controlled through the constant gate voltage on the FET.

Another consequence of the increased amplitude is a changed relative phase between the resonance and the stimulus waveform. The current and stimulus waveforms are more in-phase, drawing greater power to maintain the higher resonance amplitude. The process of increasing the amplitude through the voltage on the FET gate may continue until the stimulus and resonance are in phase.

Figure 4 shows a circuit with the same resonance circuit as figure 1 but with added control circuitry. The functions of the main blocks are indicated on the figure and are now described: Resonance circuit: This is as described for figure 1.

Negative voltage rail This block takes as its input the resonance voltage from the FETI drain and stores the peak negative voltage on C4 (less one diode drop). This stored negative voltage is subsequently used by the gate voltage control block to set the FETI gate voltage.

Gate voltage control This block has two digital control lines, Vg_ZERO and Vg_DEC. A voltage pulse on Vg_ZERO turns FET2 on and the storage capacitor C5 is connected to ground. A voltage pulse on Vg_DEC gives rise to conduction through TI to the negative voltage rail; the voltage on C5 is made more negative. Using the digital control lines the voltage on C5 may be gradually made more negative or may be zeroed. This is connected to FETI gate and so controls the amplitude of the oscillation of the resonance circuit. The amplitude of the resonance may either be set with a fixed number of pulses on Vg_DEC or alternatively the gate voltage may be reduced until the amplitude, e.g. measured with an ADC, reaches the desired value.

Stimulus pulse generator An n&p type MOSFET pair are used to control the stimulus voltage applied to the resonance circuit. The positive voltage (Vi) is coupled in via a shottkey diode with the result that the transient current into the resonance is taken via ClO to ground. D6 and Vi supply just the end voltage of the stimulus pulse and enough current to keep the resonance at a constant amplitude. This design of stimulus pulse generator is particularly useful if the power supplied to the resonance is required to be monitored.

This may be straightforwardly monitored through a measurement of the current pulses through D6. Alternatively if there is no such requirement then Cl 0 and D6 may be removed, provided the 5V supply has sufficient capacitance and low series resistance to efficiently recycle the transient current generated by the resonance.

Deadband delay generator Lastly, the deadband delay generator takes the digital waveform Vstimulus as its input and generates a deadband delay that avoids shoot-through current in the stimulus FETs.

In summary the circuit shown in figure 4 enables an inductor current to be set over a range of frequencies determined by the inductor LI and the capacitors Cl, C2, C3, where the amplitude is set by the FET gate voltage. Furthermore the circuit will naturally adjust to different frequencies within this range and also changes in component values (provided the stimulus frequency remains within the range capability of the circuit). As such this approach is tolerant to detuning through temperature, metallic environments, and also manufacturing tolerances on component values.

The description above has been in terms of a square wave stimulus waveform.

However, this also may be a reduced duty cycle waveform, provided it can supply sufficient energy to maintain the resonance. An alternative approach is to stimulate the circuit with a waveform at a sub-harmonic of the target frequency. The main advantage of this scheme is that the n&p stimulus FETs are switched less often, leading to reduced losses in the charging/discharging of these components. A more efficient circuit operation may therefore be achieved.

The three capacitor network shown in the embodiments is not the exclusive implementation this invention. In fact the circuit may be simplified by removing C2 altogether. In this case the FET source voltage stays constant when the FET is turned off. However it has generally been found advantageous to include some capacitance for C2, which serves to increase the FET source potential when the FET is off. This acts to turn the FET off further, which minimises leakage through the FET, particularly as the gate potential can vary due to the finite gate voltage storage capacitor C5.

The task of tuning the reader antenna to a conventional transponder resonance may be carried out with one of the following methods: 1) The reader may step or sweep the stimulus frequency and monitor the power drain from the transponder. This may be carried out by monitoring the power drain from VI.

2) The reader may step or sweep the stimulus amplitude and frequency with the aim to determine the threshold amplitude as a function of frequency where the transponder modulation may be picked up. The lowest threshold amplitude will generally be at the transponder resonance frequency.

3) When the transponder modulates with a normal resistive load modulation then the transformed transponder impedance seen by the reader is also a resistive modulation when operating at the transponder resonance. When the operating frequency is different to the transponder resonance a reactive component is present in the transformed impedance. The reader may monitor the transformed impedance and adjust the frequency until a resistive impedance results.

4) The circuit may be periodically stimulated with a pulse, generating a decaying resonant waveform. The amplitude of oscillation is linked to the natural frequency of the oscillation through the duty cycle of the capacitive paths. The result is that the free decay also sweeps over a frequency range i.e. it generates a decaying chirp.

Such a signal may be used to sweep over the potential transponder frequencies.

Waveform differences with respect to a reference trace may be used to indicate the frequency of maximum absorption from the reader. This frequency would correspond to the transponder resonant frequency and the reader may be tuned accordingly. Such a system may be used as a low power method to detect the presence of a transponder and also its resonant frequency. Once detected the reader may read the transponder in the normal way.

2tid Embodiment Figure 5 shows an alternative embodiment of the invention. In addition to the n-type FET shown in figure 1, this embodiment also includes a p-type FET. The state of the n-type FET changes on the negative cycle of the FET drain, and the p-type FET changes on the positive cycle. For low amplitudes both FETs are non-conducting, whereas for high amplitudes both are conducting. The design equations for the frequency limits at 0% duty cycle and 100% duty cycle (duty cycle is the fraction of the cycle that at least one of the FETs is conducting) are as follows: 1100% -4/L.(Ci + C3)+ jL.(ci + C5)) 0% -ffJL.C1 C2 +C3' ))+JL4Cl C4 +cs)j The values shown in figure 5 give a frequency range of f100%=lO4kHz and fo%=l47kHz.

When setting the amplitude of the resonance with the gate voltages Vgatel and Vgate2, this is done in a similar manner to the single FET version of the first embodiment except that these voltages should go in opposite directions. Vgatel is reduced to negative voltages and Vgate2 is increased to positive voltages by the same amount.

One feature of this embodiment is that the effect of the FETs and capacitor networks on the waveform is more symmetric than the single PET version. This can lead to an inductor current that has reduced distortion. If this is a requirement to meet emission regulations then it may justif' the extra complexity of the circuit. In addition, the voltage swing of the FET drain, for a given inductor current, is reduced relative to the first embodiment. This property may allow the use of FETs with a lower specification of the maximum source-drain voltage. As such the cost of the FETs may be reduced andlor their properties improved through reduced turn-on resistance, etc. 3rd embodiment Figure 6 shows an alternative embodiment of an RFID transponder, or other remotely powered device. The transponder inductance is increased relative to the earlier 3OO.tH transmitting antenna example (figure 1). This is to increase the induced voltage in the transponder in response to an oscillating field. The capacitors Cl, C2, C3 are reduced by the same factor resulting in the same frequency range as the first embodiment (f50% - 106kHz, f0% = 145kHz). The Q of the transponder is set by the coil effective series resistance (RI), which has been set to 10 in this example, giving a high Q of approximately 80 at 125kHz. The n-type MOSFET has a low threshold voltage of 0.5 V. The transponder is coupled weakly to a transmitting antenna of inductance 300mH. The coupling constant between the two inductors is 1%. Figure 7A shows the current in the transmitting antenna, which is zero for the first I ins, then increasing linearly to 5OmA by lOms, then remaining constaiit amplitude. This waveform has been chosen to illustrate the operation of the circuit.

Figure 7B shows the corresponding FET drain voltage in the transponder circuit. As the current in the transmitting antenna increases, then so does the current in the transponder antenna and the FET drain voltage increases accordingly. Once the FET drain voltage reaches 0.5V, the threshold voltage of the FET, then the FET begins to conduct over part of each cycle. This allows the resonance to build up in the circuit, matching the duty cycle to the stimulus frequency of 125kHz. There is a marked jump in the FET drain voltage when this matching occurs. At this point the circuit is self-adjusted to the stimulus frequency.

Figure 8 shows a further development of the transponder circuit. Here an additional branch has been added that couples the FET drain to the gate via diode Dl. Once the circuit adjusts to the stimulus frequency and the amplitude of the voltage on the FET drain jumps above 0.5V, the diode DIstarts to conduct over part of the cycle, pulling the gate voltage low. As the gate voltage drops to negative voltage, the amplitude of the resonance increases. This process continues until the FET gate voltage reaches the limit set by reverse current flowing in the 5V zener diode (D2). The resistor R4 is included to tie the gate voltage to zero in the absence of a resonance in the transponder. As such the transponder is ready for exposure to the reading field, The operation of the circuit described above is illustrated by the waveforms in figure 9.

Figure 9A shows the same transmitter antenna current as figure 7. Figure 9B shows the transponder FET gate voltage, decreasing to negative voltage when the circuit has adjusted to the stimulus frequency (Vdrain exceeds the O.5V threshold voltage of the FET). Figure 9C shows the build up in the transponder resonance as the gate voltage drops.

In order for the transponder to adjust to the frequency of the stimulus field and ramp the amplitude of the pickup voltage, it is required that the pickup voltage first exceeds the threshold voltage of the FET. In the above embodiment a low threshold FET has been used such that this may take place at low coupling levels. An alternative approach is to increase the inductance of the transponder, which in turn generates a higher voltage in the transponder although with a higher source impedance. In this manner the specific requirement for low threshold may be reduced. However with a high transponder inductance, the resonance may be required to build up to a higher final voltage in order to transfer the same level of power from the energising field. This increased voltage may be required both for powering of the transponder and also for any modulation that the transponder carries out to communicate with the reader.

A further benefit of the above embodiment occurs when the transponder coupling to the reader is increased. At high coupling levels the conventional approach is to introduce a regulator to limit the pickup voltage and avoid damage to the rest of the circuitry. One drawback of this approach is that the transponder may absorb a significant proportion of the energising field, which is subsequently dissipated as heat in the regulator. This embodiment however limits the pickup voltage through the relative phase between the transponder resonance and the energising field i.e. the actual level of pickup is reduced, not just withstood through the use of a regulator. Through this improved behaviour in the high coupling case, this embodiment can avoid issues such as the transponder shading the energising field. This may be beneficial when reading multiple transponders.

In summary, this embodiment illustrates how the variable duty cycle technique may be applied to a remotely powered device such as an RF]D transponder. The circuit may self-adjust to the stimulus field, provided the stimulus frequency is with in the range of the transponder circuit and also that the amplitude exceeds the threshold voltage of the FET used together with the capacitor network. It has also been shown that the gate voltage may be automatically ramped such that an increased amplitude of resonance builds in the transponder.

The transponder circuit described above has several advantages over the prior as follows: 1. A high Q transponder may be used without the usual complications of precise matching of the stimulus field to the transponder frequency; the transponder adjusts to the stimulus frequency.

2. The system is more tolerant to variations in temperature and metallic environment that can affect the transponder frequency. Provided the stimulus is within the frequency range of the transponder circuit, a degree of variation in the component values will not affect the operation.

3. The system is more tolerant to variations in component values at manufacture. As such careful tuning of the component values may not be required, leading to a potentially lower cost of manufacture. This may also aid alternative manufacturing techniques that may not currently be able to achieve the tolerances required for prior art implementations. Examples of such techniques may include printed electronics and organic semiconductors.

4. The transponder can respond to more than one reader frequency. As such, transport across international borders where the regulatory frequency band changes may be tolerated.

5. The action of the circuit with a fixed gate is similar to a regulator. The gate voltage sets the amplitude at which the duty cycle matches the stimulus frequency. The main change that an increase in coupling constant has is to change the relative phase of the stimulus and transponder response. The amplitude of the transponder resonance stays approximately the same. As such, an additional regulator may not be necessary in the transponder, This also reduces the effect of a closely coupled transponder shading the reader field from additional transponders.

The circuits described in the above embodiments, while not classic LC resonances, do provide many of the usual resonance functions, while also responding to a range of frequencies. In particular, the step up of voltage is achieved through re-cycling transient energy transferred between the inductor and capacitor network.

One feature common to these circuits is that the resonance naturally turns the FET on and off through the variation in the FBI source potential. The charge required to turn the FET on and off is therefore supplied though the resonance inductor. As such the FET is turned on and off efficiently, without the normal losses associated with directly switching the FET gate with an external voltage. This benefits low power operation of the circuit and allows a high Q resonance to be set up (low Ron FET) without excessive switching losses. A further advantage is that the turn on/off is smooth and does not give rise to strong switching transients.

Although the embodiments described above are benefited by the resonance switching the FET on/off, it is also possible to set up an externally switched arrangement. This would require additional power to switch the FET and careful timing control to ensure that the duty cycle of the circuit was correct for the stimulus frequency.

The methods outlined above allow operation of a high Q LC resonance with a very stable external clock source, such as a crystal resonator. As such a field may be generated efficiently that is very constant with time, both in and frequency. Such a task is advantageous when implementing a read function of an RFID transponder, as any noise on the reading waveform may translate to noise in the output waveform.

The 125kHz frequency band chosen for the embodiments is purely by way of example.

The invention should not be limited to frequencies around this band, rather it should extended to include all oscillator frequencies ranging from sub-sonic to microwave frequencies and beyond. More specifically to RFID, all common RFID frequency bands are included, such as 125kHz, 134kHz, 13.56MHz, 869MHz, 915MHz, etc. No doubt many other effective alternatives will occur to the skilled person. It will be understood that the invention is not limited to the described embodiments and encompasses modifications apparent to those skilled in the art lying within the spirit and scope of the claims appended hereto.

Claims

CLAIMS: 1. A controllable electric resonator comprising an inductor

coupled to a first capacitor to form a resonant circuit, the resonator further comprising a controllable element, a second capacitor controllable coupled across said first capacitor by said controllable element, and a control device to control said controllable element such that a total effective capacitance of said first and second capacitor varies over a duty cycle of an oscillatory signal on said resonator.

2. A controllable electronic resonator as claimed in claim 1 wherein said controllable device comprises a transistor, and wherein said control device comprises a bias circuit for said transistor.

3. A controllable electronic resonator as claimed in claim 2 wherein said transistor comprises a MOS transistor.

4. A controllable electronic resonator as claimed in claim 2 or 3 further comprising a power supply circuit to derive a power supply for said bias circuit from said oscillatory signal.

5. A controllable electronic resonator as claimed in claim 2, 3 or 4 wherein said bias circuit is configured to automatically adjust a bias on said transistor to increase an amplitude of said oscillatory signal.

6. A controllable electronic resonator as claimed in any preceding claim further comprising a third capacitor connected across said controllable element.

7. A controllable electronic resonator as claimed in any preceding claim wherein said inductor has a Q of greater than 50, more preferably greater than 100.

8. A controllable electronic resonator as claimed in any preceding claim further comprising a drive system to drive said oscillatory signal on said resonator.

9. A controllable electronic resonator as claimed in claim 8 wherein said drive system includes means for converting a current drawn by said resonator into a pulse having a duration depending on said current.

10. An RFID tag or tag reader including the resonator of any preceding claim.

11. A method of controlling the amplitude of oscillations in a resonant circuit driven by an oscillatory signal, the method comprising: applying a reactive element to said resonant circuit with a variable coupling; varying said coupling over a cycle of said oscillatory signal to control said amplitude of oscillations.

12. Apparatus for controlling the amplitude of oscillations on a resonant circuit driven by an oscillatory signal, the apparatus comprising: means for applying a reactive element to said resonant circuit with a variable coupling; and means for varying said coupling over a cycle of said oscillatory signal to control said amplitude of oscillations.