FI81475C - FREKVENSSTABILISERAT AUTOMATISKT FOERSTAERKNINGSREGLERAT BALLASTSYSTEM. - Google Patents

Description

81 475

Frequency stabilized current limiting system with automatic gain control

Frequency stabilization is automatically controlled by the balancing system

The invention relates mainly to an electronic frequency-stabilized, gain-controlled current limiting system for fluorescence-type gas discharge tubes with a power source for supplying at least one gas discharge tube. The system comprises a frequency control device connected to the power supply for producing a substantially constant oscillation signal having a predetermined frequency, a switching device connected to the frequency control device for providing a pulse current , which generates a voltage between the ends of said gas discharge tube corresponding to said pulsed current provided by said switching device. At the same time, the number of electronic components is minimized in the electronic current limiting system according to the invention.

Electronic current limiting systems for gas discharge tubes, and in particular fluorescent lamps, are known in the art. Electronic current limiting systems for more than one fluorescent lamp are also known.

However, many prior art current limiting systems require the gain of the switching circuits included in them to be set or adjusted, requiring additional components such as potentiometers or additional electronic circuits to make the output transistors substantially equal regardless of the current gain of each transistor. The addition of such components increases the size and cost of such prior art current limiting systems.

In addition, in many prior art current limiting systems operating at a frequency greater than twice the mains frequency, the operating frequency is not constant but varies with the supply voltage or load, or possibly both. In such prior art systems, due to this frequency variability, a flicker phenomenon may occur when one of the plurality of fluorescent lamps is electrically isolated from the overall circuit. In some cases, disconnecting one fluorescent lamp from the circuit may cause the other fluorescent lamp to turn off completely.

In the system according to the invention, the induction device includes an automatic gain control device which keeps the gain of said switching device at a predetermined level. The switching device includes first and second transistors, each having a base element, a collector element and an emitter element, such that the emitter element is connected to said automatic gain control device. The induction device comprises an inverter transformer connected to said switching device and said frequency control device and having a pair of primary windings with taps connected in series with respect to the switching device and the frequency control device, and a pair of secondary windings and a pair of switching capacitors respectively. with a first electrode of one of said gas discharge tubes.

The figure shows an electrical circuit diagram of a frequency-stabilized current limitation system with automatic gain control according to the invention.

The figure shows a frequency stabilized current limiting system 10 with automatic gain control, comprising a power supply 12 for supplying at least one of the two

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3,81475 from gas discharge pipes 40 and 40 '. The gas discharge tubes 40 and 40 'may be conventional fluorescent tubes, each having first and second filaments 42, 44 and 42', 44 '.

In addition, the current limiting system 10 provides a frequency control mechanism that allows frequency stabilization, as described below, which allows the system 10 to operate normally without disturbing flicker in the event that one of the gas discharge tubes 40 and 40 'is electrically separated from the system 10.

The electronic current limiting system 10 shown in the figure, which includes a power supply 12 for supplying at least one of the two gas discharge tubes 40 and 40 ', includes a frequency control circuit 11 connected to the power supply 12 to provide a substantially constant oscillation signal at a predetermined frequency. The current limiting system 10 includes a switch circuit 13 electrically connected to the frequency control circuit 11, by means of which a pulsed current is provided at a substantially constant rate of a constant oscillation signal at a predetermined frequency. An induction circuit 15 is connected to the frequency control circuit 11 and the switching circuit 13, the purpose of which is to generate a predetermined voltage between the terminals of the gas discharge tubes 40 and 40 'corresponding to the pulsed current produced by the switching circuit 13. The induction circuit 15 includes, as will be described in more detail below, an automatic gain control circuit 17 which keeps the gain of the switching circuit 13 at a predetermined level.

The figure shows a power supply 12 which supplies electrical power to a frequency stabilized current limitation system 10 with automatic gain control. In the embodiment shown in the figure, the power supply 12 is a conventional AC voltage source with a standard voltage such as 120, 240 or 270 volts or any suitable AC mains. In general, the power supply 12 may be a 4 81 475 DC power supply connected internally or externally to the system 10 in a known manner, omitting the bridge circuit and filtering elements described below. For the sake of clarity, the power supply 12 is shown below as an AC voltage source in the embodiment described herein.

Power is supplied to the system 10 from a power source 12 via a switch 14, which may be a standard switch, e.g. a single pole shut-off switch.

Power is supplied via supply line 16 (176) to rectifier circuit 18 to provide full wave rectification of AC power supply voltage. The rectifier circuit 18 may be a conventional full wave rectifier bridge known in the art. The full-wave rectifier bridge 18 can be formed by diode elements 20, 22, 24 and 26, which provide the necessary rectification of the AC voltage of the power supply 12.

The operation of the full wave rectifier bridge 18 produces a pulsating DC voltage to the output line 38 which goes to the filter circuit 36. The filter circuit 36 filters the pulsating DC voltage from the rectifier circuit 18 and is electrically connected to the rectifier circuit 18 by the output line 38.

As described in more detail below, the filtering circuit or network 36 equalizes the pulsating DC voltage so that a substantially constant continuous voltage is obtained for the system 10. The rectifier circuit 18 is connected to ground 30, which acts as the return line of the DC supply, while the opposite end of the bridge circuit 18 supplies DC power to the output line 38.

The filter circuit 36 has a choke member 32 connected in series with the rectifier circuit 18 and two filter capacitors 28 and 34 connected to opposite ends of the choke member 32. The first filter capacitor 28 is electrically connected at one end to the output line 38 and the choke member

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5,81475 32 and from the other end to the ground 30, as shown in the figure. As can be seen, the choke member 32 is connected in series with the full-wave rectifier bridge 18 and the power supply line 41. The choke member 32 is further connected at one end to the first filter capacitor 28 and output line 38 and at the other end to the second filter capacitor 34. The second filter capacitor 34 is connected to the choke member 32 and the power supply line 41 and ground 30. The second filter capacitor 34 together with the choke member 32 smooths the full wave 18 produced by a pulsating DC voltage at a frequency of 120 Hz. In addition, this combination keeps the current drawn by the system 10 at an average value without causing a too capacitive or too inductive power factor. An unfavorable capacitive or inductive power factor could occur if high inductance or high capacitance alone was used to filter the pulsating DC voltage.

If the system 10 did not have a choke member 32, the second filter capacitor 34 would take a large current, often referred to as a charging current peak, at the beginning of each cycle when the capacitor 34 begins to charge. When the choke member 32 is used, its inductance stores energy during each cycle that provides current for charging the second filter capacitor 34 so that the average load current seen by the power source 12 is uniform.

In the embodiment shown here, the values of the first filtering capacitor 28 and the choke member 32 are selected so that the combination of the members 28 and 32 is tuned to 360 Hz to form a third harmonic filter for the 120 Hz pulsating DC voltage produced by the full bridge circuit 18. in the system, the first filtering capacitor 28 may be a 250-volt Polari-unopened 2.0 microfarad capacitor. In addition, the choke member 32 may be an inducer of about 97.0 milligrams. The second filter capacitor 34 is a standard commercially available 100.0 6 81475 microfarad, 250 volt capacitor.

The current flowing in the power supply line 41 passes through a bias resistor 52 and a bias capacitor 54 connected in parallel therewith. The bias resistor 52 and the bias capacitor 54 are electrically connected to the center input conductor 60 of the vibration control transformer 43 of the frequency control circuit 11. As can be seen, the vibration control transformer 43 has a primary winding 45 and two secondary windings 45 and 48 connected to the power supply 12 a power supply line 41 is connected. The center input of the secondary winding 47 of the vibration control transformer 43 is connected to the center input line 60. By means of the center input connected to the center input conductor 60, an oscillation signal of the opposite polarity to the center input is provided.

The bias resistor 52 and bias capacitor 54 generate a bias voltage that causes the oscillation to begin when power is initially applied to the system 10. In a particular operating system, the bias resistor 52 may be about 470 x 103 ohms and the bias capacitor 54 may be about 1.0 micro-farad.

The current limiting resistor 56 and the blocking diode 58 are connected in series to the center input line 60. Although not essential to the invention, the current limiting resistor 56 may have a resistance of about 15.0 ohms and a power average of about 0.5 watts. The current limiting resistor 56 and the blocking diode 58 connected in series form a return path to ground for the oscillation signal generated in the second secondary winding 47 of the oscillation control transformer 43 when the oscillation of the system 10 has started. As can be clearly seen, the bias circuit of the secondary winding 47 includes a bias

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7,81475 a resistor 52 and a bias capacitor 54 connected in parallel. This combination is connected in series with the second secondary winding 47 of the oscillation control transformer 43 via the center input conductor 60, and initiates the total oscillation.

The combination described above mainly causes the current limiting system 10 to start operating when the switch 14 is closed. In addition, the current limiting resistor 56 is connected to the center input conductor 60 and in series with the blocking diode 58, providing a return path to the oscillation signal when the oscillation of the system 10 has begun.

The frequency control circuit 11, which includes a vibration control transformer 43, also includes a vibration control capacitor 50 connected in parallel with the first secondary winding 48. The vibration control capacitor 50 may be, for example, about 0.001 microfarads.

The value of the oscillation control capacitor 50 and the inductance formed by the first secondary winding 48 predetermine the frequency of the oscillation. The inductance of the first secondary winding 48 of the vibration control transformer 43 and the vibration control capacitor form a resonant circuit which gives the system a stable oscillation frequency. The inductance of the first secondary winding 48 is determined by the number of turns of the winding and the properties of the core material to which the winding is wound. Since the oscillation frequency is determined by the means 48 and 50, it is obvious that the frequency is independent of the voltage at which the system 10 operates, as well as independent of the current absorbed by the load. Thus, the system 10 operates at a stable frequency determined by the members 48 and 50 and independent of variations that may occur elsewhere in the system 10, in the load, or in the external distribution network.

In a functional embodiment of the invention, the primary winding 45 of the oscillation control transformer may be 7 turns of No. 26 8 81 475 wire so that the central input is a power supply line 41 with 3.5 turns of winding on each side. The second secondary winding 47 of the vibration control transformer may be 5 turns of No. 26 wire so that the center input is a center input conductor 60 having 2.5 turns of winding on each side. The first secondary winding 48 may be 150 turns of No. 28 wire, and the vibration control transformer 43 has successfully operated as a ferrite core transformer that can be used in the impregnation mode to feed the gas discharge tubes 40 and 40 '.

The electronic current limiting system 10 includes a switching circuit 13 having a pair of transistors 70 and 70 'connected to the oscillation control transformer 43 in a feedback manner. In this way, a current signal can be switched in step with the generated oscillation signal. In the second secondary winding 47 of the oscillation control transformer 43, the current coming through the center input 60 is distributed and flows to both the first transistor line 62 and the second transistor line 64. The first and second transistors 70 and 70 'include bases 72, 72', emitters 76, 76 'and collectors 74, 74', respectively. The first and second transistors 70 and 70 'may be of the NPN type.

The current of the first and second transistor lines 62 and 64 goes to the bases 72 and 72 'of the first and second transistors 70 and 70', respectively. The gain of either the first or the second transistor 70 or 70 'is undoubtedly higher than the second, so that it enters the conducting state first. When either the first or second transistor 70 or 70 'enters the conductive state, it keeps the second transistor in the non-conductive state as long as one of the transistors 70 or 70' is in the conductive state.

Assuming, for example, that the second transistor 70 'enters a conductive state, the voltage level of the collector 74' of the second transistor comes close to the voltage of the emitter 76 'of the second transistor with an accuracy of about 1 volt. As shown in the circuit diagram of the figure, the emitter 76 'is electrically connected to the inverter

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9 81475 to the gain control secondary winding 81 of the transformer, which in turn is connected to ground 30. In this way, the base control current path is completed. The emitter 76 of the first transistor 70 is connected to the gain control secondary winding 80 of the inverter transformer, which, like the secondary winding 81, is connected to ground 30.

The induction circuit 15 includes an inverter transformer 78 connected to the switch circuit 13 as described above. In addition, the inverter transformer 78 is connected to the frequency control circuit 11 having two tap-through primary windings 82 and 84 and a plurality of secondary windings 102, 104, 106 and an inverter transformer gain The induction circuit 15 further includes a pair of switching capacitors 86 and 88 connected in series with the respective primary windings 82 and 84 and the gas discharge tubes 40 and 40 '.

Thus, the inverter transformer 78 includes a first inverter transformer primary winding 82 and a second inverter transformer primary winding 84, so that the other end of each primary winding 82 and 84 is connected to opposite ends of the vibration control transformer primary winding 45 by conductors 66 and 68, respectively.

The primary windings 82 and 84 of the inverter transformer 78 are provided with taps so as to form a saving transformer connection. The first inverter transformer primary winding 82 has a transistor tap 90 that electrically connects a portion of the first inverter transformer primary windings 82 to the collector 74 of the first transistor 70. Thus, the primary winding 45 of the vibration control transformer similarly, the opposite end of the primary winding 45 of the oscillation control transformer is connected by a conductor 68 to a first portion 98 of the primary winding 84 10 81 475 of the second inverter transformer, which is then connected to a second transistor output 92 which goes to the second transistor collector 74 '.

In contrast to some prior art systems in which a saturable transformer is controlled by the amount of feedback voltage, the electronic current limiting system 10 of the invention is current controlled. During one half cycle, the collector current of the first transistor 70 is in feedback with the primary winding 45 of the oscillation control transformer as it flows from the power supply line 41 through the second half of the primary winding 45 of the vibration control transformer. through the collector 74 of the first transistor.

Similarly, during the opposite half cycle, the collector current of the second transistor 70 'is in feedback to the primary winding 45 of the vibration control transformer because the collector current passes from the power supply line 41 through the primary winding 92 of the vibration control transformer 98 through the second transistor 70 'to the collector 74'.

The current flowing through the primary winding 45 of the vibration control transformer during each half-cycle generates a magnetic flux which saturates the core of the vibration control transformer 43. The maximum current that can pass through the primary winding 45 is determined by the effective voltage divided by the sum of the impedances of the second half of the primary winding 45, the first portion 94 of the first transformer primary winding 94 or the first portion 98 of the second inverter transformer primary winding 84, and the first or second gain control secondary winding 80, 81.

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11 81475

When the current stops increasing, the direction of the magnetic flux reverses, which determines the oscillation frequency, which is very little dependent on the input voltage. When the magnetic flux disappears after impregnation, the frequency of the voltage induced in the first secondary winding 48 of the vibration control transformer 43 (48) is equal to the square root of the capacitance input of the inductance and the vibration control capacitor 50 of the first secondary winding 48 multiplied by two and g. Since it is a forced oscillation caused by a current impulse flowing in the primary winding 45, the voltage generated in the first secondary winding 48 is lower in a ratio determined by the ratio of the speeds of the second secondary winding 47 and the first secondary winding 48. Although the voltage waveform of the first secondary winding 48 is substantially sinusoidal, the voltage from the second secondary winding 47 to the base control line 62 or 64 is cut due to the diode effect of the base-emitter connection of the first transistor 70 or the second transistor 70 '. Thus, it is substantially constant in amplitude when the transistor is conductive and approximately linear when the transistor is non-conductive. The useful time ratio remains constant within the operating range determined by the mains voltage.

When the second transistor 70 'is guided to the conductive state, current flows from the power supply line 41 from the center of the oscillation control transformer primary winding 45 through half of the winding 45 to the conductor 68. The current then passes through the first portion 98 of the second inverter primary winding 84 to the second transistor . Since the second transistor 70 'is conductive, current flows from the collector 74' (70 ') to the emitter 76' connected to the second gain control secondary winding 81 of the inverter transformer and further to ground 30, which complements the current path through the transistor element 70 '. The current drawn by the second transistor 70 'forces current to flow in a portion of the primary windings 45 of the oscillation control transformer and induces voltages in the secondary windings 47 and 48 of the oscillation control transformer. The voltage induced in the first secondary windings 48 of the oscillation control transformer 12 A voltage having a predetermined phase is induced in the second secondary winding 47 of the oscillation control transformer so that it is controlled from the transistors 70 and 70 'in the non-conductive state to enter the conducting state. The one of the transistors 70 and 70 'which was in the conductive state is connected to the opposite end of the winding 47 and receives such a control that it enters the non-conductive state.

In the exemplary situation, current can now flow from one end of the second secondary winding 47 of the oscillation control transformer through a conductor 62 to a base 72 of the first transistor 70, directing the transistor to a conductive state. The current of the first transistor 70 comes from the power supply line 41 through the center input of the vibration control transformer primary winding 45 through the coil 45 half to the conductor 66 which connects it to the first portion 94 of the first inverter transformer primary winding 82. The current then passes through the first transistor tap 90 to 70 is conductive, current flows from the collector 74 to the emitter 76 and further through the first gain-control secondary winding 80 of the inverter transformer to ground 30.

As shown, this sequence of events is repetitive and thus causes continuous oscillation as long as the power supply 12 is connected to the system 10 via the switch 14.

As is known from classical transistor theory, the emitter current of a transistor is a combination of a base current and a koilector current. In the current limiting system 10, for example, the base current portion of the emitter current of transistor 70 when the transistor is conductive passes from ground 30 to blocking diode 58 and further through current limiting resistor 56 to center conductor 60, vibration control transformer secondary coil 47 half, and conductor 62 to base 72.

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13 81 475 through the emitter 76 of the resistor to the gain control secondary windings 80 of the inverter transformer and then back to ground 30. During the next half cycle, when the second transistor 70 'is conductive, the main current flows from ground 30 through the blocking diode 58, current limiting resistor 56 and center conductor 60 to the secondary winding 47. The current in the winding 47 passes through the conductor 64 to the base 72 'of the second transistor 70' and then through the base-emitter connection 72 ', 76' to the second gain control secondary windings 81 and then to the ground 30 of the inverter transformer. mode of oscillation.

The center input of the secondary winding 47 of the oscillation control transformer is negative with respect to ground 30, but positive with respect to the emitter voltage of either the first or second transistor 70 or 70 '. In general, to ensure proper operation of the oscillator circuit, the transistors used in the prior art systems must either be matched very precisely to each other or the gains of the transistors must be set by external components so as to achieve a gain gain of the transistors. Such methods naturally increase the cost and complexity of these circuits.

The frequency stabilized current limitation system 10 with automatic gain control uniquely provides for gain control without requiring transistors to be matched or manually adjusted with external components, as is common in the art. The current limiting system 10 includes an automatic gain control circuit 17 that includes a pair 80 and 81 of a plurality of secondary windings of the inverter transformer 78. The gain control secondary windings 80 and 81 of the inverter transformer are connected to the emitter elements 76 and 76 'of the first and second transistors 70 and 70', respectively, as shown in the figure. As shown below, the secondary windings 80 and 81 included in the automatic gain control circuit 17 are wound at a predetermined frequency 81475 so as to cause a negative feedback voltage to the emitter elements 76 and 76 'of the first and second transistors 70 and 70'. As collector current flows in the first portion 94 of the primary winding 82, a voltage is induced in the gain control secondary winding 80 of the inverter transformer in a direction such that it causes a negative bias voltage to ground 30 on the emitter 76 of the first transistor 70, thereby producing negative feedback to the first transistor 70. This feedback voltage is proportional to the current flowing through the first portion 94 of the primary winding 82, which is the collector current of the first transistor 70. Similarly, during opposite half-cycles, the collector current of the second transistor 70 'passes through the first portion 98 of the second primary winding 84, providing negative feedback to the second transistor 70'.

Since the collector current of the first and second transistors 70 and 70 'is a function of the base current and gain of the respective transistors 70 and 70', and assuming that the base currents of both transistors 70 and 70 'are substantially equal, the difference in collector currents is proportional to the gain of each transistor 70 and 70' . By arranging a negative feedback proportional to the collector current, the gain of both transistors 70 and 70 'is adjusted to a predetermined value. Since the negative feedback limits the gain of both transistors to a predetermined value that is less than the minimum gain of the transistor specified by the manufacturer, the gain of both transistors is substantially equal to the circuit.

The base current flowing through the current limiting resistor 56 and the center input conductor 60 travels symmetrically along even one km of transistor circuit, so that the currents are practically equal. The apparent gain of both transistors 70 and 70 'is the same and is automatically controlled in the first gain control section of the inverter transformer.

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is 81475 negative feedback in winding 80 and second gain control secondary winding 81.

The mutual polarity of the emitter-feedback voltages between the first inverter transformer control secondary winding 80 or the second inverter transformer control secondary winding 81 and ground 30 and the base control voltages between base 72 or 72 'and ground 30 is negative, but the mutual polarities are negative, but their mutual with respect to the feedback voltage while the transistor 70 or 70 'is in the conductive state. When the transistor is in a non-conducting state, both the carrier voltage and the emitter feedback voltage are positive with respect to ground potential, but the voltage difference between them is such that the base 72 or 72 'is about 2.5 volts negative to the corresponding emitter 76 or 76', resulting in a fast landing time and short charge retention time and thus low power loss in transistors 70 and 70 '. As the DC voltage from the power supply line 41 increases as the AC voltage from the power supply 12 increases, both the carrier voltage and the emitter feedback voltage increase, but the difference between them remains constant, being substantially equal to 0.7 volts for the selected transistor type and output power.

In the illustrated and described embodiment, the inverter transformer 78 is a ferrite type core transformer having an air gap of 0.040 inches (about 1 mm) that increases reluctance to the magnetic circuit of the transformer 78 and prevents saturation of the magnetic substance. In one operating system 10, the primary windings 82 and 84 of the inverter transformer have first portions 94 and 98 having 76.5 turns and second portions having 16 turns, respectively. The gain control secondary windings 80 and 81 each have 3.0 turns. The glow-secondary windings 102, 104 and 106 of the inverter converter 78 may have 2.0 turns each of the 81475.

As described above, the taps of the primary windings 82 and 84 of the inverter-equipped transformer with taps are made so as to form a saving transformer circuit, where the first parts 94 and 98 are the primary windings of this saving transformer circuit and the second parts 96 and 100 are secondary windings. In this connection, the voltages of the primary parts 94 and 98 add up to the voltages of the secondary parts 96 and 100.

A current flows through the primary winding portion 98 of the inverter transformer 78, for example, if the transistor 70 'is conductive, from its collector 74'. When the switching operation takes place, the transistor 70 'enters a non-conductive state, which causes a rapid current change and generates a high voltage in the primary winding section 98, where it is about 240 volts, and in the primary winding section 100, where it is about 80 volts, and these voltages are summed. . This summed voltage then occurs at the second switching capacitor 88. Similarly, a high voltage is generated at the primary winding portion 94, which corresponds to the voltage generated at the primary winding portion 98. A voltage corresponding to the voltage generated at the primary winding portion 100 is generated at the primary winding portion 96, and due to the power transformer connection, the voltages generated at the primary winding portions 94 and 96 add up and act through the first coupling capacitor 86 to the gas discharge tube 40.

The voltage induced in the first primary winding 82 when the first transistor 70 is connected to the non-conductive state is substantially equal but opposite in polarity to the voltage induced in the winding 84 (82) when the second transistor 70 'is connected to the non-conductive state. Thus, an alternating voltage is generated, the frequency of which is predetermined by the frequency control circuit 11. Similarly, the voltage induced in the second primary winding 84 is an alternating voltage of a predetermined frequency of about 180 'in phase shift induced in the first primary winding 82.

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17 81 475 relative to the incoming voltage, since only one of the transistors 70 and 70 'is in each case conductive or non-conductive.

The first and second switching capacitors 86 and 88 are connected to the primary windings 82 and 84 of the inverter transformer 78. The capacitors 86 and 88 are also connected to the first filaments 42 and 42 'of the gas discharge tubes 40 and 40' to discharge the inducible voltage. The annealing windings 102 and 106 are connected in series with the first and second switching capacitors 86 and 88, respectively, to the first winding portions 94 and 98 and the second portions 96 and 100 to discharge the sum of induced voltages into the gas discharge tubes 40 and 40 '.

As will be appreciated, the filament windings 102 and 104 of the inverter transformer 78 heat the filaments 42 and 44 of the gas discharge tube 40. Similarly, the filament windings 104 and 106 of the inverter transformer 78 heat the filaments 44 'and 42' of the gas discharge tube 40 '.

The inducible voltage that causes the discharge in the fluorescent lamps 40 and 40 'causes current to flow from the filaments 42 and 42' to the filaments 44 and 44 ', respectively. The filaments 44 and 44 'are connected to ground 30 via a filament conductor 108. The second filaments 44 and 44 'of the gas discharge tubes 40 and 40' are connected in parallel by conductors 108 and 110.

The filament winding 104 is connected in parallel with the second filaments 44 and 44 'of the gas discharge tubes 40 and 40'. Similarly, the filament windings 102 and 106 are connected in parallel with the first filaments 42 and 42 ', respectively. Thus, the first filaments 42 and 42 'are heated by filament windings 102 and 106, and the second filaments 44 and 44' are each powered by filament windings 104 connected to ground 30 to provide return paths for the inducible discharge current.

Claims (8)

A frequency stabilized gain-controlled ballast system (10) comprising a current source (12) for operating at least one gas discharge tube (40, 40 '), further comprising a frequency control device (11) connected to the current source (12) to provide a substantially constant oscillation signal a predetermined frequency, a switching device (13) connected to the frequency control device (11) to produce a pulsating current in line with said substantially constant oscillation signal at said predetermined frequency, and an induction device (13) connected to the frequency control device (11) between the ends of said gas discharge tube (40, 40 '), a voltage corresponding to the pulsating current produced by means of said switching device (13) is characterized in that the induction device (15) comprises an automatic gain control device (17) for keep the reinforcement of said switching device (13) at a predetermined level, said switching device (13) comprising a first and second transistors (70, 70 ') each comprising a base element (72, 72'), a collector element (74, 74 ') and an emitter element (76, 76 ') so that the emitter element (76, 76') is coupled to said automatic gain control device (17), said induction device (15) comprises an inverter transformer (78) coupled to said switching device (13), and to said frequency control device (11), and comprising a pair of intermediate terminal primary windings (82, 84) connected in series with respect to the switching device (13) and the frequency control device (11), and a pair of secondary windings (80, 81) and a a pair of coupling capacitors (86, 88) each connected in series with one of said primary windings (82, 84), respectively, and with a first electrode of one of said gas discharge tubes (40, 40 '). 22 81475 2. A system according to claim 1, characterized in that said frequency control device (11) comprises an oscillation control transformer (43) comprising a primary winding (45) and a pair of secondary winding (47, 48), such that the primary winding (45) is connected to the power source. (12) and to the induction device (15). 3. System according to claim 2, characterized in that said frequency control device (11) comprises an oscillation control capacitor (50) coupled to the first secondary winding (48) of the oscillation control transformer (43), the capacitor (50) having a predetermined capacitance value for said predetermined frequency of said oscillation signal. 4. A system according to claim 3, characterized in that the oscillation control transformer (43) comprises a second secondary winding (47) with a central outlet connected to the current source (12) and which at its opposite ends is connected to said switching device (13). System according to claim 2, characterized in that the oscillation control transformer (43) has a core of ferrite material. System according to claim 5, characterized in that the oscillation control transformer (43) operates in a saturated state as said gas discharge tube (40, 40 ') is in operation. 7. A system according to claim 1, characterized in that said switching device (13) comprises at least a pair of transistors (70, 70 ') connected to said frequency control device (11) and said induction device (15). li 23 8 1 475 System according to claim 7, characterized in that the frequency control device (11) comprises an oscillation control transformer (43) having a primary winding (45), a first secondary winding (48) and a second secondary winding (47), thus, said second winding ( 47) at a center outlet is connected to the power source (12) and the base elements (72, 72 ') of said first and second transistors are coupled to opposite ends of said second secondary winding (47).