FI69540B - FOERKOPPLINGSANORDNING FOER EN URLADDNINGSLAMPA - Google Patents

Description

1 69540

Discharge lamp ballast

The invention relates to a pre-switching device according to the preamble of claim 1.

The object of the invention is to protect the rectifier against overload. The solution to the problem is given in the characterizing part of claim 1.

In the invention, the oscillation of the inverter is thus interrupted depending on the load, whereby one winding of the transformer supplying the control voltage to the inverter switch is short-circuited. This additional winding preferably also supplies a current-dependent voltage to the RC member at the input of the bistable switching device.

15 In this case, however, the bistable switching device may react accidentally and switch off the pre-switching device if the pre-switching device is restarted very quickly after the previous disconnection and fault rectification. If this time interval is too short, there may still be so much charge in the RC member in the input circuit of the bistable switching device-20 that the switching device reacts again. To avoid this, according to the invention, the RC element is connected via a disconnected field diode in parallel with the transistor of the bistable switching device, whereby its charge is discharged when the switching device reacts.

The current-dependent cut-off of the pre-switching device also works in the absence of a lamp, if it is operated according to the invention that the electrode connections of the lamp are bypassed by a capacitor dimensioned so that a sufficiently large current flows in the series resonant circuit.

When the inverter is supplied by means of a DC regulator from a DC voltage source, it is expedient that this DC regulator, preferably directly via the transistor of the bistable switching device, also switches off when the bistable switching device is in the closed state.

One embodiment of the invention relates to a pre-switching device in which a single DC regulator supplies two inverters _ _ - __ i ...__ 2 69540 rectifiers. This involves the selective disconnection of the inverter in question in each case, in which case the other inverter continues to operate and the lamp supplied by it can still be lit.

The invention will be described in more detail by means of an embodiment shown in the figures.

Fig. 1 shows a circuit diagram of a DC regulator and an inverter connected thereto and its monitoring part, and Fig. 2 shows the structure of an inverter switch having a monitoring part.

DC regulator according to Figure 1 includes a storage capacitor C18, which on the other side has a blocking choke-men L1, via the charging through D27 and varauskuristimen L4 and 15 on one side of the R33-connected measuring resistor päätasa-inverter G1 kaksitiekytkennässä by a rectifier fed with an alternating voltage network N and which napoihinsa 1, 4 produces a substantially unbalanced voltage.

The voltage control of the storage capacitor C18 is served by the controller X together with the main transistor V6, through which the charging choke L4 can be connected to the main rectifier and can be charged here. The signal from the other side of the päätransistorin V6 and the storage capacitor mittavastuksessa G1 between the C18 and the other side päätasasuuntaajän R 33 is passed viivytyseiimen 25 (resistor R27 and capacitor 14) through the inlet 7 of the controller X, which controls the päätransis-V6 monitor switch.

The inverter W has two transistors V7, V8 30 in series between the terminals 11, 8 of the storage capacitor C18. In parallel with the switching interval of the transistor V7 there is a series resonant circuit with a choke L7 and a capacitor C23, in series with the primary winding of the oscillation capacitor C22 and the saturation transformer T8, the secondary windings of which are L82, L83, L84. A discharge lamp L is arranged in parallel with the capacitor C23 so that its electrodes are in series with the series resonant circuit.

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The alternating operation of the transistors V7, V8 is served by a circuit known per se, from which only the secondary windings L83, L84 of the saturated transformer T8 are shown: from this the charge of the oscillation capacitor C22 is alternately obtained via C18 and the discharge via V7. The operating frequency of the inverter determined by the impregnation converter is then slightly above the resonant frequency of the series resonant circuit: thus a holiday is created between the switching control of V7 and V8.

10 The decelerated travel of the inverter is ensured by a capacitor C19, which is charged only when the controller X is operating: when the C19 voltage is sufficient, V8 is passed through the trigger diode D34 and then C19 is discharged again via D33 (sufficient energy for cold start).

C19, in turn, is connected via capacitor C20 and diode D32 in parallel with the coupling gap of main transistor V6 and can therefore only prepare for a sufficient value for the oscillation of the inverter when the controller synchronizes properly. For periodic discharge when the main transistor V6 is through-pass, capacitor C20 is connected in parallel with protection capacitor C17 via resistor R40: the energy of C20 therefore works like that of C17 to charge subcapacitors C8 and C9, which supply 25 operating voltages to the electronics.

The saturation transformer T8 has one additional second-winding L82, through which a current-dependent cut-off is provided via a bistable switching device K. The switching device K comprises a comparator 30 V12 and a transistor V10 controlled by it via a diode D16.

In normal mode, the output of comparator 12 - terminal 16 - has a negative potential. The comparator V12 only switches (positive potential at the output) when the reference voltage at its positive input-35 becomes higher than the voltage at its negative input-35. In this case, V17 receives the through current via R60 and puts transistors V2, V3 and, through these, 4 69540 also V5 and the main transistor V6.

At the same time, in normal use, the transistor V10 in the blocked state - the positive potential at terminal 15 - is controlled via resistor R13 by a conductive resistor - zero potential at 15 - and thus the winding L82 of transformer T8 is short-circuited via diode D41. In this case, the transistors V7, V8 in the inverter no longer receive any control voltage and enter the inhibit state. Through the through-transistor V10 and the resistor R41, the capacitor 10 C11 is at the same time discharged to such an extent that no disconnection occurs immediately in the re-commissioning.

The described cut-off is dependent on the voltage in capacitors C11 and C10, which are discharged through R14 and charged via diode D41 and resistor R41 to the voltage produced by the second-winding L82 of the inverter transformer T8, which is proportional to the current through L2 (particularly high when the lamp is non-ignited, the series resonant circuit is not certified).

After switching off, the positive input potential of the comparator V12 (voltage-20 divider R12, D16 and R14) is above the negative input potential (Zener diode D13): re-commissioning is therefore only possible by switching off the mains voltage and performing a new start.

The cut-out operates not only at overload but also when 25 lamps are constantly reluctant to light up. Capacitor C10 and resistor R14 are then dimensioned so that a rapid series of small starting pulses - starting a new lamp - leads to a cut-off as little as a larger number of pulses at larger intervals (starting an old lamp).

30 Finally, switching off is also necessary in the absence of a lamp.

In order to use the same current-dependent cut-off, capacitors C26, C27 are arranged in parallel with the electrodes 11, 12 and are dimensioned so that the current flowing in the absence of the lamp is above the operating limit value. For this purpose, the parasitic resistance of capacitors C26 and C27 at an average operating frequency of about 40 kHz is approximately ten-disc compared to the resistance of a single electrode.

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The controller shown in Figure 2 is constructed as a two-point controller and has a comparator V13, the output of which via a resistor R25 is at a terminal P to which a second positive operating voltage can be connected via a transistor V3.

5 V13 controls a transistor V4, the collector of which is connected to the base in the second transistor V5, through which the control current of the main transistor V6 is then conducted.

The base of V5 is further at the tap of the voltage divider connected between the positive terminal P and the terminal 4 at zero potential and producing a setpoint for the comparator V13; for this purpose, a signal dependent on the rectified mains voltage is applied to the voltage divider resistor R2 via terminal 1 and resistor R1. The hysteresis of the two-point controller is determined by the connection of the collector of the transistor V4 to the tap of this voltage divider. The setpoint of the controller can be reduced by connecting transistor V16 in parallel with resistor R2. In this case, V16 is controlled by comparator V15, which normally has a negative output potential so that V16 closes. However, V15 controls V16 to conduct when the switching device K moves the monitoring section ϋ to the disconnection state. In this case, the negative input of V15 is at zero potential through D52, terminal 15 and transistor V10.

The disconnection of the controller under critical operating conditions 25 is served by transistor V3, through which a positive operating voltage can be connected to the controller (terminal P). This transistor is controlled by another transistor V2, which becomes controlled only when the operating voltage has reached the minimum value required for operation. In this case, the potential in the base of V2 is sufficiently higher than in the emitter, which is determined by the Zener diode D13, then the transistors V2 and V3 become conductive and the supply voltage for the regulator is in P.

V3 remains conductive as long as the monitoring section ϋ is in the normal state, i.e. at the output of the comparator V12, and thus the terminal 15 has a negative potential, and as a result, the transistor V17 is in the inhibit state through the diode D62.

___ - ___ - ΊΓΖ- - 6 69540

Another identical inverter (not shown) may be connected in parallel with the terminals similarly marked in Fig. 1 by means of terminals 11, 13, 14, 8 and 4. The terminals 15 and 16 of the second inverter must be connected to the terminals 15 'of the controller X na-5, respectively. 16 '.

The inputs 15, 15 'belong to the OR gate (R5Q, D51, D52), so the state of V15 is changed and the setpoint of the controller X decreases when one of the inverters is disconnected from its control part.

Inputs 16, 16 ', on the other hand, belong to the OR gate (R60, D61, D62) by transistor V17, which is only then controlled to conduct when both inverters are disconnected. As long as one of the two inverters is operating, the terminal 16, 16 'of the monitoring part has a negative potential of 15, as a result of which the diode D62 resp. D61 is conductive, with V17 in inhibit state.

Claims (9)

A ballast for a discharge lamp (L) which in series with a series resonant circuit (L7, C21) forms 5 load circuits for an inverter, the switches (V7, V8) of which are controlled depending on the voltages of the secondary windings (L83, L84) of the transformer (T8). likewise in the load circuit, characterized in that the transformer (T8) has another secondary winding (L82) parallel to the transistor (V10) of the bista-10 carbon switching device (K), and that this switching device (K) is controlled via an RC element in the inverter load circuit depending on the current present, at which point it enters a closed state at a certain limit value of its input voltage, where its transistor (V10) is in a conductive state. Pre-switching device according to Claim 1, characterized in that the auxiliary winding (L82) of the transformer (T8) supplies the RC element of the bistable switching device (K) via a diode (D41). Pre-switching device according to Claim 2, characterized in that the RC element (C11, R14, C10) in the input circuit of the bistable switching device (K) is parallel to the transistor (V10) of the bistable switching device (K). Pre-switching device according to one of Claims 1 to 3, with a direct current controller (L4, V6, D27, C18, X) between the rectifier and the direct current source G1), characterized in that the direct current controller is in the closed state when the bistable switching device (K) the transistor (V10) is in the conducting state. Pre-switching device according to Claim 4, characterized in that there is only one direct current controller for supplying the two inverters, which is switched off via the AND gate (R60, D61, D62) when the bistable switching devices (K) of each inverter (W) are in the closed state. Pre-switching device according to Claim 5, characterized in that the setting value of the direct current controller (X) is reduced via the OR gate (R50, D51, D52) when the bistable switching device (K) of the second inverter is in the closed state. Pre-switching field device according to one of Claims 1 to 6, characterized in that the discharge lamp (L) has heated electrodes, that the capacitor (C23) of the series resonant circuit (C23, L7) is arranged between the electrodes and that each electrode is bypassed with a capacitor (C26, Q.21) that the bistable switching device (K) reacts even in the absence of the lamp (L). 7 69540 Ballast according to Claim 7, characterized in that the impedance of the capacitors (C26, C27) connected in parallel with the electrodes (11, 12) at the average operating frequency of the inverter corresponds approximately to ten times the value of the resistance of the electrode. 9 69540