ES2305824T3 - METHOD AND DEVICE FOR DEMODULAR BEARING SIGNALS OF ALTERNATE BINARY DISPLACEMENT OF GALILEO (ALTBOC). - Google Patents

Abstract

A method for demodulating alternate binary offset carrier signals includes at least two subcarriers, each having an in-phase and a quadrature component modulated by pseudo-random codes. The quadrature components are modulated by dataless pilot signals. The in-phase components are modulated by data signals. The method includes converting the alternate binary offset carrier signals into an intermediate frequency, band-pass filtering the converted signals and sampling the filtered signals, generating a carrier phase and carrier phase-rotating the sampled signals by the carrier phase, correlating the rotated sampled signals, and generating, for each subcarrier, pseudo-random binary codes and a subcarrier phase, which are used to correlate the rotated sampled signals.

Description

Method and device for demodulating signals carriers of binary alternating displacement of galileo (AltBOC).

Field and background of the invention

In general, the present invention relates to Recipients of Satellite Navigation Coverage Systems Global (GNSS, "Global Navigation Satellite System") and, in specifically, to receivers that operate with carrier signals from Alternate binary displacement of Galileo by satellite (AltBOC).

Global coverage satellite navigation (GNSS) receivers, such as Galileo receivers, determine their overall position based on the signals received from GNSS satellites in orbit. GNSS satellites transmit the signals using at least one carrier, each carrier being modulated by at least one binary pseudorandom code (PRN), which consists of a seemingly random sequence of ones and zeros that are repeated periodically. The ones and zeros of the PRN code are called "code chips" and the transitions in the code ranging from one to zero or from zero to one, which occur in "code chip times", are called "chip transitions of code. " Each GNSS satellite uses a single PRN code and, therefore, a GNSS receiver can associate a received signal to a specific satellite by determining which PRN code is included in the
signal.

The GNSS receiver calculates the difference between the time it takes for a satellite to transmit its signal and the time it takes It takes the receiver to receive it. The receiver then calculates the distance or "pseudo-range" from the satellite in based on the difference of associated times. Using pseudo-ranges of at least four satellites the receiver determines its global position.

To determine the time difference, the GNSS receiver synchronizes a locally generated PRN code with the PRN code of the received signal by aligning the chips of code of each of the codes. The GNSS receiver determines then how much the generated code has moved over time locally from the known synchronization of the PRN code of the satellite at the time of transmission and calculates the associated pseudo-range. How much more accurate is the GNSS receiver alignment of the PRN code generated locally with The PRN code of the received signal, the GNSS receiver will determine more accurately the difference in associated times and the pseudorangol and, with it, its global position.

Code sync operations include satellite PRN code acquisition and tracking  Of code. To acquire the PRN code, usually the receiver GNSS performs a series of separate correlation measures in the Time using a code chip. After the acquisition, the GNSS receiver tracks the received code. Correlation measures are generally taken "advance minus delay ", that is, measures of the difference between (i) a measure correlation associated with the PRN code of the received signal and a initial version of the locally generated PRN code, and (ii) a correlation measure associated with the PRN code of the received signal and a later version of the local PRN code. The GNSS receiver uses then the forward minus delay measurements in a blocking loop of delay (DLL), which produces an error signal that is proportional to misalignment between local PRN codes and received. The error signal is used in turn to control the PRN code generator, which basically shifts the PRN code local to minimize the DLL error signal.

Normally, the GNSS receiver also aligns the satellite carrier with a local carrier using measures of correlation associated with a specific version of the local PRN code. To do this, the receiver uses a phase lock loop of carrier tracking (PLL).

The European Commission and the Space Agency European (ESA) are developing a GNSS known as Galileo. Galileo satellites will transmit two signals in the E5a band (1,176.45 MHz) and two signals in the E5b band (1,207.14 MHz) in form of a composite signal with a center frequency of 1,191,795 MHz and with a bandwidth of at least 70 MHz, using a AltBOC modulation. The AltBOC signal generation is described in the document of the Galileo Signal Task Force of the European Commission, "Status of Galileo Frequency and Signal Design", G.W. Hein, J. Godet, J.L. Issler, J.C. Martin, P. Erhard, R. Lucas-Rodríguez and T. Pratt, 09-25-2002, published in the following email address: http://europa.eu.int/comm/dgs/energy_transport/galielo/documents/technical_en.htm.  Like GPS satellites, each Galileo satellite transmits unique PRN codes and thus a Galileo receiver can associate a signal received to a particular satellite. Therefore, the receiver Galileo determines the corresponding pseudo-ranges based on the difference between the time it takes for the satellite to transmit the signals and the time it takes for the receiver to Receive the AltBOC signals.

A standard linear displacement carrier (LOC) modulates a time domain signal using a wave sinusoidal sin (\ omega_ {0} t), which shifts the frequency of the signal to both an upper side band and a band corresponding lower side. BOC modulation performs the frequency shift using a square wave, or signal (without (\ omega_ {0} t)), and is usually called BOC (f_ {s}, f_ {c}), where f_ {s} is the frequency subcarrier (square wave) and f_ {c} is the bit rate Redundant broadcast Normally, the 1,023 MHz factors are omitted from the notation for reasons of clarity, so that a BOC modulation (15,345 MHz, 10.23 MHz) is called BOC (15.10).

Modulation of a temporal domain signal using an exponential complex e i \ omega_ {t} displaces the signal frequency only to the upper sideband. The The purpose of the AltBOC modulation is to consistently generate the E5a and E5b bands, which are respectively modulated by complex exponentials, or subcarriers, so that the signals can be received as a "BOC type signal" band wide Each E5a and E5b band has broadcast codes associated in-phase (I) and quadrature (Q), or PRN, at E5a codes shifted to the lower sideband and codes E5b displaced to the upper lateral band. The carriers in corresponding E5a and E5b quadrature are modulated by pilot signals without data and corresponding phase carriers They are modulated by PRN codes and data signals.

AltBOC modulation offers the advantage that signals E5a (I and Q) and E5b (I and Q) can be processed in a manner independent in the same way as BPSK signals (10) traditional (Binary Phase-Shift Keying signals) or together, resulting in performance extraordinary when it comes to noise tracking and multipath.

To calculate the demodulation principle of the AltBOC modulation is sufficient to approximate the AltBOC signal of the baseband to its equivalent AltLOC (Alternate Linear Offset Carrier):

one

where:

-

c_ {1} (t) is the PRN code of the data component E5b (E5bl) and d_ {1} (t) is the corresponding bit modulation;

-

c_ {2} (t) is the PRN code of the data component E5a (E5al) and d_ {2} (t) is the corresponding bit modulation;

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c_ {3} (t) is the PRN code of the pilot component E5b (E5bQ);

-

c_ {4} (t) is the PRN code of the pilot component E5a (E5aQ);

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the exponential factors represent subcarrier modulation of E5a and E5b;

-

\ omega_ {s} is the pulse of sideband offset: \ omega_ {s} = 2 \ pif_ {s}, with f_ {s} = 15,345 MHz.

Actually, s (t) contains terms of Additional product and exponential subcarrier are quantified This effect will not be explicitly included in the equations for reasons of clarity. s (t) is modulated in the E5 carrier at 1,191,795 MHz.

Most previous publications present AltBOC from a satellite payload perspective, that is, from the point of view of a transmitter. So far, the receiver side processing has received very little Attention.

The publication "Comparison of AWGN Code Tracking Accuracy for Alternative BOC, Complex-LOC and Complex BOC Modulation Options in Galileo E5-Band ", M. Soellner and Ph, Erhard, GNSS 2003, April 2003, describes the principle of a receiver architecture AltBOC for monitoring the pilot component AltBOC, as shown in Figure 1.

In Figure 1, the AltBOC receiver receives through an antenna 1 a signal that includes composite codes AltBOCs transmitted by all satellites in sight. The received signal is applied in an RF / IF 2 phase that, in a way conventional, transforms the received RF signal into an IF signal of intermediate frequency that has a frequency compatible with the other components of the receiver, filters the IF signal through a IF band pass filter, with a frequency band pass desired central carrier, and samples the filtered IF signal to a speed that satisfies Nyquist's theorem in order to produce the digital phase (I) and quadrature (Q) signal samples corresponding in a known way. Filter bandwidth it must be wide enough to allow the passage of primary harmonic of the AltBOC composite pilot code, or of approximately 51 MHz. The bandwidth amplitude gives as result code chip transitions relatively in the received code and therefore correlation peaks quite well defined.

The AltBOC receiver comprises an oscillator of local carrier 4, for example of the NCO type (Numerically Controlled Oscillator - Numerically Controlled Oscillator), synchronized with the IF frequency to generate a phase rotation angle in M bits, which is applied to a phase 3 rotator that receives the samples of the IF signal in N bits. Signal samples with rotation of N-bit phase supplied by the rotator are applied to three complex correlators, each comprising a multiplier signal 10, 11, 12 and an integrator 13, 14, 15. The integrators sum the signal samples received during a time of predefined T_ {int} integration.

The AltBOC receiver also comprises another local oscillator 5 of the NCO type synchronized with the speed of redundant bit code f_ {c} and that directs a generator of complex AltBOC codes 6 to generate pilot codes locally PRN complexes for a given satellite. The pilot codes generated cross a multi-bit 7 delay line that it comprises three cells E, P, L, which produce, respectively, replicas of start, immediacy and delay of local PRN codes, which apply to an entry of multipliers 10, 11, 12 respectively.

Then, the signals C_ {E}, C_ {p} and C_ {L} sent by integrators 13, 14, 15 are used to generate a carrier phase and code error signals, being used to control the oscillators NCO 4, 5.

The AltBOC 6 code generator features the inconvenient of being complex and multi-bit. This is, produces a quantified version of the baseband signal Alt-LOC (assuming that only the pilot component) as follows:

2

Produce such a complex baseband signal It is uncomfortable. The architecture shown in Figure 1 also implies that all operators (delay line, multipliers and integrators) operate on complex numbers multi-bit

Summary of the invention

An object of the present invention is provide a simplified method and device for Galileo signal demodulation.

This object is achieved by a method to demodulate carrier signals of alternating binary displacement comprising at least two subcarriers, each with a phase component and a quadrature component modulated by pseudo-random codes, where the quadrature components are modulated by pilot signals without data, and the components in phase are modulated by data signals, said said Method the stages of:

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convert the carrier signals of alternate binary shift at an intermediate frequency, filter with a pass filter band the transformed signals and sample the filtered signals,

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generate a carrier phase and submit a carrier phase rotation the signals sampled by said carrier phase, and

-

establish a correlation between Sampled signals rotated.

According to the invention, this method also includes the steps of generating codes for each subcarrier pseudo-random binaries and a sub-carrier phase, which are used to correlate the sampled signals rotated.

According to a preferred embodiment of the invention, the method further comprises a step of translating said pseudo-random codes of said subcarriers at phase angles, which are combined respectively with the subcarrier phases to obtain the resulting phase angles for each subcarrier, said resulting phase angles are offset to obtain at least one initial, immediate phase angle and at least one delay phase angle for each subcarrier, comprising said correlation stage the phases of subjecting a rotation of phase such sampled signals rotated by said angles initial, immediate and delay phase of each subcarrier, in order to obtain start, immediate and delay replicas of said sampled signals for each subcarrier, and integrate respectively the start, immediate and delay replicas for each subcarrier for a predefined period of time.

According to a preferred embodiment of the invention, the method further comprises a stage of phase rotation of said displayed signals rotated by the phases of subcarrier, in order to obtain sampled phase rotated signals for each subcarrier before establishing a correlation between said sampled signals rotated.

According to a preferred embodiment of the invention, the method further comprises a step of displacement of  bits of said pseudo-random codes to obtain at least one pseudo-random start, immediate and delay code, said correlation stage comprising the phases of combining the sampled phase rotated signals for each subcarrier with said pseudo-random start, immediate and delay codes, and integrate the resulting signals over a period of time predefined, in order to obtain start correlation signals, immediate and delay for each subcarrier, also comprising said method a low post correlation stage speed comprising the stages of phase rotation the signals of start and delay correlation of each subcarrier, respectively by opposite constant phase angles, and add the start correlation signals respectively as well obtained from said subcarriers, the correlation signals immediate of said subcarriers and the correlation signals of delay obtained from said subcarriers, in order to obtain Start, immediate and delay correlation signals resulting respectively.

According to a preferred embodiment of the invention, the method also comprises a step of determining a combined carrier and subcarrier frequency for each subcarrier, combining the stages of phase rotation by said carrier phase and said subcarrier phases in a single rotation stage for each subcarrier, using said carrier and subcarrier frequencies.

According to a preferred embodiment of the invention, said correlation step comprises the phases of combine said sampled phase rotated signals for each subcarrier, respectively, with the pseudo-random codes of such subcarriers and to integrate over a period of time predetermined the resulting signals, in order to obtain a signal correlation for each subcarrier.

According to a preferred embodiment of the invention, the method further comprises a step of low speed post correlation comprising the phases of combining the correlation signals for said subcarriers, in order to obtain an immediate correlation signal used as an input of a PLL discrimination unit that conditions an oscillator that controls said rotation stage of the carrier and a minus delay start correlation signal used as an input of a DLL discrimination unit that conditions an oscillator that controls said code generation and said generation of carrier phases.

According to a preferred embodiment of the invention, the start minus delay correlation signal is obtained from the correlation signals for said subcarriers E5a, E5b using the following formula:

3

According to a preferred embodiment of the invention, DLL discrimination is of the type of discrimination of scalar product power and develops the following operation:

4

where Real () is a function that returns the real part of a number complex,

using signal D to direct the oscillator that controls said code generation and said phase generation of subcarrier

According to a preferred embodiment of the invention, DLL discrimination develops the following operation:

5

where Imag () is a function that returns the imaginary part of a number complex.

The invention also relates to a device. to demodulate carrier signals of alternating binary displacement comprising at least two subcarriers each having a phase component and a quadrature component modulated by pseudo-random codes, where the quadrature components are modulated by pilot signals without data and components in phase They are modulated by data signals. According to the invention, this device comprises means to implement the defined method previously.

Brief description of the figures

The invention will be more easily understood and other features and advantages of the invention in the following description and with reference to the figures They accompany each other.

Figure 1: Functional block diagram of a AltBOC demodulator channel corresponding to the state of the technique.

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Figure 2: represents the curve of a function of single component correlation to demodulate each component of an AltBOC signal.

Figure 3: Fresnel diagram of the functions single component correlation E5aQ and E5bQ.

Figure 4: represents the curve of a function of the combined correlation peak that combines the functions of Individual component correlation E5aQ and E5bQ.

Figure 5: Functional block diagram of a AltBOC demodulator channel according to a first embodiment of the present invention

Figure 6: Functional block diagram of a AltBOC demodulator channel according to a second embodiment of the present invention

Figure 7: Functional block diagram of a AltBOC demodulator with two channels as shown in Figure 6.

Figure 8: Functional block diagram of a two channel AltBOC demodulator according to a third embodiment of The present invention.

Figure 9: Fresnel diagram of the functions of individual component correlation E5aQ and E5bQ obtained with the AltBOC demodulator of Figure 8.

Figure 10: Functional block diagram of a first embodiment of a receiver comprising the demodulator AltBOC of Figure 8.

Figure 11: Functional block diagram of a second embodiment of a receiver comprising the demodulator AltBOC of Figure 8.

Figures 12 and 13: block diagrams functionalities of a third and fourth embodiment of a receiver AltBOC derived from the receptor of Figure 11.

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Detailed description of the preferred embodiments

The characteristics are detailed below. main of the invention. According to the demodulation principle AltBOC, the pilot channel is formed by the combination of E5aQ and E5bQ signals. The AltBOC pilot signal is composed of C3 and C4 components:

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6

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where \ omega_ {s} is the sideband shift pulse: \ omega_ {s} = 2 \ pif_ {s}, with f_ {s} = 15,345 MHz

In principle, each component could be demodulated by correlating s_ {p} (t) with the code chip sequence, sequence c_ {i}, multiplied by the complex conjugate of the corresponding subcarrier exponential, for example to make a component tracking c_ {3} (t) the receiver must be correlated with c_ {3} (t) \ cdote ^ {- i (\ omega_ {s} t + \ pi / 2)}. The corresponding correlation function C_ {E5bQ} (\ tau) can be easily calculated (assuming a bandwidth
infinite):

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7

where:

-

he sign "\ propto" means "proportional to";

-

\ propto triangle (\ tau) = 100

-

\ tau is the delay between the input signal and the local code and the subcarrier replicas,

-

T_ {int} is the integration time; Y

-

T_ {c} is the chip length in time units

Signal variations C_ {E5b} (\ tau) as a function of the tracking error of code are shown in Figure 2. Curves 17 and 18 are the real (I) and imaginary (Q) components of this function respectively, while curve 16 is the magnitude. It can note that it is a complex function of \ tau: if the local code and subcarrier replicas are misaligned, energy shifts from branch I to Q. You cannot track such a peak correlation while code misalignments and of carrier are not clearly separated: any misalignment of code results in a tracking error of the carrier phase. Because the carrier loop is usually much faster that of the code, the energy will tend to zero in the Q branch, giving as a result that the code loop describes a peak of pure BPSK correlation.

The additional information needed to apply the BOC principle is the fact that the other sideband is transmits consistently at a frequency distance of exactly 2f_ {s} = \ omega_ {s} / \ pi. The function of correlation C_ {E5aQ} (\ tau) is given by the correlation between sp (t) and c_ {4} (t) \ cdote ^ {i (\ omega_ {s} t- \ pi / 2)}:

8

A Fresnel diagram like the one in Figure 3 provides an intuitive view of complex correlations C_ {E5aQ} (\ tau) and C_ {E5bQ} (\ tau). In this diagram the two correlations are represented as vectors in the plane I, Q. TO as the code delay increases \ tau, C_ {E5bQ} (\ tau) and C_ {E5abQ} (\ tau) rotate with an angle + \ omega_ {s} \ tau and - \ omega_ {s} \ tau respectively and its amplitude decreases from according to the triangle function, leading to the two propellers that They show in the figure.

A peak function of combined correlation by adding the correlations C_ {E5aQ} and C_ {E5bQ}, which corresponds to adding the vectors in Figure 3:

9

As shown in Figure 4, the function C_ {E5Q} (\ tau) corresponding to the peak function of AltBOC correlation is real (curve 36) for all delays of code, the imaginary part (curve 37) is null and therefore You can use to track code.

For the pilot channel, the correlation peak combined E5a / E5b is simply the sum of the individual peaks E5a and E5b. The same can be applied to the data channel principle, but the data bits have to be deleted before the combination: the E5 data correlation peak is given by:

10

This bit estimation process makes the tracking channel be less hard, specifically in the relationship signal-noise (C / N_ {0}), where the probability of Bit error is high.

From this principle five embodiments are derived preference of an AltBOC demodulator according to the invention. With a intelligent partition of the processes of pre and post-correlation, processing can be performed  AltBOC baseband with little expense with respect to signals Traditional BPSK

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The AltBOC demodulators presented Subsequently they are calculated assuming a follow-up of the pilot channel, although the extension of the channel tracking Data be direct.

It has been shown that obtaining the peak of AltBOC correlation implies establishing a correlation between the incoming signal and c_ {3} (t) \ cdote ^ {- i (\ omega_ {s} t + \ pi / 2)} and c_ {4} (t) \ cdote ^ {i (\ omega_ {s} t- \ pi / 2)} and adding These two complex correlations. In the receiver this leads to out on two identical channels, sharing the same local code and carrier oscillators.

As explained above, the demodulation of component c_ {3} implies correlating the signal incoming with c_ {3} (t) \ cdote ^ {- i (\ omega_ {s} t + \ pi / 2)}. This operation is equivalent to rotating the incoming signal an angle - \ omega_ {s} t- \ pi / 2, and then multiply by PRN c_ {3} and integration chips. The multiplication by the chips of code can be seen as an additional rotation of 0 ° when the chip is +1 or 180º when the chip is -1. This observation we leads to the first AltBOC demodulator channel architecture such as depicted in Figure 5.

In Figure 5, the AltBOC demodulator channel receives through a antenna 1 a signal that includes the codes AltBOC compounds transmitted by all satellites that are at the view The received signal is applied in an RF / IF 2 phase that transforms the received RF signal into a frequency signal intermediate IF with a frequency compatible with the others receiver components, filters the IF signal using a filter band pass with a band pass at the center carrier frequency desired and samples the filtered IF signal at a rate that satisfies Nyquist's theorem, in order to produce the signals of corresponding digital phase (I) and quadrature (Q) sampling in N bits in a known way. The bandwidth of the filter is what wide enough to allow the harmonic to pass primary of the AltBOC composite pilot code, or approximately 51 MHz. The bandwidth amplitude results in transitions of relatively conformed code chips in the received code and, therefore, correlation peaks quite well defined.

The AltBOC demodulator comprises an oscillator local 4, for example of the NCO type (Numerically Controlled Oscillator), synchronized with the IF frequency, to generate a phase rotation angle in M bits, which is applied to a rotator phase 3 that receives the IF signal samples in N bits. The rotated phase signal samples supplied by the rotator of Phase 3 are applied in parallel to three phase rotators 25, 26, 27 before being integrated into three corresponding integrators 28, 29, 30, which add their input signal samples during the integration time T_ {int}.

The AltBOC demodulator also comprises another local oscillator 5 of the NCO type synchronized with the redundant bit code rate f_ {c} and which generates the redundant bit code frequency and the subcarrier frequency f_ {s} = 1.5f_ {c} , to control a subcarrier phase generator 20 and a E5b 21 code generator. The output of the E5b 21 code generator is connected to a PRN 22 phase detector. The subcarrier phase generator 20 generates the subcarrier phase in M bits at the speed f_ {s} provided by the NCO code oscillator 5. The E5bQ 21 code generator generates code chips E5bQ (0 or 1) at the speed f_ {c} provided by the NCO 5 code oscillator. The PRN phase detector translates code chips (0 or 1) into a phase rotation angle
0 or \ pi.

The respective generator output signals of subcarrier phase and PRN phase detector are summed in a additive 23, the additive output signal being a signal outdated (real number encoded in M bits) that controls a line 24-bit delay with three cells E, P, L, that produce start, immediate and delay replicas respectively of the received PRN codes, which are applied as phase shifts to phase rotators 25, 26, 27 respectively.

The correlation signals C_ {E5b, -1}, C_ {E5b, 0} and C_ {E5b, 1} supplied by integrators 28, 29, 30 are then used as input to the discriminators who they are sensitive to code and carrier phase misalignments, used to control oscillators 4, 5.

The demodulator channel of Figure 5 presents two important differences with respect to a demodulation channel Traditional AltBOC like the one shown in Figure 1:

-

the entry to the delay line 7 is a signal delay of real values;

-

the multiplication with the chip before integration is replaced by a phase rotation.

Although the number counters needed for this architecture is smaller than that of the standard architecture of the Figure 1, is still large compared to the traditional line of 1 bit wide delay.

The architecture described in reference to the Figure 5 can be greatly improved considering that all rotators E, P and L 25, 26, 27 rotate with it frequency, although with a fixed phase difference. That is to say, when the P 26 rotator applies a lag of - \ omega_ {s} t- \ pi / 2, the rotator E 25 applies a lag of - \ omega_ {s} (t + dT_ {c} / 2) - \ pi / 2 and the rotator L 27 of - \ omega_ {s} (t + dT_ {c} / 2) - \ pi / 2, where d the initial separation-delay in units of chip and T_ {c} is the duration of the chip. This phase difference constant of \ pm \ omega_ {s} dT_ {c} / 2 can be extracted from the integral and be carried out at slow speed after the correlation (after integration).

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This leads to the optimized architecture that is presented in Figure 6. Compared to the architecture of the Figure 5:

-

every one of the three rotators 25, 26, 27 is replaced by a corresponding signal multiplier 33, 34, 35,

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be Insert a E5bQ 31 subcarrier rotator between the output of the carrier rotator 3 and the corresponding inputs of the signal multipliers 33, 34, 35, and performs a phase rotation of e ^ {- i (\ omega_ {s} t + \ pi / 2)},

-

the multi-bit delay line 24 is replaced by a 32-bit wide code delay line (removing the PRN phase detector) and is directly controlled by the E5b 21 code generator and

-

be insert two signal multipliers 36, 37, with e-α and e <i} respectively at the output of integrators E and L, 28 and 30.

The two signal multipliers 36, 37 belong to a low post correlation stage speed (after integration), while the other party of this architecture belongs to a stage of high speed pre-correlation.

With this architecture, the only additional block with respect to a traditional BPSK demodulator is the rotator of subcarrier 31, whose phase is controlled by an oscillator of NCO code 5. This architecture is mathematically equivalent to the architecture of Figure 5 when? fits \ omega_ {s} dT_ {c} / 2. However, you can choose others α values to obtain virtually any other lag between start and delay replicas.

For reasons of clarity, the architectures of AltBOC demodulator described with reference to Figures 5 and 6 only show three complex correlators (start, point and delay). Actually, the tracking detection of globules may require at least two other correlators (very advanced and very delayed), although this is an extension Direct structure.

Thus, the architecture that is represented in the Figures 5 or 6 can be extended to any number of correlators. For example, n can be used Start and m delay correlators, feeding each of a corresponding cell of a delay line. C_ {E5b, 0} corresponds to the immediate correlation. Normally, the Start and delay correlators are calculated with a delay of a cell with respect to the immediate correlation, that is to say correspond to C_ {E5b, 1} and C_ {E5b, -1} respectively. Without However, they can be adjusted to any other delay. an app typical of additional correlations is tracking detection of secondary peaks.

Figures 5 and 6 illustrate the architecture of a individual channel On the AltBOC receiver, two of these channels for the E5 signal (one for E5a and one for E5b) are placed together and add the correlations to produce a correlation signal AltBOC Such a combined channel derived from the architecture of the Figure 6 is represented in Figure 7.

In Figure 7, the architecture comprises a Common RF / IF phase 2, a carrier rotator 3, an NCO 4 carrier and an NCO 5 code.

Each channel E5a, E5b comprises a phase rotator of subcarrier 31a, 31b, a code generator E5a / E5b 21a, 21b that feeds the corresponding delay line 32a, 32b, three corresponding correlators E, P, L, each including a signal multiplier 33a, 34a, 35a, 33b, 34b, 35b and an integrator 28a, 29b, 30a, 28b, 29b, 30b. The start and delay branches of each channel E5a, E5b also comprise two signal multipliers corresponding 36a, 37a, 36b, 37b by a factor equal to e <-1> and e <alpha] respectively. The phase rotator of subcarrier 31b executes a phase rotation of e ^ {- i (\ omega_ {s} t + \ pi / 2}, while the phase rotator of carrier 31a executes a phase rotation of e ^ (\ omega_ {s} t + \ pi / 2}.

The E5a channel also comprises a multiplier of additional signal 41a by a factor equal to -1 inserted between the code NCO 5 and the subcarrier rotator E5aQ 31a. The outputs of the two channels are added by three additives 42, 43, 44, issuing the corresponding correlation signals C_ {E5,1} C_ {E5.0} and C_ {E5, -1}.

Delving into formulas (4) and (5), it you can deduce that the correlations C_ {E5b, k} and C_ {E5a, k} They are given as follows:

eleven

12

13

14

fifteen

16

where α = \ omega_ {s} dT_ {c} / 2 = 2 \ pif_ {s} dT_ {c} / 2. The start-delay separation d is determined by the synchronization frequency of the delay line 32. Normally, d ranges from 0.1 to one.

For monitoring, the receiver uses the correlations C_ {E5, k} in order to build discriminators of codes and carrier phase whose output is proportional to the error code tracking and carrier phase respectively.

The amount of base used in the discriminator PLL is the point correlation C_ {E5.0}. Basic amount used in the DLL discriminator is the difference between Start and Delay correlations, also called correlation Start minus delay and indicated as C_ {E5, EmL}. This difference is:

17

In the special case of d = 1 / (2f_ {s} T_ {c}) = 1 / (2x15,345 / 10.23) = 1/3, α is equal to \ pi / 2 and can be demonstrate that C_ {E5, EmL} is proportional to i (C_ {E5a, 0} - C_ {E5b, 0}) for small tracking errors. This fact leads to a dramatic reduction in the complexity of channels, since you only have to calculate the point correlations (C_ {E5a, 0} and C_ {E5b, 0}) needed to compute both the code tracking as carrier.

This property can be demonstrated by remaking the expression for C_ {E5, EmL} as follows, taking into notice that \ alpha = \ pi / 2. When formulas (8) are inserted into (13) in formula (14), we have:

18

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twenty

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On the other hand, for tracking errors small (\ tau << 1), i (C_ {E5a, 0} - C_ {E5b, 0}) is simply:

twenty-one

This relationship shows that C_ {E5, EmL} is proportional to i (C_ {E5a, 0} - C_ {E5b, 0}). The factor (2-d) is irrelevant since it is simply a compensated amplification factor for normalization of discriminator

This leads to an architecture like the one represented in Figure 8, which is equivalent to that in Figure 7 for the case d = 1/3, although much simpler.

With respect to the architecture of Figures 6 and 7, this architecture does not include lines of code delay 32a, 32b and has a single correlator for each E5a code and E5b. Each correlator comprises a single signal multiplier 51a, 51b receiving the output of the subcarrier rotator corresponding E5a and E5b 31a, 31b and codes from the generator of corresponding E5a and E5b codes 21a, 21b and a single integrator 52a, 52b. The output signals C_ {E5a, 0} and C_ {E5b, 0} of the integrators 52a, 52b are applied to an additer 63 to obtain the point correlation signal C_ {E5.0} and to a comparator 64 and a multiplier by i 65 to get the start correlation signal minus delay C_ {E5, EmL} = i (C_ {E5a, 0} - C_ {E5b, 0}).

It can be seen that this last architecture It is extremely simple, since you only need a single channel correlator. Surprisingly, this brings us to the conclusion that the AltBOC demodulator can be implemented from very effective way in terms of accountants, despite their apparent complexity.

This latest architecture shows that you can track the AltBOC signal without any correlator of Start or delay. This surprising result can be understood as intuitive way by drawing another Fresnel diagram, just like in the Figure 9. As shown above, misalignment of code \ tau is proportional to the angle \ varphi between the correlation vectors C_ {E5a, 0} and C_ {E5b, 0}: \ varphi = 2 \ omega_ {s} \ tau. You can also see in the diagram that the vector i (C_ {E5a, 0} - C_ {E5b, 0}), marked as "E-L corr" in the diagram, obtained by subtraction of the vector C_ {E5b, 0} and the vector C_ {E5a, 0}, and by rotation of the resulting vector 90 degrees, is real, and has a amplitude proportional to the angle var. This is the reason fundamental why tracking AltBOC code does not need Start and delay code replicas: misalignment of code can be calculated only from the correlators exact.

Figure 10 represents a receiver that comprises the AltBOC demodulator of Figure 8 and PLL (Phase-Lock Loop) and DLL (Delay-Lock Loop) that respectively control the NCO 4 carrier and the Code NCO 5. The PLL comprises a discriminator 71 whose output P is filtered by a PLL 72 filter before being applied to an input of carrier control NCO 4. The discriminator PLL 71 is the arcotan discriminator, which consists in calculating the angle of the complex number C_ {E5a, 0}:

101

The DLL comprises a discriminator DLL that receives the correlation signal C_ {E5a, EmL} and a connected DLL 76 filter to a control input of the NCO 5 code. The DLL discriminator is of the discriminant type of scalar product power, which calculates the signal D = Real (C_ {E5a, EmL \ Cdot} C * _ E5,0}). Of this mode, the DLL discriminator comprises a complex conjugate function 73 to which the C_ {E5,0} signal and a multiplier of signal 74 to multiply the signals provided by the multiplier by i 65 and the complex conjugate function 73. The signal D is then obtained by a function 75, subtracting the real part of the complex signal sent by signal multiplier 74.

After some algebraic manipulations, a simplified architecture can be derived like the one represented in Figure 11 from the architecture of Figure 10, which needs fewer operations to calculate the same DLL discriminator. According to the discriminator of Figure 10:

22

Thus, in Figure 11, the DLL discriminator it comprises a complex conjugate function 81 to which the correlation signal C_ {E5a, 0} and a signal multiplier 82 for multiply the signal provided by the conjugate function complex and the correlation signal C_ {E5b, 0}. The signal D is obtained later by an Imag () 83 function, subtracting the part imaginary of the complex signal sent by the multiplier of signal 82.

Another modification of the architecture of the Figure 11 would be the substitution of the Imag () operator with an operator Angle () (that is, a block that provides the same functionality that the discriminator arches 71).

The architecture of Figure 11 can be further optimize as shown in Figure 12, taking into account  that the phase rotation in the carrier rotator 3 followed by the phase rotation in subcarrier rotators 31a, 31b se they can combine in a single phase of rotation through a phase corresponding to the sum of the phases of the carrier and the subcarrier

Thus, in Figure 12, the carrier rotator 3, the two subcarrier rotators 31a, 31b and the multiplier 41a of Figure 11 are replaced by two phase rotators 92a and 92b (one for each channel E5a and E5b) that receive the transformed signal RF to IF of the RF / IF stage 2. In addition, the subcarrier phase that provides the code NCO 4 is added with an additive 93a to the phase provided by the NCO 3 carrier and subtracted from it by an additive 93b, applying the results of the addition to the phase rotors 92a, 92b respectively of channels E5a and E5b.

The architecture shown in Figure 13 can be derived from the previous architecture, replacing the Code NCO for a simpler NCO 95 that supplies only the speed of redundant bits f_ {c} and a frequency multiplier 96 by 1.5 applied to the redundant bit rate f_ {c}, so that the subcarrier frequency f_ {s} is obtained, which is applied as an entry to the additives 93a, 93b. This requires duplicate the NCO 4 carrier, one for each channel E5a, E5b. The carrier frequency followed by the PLL applies to additives 93a, 93b, whose corresponding outputs direct the carrier NCO 91a, 91b of the two channels E5a, E5b, in order to continue the corresponding combined frequencies carrier + subcarrier  of the two channels E5a, E5b.

In this architecture, the stages of high-speed pre-correlation of E5a channels and E5b remain identical. Both comprise a phase rotator 92a, 92b, two NCO 91a, 91b, a code generator 21a, 21b and a correlator Also, if you double the NCO code to have a NCO per channel, each of the stages of high-speed pre-correlation of E5a channels and E5b is identical to a traditional BPSK channel (Binary Phase-Shift Keying), which offers great benefits in the design of a combined AltBOC / BPSK receiver.

Naturally, the optimizations made in the architecture of Figures 12 and 13 can also be applied to the architectures of Figures 5, 6 or 7.

Claims (11)

1. Method for demodulating carrier signals from alternate binary displacement comprising at least two subcarriers (E5a, E5b) each having a component in phase and a quadrature component modulated by codes pseudo-random, the quadrature components being modulated by means of pilot signals without data and being modulated the components in phase by means of data signals, said comprising Method the stages of: transform the carrier signals of alternate binary shift at an intermediate frequency, filter  through a bandpass filter the transformed signals and sample the filtered signals, generate a carrier phase and a rotation of carrier phase of the signals sampled by said phase of carrier, and correlate the sampled signals rotated, characterized in that it also comprises the steps of generating for each subcarrier (E5a, E5b) pseudo-random binary codes and a subcarrier phase, which are used to correlate the rotated sampled signals. 2. Method according to claim 1, further it comprises a step of translating said binary codes pseudorandom of said subcarriers at phase angles, the which are combined respectively with the subcarrier phases to obtain resulting phase angles for each subcarrier, said resulting phase angles are offset to obtain at least one start phase angle, one phase angle immediately and at least one phase delay angle for each subcarrier, said correlation stage comprising the phases of rotate said sampled signals rotated by said phase start, immediate and delay phase angles of each subcarrier, to obtain start, immediate and delay of said sampled signals for each subcarrier, and of integrate respectively the start, immediate and delay for each subcarrier for a period of time predefined 3. Method according to claim 1, further it comprises a phase of phase rotation of said signals shown  rotated by said subcarrier phases in order to obtain signals sampled rotated phase for each subcarrier (E5a, E5b) before of correlating said rotated sampled signals. 4. Method according to claim 3, which further it comprises a stage of binary displacement of said codes pseudo-random to obtain at least one pseudo-random code of start, immediately and at least one pseudorandom delay code, said correlation stage comprising the phases of combining said sampled phase rotated signals for each subcarrier with these pseudo-random start, immediate and delay, and of integrating the resulting signals over a period of predefined time, in order to obtain correlation signals from start, immediate and delay (C_ {E5a, -1}; C_ {E5a, 0}; C_ {E5a, 1}; C_ {E5b, -1}; C_ {E5b, 0}; C_ {E5b, 1}) for each subcarrier (E5a, E5b), said method further comprising a step of post-correlation at low speed comprising phase rotation stages of the start correlation signals and delay of each subcarrier respectively by angles of opposite constant phase (i \ alpha, i \ alpha) and add respectively the start correlation signals thus obtained of said subcarriers, the immediate correlation signals of said subcarriers and delay correlation signals as well obtained from said subcarriers, in order to obtain respectively Start, immediate and delay correlation signals resulting (C_ {E5, -1}; C_ {E5,0}; C_ {E5,1}). 5. Method according to claim 3, which further it comprises a step of determining a carrier frequency and combined subcarrier for each subcarrier, combining the phase rotation stages with said carrier phases and said subcarrier phases in a single phase of phase rotation for each subcarrier using said carrier frequencies and combined subcarriers. Method according to claims 3 or 5, characterized in that said correlation step comprises the phases of combining said sampled phase rotated signals for each subcarrier (E5a, E5b) respectively with the pseudo-random codes of said subcarrier and of integrating during a period of predetermined time the resulting signals in order to obtain a correlation signal (C_ {E5a, 0}; C_ {E5b, 0}) for each subcarrier. 7. Method according to claim 6, which further it comprises a low post correlation stage speed comprising the steps of combining the signals of correlation (C_ {E5a, 0}; C_ {E5b, 0}) for these subcarriers (E5a, E5b) in order to obtain an immediate correlation signal (C_ {E5,0}) used as an input of a PLL discrimination which directs an oscillator (4) that controls said rotation stage of carrier and a correlation signal start minus delay (C_ {E5, EmL}) used as input of a DLL discrimination that directs an oscillator (5) that controls said code generation and said generation of subcarrier phase. Method according to claim 7, characterized in that the start minus delay correlation signal (C_ {E5, EmL}) is obtained from the correlation signals (C_ {E5a, 0}; C_ {E5b, 0}) for said subcarriers (E5a, E5b) by the following formula: 102 where C_ {E5, EmL} is the signal of correlation start minus delay and C_ {E5a, 0}; C_ {E5b, 0} are the correlation signals for said subcarriers 9. Method according to any of claims 7 to 8, characterized in that the DLL discrimination is of the scalar product power discrimination type and performs the following operation: 103 where C_ {E5, EmL} is the start minus minus delay correlation signal, C_ {E5a, 0} is the immediate and Real () correlation signal
It is a function that returns the real part of a complex number, using signal D to direct the oscillator (5) that controls said code generation and said subcarrier phase generation. 10. Method according to any of claims 7 to 8, characterized in that the DLL discrimination performs the following operation: 104 where C_ {E5a, 0,} C_ {E5b, 0} are the correlation signals for said subcarriers and Imag () is a function that returns the imaginary part of a number complex. 11. Device for demodulation of carrier signals of alternating binary displacement comprising at least two subcarriers (E5a, E5b) each having a phase component and a quadrature component modulated by pseudo-random codes, the quadrature components are modulated by pilot signals without data and phase components are modulated by data signals, characterized in that it comprises means for implementing the method according to any one of claims 1 to 10.