EP3917012A1 - Circuit d'interface de capteur à base d'oscillateur - Google Patents

Circuit d'interface de capteur à base d'oscillateur Download PDF

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Publication number
EP3917012A1
EP3917012A1 EP20176683.9A EP20176683A EP3917012A1 EP 3917012 A1 EP3917012 A1 EP 3917012A1 EP 20176683 A EP20176683 A EP 20176683A EP 3917012 A1 EP3917012 A1 EP 3917012A1
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EP
European Patent Office
Prior art keywords
signal
oscillator
interface circuit
sensor interface
based sensor
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EP20176683.9A
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German (de)
English (en)
Inventor
Johan Vergauwen
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Melexis Technologies NV
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Melexis Technologies NV
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Priority to EP20176683.9A priority Critical patent/EP3917012A1/fr
Priority to EP21175168.0A priority patent/EP3917013A1/fr
Priority to US17/329,814 priority patent/US11632118B2/en
Priority to CN202110576669.5A priority patent/CN113726328B/zh
Publication of EP3917012A1 publication Critical patent/EP3917012A1/fr
Withdrawn legal-status Critical Current

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/12—Analogue/digital converters
    • H03M1/60—Analogue/digital converters with intermediate conversion to frequency of pulses
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
    • H03L7/00—Automatic control of frequency or phase; Synchronisation
    • H03L7/06—Automatic control of frequency or phase; Synchronisation using a reference signal applied to a frequency- or phase-locked loop
    • H03L7/08—Details of the phase-locked loop
    • H03L7/099—Details of the phase-locked loop concerning mainly the controlled oscillator of the loop
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K19/00—Logic circuits, i.e. having at least two inputs acting on one output; Inverting circuits
    • H03K19/0175—Coupling arrangements; Interface arrangements
    • H03K19/017509—Interface arrangements
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
    • G01D11/00—Component parts of measuring arrangements not specially adapted for a specific variable
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
    • G01D18/00—Testing or calibrating apparatus or arrangements provided for in groups G01D1/00 - G01D15/00
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
    • G01D21/00—Measuring or testing not otherwise provided for
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
    • G01D3/00—Indicating or recording apparatus with provision for the special purposes referred to in the subgroups
    • G01D3/028—Indicating or recording apparatus with provision for the special purposes referred to in the subgroups mitigating undesired influences, e.g. temperature, pressure
    • G01D3/036—Indicating or recording apparatus with provision for the special purposes referred to in the subgroups mitigating undesired influences, e.g. temperature, pressure on measuring arrangements themselves
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
    • G01R19/25—Arrangements for measuring currents or voltages or for indicating presence or sign thereof using digital measurement techniques
    • G01R19/252—Arrangements for measuring currents or voltages or for indicating presence or sign thereof using digital measurement techniques using analogue/digital converters of the type with conversion of voltage or current into frequency and measuring of this frequency
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/06—Continuously compensating for, or preventing, undesired influence of physical parameters
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/06—Continuously compensating for, or preventing, undesired influence of physical parameters
    • H03M1/0602—Continuously compensating for, or preventing, undesired influence of physical parameters of deviations from the desired transfer characteristic
    • H03M1/0604—Continuously compensating for, or preventing, undesired influence of physical parameters of deviations from the desired transfer characteristic at one point, i.e. by adjusting a single reference value, e.g. bias or gain error
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/10—Calibration or testing
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00—Analogue/digital conversion; Digital/analogue conversion
    • H03M1/10—Calibration or testing
    • H03M1/1009—Calibration
    • H03M1/1014—Calibration at one point of the transfer characteristic, i.e. by adjusting a single reference value, e.g. bias or gain error
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
    • G01D3/00—Indicating or recording apparatus with provision for the special purposes referred to in the subgroups
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
    • G01D5/00—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable
    • G01D5/12—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means
    • G01D5/14—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage
    • G01D5/16—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage by varying resistance

Definitions

  • the present invention is generally related to the field of sensor interface circuits for sensor systems.
  • Sensors are increasingly important in any field where finer and ever more intelligent control is needed. Examples are found in the growing fields of automotive applications or wireless sensor networks (WSN). In the automotive industry sensors are essential for applications ranging from increased safety to road stability as well as to improve car performance and reliability demanded by customers. Further, compact and low-power sensor interfaces are needed to be competitive on the growing market and to enable new applications for the 'Internet of Things'.
  • WSN wireless sensor networks
  • the silicon area is a main contributor to the cost of the sensor interface, therefore the interface circuit has to be made as small as possible. This should not only be valid for the technology nodes that are used today (and which are still relative big for the automotive industry), but also in more advanced technologies.
  • the sensor signal is continuous in time and amplitude.
  • this analog signal is amplified, sampled and converted to the digital domain by an analog to digital converter (ADC).
  • a well-known ADC type is a Delta-Sigma ADC, which exploits an oversampling of the input signal and a noise shaping technique to obtain an improved precision.
  • the sensor signal frequency varies from DC up to 10 - 100 kHz, which allows for the oversampling needed for a Delta-Sigma converter.
  • Time/frequency-based conversion mechanisms quantize the continuous input signal by using a known time/frequency signal as a reference instead of a voltage.
  • a time/frequency-based conversion circuit contains two building blocks : a Voltage-to-Time Converter (VTC) transforms the analog signal c(t) into time or frequency information f(t), while a Time-to-Digital Converter (TDC) digitizes this information with the help of a reference frequency.
  • VTC Voltage-to-Time Converter
  • TDC Time-to-Digital Converter
  • Closed-loop oscillator-based sensor interfaces as in Fig.1 combine the advantages of time based converters (small and scaling with technology) and sigma-delta ADCs (high accuracy due to oversampling and noise shaping).
  • This architecture is basically a Phase-Locked Loop (PLL) structure, but it has also similarities with a sigma-delta converter as explained in the paper " A novel PLL-based sensor interface for resistive pressure sensors" (H. Danneels et al., Procedia Engineering, Eurosensors 2010, vol. 5, 2010, pp. 62-65 ). It has the same noise-shaping properties which contribute in increasing the accuracy (expressed e.g. in terms of SNR).
  • a typical set-up of a closed-loop oscillator-based sensor read-out circuit contains two controlled oscillators, for example two voltage controlled oscillators (VCOs) which are matched, a binary phase detector to compare the phase difference between the two oscillator outputs and a feedback mechanism towards the sensing means.
  • the digital output signal of the interface circuit is also derived from the phase detector output.
  • a digital filter is provided after the phase detector to filter the phase detector output before it is fed back and made available as output signal of the interface circuit.
  • a conventional oscillator based sensor interface circuit used in a closed loop as illustrated in Fig.1 operates as follows.
  • a physical quantity is converted by the sensor (1) into electrical signals (11, 12) that influence the oscillators (21, 22) connected to it.
  • the phase of the oscillator outputs (41, 42) is compared in the phase detector (3).
  • the phase detector output signal (31) is fed back through a feedback element (4) to the sensor (1) in order to keep the phase difference between both oscillators small and on average close to zero.
  • the closed-loop ensures that the averaged phase detector output (31) is a digital representation of the physical quantity.
  • the input signal containing the physical quantity to be converted in the sensor typically represents a pressure, temperature or magnetic field. Also other types of physical signals can be used as input to the interface.
  • an oscillator-based sensor interface circuit comprising at least two oscillators.
  • a feedback element receives the digital output signal and provides a feedback signal to maintain a given relation between oscillator frequencies of the at least two oscillators.
  • An error like e.g. an oscillator mismatch can be detected and a tuning element is provided to tune at least one characteristic of the sensor interface circuit in order to cause a change in oscillator frequency of at least one of the oscillators to reduce the detected error.
  • Chopping circuitry may be provided for applying modulation and demodulation by means of a chopping signal. In this way offset errors can be reduced or even completely removed.
  • Patent US10574244 B2 relates to an oscillator-based sensor interface circuit with only one oscillator, wherein a switching element is provided to switch between at least two signals to be applied to the oscillator.
  • a counter counts the number of cycles produced by the oscillator when receiving from the oscillator a signal derived from one of the applied signals.
  • a control output signal is derived from a number of counted oscillator cycles when a first signal is applied to the oscillator and a number of counted oscillator cycles when a signal different from that first signal is applied. The control output signal is then used to derive a feedback signal that aims to maintain a fixed relation between the counted numbers of oscillator cycles.
  • the invention relates to an oscillator-based sensor interface circuit comprising
  • the proposed solution indeed allows for an analogue filtering of the electrical signal prior to applying that signal to the oscillation means without jeopardizing the loop stability.
  • This is achieved by splitting the feedback path in two parts.
  • One part comprises the digital signal coming from the comparator means or a representation thereof, which is in a first feedback element converted into a first feedback signal and fed to the oscillation means.
  • the other part comprises a second feedback element which receives the signal output by the digital filter and feeds a signal back to the input node and the subsequent analogue filter.
  • only the feedback signal obtained via the digital filter and the second feedback element is fed to the analog filter. Adding the first feedback signal only after the analog filter is beneficial for the stability of the sensor interface circuit.
  • said oscillator of the oscillation means is arranged to receive the first feedback signal.
  • the sensor interface circuit is arranged to combine the first feedback signal with the filtered signal received from the analog filter.
  • the oscillation means comprises a switching device arranged to switch between at least two signals to be applied alternately to said oscillator.
  • the sensor interface circuit comprises at least one further oscillator arranged to receive the filtered signal from the analog filter.
  • the first feedback element is arranged to multiply the digital comparator output signal with a scaling factor.
  • the comparator means comprises storage means for storing the digital comparator output signal.
  • the comparator means is arranged to add dither before comparing the signals from the oscillation means.
  • the first feedback element comprises a digital-to-analog converter to convert the representation of the digital comparator output signal into the first feedback signal.
  • the digital filter comprises an integrator arranged to yield the integrated version of the digital comparator output signal.
  • the integrator is in some embodiments implemented as a counter.
  • the feedback path comprising the integrator advantageously also contains a finite impulse response filter.
  • the finite impulse response filter may also be present in embodiments wherein there is no integrator in the digital filter.
  • the first feedback element is implemented as a voltage divider.
  • the voltage divider is selectable, i.e. the voltage division can be adapted so that a different output signal is obtained.
  • the sensor interface circuit comprises chopping circuitry arranged for modulating the filtered electrical signal representative of said electrical quantity with a chopping signal and for feeding the modulated signal to the oscillation means and arranged for demodulating.
  • the oscillation means comprises a ring oscillator.
  • the comparator means is implemented as a phase detector.
  • the phase detector is realized with a single flip-flop.
  • the comparator may comprise a counter in certain embodiments.
  • the oscillator-based sensor interface circuit comprises a sensing means arranged to convert the physical quantity comprised in a received signal into the electrical quantity and to output the electrical signal representative of the electrical quantity.
  • the received signal may be one of a pressure, a temperature, a force, an optical signal or a magnetic signal.
  • the sensing means is arranged to receive the second feedback signal.
  • the present invention discloses a closed-loop sensor interface circuit with improved EMI robustness. More in particular, a sensor interface circuit is proposed so designed that a low-pass filter to combat EMI disturbance can be added between sensing and oscillation means without rendering the loop unstable. This filtering also reduces the sensor noise and the noise created by the feedback to the sensor or sensor nodes.
  • a sensor converts a physical quantity into an analog electrical quantity.
  • the sensor output signal is too small to be used directly. Therefore a sensor interface circuit amplifies the signal comprising the electrical quantity received from the sensor to obtain a more useful signal.
  • the signal is often also further processed in the sensor interface. Additionally the signal can be converted to a digital signal and further processed in the digital domain.
  • a sensor interface circuit is considered to be a structure to transfer the electrical quantity coming out of the sensor into a digital signal.
  • the input signal containing the physical quantity to be converted in the sensor often represents a pressure, temperature or magnetic field, also other types of physical signals can be used as input to the interface circuit of this invention.
  • the sensor interface circuit of the invention may have the sensor as a part of the interface circuit, this is not strictly required.
  • the sensor(s) may be external to the circuit of the invention and the circuit is fed with an electrical signal representative of the electrical quantity into which the physical quantity is converted in the sensor.
  • Fig.2 illustrates two embodiments of a block scheme of a closed-loop sensor interface circuit according to the present invention. It comprises oscillation means (110,120) and two feedback loops. Since it is highly digital-oriented, the proposed architecture is highly scalable with technology and thus more area-efficient, energy-efficient and robust against process variations and external factors. This makes the structure a good candidate for sensor readout circuitry in emerging applications such as smart sensors, wireless sensor networks, health care monitoring, etc.
  • the sensor is part of the sensor interface circuit. It is repeated that this feature is optional.
  • the sensing means (160) can be implemented in many different ways.
  • One implementation of the sensor may be with two resistors, which are varied by the signal (e.g. pressure) that needs to be sensed.
  • only one resistor is variable and the other one is fixed.
  • the number of resistors is different (e.g. four resistors in a Wheatstone bridge configuration).
  • Other types of sensors can be used in various configurations. For example, instead of a resistive sensor, a capacitive sensor can be employed.
  • the sensor and the oscillator can be merged into a sensor controlled oscillator.
  • the oscillation means comprises at least one oscillator.
  • Various kinds of controlled oscillators can be envisaged for use in the sensor interface circuit. An obvious choice is a voltage controlled oscillator or a current controlled oscillator, but other options are valid too, e.g. a capacitance controlled oscillator.
  • only one controlled oscillator can be used, for example in case there is only one sensor (e.g. a capacitor) influenced by the physical quantity.
  • Fig.2a and Fig.2b there are two controlled oscillators (110,120). There may also be more than two oscillators.
  • a third oscillator can for example be used to generate the master clock for the digital core of the chip comprising the sensor interface circuit.
  • the two other oscillators could then be changed so that on average they are running at the same frequency as the third oscillator or at a certain fraction or multiple of it.
  • the digital master clock can be directly derived from one oscillator of the oscillation means used by the sensor interface. In that case it can be advantageous to keep this oscillator at a fixed frequency and to control only the other oscillator.
  • a fixed oscillator frequency can also be generated by using a (fixed) reference signal as input of a controlled oscillator.
  • the comparator (130) compares the output signals received from the oscillation means and accordingly outputs a digital comparator output signal.
  • the comparator output can be a single bit.
  • the comparator may be just a flip-flop, e.g. a D-flip-flop wherein one signal is used as clock and the other signal as data input.
  • the output signal shows in this case only which of the two inputs toggled first.
  • the comparator can be implemented as a phase detector in some embodiments. Also multi-bit comparators can be used. In that case the output may contain further information about the time difference between the toggling of both inputs.
  • the comparator contains a memory.
  • a new comparator output value may then be based on the comparator inputs and on one or more of the previous comparator output values kept in that memory.
  • a comparator with hysteresis may be used, whereby the comparator output in between two thresholds is dependent on the previous comparator decision.
  • the comparator compares the inputs of the comparators after dither, i.e. intentionally applied noise, has been added.
  • Dither is applied to randomize the quantization error. It is often used in (first order) sigma-delta converters to avoid dead bands and limit cycles.
  • the memory function and the dither as described above can be combined, e.g. dither may be only added when a certain pattern is recognized in the comparator output.
  • the comparator output signal is applied to a feedback element (180), named feedback element 1 in Fig.2 and provided in a path towards the oscillation means.
  • the feedback signal output by the feedback element 1 (180) is in the embodiment shown in Fig.2a applied to the input to the oscillation means, where it is combined with the signal output by the filter (170).
  • the feedback signal output by the feedback element 1 (180) may directly be fed to the oscillation means.
  • the comparator output signal is converted into a feedback signal by multiplying the comparator output signal with a proportionality factor. In many embodiments the proportionality factor is equal to 1.
  • the feedback element (180) is not necessarily a pure multiplier.
  • This feedback path is referred to as proportional path since it simply acts on the comparator output signal or a scaled version of the comparator output signal.
  • the signal in this proportional path contains no real sensor information because the average signal value is fixed to the middle value, assuming that both oscillators 110, 120 are locked.
  • the main purpose of the proportional path is to make the interface circuit stable.
  • the comparator output signal is also applied to a digital filter (142).
  • the digital filter is in preferred embodiments implemented as an integrator, for example as a counter.
  • the filtered version of the comparator output signal is applied through a feedback element (150), in the embodiment shown in Fig.2a depicted as feedback element 2, to the sensor (160).
  • the feedback signal output by feedback element 2 (150) is combined with, e.g. added to, the sensor signal at an input node of the interface circuit (see Fig.2b ).
  • the combination means per se are known to the skilled person, e.g. means for summing signals.
  • the resulting signal is applied to an analog low-pass filter (170), which is added to filter out the disturbance due to EMI.
  • the inventor has made the consideration that when adding the analog low-pass filter, while still keeping the full feedback (proportional and integral) to the sensor or to the input node receiving the sensed signal, the loop becomes unstable. Adding the low-pass filter and performing the full feedback after the filter takes the sensor signal out of the loop and is fundamentally different (worse) solution. However, if a low-pass filter is added after the sensor to act on the sensed signal, the system can be made stable by having the feedback of the proportional path only after the analog filter.
  • the digital output of the sensor interface circuit can in principle be based on the sum of both proportional and integral path, but there is no real information transmitted in the proportional path. Therefore it is preferable (more stable) to use only the integral path for the data output. This confirms that there is no real need to have the proportional path fed back to the input node (or to the sensor) since is does not bring any advantage.
  • Fig.2b illustrates an embodiment with the same functional blocks as in Fig.2a , but shows one of the other options to apply the two feedback signals. Indeed, the feedback signal output by feedback element 1 (180) is directly fed to the oscillator means and the feedback signal from feedback element 2 (150) is at an input node of the sensor interface circuit combined with the sensor output signal to form the input signal to the analog low-pass filter.
  • Fig.3 shows an example implementation of the embodiments of Fig.2 .
  • the comparator (130) is implemented as a phase detector.
  • the digital PI-controller (140) the phase detector output signal is applied to two different paths.
  • One path comprises the digital filter (142) implemented as an integrator.
  • the digital filter is in preferred embodiments realized as an integrator.
  • the digital filter yields the digital output signal, which is the filtered comparator output signal.
  • the same signal is also fed to the integral feedback element (150), which in turn provides a feedback signal that is combined with (added to) the sensor signal at an input node of the interface circuit.
  • the phase detector output signal is multiplied with a proportionality factor in the optional block (144) and next applied to the proportional feedback element (180).
  • the proportionality factor is 1 and that there is no physical block for performing this multiplication.
  • a multiplication with a proportionality factor can be made in feedback element 1 itself.
  • the resulting feedback signal is combined with the analog filter output signal to obtain the input to the oscillator means.
  • FIR finite impulse response filter
  • the integral feedback element (150) is in typical embodiments already a multi-bit feedback. Still such a FIR filter can be useful to reduce the jitter of the feedback duration, especially when the feedback is determined by the period of the oscillator means, which is inherently variable.
  • Fig.4 shows in more detail an implementation of a sensor interface circuit according to an embodiment of the invention.
  • the circuit comprises two oscillators (110, 120), in this example voltage controlled oscillators (VCOs).
  • the sensor (160) consists in this embodiment of two resistors of which the resistance value for one increases and for the other decreases if the physical quantity increases.
  • the resistors are biased by current sources, which function as a feedback element (150).
  • the current sources contain a fixed part and a variable part controlled by the output signal of the digital filter.
  • the change in resistance due to the presence of the physical quantity is converted through the current biasing in a voltage difference at the input nodes of the VCOs. If the physical quantity increases, the voltage signal at first instance increases while the voltage at the other node decreases.
  • the period of a VCO is approximately inversely proportional to the input voltage. Therefore the increase of the physical quantity makes VCO (110) faster and VCO (120) slower, which leads after one or more VCO periods to the situation that one oscillator output signal has its rising edge before the other signal oscillator output signal. At that moment the output of the D flip-flop (130) which is used as phase detector becomes high (assuming the output was low before). This increases the output of the digital filter (142), which is updated every VCO period. This signal is fed back to the current sources used as feedback element (150). The feedback is done in such a way that the phases of oscillator output signals are locked to each other. This means the average frequency of both oscillators is the same.
  • the two VCOs are perfectly matched, it also means that the average voltage of the signals applied to the oscillating means is regulated to the same value. So the current at the left side is increased while the current at the right side is decreased to compensate for the resistance change caused by the increased physical quantity.
  • the feedback loop ensures the ratio of the current difference and the common mode current is on average equal to the ratio of the resistance difference and the common mode resistance.
  • the analog filter (170) in the embodiment of Fig.4 is realised with a network of resistors and capacitors.
  • the skilled person will readily understand that the invention is not limited to the filtering structure shown in Fig.4 , but that any suitable low-pass filtering configuration may be used instead.
  • the oscillators (110, 120) are in this embodiment implemented as a multi-stage ring oscillator. Ring oscillators as such are well known in the art. Their use offers various advantages, like e.g. a wide and easily controllable tuning range, low power consumption, etc.
  • the proportional feedback element (180) is implemented as a voltage divider. One of the outputs of the voltage divider is going to one oscillator and the other output is going to the other oscillator. So, the proportional feedback signal goes to the first stage of the ring oscillators.
  • the filtered sensor signal is applied to the other stages of the ring oscillators. It depends on the comparator 130 output which output is connected to which oscillator. This is achieved with a switching block 190 which may comprise e.g. four switches controlled by the comparator output signal.
  • chopping is a well-known technique to reduce offset and low frequency noise.
  • the sensor (160) is shown to be part of the interface circuit.
  • Two chopping blocks (115,125) have been added before and after the oscillators (110, 120), for modulation (i.e. chopping with a signal with a frequency f chop ) and demodulation, respectively.
  • the chopping operation can be divided into two phases. Chopping can facilitate the detection of an offset mismatch between the two oscillators, because such mismatch leads to a difference in the (averaged) digital output in each chopping phase.
  • the signals output by the analog filter (170) are fed via switches to a respective one of the oscillators. Simultaneously the signals output by the oscillators are each connected to an input of the comparator (130). In the other chopping phase the respective oscillators receive at their input the signal they did not receive in the first chopping phase, in other words the inputs are switched compared to the first phase. The same holds for the outputs.
  • the phase of the chopping blocks is controlled by a chopping control signal. After dechopping (demodulation) the signals are compared in the comparator (130) and a digital comparator output signal is output accordingly.
  • the average digital output during one chopping phase or during the other chopping phase is different when there is an offset mismatch between the oscillators.
  • the difference of the average digital output of one chopping phase from the average digital output of the other chopping phase is a measure for the mismatch.
  • This difference can be detected in the error detection block and used to tune the oscillators by the tuning element in such a way that this difference is compensated.
  • the tuning element at least one parameter of the sensor interface circuit is tuned, e.g. the gain of the oscillators, so as to reduce the offset error.
  • the tuning causes directly or indirectly in at least one of the oscillators a change in oscillator frequency.
  • the inputs of the modulating chopping block all come from the sensor. Some of the inputs can also come from a reference signal or from another signal (e.g. a signal used for tuning).
  • FIG.6 Another embodiment of an oscillator-based sensor interface circuit according to the invention is illustrated in Fig.6 .
  • a single oscillator (110) is preceded by a switching/multiplexing means so that the input to the oscillator can be selected to come from one of two or more sources.
  • the switching/multiplexing is controlled by means of a digital controller (not shown in Fig.6 ), whereby there is a phase wherein a first signal is applied to the oscillator and a phase wherein a second signal is applied, resulting in a different oscillator frequency. If the sensor generates a differential signal, the multiplexer can switch between both signals forming the differential signal.
  • the multiplexer can switch between the sensor signal and a reference signal.
  • a reference voltage signal is created by a fixed current flowing through a fixed resistor.
  • the sensor comprises a variable resistor.
  • the current source on top of the sensor resistor acts as feedback element 2 (150) in this embodiment and is adjusted to get the resulting sensor voltage equal to the reference voltage.
  • the multiplexer can also switch between different sensor signals (or other signals to be monitored) if multiple sensors or other channels have to be converted to the digital domain.
  • a further oscillator is still part of the chip, namely for generating the master clock that determines e.g. the sampling instants, as already mentioned above.
  • a practical example of a use case for a sensor interface circuit embodied as in Fig.6 is where a sensor with only one output is applied, e.g. a Negative Temperature Coefficient (NTC) sensor.
  • NTC Negative Temperature Coefficient
  • Fig.6 The benefit of embodiments as in Fig.6 is that mismatch between the oscillators, e.g. voltage controlled oscillators, or coupling between the oscillator outputs is avoided.
  • voltage controlled oscillators or other controlled oscillators, like current controlled oscillators or sensor controlled oscillators.
  • the mismatch also varies over time and over temperature. As they continuously toggle, two VCOs oscillating at similar frequencies can interfere with each other, leading to artefacts in the digital output.
  • Embodiments like in Fig.6 may further also have a better energy efficiency, as the oscillators are the main contributors to the power consumption.
  • Such embodiments of the sensor interface circuit may be useful in application where compactness and energy efficiency are crucial.
  • the comparator block (130) comprises an upcounter that counts the periods in the oscillator output signal, a differentiator that determines the difference of the newly counted value with the previous counter value in order to get the amount of oscillator periods between two sampling moments. Afterwards another difference is made versus a reference value (that was stored e.g. in a register). The resulting difference is further integrated in an integrator block to which the difference is input. The integrator accumulates these received differences until a new phase is started, i.e. when a switch is made to the other input. The output of the comparator block is then applied to the digital filter (142) as already described above.
  • FIG.7 A further exemplary embodiment is shown in Fig.7 .
  • a number of building blocks are combined that were already shown in one or more of the embodiments of the preceding figures.
  • the resistance sensing means are implemented in a Wheatstone bridge configuration.
  • the integral feedback element (150) connects a pull-up current to one output node of the Wheatstone bridge and a pull-down current to the other output node of the sensor bridge.
  • the switches are controlled by the digital filter output signal that is fed back. Current is supplied via the switches by means of current sources.
  • the analog filter (170) is represented as a RC network.
  • the comparator (130) is shown to comprise an up/down counter to count the periods in the signals output by oscillator (110) and oscillator (120), respectively.
  • a resulting signal indicative of the difference in the number of periods of both oscillators is sampled and applied to the proportional feedback element (180) and to an integrator (142), where the number of periods in that resulting signal is summed. That information is then used to control the switches in the integral feedback element (150).
  • Fig.8 More in particular, simulation results of the frequency spectrum when the interface circuit measures a sensor signal of 10kHz.
  • Fig.8a and Fig.8b are the results obtained with a circuit like in Fig.2b
  • Fig.8c and Fig.8d depict the results when the proportional and integral feedback are both performed on to the sensor signal.
  • Fig.8b and Fig.8d use a filter with a time constant of 1ms (so, 10 times bigger than the average oscillator period). There is almost no difference between Fig.8a and Fig.8c , which is normal, because the filter is negligible.
  • Fig.8d illustrates the system does not function properly if the proportional and integral feedback are both done to the sensor signal when there is a lot of filtering.
  • Fig.8c however shows still a good noise shaping, illustrating that the circuit designed according to the invention remains operational when a lowpass filter is added (but with a higher noise floor).
  • a computer program may be stored/distributed on a suitable medium, such as an optical storage medium or a solid-state medium supplied together with or as part of other hardware, but may also be distributed in other forms, such as via the Internet or other wired or wireless telecommunication systems. Any reference signs in the claims should not be construed as limiting the scope.

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EP20176683.9A 2020-05-26 2020-05-26 Circuit d'interface de capteur à base d'oscillateur Withdrawn EP3917012A1 (fr)

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EP20176683.9A EP3917012A1 (fr) 2020-05-26 2020-05-26 Circuit d'interface de capteur à base d'oscillateur
EP21175168.0A EP3917013A1 (fr) 2020-05-26 2021-05-21 Circuit d'interface de capteur à base d'oscillateur
US17/329,814 US11632118B2 (en) 2020-05-26 2021-05-25 Closed-loop oscillator based sensor interface circuit
CN202110576669.5A CN113726328B (zh) 2020-05-26 2021-05-26 基于闭环振荡器的传感器接口电路

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US20210376839A1 (en) 2021-12-02
CN113726328A (zh) 2021-11-30
US11632118B2 (en) 2023-04-18
CN113726328B (zh) 2024-02-02

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