EP3330966B1 - Improved frequency band extension in an audio frequency signal decoder - Google Patents

Description

The present invention relates to the field of the coding/decoding and processing of audio frequency signals (such as speech, music or other signals) for their transmission or their storage.

More particularly, the invention relates to a method and a device for frequency band extension in a decoder or a processor performing an audio frequency signal enhancement.

Many techniques exist to compress (with loss) an audio frequency signal such as speech or music.

Conventional coding methods for conversational applications are generally classified as waveform coding (PCM for "Pulse Modulation and Coding", ADPCM for "Pulse Modulation and Adaptive Differential Coding", transform coding, etc.), parametric coding (LPC for "Linear Predictive Coding", sinusoidal coding, etc.) and hybrid parametric coding with quantification of the parameters by "analysis by synthesis", including CELP coding (for "Code Excited Linear Prediction" in English). English) is the best known example.

For non-conversational applications, the state of the art in audio signal coding (mono) consists of perceptual coding by transform or in sub-bands, with parametric coding of high frequencies by band replication (SBR for Spectral Band Replication in English).
A review of classical speech and audio coding methods can be found in the books WB Kleijn and KK Paliwal (Eds.), Speech Coding and Synthesis, Elsevier, 1995 ; M. Bosi, RE Goldberg, Introduction to Digital Audio Coding and Standards, Springer 2002 ; J. Benesty, M. M. Sondhi, Y. Huang (Eds.), Handbook of Speech Processing, Springer 2008 .

We are particularly interested here in the codec (coder and decoder) standardized 3GPP AMR-WB (for "Adaptive Multi-Rate Wideband") which operates at an input/output frequency of 16 kHz and in which the signal is divided into two sub-bands, the low band (0-6.4 kHz) which is sampled at 12.8 kHz and coded by CELP model and the high band (6.4-7 kHz) which is reconstructed para metric by “Bandwidth Extension ” (or BWE) with or without additional information depending on the mode of the current frame. It can be noted here that the limitation of the coded band of the AMR-WB codec to 7 kHz is essentially linked to the fact that the frequency response in transmission of wideband terminals was approximated at the time of standardization (ETSI/3GPP then UIT-T) according to the frequency mask defined in the UIT-T P.341 standard and more precisely by using a so-called "P341" filter defined in the UIT-T G.191 standard. which cuts frequencies above 7 kHz (this filter respects the mask defined in P.341). However, in theory, it is well known that a signal sampled at 16 kHz can have a defined audio band from 0 to 8000 Hz; the AMR-WB codec therefore introduces a limitation of the high band in comparison with the theoretical bandwidth of 8 kHz.

The 3GPP AMR-WB speech codec was standardized in 2001 primarily for circuit mode (CS) telephony applications on GSM (2G) and UMTS (3G). This same codec was also standardized in 2003 at the ITU-T as recommendation G.722.2 "Wideband coding speech at around 16kbit/s using Adaptive Multi-Rate Wideband (AMR-WB)".

It includes nine bit rates, called modes, from 6.6 to 23.85 kbit/s, and includes mechanisms for continuous transmission (DTX for "Discontinuous Transmission") with voice activity detection (VAD for "Voice Activity Detection") and generation of comfort noise (CNG for "Comfort Noise Generation") from silence description frames (SID for "Silence Insertion Descriptor"), as well as mechanisms for correction of lost frames (FEC for "Frame Erasure Con cealment", sometimes called PLC for "Packet Loss Concealment").

The details of the AMR-WB coding and decoding algorithm are not repeated here, a detailed description of this codec can be found in the 3GPP specifications (TS 26.190, 26.191, 26.192, 26.193, 26.194, 26.204) and UIT-TG.722.2 (and the corresponding Annexes and Appendix) as well as in the article of B. Bessette et al. entitled “The adaptive multirate wideband coded speech (AMR-WB)”, IEEE Transactions on Speech and Audio Processing, vol. 10, no. 8, 2002, p. 620-636 and the source codes of the associated 3GPP and ITU-T standards.

The principle of band extension in the AMR-WB codec is quite rudimentary. Indeed, the high band (6.4-7 kHz) is generated by shaping a white noise by means of a temporal envelope (applied in the form of gains per subframe) and frequency (by the application of a linear prediction synthesis filter or LPC for "Linear Predictive Coding"). This band extension technique is illustrated in figure 1 .

• Un premier facteur est calculé (bloc 101) pour mettre le bruit blanc u _HB1(n) (bloc 102) à un niveau semblable à celui de l'excitation, u(n), n = 0,...,63, décodée à 12.8 kHz dans la bande basse : $$u_{\mathit{HB}2}\left(n\right)=u_{\mathit{HB}1}\left(n\right)\sqrt{\frac{{\displaystyle \sum _{l=0}^{63}u{\left(l\right)}^2}}{{\displaystyle \sum _{l=0}^{79}{u}_{\mathit{HB}1}{\left(l\right)}^2}}}$$
  

A white noise, u _{HB 1} ( n ), n = 0.79, is generated at 16 kHz per 5 ms subframe by a linear congruent generator (block 100). This noise u _{HB 1} ( n ) is shaped over time by applying gains per subframe; this operation is broken down into two processing steps (blocks 102, 106 or 109):

• A first factor is calculated (block 101) to put the white noise u _{HB 1} ( n ) (block 102) at a level similar to that of the excitation, u ( n ), n = 0,...,63, decoded at 12.8 kHz in the low band: $$a_{\mathit{HB}2}\left(\mathrm{not}\right)=a_{\mathit{<figure-callout\; id="1"\; label="HB"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">HB</figure-callout>}1}\left(\mathrm{not}\right)\sqrt{\frac{{\displaystyle \sum _{I=0}^{63}a{\left(I\right)}^2}}{{\displaystyle \sum _{I=0}^{79}{a}_{\mathit{HB}1}{\left(I\right)}^2}}}$$
  

• L'excitation dans la bande haute est ensuite obtenue (bloc 106 ou 109) sous la forme : $$u_{\mathit{HB}}\left(n\right)={\widehat{g}}_{\mathit{HB}}{u}_{\mathit{HB}2}\left(n\right)$$
  
  où le gain ĝ_HB est obtenu différemment selon le débit. Si le débit de la trame actuelle est <23.85 kbit/s, le gain ĝ_HB est estimé « en aveugle » (c'est-à-dire sans information supplémentaire); dans ce cas, le bloc 103 filtre le signal décodé en bande basse par un filtre passe-haut ayant une fréquence de coupure à 400 Hz pour obtenir un signal ŝ_hp (n), n = 0,···,63 - ce filtre passe-haut élimine l'influence des très basses fréquences qui peuvent biaiser l'estimation faite dans le bloc 104 - puis on calcule le « tilt » (indicateur de pente spectrale) noté e_tilt du signal ŝ_hp (n) par autocorrélation normalisée (bloc 104): $$e_{\mathit{tilt}}=\frac{{\displaystyle \sum _{n=1}^{63}{\widehat{s}}_{\mathit{hp}}\left(n\right){\widehat{s}}_{\mathit{hp}}\left(n-1\right)}}{{\displaystyle \sum _{n=0}^{63}{\widehat{s}}_{\mathit{hp}}{\left(n\right)}^2}}$$
  
  et enfin on calcule ĝ_HB sous la forme : $${\widehat{g}}_{\mathit{HB}}=w_{\mathit{SP}}{g}_{\mathit{SP}}+\left(1-w_{\mathit{SP}}\right)g_{\mathit{BG}}$$
  
  où g_SP =1-e_tilt est le gain appliqué dans les trames actives de parole (SP pour speech), g_BG =1.25g_SP est le gain appliqué dans les trames inactives de parole associées à un bruit de fond (BG pour Background) et w_SP est une fonction de pondération qui dépend de la détection d'activité vocale (VAD). On comprend que l'estimation du tilt ( e_tilt ) permet d'adapter le niveau de la bande haute en fonction de la nature spectrale du signal ; cette estimation est particulièrement importante quand la pente spectrale du signal décodé CELP est telle que l'énergie moyenne décroît quand la fréquence augmente (cas d'un signal voisé où e_tilt est proche de 1, donc g_SP =1-e_tilt est ainsi réduit). A noter aussi que le facteur ĝ_HB dans le décodage AMR-WB est borné pour prendre des valeurs dans l'intervalle [0.1, 1.0]. En fait, pour les signaux dont le spectre a plus d'énergie en hautes fréquences (e_tilt proche de -1, g_SP proche de 2), le gain ĝ_HB est d'habitude sous-estimé.

It can be noted here that the normalization of the energies is done by comparing blocks of different sizes (64 for u ( n ) and 80 for u _{HB 1} ( n )), without compensation for the differences in sampling frequencies (12.8 or 16 kHz).

• The excitation in the high band is then obtained (block 106 or 109) in the form: $$a_{\mathit{HB}}\left(\mathrm{not}\right)={\widehat{g}}_{\mathit{HB}}{a}_{\mathit{HB}2}\left(\mathrm{not}\right)$$
  
  where the gain ĝ _HB is obtained differently according to the flow. If the bit rate of the current frame is <23.85 kbit/s, the gain ĝ _HB is estimated “blind” (ie without additional information); in this case, block 103 filters the low-band decoded signal by a high-pass filter having a cut-off frequency of 400 Hz to obtain a signal ŝ _hp ( n ), n = 0.63 - this high-pass filter eliminates the influence of very low frequencies which can bias the estimate made in block 104 - then the "tilt" (indicator of spectral slope) denoted e _tilt of the signal ŝ _hp ( n ) by autocor is calculated normalized relationship (block 104): $$e_{\mathit{tilt}}=\frac{{\displaystyle \sum _{\mathrm{not}=1}^{63}{\widehat{s}}_{\mathit{hp}}\left(\mathrm{not}\right){\widehat{s}}_{\mathit{hp}}\left(\mathrm{not}-1\right)}}{{\displaystyle \sum _{\mathrm{not}=0}^{63}{\widehat{s}}_{\mathit{hp}}{\left(\mathrm{not}\right)}^2}}$$
  
  and finally we calculate ĝ _HB in the form: $${\widehat{g}}_{\mathit{HB}}=w_{\mathit{MS}}{g}_{\mathit{MS}}+\left(1-w_{\mathit{MS}}\right)g_{\mathit{BG}}$$
  
  where g _SP =1- e _tilt is the gain applied in active speech frames (SP for speech), g _BG =1.25 g _SP is the gain applied in inactive speech frames associated with background noise (BG for Background) and w _SP is a weighting function that depends on voice activity detection (VAD). It is understood that the estimation of the tilt ( e _tilt ) makes it possible to adapt the level of the high band according to the spectral nature of the signal; this estimate is particularly important when the spectral slope of the decoded signal CELP is such that the average energy decreases when the frequency increases (case of a voiced signal where e _tilt is close to 1, therefore g _SP = 1- e _tilt is thus reduced). Note also that the factor ĝ _HB in the AMR-WB decoding is bounded to take values in the interval [0.1, 1.0]. In fact, for signals whose spectrum has more energy at high frequencies ( e _tilt close to -1, g _SP close to 2), the gain ĝ _HB is usually underestimated.

At 23.85 kbit/s, correction information is transmitted by the AMR-WB coder and decoded (blocks 107, 108) in order to refine the estimated gain per subframe (4 bits every 5 ms, ie 0.8 kbit/s).

• A 6.6 kbit/s, le filtre 1/ A_HB (z) est obtenu en pondérant par un facteur γ=0.9 un filtre LPC d'ordre 20, 1 Â^ext (z) qui « extrapole » le filtre LPC d'ordre 16, 1/ Â(z), décodé dans la bande basse (à 12.8 kHz) - les détails de l'extrapolation dans le domaine des paramètres ISF (pour "Imittance Spectral Frequency" en anglais) sont décrits dans la norme G.722.2 à la section 6.3.2.1; dans ce cas, $$1/A_{\mathit{HB}}\left(z\right)=1/{\widehat{A}}^{\mathit{ext}}\left(z/\gamma \right)$$
  
• Aux débits > 6.6 kbit/s, le filtre 1/ A_HB(z) est d'ordre 16 et correspond simplement à : $$1/A_{\mathit{HB}}\left(z\right)=1/\widehat{A}\left(z/\gamma \right)$$
  
  où γ=0.6. A noter que dans ce cas le filtre 1/ Â(z / γ) est utilisé à 16 kHz, ce qui résulte en un étalement (par homothétie) de la réponse en fréquence de ce filtre de [0, 6.4 kHz] à [0, 8 kHz].

The artificial excitation u _HB ( n ) is then filtered (block 111) by an LPC synthesis filter with transfer function 1/ A _HB (z) and operating at the sampling frequency of 16 kHz. The realization of this filter depends on the rate of the current frame:

• At 6.6 kbit/s, the 1/ A _HB ( z ) filter is obtained by weighting by a factor γ = 0.9 an LPC filter of order 20, 1 Â ^ext (z) which "extrapolates" the LPC filter of order 16, 1/ Â ( z ), decoded in the low band (at 12.8 kHz) - the details of the extrapolation in the domain of the ISF parameters (for "Imittance Spectral Frequency" in English) are described in the G.722.2 standard in section 6.3.2.1; in this case, $$1/{\mathrm{AT}}_{\mathit{HB}}\left(z\right)=1/{\widehat{\mathrm{AT}}}^{\mathit{outside}}\left(z/\gamma \right)$$
  
• At bit rates > 6.6 kbit/s, the 1/ A _HB (z) filter is of order 16 and simply corresponds to: $$1/{\mathrm{AT}}_{\mathit{HB}}\left(z\right)=1/\widehat{\mathrm{AT}}\left(z/\gamma \right)$$
  
  where γ =0.6. Note that in this case the 1/ Â ( z / γ ) filter is used at 16 kHz, which results in a spreading (by homothety) of the frequency response of this filter from [0.6.4 kHz] to [0.8 kHz].

The result, s _HB ( n ) , is finally processed by a band-pass filter (block 112) of the FIR ("Finite Impulse Response") type, to keep only the 6-7 kHz band; at 23.85 kbit/s, a low-pass filter also of the FIR type (block 113) is added to the processing to further attenuate the frequencies above 7 kHz. The high frequency (HF) synthesis is finally added (block 130) to the low frequency (LF) synthesis obtained with blocks 120 to 123 and resampled at 16 kHz (block 123). Thus even if the high band extends in theory from 6.4 to 7 kHz in the AMR-WB codec, the HF synthesis is rather included in the 6-7 kHz band before addition with the LF synthesis.

• Le signal dans la bande haute est un bruit blanc mis en forme (par gains temporels par sous-trame, par filtrage par 1/ A_HB (z) et filtrage passe-bande), ce qui n'est pas un bon modèle général du signal dans la bande 6.4-7 kHz. Il existe par exemple des signaux de musique très harmoniques pour lesquels la bande 6.4-7 kHz contient des composantes sinusoïdales (ou tones) et aucun bruit (ou peu de bruit), pour ces signaux l'extension de bande du codée AMR-WB dégrade fortement la qualité.
• Le filtre passe-bas à 7 kHz (bloc 113) introduit un décalage de près de 1 ms entre les bandes basses et hautes, ce qui peut potentiellement dégrader la qualité de certains signaux en désynchronisant légèrement les deux bandes à 23.85 kbit/s - cette désynchronisation peut également poser problème lors d'une commutation de débit de 23.85 kbit/s à d'autres modes.
• L'estimation de gains par sous-trame (bloc 101, 103 à 105) n'est pas optimale. Pour partie, elle se base sur une égalisation de l'énergie « absolue » par sous-trame (bloc 101) entre des signaux à des fréquences différentes : l'excitation artificielle à 16 kHz (bruit blanc) et un signal à 12.8 kHz (excitation ACELP décodée). On peut noter en particulier que cette approche induit implicitement une atténuation de l'excitation bande haute (par un ratio 12.8/16=0.8) ; en fait, on notera également qu'aucune désaccentuation (ou déemphase) n'est effectuée sur la bande haute dans le codée AMR-WB, ce qui induit implicitement une amplification relative proche de 0.6 (qui correspond à la valeur de la réponse en fréquence de 1 / (1-0.68z^-1) à 6400 Hz). En fait, les facteurs de 1/0.8 et de 0.6 se compensent approximativement.
• Sur la parole, les tests de caractérisation du codée 3GPP AMR-WB documentés dans le rapport 3GPP TR 26.976 ont montré que le mode à 23.85 kbit/s a une qualité moins bonne qu'à 23.05 kbit/s, sa qualité est en fait similaire à celle du mode à 15.85 kbit/s. Ceci montre en particulier que le niveau du signal HF artificiel doit être contrôlé de façon très prudente, car la qualité est dégradée à 23.85 kbit/s alors que les 4 bits par trame sont sensés permettre de mieux approcher l'énergie des hautes fréquences originales.
• La limitation de la bande codée à 7 kHz résulte de l'application d'un modèle strict de la réponse en émission des terminaux acoustiques (filtre P.341 dans la norme UIT-T G.191). Or, pour une fréquence d'échantillonnage de 16 kHz, les fréquences dans la bande 7-8 kHz restent importantes, en particulier pour les signaux de musique, pour assurer un bon niveau de qualité.

We can identify several drawbacks to the band extension technique of the AMR-WB codec:

• The signal in the high band is shaped white noise (by temporal gains per subframe, by filtering by 1 / A _HB ( z ) and band-pass filtering), which is not a good general model of the signal in the 6.4-7 kHz band. There are for example very harmonic music signals for which the 6.4-7 kHz band contains sinusoidal components (or tones) and no noise (or little noise), for these signals the band extension of the AMR-WB codec strongly degrades the quality.
• The low pass filter at 7 kHz (block 113) introduces an almost 1 ms lag between the low and high bands, which can potentially degrade the quality of some signals by slightly desynchronizing the two bands at 23.85 kbit/s - this desynchronization can also cause problems when switching from 23.85 kbit/s bitrate to other modes.
• The estimation of gains per subframe (block 101, 103 to 105) is not optimal. In part, it is based on an “absolute” energy equalization per subframe (block 101) between signals at different frequencies: the artificial excitation at 16 kHz (white noise) and a signal at 12.8 kHz (decoded ACELP excitation). It can be noted in particular that this approach implicitly induces an attenuation of the high band excitation (by a ratio 12.8/16=0.8); in fact, it should also be noted that no de-emphasis (or de-emphasis) is performed on the high band in the AMR-WB codec, which implicitly induces a relative amplification close to 0.6 (which corresponds to the value of the frequency response of 1 / (1-0.68z ^-1 ) at 6400 Hz). In fact, the factors of 1/0.8 and 0.6 approximately compensate each other.
• On speech, the characterization tests of the 3GPP AMR-WB codec documented in the 3GPP report TR 26.976 showed that the 23.85 kbit/s mode has a lower quality than at 23.05 kbit/s, its quality is in fact similar to that of the 15.85 kbit/s mode. This shows in particular that the level of the artificial HF signal must be controlled in a very careful way, because the quality is degraded at 23.85 kbit/s whereas the 4 bits per frame are supposed to make it possible to better approach the energy of the original high frequencies.
• The limitation of the coded band to 7 kHz results from the application of a strict model of the transmission response of acoustic terminals (P.341 filter in the ITU-T G.191 standard). However, for a sampling frequency of 16 kHz, the frequencies in the 7-8 kHz band remain important, in particular for music signals, to ensure a good level of quality.

The AMR-WB decoding algorithm was improved in part with the development of the ITU-T G.718 scalable codec which was standardized in 2008.

The ITU-T G.718 standard includes a so-called interoperable mode, for which the core coding is compatible with the G.722.2 (AMR-WB) coding at 12.65 kbit/s; moreover, the G.718 decoder has the particularity of being able to decode an AMR-WB/G.722.2 binary train at all possible rates of the AMR-WB codec (from 6.6 to 23.85 kbit/s).

The interoperable G.718 decoder in low-delay mode (G.718-LD) is shown in Fig. figure 2 . Listed below are enhancements to the AMR-WB bitstream decoding functionality in the G.718 decoder, with references to the figure 1 when necessary:
The band expansion (described for example in clause 7.13.1 of recommendation G.718, block 206) is identical to that of the AMR-WB decoder, except that the 6-7 kHz bandpass filter and the 1/A _HB (z) synthesis filter (blocks 111 and 112) are in reverse order. Moreover, at 23.85 kbit/s the 4 bits transmitted by subframes by the AMR-WB coder are not used in the interoperable G.718 decoder; the synthesis of high frequencies (HF) at 23.85 kbit/s is therefore identical to 23.05 kbit/s, which avoids the known quality problem of AMR-WB decoding at 23.85 kbit/s. A fortiori, the low-pass filter at 7 kHz (block 113) is not used, and the specific decoding of the mode at 23.85 kbit/s is omitted (blocks 107 to 109).

A post-processing of the synthesis at 16 kHz (see clause 7.14 of G.718) is implemented in G.718 by " noise gate " in block 208 (to "improve" the quality of silences by level reduction), high-pass filtering (block 209), low-frequency post-filter (called " bass posfilier ") in block 210 attenuating inter-harmonic noise at low frequencies and a conversion to 16-bit integers with saturation control (with gain control or AGC) in block 211.

However, the band extension in the AMR-WB and/or G.718 codecs (interoperable mode) still remains limited in several aspects.

In particular, the synthesis of high frequencies by shaped white noise (by a temporal approach of the LPC source-filter type) is a very limited model of the signal in the band of frequencies above 6.4 kHz.

Only the 6.4-7 kHz band is re-synthesized artificially, while in practice a wider band (up to 8 kHz) is theoretically possible at the 16 kHz sampling frequency, which can potentially improve the quality of the signals, if they are not preprocessed by a P.341 type filter (50-7000 Hz) as defined in the Software Tool Library (G.191 standard) of the ITU-T.

The article " New Enhancements to the Audio Bandwidth Extension Toolkit (ABET)"by Anndana et al. describes a series of improvements to the frequency band extension tools (ASR, FSSM and MBTAC).

There is therefore a need to improve the band extension in an AMR-WB type codec or an interoperable version of this codec or more generally to improve the band extension of an audio signal, in particular to improve the frequency content of the band extension.

The present invention improves the situation.

The invention proposes for this purpose a method for extending the frequency band of an audio frequency signal during a decoding or improvement process comprising a step of obtaining the decoded signal in a first so-called low band frequency band.

The method is such that it includes the steps of claim 1.

It will be noted that subsequently the "band extension" will be taken in the broad sense and will include not only the case of the extension of a sub-band at high frequencies but also the case of a replacement of sub-bands set to zero (of the "noise filling" type in transform coding).

Thus, taking into account both tonal components and an ambient signal extracted from the signal resulting from the decoding of the low band makes it possible to carry out the band extension with a signal model adapted to the true nature of the signal unlike the use of artificial noise. The quality of the band extension is thus improved and in particular for certain types of signals such as music signals.

Indeed, the signal decoded in the low band comprises a part corresponding to the sound environment which can be transposed at high frequency in such a way that a mixing of the harmonic components and of the existing atmosphere makes it possible to ensure a coherent reconstructed high band.

It will be noted that even if the invention is motivated by the improvement of the quality of the band extension in the context of the interoperable AMR-WB coding, the various embodiments apply to the more general case of the band extension of an audio signal, in particular in an improvement device performing an analysis of the audio signal to extract the parameters necessary for the band extension.

The different particular embodiments mentioned below can be added independently or in combination with each other, to the steps of the extension method defined above.

In one embodiment, the band expansion is performed in the excitation domain and the decoded low band signal is a decoded low band excitation signal.

The advantage of this embodiment is that a transformation without windowing (or equivalently with an implicit rectangular window of the frame length) is possible in the excitation domain. In this case no artefact (block effects) is then audible.

• détection des composantes tonales dominantes du signal bande basse décodé ou décodé et étendu, dans le domaine fréquentiel ;
• calcul d'un signal résiduel par extraction des composantes tonales dominantes pour obtenir le signal d'ambiance.

In a first embodiment not covered by the text of the claims, the extraction of the tonal components and of the ambient signal is carried out according to the following steps:

• detection of the dominant tonal components of the decoded or decoded and extended low-band signal, in the frequency domain;
• calculating a residual signal by extracting the dominant tonal components to obtain the ambient signal.

This embodiment allows precise detection of the tonal components.

• obtention du signal d'ambiance par calcul d'une valeur moyenne du spectre du signal bande basse décodé ou décodé et étendu ;
• obtention des composantes tonales par soustraction du signal d'ambiance calculé au signal bande basse décodé ou décodé et étendu.

In a second embodiment, of low complexity, the extraction of the tonal components and of the ambient signal is carried out according to the following steps:

• obtaining the ambient signal by calculating an average value of the spectrum of the decoded or decoded and extended low-band signal;
• obtaining the tonal components by subtracting the calculated ambient signal from the decoded or decoded and extended low-band signal.

In one embodiment of the combining step, an energy level control factor used for the adaptive mixing is calculated based on the total energy of the decoded or decoded and extended low-band signal and the tonal components.

Applying this control factor allows the combining stage to adapt to signal characteristics to optimize the relative proportion of ambient signal in the mix. The energy level is thus controlled so as to avoid audible artifacts.

In a preferred embodiment, the decoded low-band signal undergoes a step of decomposition into sub-bands by transform or by filter bank, the extraction and combination steps then being performed in the frequency domain or in sub-bands.

The implementation of the band extension in the frequency domain makes it possible to obtain a precision of frequency analysis which is not available with a temporal approach, and also makes it possible to have a frequency resolution sufficient to detect the tonal components.

avec k l'indice de l'échantillon, U(k) le spectre du signal obtenu après une étape de transformée U _HB1(k) le spectre du signal étendu, et start_band une variable prédéfinie. Ainsi, cette fonction comprend un ré-échantillonnage du signal en ajoutant des échantillons au spectre de ce signal. D'autres façons d'étendre le signal sont cependant possibles, par exemple par translation dans un traitement sous-bandes.In a detailed embodiment, the decoded and extended low band signal is obtained according to the following equation: $$U_{\mathit{HB}1}\left(k\right)=\{\begin{array}{ll}0& k=0,\cdots ,199\\ U\left(k\right)& k=200,\dots ,239\\ U\left(k+\mathit{start}\_\mathit{band}-240\right)& k=240,\dots ,319\end{array}$$

with k the index of the sample, U(k) the spectrum of the signal obtained after a U _{HB 1} transform step ( k ) the spectrum of the extended signal, and start_band a predefined variable. Thus, this function includes a resampling of the signal by adding samples to the spectrum of this signal. Other ways of extending the signal are however possible, for example by translation in sub-band processing.

• un module d'extraction de composantes tonales et d'un signal d'ambiance à partir d'un signal issu du signal bande basse décodé;
• un module de combinaison des composantes tonales et du signal d'ambiance par mixage adaptatif utilisant des facteurs de contrôle de niveau d'énergie pour obtenir un signal audio, dit signal combiné;
• un module d'extension sur au moins une deuxième bande de fréquence supérieure à la première bande de fréquence mis en oeuvre sur le signal décodé bande basse avant le module d'extraction.

The present invention also relates to a device for extending the frequency band of an audio frequency signal, the signal having been decoded in a first frequency band called the low band. The device is such that it comprises:

• a module for extracting tonal components and an ambient signal from a signal coming from the decoded low-band signal;
• a module for combining the tonal components and the ambient signal by adaptive mixing using energy level control factors to obtain an audio signal, called the combined signal;
• an extension module on at least a second frequency band higher than the first frequency band implemented on the low band decoded signal before the extraction module.

This device has the same advantages as the method described previously, which it implements.

The invention relates to a decoder comprising a device as described.

It relates to a computer program comprising code instructions for implementing the steps of the band extension method as described, when these instructions are executed by a processor.

Finally, the invention relates to a storage medium, readable by a processor, integrated or not in the tape extension device, optionally removable, storing a computer program implementing a tape extension method as described previously.

• la figure 1 illustre une partie d'un décodeur de type AMR-WB mettant en oeuvre des étapes d'extension de bande de fréquence de l'état de l'art et tel que décrit précédemment;
• la figure 2 illustre un décodeur de type interopérable G.718-LD à 16kHz selon l'état de l'art et tel que décrit précédemment;
• la figure 3 illustre un décodeur interopérable avec le codage AMR-WB et intégrant un dispositif d'extension de bande selon un mode de réalisation de l'invention;
• la figure 4 illustre sous forme d'organigramme, les étapes principales d'un procédé d'extension de bande selon un mode de réalisation de l'invention;
• la figure 5 illustre un mode de réalisation dans le domaine fréquentiel d'un dispositif d'extension de bande selon l'invention intégré dans un décodeur; et
• la figure 6 illustre une réalisation matérielle d'un dispositif d'extension de bande selon l'invention.

Other characteristics and advantages of the invention will appear more clearly on reading the following description, given solely by way of non-limiting example, and made with reference to the appended drawings, in which:

• there figure 1 illustrates part of an AMR-WB type decoder implementing state-of-the-art frequency band extension steps and as described previously;
• there figure 2 illustrates a decoder of the interoperable G.718-LD type at 16 kHz according to the state of the art and as previously described;
• there picture 3 illustrates a decoder interoperable with AMR-WB coding and integrating a band extension device according to one embodiment of the invention;
• there figure 4 illustrates in the form of a flowchart, the main steps of a band extension method according to an embodiment of the invention;
• there figure 5 illustrates an embodiment in the frequency domain of a band extender device according to the invention integrated in a decoder; And
• there figure 6 illustrates a hardware realization of a tape extender according to the invention.

There picture 3 illustrates an example of a decoder, compatible with the AMR-WB/G.722.2 standard in which there is a post-processing similar to that introduced in G.718 and described with reference to figure 2 and an improved tape extension according to the extension method of the invention, implemented by the tape extension device illustrated by block 309.

Unlike AMR-WB decoding which works with an output sampling frequency of 16 kHz and G.718 decoding which works at 8 or 16 kHz, we consider here a decoder which can work with an output signal (synthesis) at the frequency fs = 8, 16, 32 or 48 kHz. Note that it is assumed here that the coding was carried out according to the AMR-WB algorithm with an internal frequency of 12.8 kHz for the CELP coding in the low band and at 23.85 kbit/s a gain coding per subframe at the frequency of 16 kHz, but interoperable variants of the AMR-WB coder are also possible; even if the invention is described here at the decoding level, it is assumed here that the coding can also operate with an input signal at the frequency fs = 8, 16, 32 or 48 kHz and appropriate resampling operations, going beyond the scope of the invention, are implemented at the coding according to the value of fs. It can be noted that when fs=8 kHz at the decoder, in the case of a decoding compatible with AMR-WB, it is not necessary to extend the low band 0-6.4 kHz, because the audio band reconstructed at the frequency fs is limited to 0-4000 Hz.

To the picture 3 , the CELP decoding (LF for low frequencies) always operates at the internal frequency of 12.8 kHz, as in AMR-WB and G.718, and the band extension (HF for high frequencies) which is the subject of the invention operates at the frequency of 16 kHz, the LF and HF synthesis are combined (block 312) at the frequency fs after adequate resampling (blocks 307 and 311). In variants of the invention, the combination of the low and high bands could be done at 16 kHz, after having resampled the low band from 12.8 to 16 kHz, before resampling the combined signal at the frequency fs.

• Démultiplexage des paramètres codés (bloc 300) en cas de trame correctement reçue (bfi=0 où bfi est le « bad trame indicator» valant 0 pour une trame reçue et 1 pour une trame perdue).
• Décodage des paramètres ISF avec interpolation et conversion en coefficients LPC (bloc 301) comme décrit dans la clause 6.1 de la norme G.722.2.
• Décodage de l'excitation CELP (bloc 302), avec une partie adaptative et fixe pour reconstruire l'excitation (exc ou u '(n)) dans chaque sous-trame de longueur 64 à 12.8 kHz: $$u\prime \left(n\right)={\widehat{g}}_p\nu \left(\mathrm{n}\right)+{\widehat{g}}_{c}c\left(n\right),n=0,\cdots ,63$$
  
  en suivant les notations de la clause 7.1.2.1 de G.718 concernant le décodage CELP, où v(n) et c(n) sont respectivement les mots de code des dictionnaires adaptatif et fixe, et ĝ_p et ĝ_c sont les gains décodés associés. Cette excitation u'(n) est utilisée dans le dictionnaire adaptatif de la sous-trame suivante ; elle est ensuite post-traitée et on distingue comme dans G.718 l'excitation u'(n) (aussi notée exc) de sa version post-traitée modifiée u(n) (aussi notée exc2) qui sert d'entrée au filtre de synthèse, 1/ Â(z), dans le bloc 303. Dans des variantes qui peuvent être mises en oeuvre pour l'invention, les post-traitements appliqués à l'excitation peuvent être modifiés (par exemple, la dispersion de phase peut être améliorée) ou ces post-traitements peuvent être étendus (par exemple, une réduction du bruit inter-harmonique peut être mise en oeuvre), sans affecter la nature du procédé d'extension de bande selon l'invention.
• Filtrage de synthèse par 1/ Â(z) (bloc 303) où le filtre LPC décodé A(z) est d'ordre 16
• Post-traitement bande étroite (bloc 304) selon la clause 7.3 de G.718 si fs=8 kHz.
• Désaccentuation (bloc 305) par le filtre 1/ (1 - 0.68z ^-1)
• Post-traitement des basses fréquences (bloc 306) tel que décrit à la clause 7.14.1.1 de G.718. Ce traitement introduit un retard qui est pris en compte dans le décodage de la bande haute (>6.4 kHz).
• Ré-échantillonnage de la fréquence interne de 12.8 kHz à la fréquence de sortie fs (bloc 307). Plusieurs réalisations sont possibles. Sans perte de généralité, on considère ici à titre d'exemple que si fs=8 ou 16 kHz, le ré-échantillonnage décrit dans la clause 7.6 de G.718 est repris ici, et si fs=32 ou 48 kHz, des filtres à réponse impulsionnelle finie (FIR) supplémentaires sont utilisés.
• Calcul des paramètres du "noise gate" (bloc 308) qui est réalisé de façon préférentielle comme décrit dans la clause 7.14.3 de G.718.

Decoding according to picture 3 depends on the AMR-WB mode (or rate) associated with the current frame received. As an indication and without this impacting the block 309, the decoding of the CELP part in low band comprises the following steps:

• Demultiplexing of the coded parameters (block 300) in the event of a correctly received frame ( bfi =0 where bfi is the “ bad frame indicator ” equal to 0 for a received frame and 1 for a lost frame).
• Decoding the ISF parameters with interpolation and conversion to LPC coefficients (block 301) as described in clause 6.1 of the G.722.2 standard.
• Decoding the CELP excitation (block 302), with an adaptive and fixed part to reconstruct the excitation (exc or u' ( n )) in each subframe of length 64 at 12.8 kHz: $$a\prime \left(\mathrm{not}\right)={\widehat{g}}_p\nu \left(\mathrm{not}\right)+{\widehat{g}}_{\mathrm{vs}}\mathrm{vs}\left(\mathrm{not}\right),\mathrm{not}=0,\cdots ,63$$
  
  following the notations of clause 7.1.2.1 of G.718 regarding CELP decoding, where v(n) and c ( n ) are the codewords of the adaptive and fixed dictionaries respectively, and ĝ _p and ĝ _c are the associated decoded gains. This excitation u' ( n ) is used in the adaptive dictionary of the next subframe; it is then post-processed and we distinguish, as in G.718, the excitation u'(n) (also denoted exc) from its modified post-processed version u(n) (also denoted exc2) which serves as input to the synthesis filter, 1/ Â(z), in block 303. In variants which can be implemented for the invention, the post-processings applied to the excitation can be modified (for example, the phase dispersion can be post-processings can be extended (for example, a reduction of the inter-harmonic noise can be implemented), without affecting the nature of the band extension method according to the invention.
• Synthesis filtering by 1/ Â (z) (block 303) where the decoded LPC filter A(z) is of order 16
• Narrowband post-processing (block 304) according to clause 7.3 of G.718 if fs=8 kHz.
• De-emphasis (block 305) by filter 1/ (1 - 0.68 z ^-1 )
• Low frequency post-processing (block 306) as described in clause 7.14.1.1 of G.718. This processing introduces a delay which is taken into account in the decoding of the high band (>6.4 kHz).
• Resampling the internal frequency of 12.8 kHz to the output frequency fs (block 307). Several realizations are possible. Without loss of generality, it is considered here as an example that if fs=8 or 16 kHz, the resampling described in clause 7.6 of G.718 is repeated here, and if fs =32 or 48 kHz, additional finite impulse response (FIR) filters are used.
• Calculation of the parameters of the " noise gate " (block 308) which is carried out preferentially as described in clause 7.14.3 of G.718.

In variants which can be implemented for the invention, the post-processings applied to the excitation can be modified (for example, the phase dispersion can be improved) or these post-processings can be extended (for example, a reduction of the inter-harmonic noise can be implemented), without affecting the nature of the band extension. The case of the decoding of the low band when the current frame is lost (bfi=1) which is informative in the 3GPP AMR-WB standard is not described here; in general, whether it is the AMR-WB decoder or a general decoder based on the source-filter model, it is typically a question of best estimating the LPC excitation and the coefficients of the synthesis LPC filter in order to reconstitute the lost signal while keeping the source-filter model. When bfi=1 it is considered here that the band extension (block 309) can operate as in the case bfi =0 and bit rate <23.85 kbit/s; thus, the description of the invention will hereafter assume and without loss of generality that bfi =0 .

It may be noted that the use of blocks 306, 308, 314 is optional.

It will also be noted that the decoding of the low band described above assumes a current so-called “active” frame with a bit rate between 6.6 and 23.85 kbit/s. In fact, when the DTX mode (continuous transmission in French) is activated, some frames can be coded as "inactive" and in this case you can either transmit a silence descriptor (on 35 bits) or transmit nothing. In particular, it is recalled that the SID frame of the AMR-WB coder describes several parameters: ISF parameters averaged over 8 frames, average energy over 8 frames, "dithering flag" for the reconstruction of non-stationary noise. In all cases, at the decoder, the same decoding model is found as for an active frame, with reconstruction of the excitation and of an LPC filter for the current frame, which makes it possible to apply the invention even to inactive frames. The same observation applies for the decoding of “lost frames” (or FEC, PLC) in which the LPC model is applied.

This exemplary decoder operates in the excitation domain and therefore includes a step for decoding the low-band excitation signal. The band extension device and the band extension method within the meaning of the invention also operate in a domain different from the domain of excitation and in particular with a direct signal decoded in low band or a signal weighted by a perceptual filter.

Unlike AMR-WB or G.718 decoding, the decoder described makes it possible to extend the decoded low band (50-6400 Hz taking into account the high-pass filtering at 50 Hz at the decoder, 0-6400 Hz in the general case) to an extended band whose width varies, ranging approximately from 50-6900 Hz to 50-7700 Hz depending on the mode implemented in the current frame. We can thus speak of a first frequency band from 0 to 6400Hz and of a second frequency band from 6400 to 8000Hz. In reality, in the preferred embodiment, the excitation for high frequencies is generated in the frequency domain in a band from 5000 to 8000 Hz, to allow a bandpass filtering of width 6000 to 6900 or 7700 Hz whose slope is not too steep in the upper rejected band.

The high band synthesis part is carried out in the block 309 representing the band extension device according to the invention and which is detailed in figure 5 in one embodiment.

In order to align the decoded low and high bands, a delay (block 310) is introduced to synchronize the outputs of blocks 306 and 309 and the synthesized 16 kHz high band is resampled by 16 kHz at frequency fs (output of block 311). The value of the delay T will have to be adapted for the other cases ( fs =32.48 kHz) according to the processing implemented. Remember that when fs =8 kHz, it is not necessary to apply blocks 309 to 311 because the band of the signal at the output of the decoder is limited to 0-4000 Hz.

It should be noted that the extension method of the invention implemented in block 309 according to the first embodiment preferentially does not introduce any additional delay with respect to the reconstructed low band at 12.8 kHz; however, in variants of the invention (for example using a time/frequency transformation with overlap), a delay may be introduced. Thus, generally the value of T in block 310 will need to be adjusted depending on the specific implementation. For example, in the case where the post-processing of low frequencies (block 306) is not used, the delay to be introduced for fs =16 kHz could be fixed at T =15.

The low and high bands are then combined (added) in block 312 and the synthesis obtained is post-processed by high-pass filtering at 50 Hz (of IIR type) of order 2 whose coefficients depend on the frequency fs (block 313) and output post-processing with optional application of the " noise gate " in a manner similar to G.718 (block 314).

The band extender according to the invention, illustrated by block 309 according to the decoder embodiment of the figure 5 , implements a band extension method (in the broad sense) described now with reference to the figure 4 .

This extension device can also be independent of the decoder and can implement the method described in figure 4 to carry out a band extension of an existing audio signal stored or transmitted to the device, with an analysis of the audio signal to extract therefrom for example an excitation and an LPC filter.

This device receives as input a decoded signal in a first frequency band called the low band u ( n ) which can be in the field of excitation or in that of the signal. In the embodiment described here, a step of decomposition into sub-bands (E401b) by time-frequency transform or bank of filters is applied to the decoded low-band signal to obtain the spectrum of the decoded low-band signal U ( k ) for implementation in the frequency domain.

A step E401a of extending the low band decoded signal into a second frequency band higher than the first frequency band, to obtain an extended low band decoded signal U _{HB 1} ( k ), can be performed on this low band decoded signal before or after the analysis step (decomposition into sub-bands). This extension step can comprise both a resampling step and an extension step or simply a frequency translation or transposition step as a function of the signal obtained as input. Note that in variants, step E401a may be performed at the end of the processing described in figure 4 ,, that is to say on the combined signal, this processing then being mainly carried out on the low band signal before extension, the result being equivalent.

This step is detailed later in the embodiment described with reference to the figure 5 .

A step E402 of extracting an ambient signal ( U _HBA ( k )) and tonal components (y(k)) is performed from the decoded low band signal ( U ( k )) or decoded and extended ( U _{HB 1} ( k )) . Ambiance is defined here as the residual signal which is obtained by removing the main (or dominant) harmonics (or tonal components) from the existing signal.

In most wideband signals (sampled at 16 kHz), the high band (>6 kHz) contains ambient information that is generally similar to that present in the low band.

• détection des composantes tonales dominantes du signal bande basse décodé (ou décodé et étendu), dans le domaine fréquentiel; et
• calcul d'un signal résiduel par extraction des composantes tonales dominantes pour obtenir le signal d'ambiance.

The step of extracting the tonal components and the ambient signal comprises for example the following steps:

• detection of the dominant tonal components of the decoded (or decoded and extended) low-band signal, in the frequency domain; And
• calculating a residual signal by extracting the dominant tonal components to obtain the ambient signal.

• obtention du signal d'ambiance par calcul d'une moyenne du signal bande basse décodé (ou décodé et étendu); et
• obtention des composantes tonales par soustraction du signal d'ambiance calculé au signal bande basse décodé (ou décodé et étendu).

This step can also be achieved by:

• obtaining the ambient signal by calculating an average of the decoded (or decoded and extended) low-band signal; And
• obtaining the tonal components by subtracting the calculated ambient signal from the decoded (or decoded and extended) low-band signal.

The tonal components and the ambient signal are then combined adaptively using energy level control factors in step E403 to obtain a so-called combined signal ( U _{HB 2} ( k )). The extension step E401a can then be implemented if it has not already been performed on the decoded low band signal.

Thus, the combination of these two types of signals makes it possible to obtain a combined signal with characteristics more suited to certain types of signals such as musical signals and richer in frequency content and in the extended frequency band corresponding to the entire frequency band including the first and the second frequency band.

Band extension according to the method improves the quality for this type of signal compared to the extension described in the AMR-WB standard.

The fact of using a combination of ambient signal and tonal components makes it possible to enrich this extension signal to make it closer to the characteristics of the real signal and not to an artificial signal.

This combination step will be detailed later with reference to the figure 5 .

A synthesis step, which corresponds to the analysis at 401b, is carried out at E404b to bring the signal back into the time domain.

Optionally, a step of adjusting the energy level of the high band signal can be performed in E404a, before and/or after the synthesis step, by applying a gain and/or by suitable filtering. This step will be explained in more detail in the embodiment described in figure 5 for blocks 501 to 507.

In an exemplary embodiment, tape expander 500 is now described with reference to figure 5 illustrating both this device but also processing modules suitable for implementation in a decoder of the interoperable type with AMR-WB coding. This device 500 implements the band extension method previously described with reference to the figure 4 .

Thus, processing block 510 receives a decoded low band signal ( u ( n )). In a particular embodiment, the band extension uses the decoded excitation at 12.8 kHz (exc2 or u ( n ) ) at the output of block 302 of the picture 3 .

This signal is decomposed into frequency sub-bands by the sub-band decomposition module 510 (which implements step E401b of the figure 4 ) which generally performs a transform or applies a bank of filters, to obtain a decomposition into sub-bands U(k) of the signal u(n).

où N = 256 et k = 0,···,255. In a particular embodiment, a DCT-IV type transform (for " Discrete Cosine Transform " - Type IV in English) (block 510) is applied to the current frame of 20 ms (256 samples), without windowing, which amounts to directly transforming u(n) with n = 0.255 according to the following formula: $$U\left(k\right)={\displaystyle \sum _{\mathrm{not}=0}^{\mathrm{NOT}-1}a\left(\mathrm{not}\right)\cos \left(\frac{\pi }{\mathrm{NOT}}\left(\mathrm{not}+\frac{1}{2}\right)\left(k+\frac{1}{2}\right)\right)}$$

where N = 256 and k = 0.255 .

A windowless transformation (or equivalently with an implicit rectangular window of the frame length) is possible when the processing is done in the excitation domain, not the signal domain. In this case, no artefact (block effects) is audible, which constitutes an important advantage of this embodiment of the invention.

In this embodiment, the DCT-IV transformation is implemented by FFT according to the so-called “ Evolved DCT (EDCT)” algorithm described in the article by DM Zhang, HT Li, A Low Complexity Transform - Evolved DCT, IEEE 14th International Conference on Computational Science and Engineering (CSE), Aug. 2011, p. 144-149 , and implemented in ITU-T G.718 Annex B and G.729.1 Annex E.

In variants of the invention and without loss of generality, the DCT-IV transformation may be replaced by other short-term time-frequency transformations of the same length and in the excitation domain or in the signal domain, such as an FFT (for " Fast Fourier Transform" ) or a DCT-II ( Discrete Cosine Transform - Type II). Alternatively, the DCT-IV on the frame can be replaced by a transformation with overlap-addition and windowing of length greater than the length of the current frame, for example by using an MDCT (for “ Modified Discrete Cosine Tranform ”). In this case the delay T in block 310 of the picture 3 , will have to be adjusted (reduced) in an adequate way according to the additional delay due to the analysis/synthesis by this transform.

In another embodiment, the decomposition into sub-bands is performed by applying a bank of filters, for example of real or complex PQMF (Pseudo-QMF) type. For certain banks of filters, one obtains, for each sub-band in a given frame, not a spectral value but a series of temporal values associated with the sub-band; in this case, the preferred embodiment in the invention can be applied by carrying out for example a transform of each sub-band and by calculating the ambient signal in the domain of absolute values, the tonal components always being obtained by difference between the signal (in absolute value) and the ambient signal. In the case of a complex filter bank, the complex modulus of the samples will replace the absolute value.

In other embodiments, the invention will be applied in a system using two sub-bands, the low band being analyzed by transform or by bank of filters.

où on prend de façon préférentielle start_band = 160.In the case of a DCT, the DCT spectrum, U(k), of 256 samples covering the band 0-6400 Hz (at 12.8 kHz), is then extended (block 511) into a spectrum of 320 samples covering the band 0-8000 Hz (at 16 kHz) in the following form: $$U_{\mathit{HB}1}\left(k\right)=\{\begin{array}{ll}0& k=0,\cdots ,199\\ U\left(k\right)& k=200,\dots ,239\\ U\left(k+\mathit{start}\_\mathit{band}-240\right)& k=240,\dots ,319\end{array}$$

where we preferentially take start_band = 160.

Block 511 implements step E401a of the figure 4 , that is to say the extension of the low band decoded signal. This step can also include resampling from 12.8 to 16 kHz in the frequency domain, by adding ¼ samples (k = 240,···,319) to the spectrum, the ratio between 16 and 12.8 being 5/4.

In the frequency band corresponding to the samples ranging from the indices 200 to 239, the original spectrum is kept, in order to be able to apply a progressive attenuation response of the high-pass filter in this frequency band and also so as not to introduce audible defects during the step of adding the low-frequency synthesis to the high-frequency synthesis.

It will be noted that in this embodiment, the generation of the oversampled extended spectrum is carried out in a frequency band ranging from 5 to 8 kHz, therefore including a second frequency band (6.4-8 kHz) higher than the first frequency band (0-6.4 kHz).

Thus, the extension of the decoded low band signal takes place at least on the second frequency band but also on part of the first frequency band.

Obviously, the values defining these frequency bands can be different depending on the decoder or the processing device in which the invention applies.

In addition, block 511 performs implicit high-pass filtering in the 0-5000 Hz band since the first 200 samples of U _{HB 1} ( k ) are set to zero; as explained later, this high-pass filtering can also be supplemented by a part of progressive attenuation of the spectral values of indices k =200,····,255 in the 5000-6400 Hz band, this progressive attenuation is implemented in block 501 but could be carried out separately outside of block 501. Equivalently and in variants of the invention, the implementation of the high-pass filtering separated into blocks of coefficients of index k=0, ···,199 set to zero, with coefficients k =200,····,255 attenuated, in the transformed domain, can therefore be carried out in a single step.

In this embodiment and according to the definition of U _{HB 1} ( k ), we note that the 5000-6000 Hz band of U _{HB 1} ( k ) (which corresponds to the indices k = 200, 239) is copied from the 5000-6000 Hz band of U(k). This approach preserves the original spectrum in this band and it avoids introducing distortions in the 5000-6000 Hz band when adding HF synthesis with LF synthesis - in particular the signal phase (implicitly represented in the DCT-IV domain) in this band is preserved.

The 6000-8000 Hz band of U _{HB 1} ( k ) is here defined by copying the 4000-6000 Hz band of U(k) since the value of start_band is preferentially fixed at 160.

In a variant of the embodiment, the value of start_band could be made adaptive around the value of 160, without modifying the nature of the invention. The details of the adaptation of the start_band value are not described here because they exceed the scope of the invention without changing the scope thereof.

In most wideband signals (sampled at 16 kHz), the high band (>6 kHz) contains ambient information that is naturally similar to that present in the low band. Ambiance is defined here as the residual signal which is obtained by removing the main (or dominant) harmonics from the existing signal. level of harmonicity in the 6000-8000 Hz band is generally correlated with that of lower frequency bands.

This decoded and extended low-band signal is supplied at the input of the extension device 500 and in particular at the input of the module 512. Thus the block 512 for extracting tonal components and an ambient signal implements step E402 of the figure 4 in the frequency domain. The ambient signal, U _HBA ( k ) for k =240.319 (80 samples) is thus obtained for a second so-called high frequency frequency band in order to then combine it adaptively with the extracted tonal components y(k), in the combination block 513.

• Calcul de l'énergie totale du signal bande basse décodé étendu ener_HB : $${\mathit{ener}}_{\mathit{HB}}={\displaystyle \sum _{k=240}^{319}{U}_{\mathit{HB}1}{\left(k\right)}^2+\epsilon }$$
  
  où ε=0.1 (cette valeur peut être différente, elle est fixée ici à titre d'exemple).
• Calcul de l'ambiance (en valeur absolue) qui correspond ici au niveau moyen du spectre lev(i) (raie par raie) et calcul de l'énergie ener_tonal des parties tonales dominantes (dans le spectre hautes fréquences)

In a particular embodiment, the extraction of the tonal components and of the ambient signal (in the 6000-8000 Hz band) is carried out according to the following operations:

• Calculation of the total energy of the extended decoded low band signal ener _HB : $${\mathit{ener}}_{\mathit{HB}}={\displaystyle \sum _{k=240}^{319}{U}_{\mathit{HB}1}{\left(k\right)}^2+\epsilon }$$
  
  where ε=0.1 (this value can be different, it is set here as an example).
• Calculation of the ambience (in absolute value) which here corresponds to the average level of the lev ( i ) spectrum (line by line) and calculation of the ener _tonal energy of the dominant tonal parts (in the high frequency spectrum)

For i = 0... L - 1, this average level is obtained by the following equation: $$\mathit{up}\left(I\right)=\frac{1}{\mathit{fn}\left(I\right)-\mathit{fb}\left(I\right)+1}{\displaystyle \sum _{I=\mathit{fb}\left(I\right)}^{\mathit{fn}\left(I\right)}\left|U_{\mathit{HB}1}\left(I+240\right)\right|}$$

This corresponds to the average level (in absolute value) and therefore represents a kind of envelope of the spectrum. In this embodiment, L =80 and represents the length of the spectrum and the index i from 0 to L -1 corresponds to the indices j +240 from 240 to 319, ie the spectrum from 6 to 8 kHz.

$$\mathit{fb}\left(i\right)=i-7\mathrm{et}\mathit{fn}\left(i\right)=L-1\mathrm{pour}i=L-7,\cdots ,\mathrm{L}-1$$

In general fb ( i ) = i -7 and fn ( i ) = i + 7, however the first and last 7 indices ( i = 0,···,6 and i = L - 7,···, L-1) require special treatment and without loss of generality we then define: $$\mathit{fb}\left(I\right)=0\mathrm{And}\mathit{fn}\left(I\right)=I+7\mathrm{For}I=0,\cdots ,6$$

$$\mathit{fb}\left(I\right)=I-7\mathrm{And}\mathit{fn}\left(I\right)=I-1\mathrm{For}I=I-7,\cdots ,\mathrm{I}-1$$

In variants of the invention, the average of | U _{HS 1} ( j +240)|, j = jb ( i ),..., fn ( i ) , can be replaced by a median value on the same set of values, i.e. lev ( i ) = median _{j=fb ( i ),..., fn ( i )} (| U _{HB 1} ( j +240)| This variant has the drawback of being more complex (in terms of number of calculations) than a moving average. In other variants, a non-uniform weighting may be applied to the averaged terms, or the median filtering may be replaced, for example, by other nonlinear filters of the “ stack filters ” type.

qui correspond (approximativement) aux composantes tonales si la valeur y(i) à une raie i donnée est positive ( y(i) >0).We also calculate the residual signal: $$\mathrm{there}\left(I\right)=\left|U_{\mathit{HB}1}\left(I+240\right)\right|-\mathit{up}\left(I\right),I=0,\dots ,I-1$$

which corresponds (approximately) to the tonal components if the value y(i) at a given line i is positive (y( i ) >0).

This calculation therefore involves an implicit detection of the tonal components. The tonal parts are therefore implicitly detected using the intermediate term y(i) representing an adaptive threshold. The detection condition being y( i ) >0. In variants of the invention, this condition may be changed for example by defining an adaptive threshold depending on the local envelope of the signal or in the form y( i )> lev ( i )+ xdB where x has a predefined value (for example x =10 dB).

The energy of the dominant tonal parts is defined by the following equation: $${\mathit{ener}}_{\mathit{tonal}}={\displaystyle \sum _{I=0\dots 7\left|\mathrm{there}\left(I\right)\right|>0}\mathrm{there}{\left(I\right)}^2}$$

Other methods of extracting the ambient signal can of course be envisaged. For example, this ambient signal can be extracted from a low frequency signal or possibly another frequency band (or several frequency bands).

The detection of the peaks or tonal components could be done differently.

The extraction of this ambient signal could also be done on the decoded excitation but not extended, that is to say before the spectral extension or translation step, that is to say for example on a portion of the low frequency signal rather than directly on the high frequency signal.

• détection des composantes tonales dominantes du signal bande basse décodé (ou décodé et étendu), dans le domaine fréquentiel ;
• calcul d'un signal résiduel par extraction des composantes tonales dominantes pour obtenir le signal d'ambiance.

In a variant embodiment, the extraction of the tonal components and of the ambient signal is carried out in a different order and according to the following steps:

• detection of the dominant tonal components of the decoded (or decoded and extended) low-band signal, in the frequency domain;
• calculating a residual signal by extracting the dominant tonal components to obtain the ambient signal.

pour i = 0,...,L - 1. Dès qu'un pic est détecté à la raie d'indice i on applique un modèle sinusoïdal afin d'estimer les paramètres d'amplitude, de fréquence et éventuellement de phase d'une composante tonale associé à ce pic. Les détails de cette estimation ne sont pas présentés ici mais l'estimation de la fréquence peut typiquement faire appel à une interpolation parabolique sur 3 points afin de localiser le maximum de la parabole approximant les 3 points d'amplitude |U _HB1(i+240)| (ramené en dB), l'estimation d'amplitude étant obtenu par le biais de cette même interpolation. Le domaine par transformée utilisé ici (DCT-IV) ne permettant pas d'obtenir directement la phase, on pourra dans un mode de réalisation, négliger ce terme, mais dans des variantes on pourra appliquer une transformée en quadrature de type DST pour estimer un terme de phase. La valeur initiale de y(i) est mise à zéro pour i = 0,...,L-1 . Les paramètres sinusoïdaux (fréquence, amplitude, et éventuellement phase) de chaque composante tonale étant estimés, on calcule alors le terme y(i) comme la somme de prototypes (spectres) prédéfinis de sinusoïdes pures transformées dans le domaine DCT-IV (ou autre si une autre décomposition en sous-bandes est utilisée) selon les paramètres sinusoïdaux estimés. Enfin, on applique une valeur absolue aux termes y(i) pour se ramener au domaine du spectre d'amplitude en valeurs absolues.This variant can for example be implemented as follows: A peak (or tonal component) is detected at a line of index i in the amplitude spectrum | U _{HB 1} ( i +240)| if the following criterion is verified: $$\left|U_{\mathit{HB}1}\left(I+240\right)\right|>\left|U_{\mathit{HB}1}\left(I+240-1\right)\right|\mathrm{And}\left|U_{\mathit{HB}1}\left(I+240\right)\right|>\left|U_{\mathit{HB}1}\left(I+240+1\right)\right|,$$

for i = 0,..., L - 1 . As soon as a peak is detected at the line of index i, a sinusoidal model is applied in order to estimate the amplitude, frequency and possibly phase parameters of a tonal component associated with this peak. The details of this estimation are not presented here but the frequency estimation can typically use a 3-point parabolic interpolation to locate the maximum of the parabola approximating the 3 amplitude points | U _{HB 1} ( i +240)| (brought back into dB), the amplitude estimate being obtained by means of this same interpolation. As the transform domain used here (DCT-IV) does not make it possible to directly obtain the phase, it is possible in one embodiment, to neglect this term, but in variants it is possible to apply a quadrature transform of the DST type to estimate a phase term. The initial value of y( i ) is set to zero for i = 0,..., L -1 . The sinusoidal parameters (frequency, amplitude, and possibly phase) of each tonal component being estimated, the term y( i ) is then calculated as the sum of predefined prototypes (spectra) of pure sinusoids transformed in the DCT-IV domain (or other if another decomposition into sub-bands is used) according to the estimated sinusoidal parameters. Finally, an absolute value is applied to the terms y( i ) to reduce to the domain of the amplitude spectrum in absolute values.

Other methods of determining the tonal components are possible, for example it would also be possible to calculate an envelope of the signal env ( i ) by interpolation by splines of the local maximum values (detected peaks) of | U _{HB 1} ( i +240)| , to lower this envelope by a certain level in dB to detect tonal components such as peaks exceeding this envelope and define y( i ) as $$\mathrm{there}\left(I\right)=\max \left(\left|U_{\mathit{HB}1}\left(I+240\right)\right|-\mathit{approx}\left(I\right),0\right)$$

In this variant, the atmosphere is therefore obtained by the equation: $$\mathit{up}\left(I\right)=\left|U_{\mathit{HB}1}\left(I+240\right)\right|-\mathrm{there}\left(I\right),I=0,\dots ,I-1$$

In other variants of the invention, the absolute value of the spectral values will be replaced, for example, the square of the spectral values, without changing the principle of the invention; in this case a square root will be necessary to return to the signal domain, which is more complex to achieve.

β étant un facteur dont un exemple de calcul est donné ci-dessous.The combining module 513 performs a combining step by adaptive mixing of the ambient signal and the tonal components. To do this, an ambient level control factor Γ is defined by the following equation: $$\mathrm{\Gamma }=\beta \frac{{\mathit{ener}}_{\mathit{HB}}-{\mathit{ener}}_{\mathit{tonal}}}{{\mathit{ener}}_{\mathit{HB}}-{\mathit{\beta ener}}_{\mathit{tonal}}}$$

β being a factor for which an example of calculation is given below.

auquel on applique les signes de U _HB1(k) : $$\mathrm{y}"\left(i\right)=\mathrm{sgn}\left(U_{\mathit{HB}1}\left(i+240\right)\right).y\prime \left(i\right)$$

où la fonction sgn (.) donne le signe : $$\mathrm{sgn}\left(x\right)=\{\begin{array}{cc}1& x\ge 0\\ -1& x<0\end{array}$$

To obtain the extended signal, we first obtain the combined signal in absolute values for i = 0...L-1: $$\mathrm{there}\prime \left(I\right)=\{\begin{array}{cc}\mathit{\Gamma y}\left(I\right)+\frac{1}{\mathrm{\Gamma }}\mathit{le\nu }\left(I\right)& \mathrm{there}\left(I\right)>0\\ \mathrm{there}\left(I\right)+\frac{1}{\mathrm{\Gamma }}\mathit{le\nu }\left(I\right)& \mathrm{there}\left(I\right)\le 0\end{array}$$

to which we apply the signs of U _{HB 1} ( k ): $$\mathrm{there}"\left(I\right)=\mathrm{sgn}\left(U_{\mathit{HB}1}\left(I+240\right)\right).\mathrm{there}\prime \left(I\right)$$

where the sgn (.) function gives the sign: $$\mathrm{sgn}\left(x\right)=\{\begin{array}{cc}1& x\ge 0\\ -1& x<0\end{array}$$

By definition the factor Γ is > 1. The tonal components, detected line by line by the condition y ( i ) > 0, are reduced by the factor Γ; the average level is amplified by the factor 1/Γ.

In adaptive mixing block 513, an energy level control factor is calculated based on the total energy of the decoded (or decoded and extended) low-band signal and the tonal components.

U _HB2(k) étant le signal combiné d'extension de bande.In a preferred embodiment of adaptive mixing, the energy adjustment is performed as follows: $$U_{\mathit{HB}2}\left(k\right)=\mathit{college}.\mathrm{there}"\left(k-240\right),k=240,\cdots ,319$$

U _{HB 2} ( k ) being the combined band-expanding signal.

The adjustment factor is defined by the following equation: $$\mathit{college}=\gamma \sqrt{\frac{{\mathit{ener}}_{\mathit{HB}}}{{\displaystyle \sum _{I=0}^{I-1}\mathrm{there}"\left(I\right)}}}$$

$$E_{N4-6}={\displaystyle \sum _{k\in \mathrm{N}\left(\mathrm{160,239}\right)}U{\prime }^2\left(k\right)}$$

$$E_{N4-6}={\displaystyle \sum _{k\in \mathrm{N}\left(\mathrm{240,319}\right)}U{\prime }^2\left(k\right)}$$

où $$U\prime \left(k\right)=\{\begin{array}{cc}\sqrt{\frac{{\displaystyle \sum _{k=160}^{239}{U}^2\left(k\right)}}{{\displaystyle \sum _{k=80}^{159}{U}^2\left(k\right)}}U\left(k\right)}& k=80,\dots ,159\\ U\left(k\right)& k=160,\dots ,239\\ \sqrt{\frac{{\displaystyle \sum _{k=160}^{239}{U}^2\left(k\right)}}{{\displaystyle \sum _{k=240}^{319}{U_{\mathit{HB}1}}^2\left(k\right)}}}{U}_{\mathit{HB}1}\left(k\right)& k=240,\dots ,319\end{array}$$

Where γ avoids an overestimation of the energy. In an exemplary embodiment, β is calculated so as to keep the same ambient signal level with respect to the energy of the tonal components in the consecutive bands of the signal. We calculate the energy of the tonal components in three bands: 2000-4000 Hz, 4000-6000 Hz and 6000-8000 Hz, with $$E_{\mathrm{NOT}2-4}={\displaystyle \sum _{k\in \mathrm{NOT}\left(80.159\right)}U{\prime }^2\left(k\right)}$$

$$E_{\mathrm{NOT}4-6}={\displaystyle \sum _{k\in \mathrm{NOT}\left(\mathrm{160,239}\right)}U{\prime }^2\left(k\right)}$$

$$E_{\mathrm{NOT}4-6}={\displaystyle \sum _{k\in \mathrm{NOT}\left(\mathrm{240,319}\right)}U{\prime }^2\left(k\right)}$$

Or $$U\prime \left(k\right)=\{\begin{array}{cc}\sqrt{\frac{{\displaystyle \sum _{k=160}^{239}{U}^2\left(k\right)}}{{\displaystyle \sum _{k=80}^{159}{U}^2\left(k\right)}}U\left(k\right)}& k=80,\dots ,159\\ U\left(k\right)& k=160,\dots ,239\\ \sqrt{\frac{{\displaystyle \sum _{k=160}^{239}{U}^2\left(k\right)}}{{\displaystyle \sum _{k=240}^{319}{{\mathrm{<figure-callout\; id="1"\; label="U\; HB"\; filenames="imgf0001.png,imgf0002.png"\; state="\{\{state\}\}">U</figure-callout>}}_{\mathit{HB}}}^2\left(k\right)}}}{U}_{\mathit{HB}1}\left(k\right)& k=240,\dots ,319\end{array}$$

And where N( k ₁ , k ₂ ) is the set of indices k for which the coefficient of index k is classified as being associated with the tonal components. This set can be for example obtained by detecting the local peaks in U' ( k ) verifying | U ( k )| > lev ( k ) or lev ( k ) is calculated as the average level of the spectrum line by line.

It may be noted that other methods of calculating the energy of the tonal components are possible, for example by taking the median value of the spectrum on the band considered.

où $$E_{N4-6}=\max \left(E_{N4-6},E_{N2-4}\right),\rho =\frac{E_{N4-6}^2}{E_{N2-4}},\rho =\max \left(\rho ,E_{N6-8}\right)$$

et max(.,.) est la fonction qui donne le maximum des deux arguments.We fix β so that the ratio between the energy of the tonal components in the 4-6 kHz and 6-8 kHz bands is the same as between the 2-4 kHz and 4-6 kHz bands: $$\beta =\frac{\rho -E_{\mathrm{NOT}6-8}}{{\displaystyle \sum _{k=160}^{239}{U}^2\left(k\right)-E_{\mathrm{NOT}6-8}}}$$

Or $$E_{\mathrm{NOT}4-6}=\max \left(E_{\mathrm{NOT}4-6},E_{\mathrm{NOT}2-4}\right),\rho =\frac{E_{\mathrm{NOT}4-6}^2}{E_{\mathrm{NOT}2-4}},\rho =\max \left(\rho ,E_{\mathrm{NOT}6-8}\right)$$

and max(.,.) is the function that returns the maximum of the two arguments.

In variants of the invention, the calculation of β could be replaced by other methods. For example, in a variant, we can extract (calculate) different parameters (or "features" in English) characterizing the signal in low band, including a "tilt" parameter similar to that calculated in the AMR-WB codec, and we will estimate the factor β as a function of a linear regression from these various parameters by limiting its value between 0 and 1. The linear regression could for example be estimated in a supervised way by estimating the factor β by giving the original high band in a learning base. It will be noted that the mode of calculation of β does not limit the nature of the invention.

on pourra calculer γ comme une fonction décroissante de α, par exemple $\gamma =b-a\sqrt{\alpha }$

, b = 1.1, a = 1.2 et γ limité de 0.3 à 1. Là encore d'autres définitions de α et γ sont possibles dans le cadre de l'invention.Then, the parameter β can be used to calculate γ taking into account that a signal with an added ambient signal in a given band is generally perceived as stronger than a harmonic signal at the same energy in the same band. If we define α as the quantity of ambient signal added to the harmonic signal: $$\alpha =\sqrt{1-\beta }$$

we can calculate γ as a decreasing function of α , for example $\gamma =b-\mathrm{To}\sqrt{\alpha }$

, b =1.1, a =1.2 and γ limited from 0.3 to 1. Here again other definitions of α and γ are possible within the framework of the invention.

At the output of the band extender 500, the block 501, in a particular embodiment, optionally performs a double operation of applying bandpass filter frequency response and de-emphasis filtering (or de-emphasis) in the frequency domain.

In a variant of the invention, the de-emphasis filtering could be performed in the time domain, after block 502 or even before block 510; however, in this case, the band-pass filtering performed in block 501 may leave some low frequency components of very low levels which are amplified by de-emphasis, which may change the decoded low band slightly perceptibly. For this reason, it is preferred here to carry out the de-emphasis in the frequency domain. In the preferred embodiment, the coefficients of index k = 0.199 are set to zero, so the de-emphasis is limited to the higher coefficients.

où G_deemph (k) est la réponse en fréquence du filtre 1/(1-0.68z ^-1) sur une bande de fréquence discrète restreinte. En prenant en compte les fréquences discrètes (impaires) de la DCT-IV, on définit ici G_deemph (k) comme: $$G_{\mathit{deemph}}\left(k\right)=\frac{1}{\left|e^{{\mathit{j\theta }}_k}-0.68\right|},k=0,\cdots ,255$$

où $${\theta }_k=\frac{256-80+k+\frac{1}{2}}{256}.$$

The excitation is first de-emphasized according to the following equation: $$U_{\mathit{HB}2}\prime \left(k\right)=\{\begin{array}{ll}0& k=0,\cdots ,199\\ G_{\mathit{deemph}}\left(k-200\right)U_{\mathit{HB}2}\left(k\right)& k=200,\cdots ,255\\ G_{\mathit{deemph}}\left(55\right)U_{\mathit{HB}2}\left(k\right)& k=256,\cdots ,319\end{array}$$

where G _deeph ( k ) is the frequency response of the 1/(1-0.68 z ^-1 ) filter over a restricted discrete frequency band. Considering the discrete (odd) frequencies of the DCT-IV, here we define G _deeph ( k ) as: $$G_{\mathit{deemph}}\left(k\right)=\frac{1}{\left|e^{{\mathit{j\theta }}_k}-0.68\right|},k=0,\cdots ,255$$

Or $${\theta }_k=\frac{256-80+k+\frac{1}{2}}{256}.$$

If a transformation other than DCT-IV is used, the definition of θ _k can be adjusted (for example for even frequencies).

Note that the de-emphasis is applied in two phases for k = 200, , 255 corresponding to the 5000-6400 Hz frequency band, where the response 1/(1-0.68 z ^-1 ) is applied as at 12.8 kHz, and for k = 256, , 319 corresponding to the 6400-8000 Hz frequency band, where the response is extended from 16 kHz here to a constant value in the 6.4-8 kHz band.

It can be noted that in the AMR-WB codec the HF synthesis is not de-emphasized. In the embodiment presented here, the high-frequency signal is on the contrary de-emphasized so as to bring it back into a coherent domain with the low-frequency signal (0-6.4 kHz) which comes out of block 305 of the picture 3 . This is important for estimating and later adjusting the energy of HF synthesis.

In a variant of the embodiment, in order to reduce the complexity, it is possible to fix G _deeph ( k ) to a constant value independent of k, by taking for example G _deeph ( k ) = 0.6 which corresponds approximately to the average value of G _deeph ( k ) for k = 200.319 under the conditions of the embodiment described above.

In another variant of the embodiment of the decoder, the de-emphasis could be carried out in an equivalent manner in the time domain after inverse DCT.

In addition to de-emphasis, band-pass filtering is applied with two separate parts: one fixed high-pass, the other adaptive low-pass (depending on the bit rate).

This filtering is performed in the frequency domain.

où N_lp =60 à 6.6 kbit/s, 40 à 8.85 kbit/s, 20 aux débits >8.85 bit/s. Ensuite on applique un filtre passe-bande sous la forme : $$U_{\mathit{HB}3}\left(k\right)=\{\begin{array}{ll}0& k=0,\cdots ,199\\ G_{\mathit{hp}}\left(k-200\right)U_{\mathit{HB}2}\prime \left(k\right)& k=200,\cdots ,255\\ U_{\mathit{HB}2}\prime \left(k\right)& k=256,\cdots ,319-N_{\mathrm{lp}}\\ G_{\mathit{lp}}\left(k-320-N_{\mathit{lp}}\right)U_{\mathit{HB}2}\prime \left(k\right)& k=320-N_{\mathrm{lp}},\cdots ,319\end{array}$$

In the preferred embodiment, the low-pass filter partial response in the frequency domain is calculated as follows: $$G_{\mathit{lp}}\left(k\right)=1-0.999\frac{k}{{\mathrm{NOT}}_{\mathit{lp}}-1}$$

where N _lp =60 at 6.6 kbit/s, 40 at 8.85 kbit/s, 20 at rates >8.85 bit/s. Then we apply a band-pass filter in the form: $$U_{\mathit{HB}3}\left(k\right)=\{\begin{array}{ll}0& k=0,\cdots ,199\\ G_{\mathit{hp}}\left(k-200\right)U_{\mathit{HB}2}\prime \left(k\right)& k=200,\cdots ,255\\ U_{\mathit{HB}2}\prime \left(k\right)& k=256,\cdots ,319-{\mathrm{NOT}}_{\mathrm{lp}}\\ G_{\mathit{lp}}\left(k-320-{\mathrm{NOT}}_{\mathit{lp}}\right)U_{\mathit{HB}2}\prime \left(k\right)& k=320-{\mathrm{NOT}}_{\mathrm{lp}},\cdots ,319\end{array}$$

The definition of G _hp ( k ), k = 0.55, is given for example in Table 1 below. <b>Table 1</b> K _ghp ( k ) K g _hp ( k) K _ghp ( k ) k _ghp ( k ) 0 0.001622428 14 0.114057967 28 0.403990611 42 0.776551214 1 0.004717458 15 0.128865425 29 0.430149896 43 0.800503267 2 0.008410494 16 0.144662643 30 0.456722014 44 0.823611104 3 0.012747280 17 0.161445005 31 0.483628433 45 0.845788355 4 0.017772424 18 0.179202219 32 0.510787115 46 0.866951597 5 0.023528982 19 0.197918220 33 0.538112915 47 0.887020781 6 0.030058032 20 0.217571104 34 0.565518011 48 0.905919644 7 0.037398264 21 0.238133114 35 0.592912340 49 0.923576092 8 0.045585564 22 0.259570657 36 0.620204057 50 0.939922577 9 0.054652620 23 0.281844373 37 0.647300005 51 0.954896429 10 0.064628539 24 0.304909235 38 0.674106188 52 0.968440179 11 0.075538482 25 0.328714699 39 0.700528260 53 0.980501849 12 0.087403328 26 0.353204886 40 0.726472003 54 0.991035206 13 0.100239356 27 0.378318805 41 0.751843820 55 1.000000000

It will be noted that in variants of the invention the values of G _hp ( k ) can be modified while keeping a progressive attenuation. Likewise, the low-pass filtering with variable bandwidth, G _lp ( k ) , can be adjusted with different values or a frequency support, without changing the principle of this filtering step.

It will also be noted that the band-pass filtering can be adapted by defining a single filtering step combining high-pass and low-pass filtering.

In another embodiment, the band-pass filtering could be carried out in an equivalent way in the time domain (as in block 112 of the figure 1 ) with different filter coefficients depending on the bit rate, after an inverse DCT step. However, it will be noted that it is advantageous to carry out this step directly in the frequency domain since the filtering is carried out in the domain of the LPC excitation and therefore the problems of circular convolution and of edge effects are very limited in this domain.

où N _16k = 320 et k = 0,···,319.The inverse transform block 502 performs an inverse DCT on 320 samples to find the high-frequency signal sampled at 16 kHz. Its implementation is identical to block 510, because the DCT-IV is orthonormal, except that the length of the transform is 320 instead of 256, and we obtain: $$a_{\mathit{HB}}\left(\mathrm{not}\right)={\displaystyle \sum _{k=0}^{{\mathrm{NOT}}_{16k}-1}{U}_{\mathit{HB}3}\left(\mathrm{k}\right)\cos \left(\frac{\pi }{{\mathrm{NOT}}_{16k}}\left(k+\frac{1}{2}\right)\left(\mathrm{not}+\frac{1}{2}\right)\right)}$$

where N _{16 k} = 320 and k = 0.319.

In the case where block 510 is not a DCT, but another transformation or decomposition into sub-bands, block 502 performs the synthesis corresponding to the analysis performed in block 510.

The 16 kHz sampled signal is then optionally scaled by defined gains per 80-sample subframe (block 504).

où $$e_1\left(m\right)={\displaystyle \sum _{n=0}^{63}u{\left(n+64m\right)}^2+\epsilon }$$

$$e_2\left(m\right)={\displaystyle \sum _{n=0}^{79}{u}_{\mathit{HB}}{\left(n+80m\right)}^2+\epsilon }$$

$$e_3\left(m\right)=e_1\left(m\right)\frac{{\displaystyle \sum _{n=0}^{319}{u}_{\mathit{HB}}{\left(n\right)}^2+\epsilon }}{{\displaystyle \sum _{n=0}^{255}u{\left(n\right)}^2+\epsilon }}$$

avec ε = 0.01. On peut écrire le gain par sous-trame g _HB1(m) sous la forme : $$g_{\mathit{HB}1}\left(m\right)=\sqrt{\frac{\frac{{\displaystyle \sum _{n=0}^{63}u{\left(n+64m\right)}^2+\epsilon }}{{\displaystyle \sum _{n=0}^{255}u{\left(n\right)}^2+\epsilon }}}{\frac{{\displaystyle \sum _{n=0}^{79}{u}_{\mathit{HB}}{\left(n+80m\right)}^2+\epsilon }}{{\displaystyle \sum _{n=0}^{319}{u}_{\mathit{HB}}{\left(n\right)}^2+\epsilon }}}}$$

ce qui montre qu'on assure dans le signal u_HB le même ratio entre énergie par sous-trame et énergie par trame que dans le signal u(n) . In a preferred embodiment, a gain g _HB1 (m) is first calculated (block 503) per subframe by energy ratios of the subframes such that in each subframe of index m =0 , 1, 2 or 3 of the current frame: $$g_{\mathit{HB}1}\left(m\right)=\sqrt{\frac{e_3\left(m\right)}{e_2\left(m\right)}}$$

Or $$e_1\left(m\right)={\displaystyle \sum _{\mathrm{not}=0}^{63}a{\left(\mathrm{not}+64m\right)}^2+\epsilon }$$

$$e_2\left(m\right)={\displaystyle \sum _{\mathrm{not}=0}^{79}{a}_{\mathit{HB}}{\left(\mathrm{not}+80m\right)}^2+\epsilon }$$

$$e_3\left(m\right)=e_1\left(m\right)\frac{{\displaystyle \sum _{\mathrm{not}=0}^{319}{a}_{\mathit{HB}}{\left(\mathrm{not}\right)}^2+\epsilon }}{{\displaystyle \sum _{\mathrm{not}=0}^{255}a{\left(\mathrm{not}\right)}^2+\epsilon }}$$

with ε = 0.01. We can write the gain per subframe g _{HB 1} ( m ) in the form: $$g_{\mathit{HB}1}\left(m\right)=\sqrt{\frac{\frac{{\displaystyle \sum _{\mathrm{not}=0}^{63}a{\left(\mathrm{not}+64m\right)}^2+\epsilon }}{{\displaystyle \sum _{\mathrm{not}=0}^{255}a{\left(\mathrm{not}\right)}^2+\epsilon }}}{\frac{{\displaystyle \sum _{\mathrm{not}=0}^{79}{a}_{\mathit{HB}}{\left(\mathrm{not}+80m\right)}^2+\epsilon }}{{\displaystyle \sum _{\mathrm{not}=0}^{319}{a}_{\mathit{HB}}{\left(\mathrm{not}\right)}^2+\epsilon }}}}$$

which shows that the same ratio between energy per subframe and energy per frame is ensured in the signal _u HB as in the signal u(n).

Block 504 performs the scaling of the combined signal (included in step E404a of the figure 4 ) according to the following equation: $$a_{\mathit{HB}}\prime \left(\mathrm{not}\right)=g_{\mathit{HB}1}\left(m\right)a_{\mathit{HB}}\left(\mathrm{not}\right),\mathrm{not}=80m,\dots ,80\left(m+1\right)-1$$

It will be noted that the realization of block 503 differs from that of block 101 of the figure 1 , because the energy at the level of the current frame is taken into account in addition to that of the subframe. This makes it possible to have the ratio of the energy of each subframe compared to the energy of the frame. Energy ratios (or relative energies) are therefore compared rather than the absolute energies between low band and high band.

Thus, this scaling step makes it possible to preserve in the high band the energy ratio between the subframe and the frame in the same way as in the low band.

où le gain g _HB2(m) est obtenu à partir du bloc 505 en exécutant les blocs 103, 104 et 105 du codée AMR-WB (l'entrée du bloc 103 étant l'excitation décodée en bande basse, u(n) ). Les blocs 505 et 506 sont utiles pour ajuster le niveau du filtre de synthèse LPC (bloc 507), ici en fonction du tilt du signal. D'autres méthodes de calcul du gain g _HB2(m) sont possibles sans changer la nature de l'invention.Optionally, block 506 then performs the scaling of the signal (included in step E404a of the figure 4 ) according to the following equation: $$a_{\mathit{HB}}"\left(\mathrm{not}\right)=g_{\mathit{HB}2}\left(m\right)a_{\mathit{HB}}\prime \left(\mathrm{not}\right),\mathrm{not}=80m,\cdots ,80\left(m+1\right)-1$$

where the gain g _{HB 2} ( m ) is obtained from block 505 by executing blocks 103, 104, and 105 of the AMR-WB codec (the input to block 103 being the low-band decoded excitation, u ( n )). Blocks 505 and 506 are useful for adjusting the level of the LPC synthesis filter (block 507), here according to the tilt of the signal. Other methods of calculating the gain g _{HB 2} ( m ) are possible without changing the nature of the invention.

Finally, the signal, u _HB '( n ) or u _HB "( n ), is filtered by the filtering module 507 which can be carried out here by taking as transfer function 1/ Â (z/ γ ), where γ =0.9 at 6.6 kbit/s and γ =0.6 at other bit rates, which limits the order of the filter to order 16.

In a variant, this filtering could be carried out in the same way as what is described for block 111 of the figure 1 of the AMR-WB decoder, however the order of the filter changes to 20 at the rate of 6.6, which does not significantly change the quality of the synthesized signal. In another variant, the LPC synthesis filtering can be performed in the frequency domain, after having calculated the frequency response of the filter implemented in block 507.

In variant embodiments of the invention, the coding of the low band (0-6.4 kHz) could be replaced by a CELP coder other than that used in AMR-WB, such as for example the CELP coder in G.718 at 8 kbit/s. Without loss of generality, other wideband coders or coders operating at frequencies above 16 kHz, in which the low band coder operates at an internal frequency of 12.8 kHz could be used. Furthermore, the invention can obviously be adapted to sampling frequencies other than 12.8 kHz, when a low-frequency coder operates at a sampling frequency lower than that of the original or reconstructed signal. When the low-band decoding does not use linear prediction, there is no excitation signal to extend, in this case it is possible to carry out an LPC analysis of the reconstructed signal in the current frame and an LPC excitation will be calculated so as to be able to apply the invention.

Finally, in another variant of the invention, the excitation or the low band signal ( u ( n )) is resampled, for example by linear interpolation or cubic "spline", from 12.8 to 16 kHz before transformation (for example DCT-IV) of length 320. This variant has the defect of being more complex, because the transform (DCT-IV) of the excitation or of the signal is then calculated over a greater length and the resampling is not carried out in the domain of the transform.

Moreover, in variants of the invention, all the calculations necessary for estimating the gains ( G _HBN , g _{HB 1} ( m ) , g _{HB 2} ( m ) , g _HBN , etc. can be performed in a logarithmic domain.

There figure 6 shows an exemplary hardware embodiment of a band extender device 600 according to the invention. This may be an integral part of an audio frequency signal decoder or of equipment receiving decoded or undecoded audio frequency signals.

This type of device comprises a processor PROC cooperating with a memory block BM comprising a storage and/or working memory MEM.

Such a device comprises an input module E capable of receiving an audio signal decoded or extracted in a first frequency band called the low band brought back into the frequency domain ( U ( k )). It comprises an output module S able to transmit the extension signal in a second frequency band ( U _{HB 2} ( k )) for example to a filter module 501 of the figure 5 .

The memory block can advantageously comprise a computer program comprising code instructions for implementing the steps of the band extension method within the meaning of the invention, when these instructions are executed by the processor PROC, and in particular the steps of extracting (E402) tonal components and of an ambient signal from a signal originating from the decoded low-band signal ( U ( k )), of combining (E403) the tonal components (y(k)) and the ambient signal ( U _HBA ( k )) by adaptive mixing using energy level control factors to obtain an audio signal, called combined signal ( U _{HB 2} ( k )), of extension (E401a) over at least a second frequency band higher than the first frequency band of the decoded low band signal before the extraction step or of the combined signal after the combining step.

Typically, the description of the figure 4 repeats the steps of an algorithm of such a computer program. The computer program can also be stored on a memory medium that can be read by a reader of the device or that can be downloaded into the memory space of the latter.

The memory MEM generally records all the data necessary for the implementation of the method.

In one possible embodiment, the device thus described may also comprise the low band decoding functions and other processing functions described for example in figure 5 And 3 in addition to the band extension functions according to the invention.

Claims (9)

1. Method for extending the frequency band of an audio frequency signal during a decoding or improvement process comprising a step for obtaining the decoded signal in a first frequency band called the low band, the method being characterized in that it includes the following steps:

   - extending (E401a) over at least a second frequency band higher than the first frequency band the decoded low band signal to form an extended decoded low band signal U_HB1(k), k representing the samples covering the U_HB1(k) spectrum;

   - extracting (E402) tonal components and of an ambient signal from the signal coming from the extended decoded low-band signal;

   - combining (E403) the tonal components and the ambient signal by adaptive mixing using energy level control factors to obtain a combined signal;

   - synthesizing (E404b) an audio signal to bring a signal from the combined signal back into the time domain; and wherein the step of extracting (E402) the tonal components and the ambient signal includes the following operations:

   (a) calculating the total energy of the extended decoded low band signal;

   (b) calculating the ambience in absolute value corresponding to the average level of the spectrum line by line and calculating the energy of the dominant tonal parts in the high frequency spectrum.
2. The method according to claim 1, wherein step (a) of calculating the total energy of the extended decoded low band signal comprises calculating: $${\mathit{ener}}_{\mathit{HB}}={\displaystyle \sum _{k=240}^{319}{U}_{\mathit{HB}1}{\left(k\right)}^2+\epsilon }$$

   

   where ε=0.1.
3. The method according to claim 1 or 2, wherein the average level of the spectrum line by line is obtained by the equation: $$\mathit{lev}\left(i\right)=\frac{1}{\mathit{fn}\left(i\right)-\mathit{fb}\left(i\right)+1}{\displaystyle \sum _{i=\mathit{fb}\left(i\right)}^{\mathit{fn}\left(i\right)}\left|U_{\mathit{HB}1}\left(j+240\right)\right|}$$

   

   Where fb(i) = 0 and fn(i)= i+7 for i=0,...,6

   fb(i)= i-7 and fn(i)= i+7 for i=7,...,L-8

   fb(i)= i-7 and fn(i)= L-l for i=L-7,. ,L-1,

   where L is the length of the spectrum.
4. The method according to claim 1, 2 or 3, wherein the calculation of the energy of the dominant tonal components comprises the calculation of the residual signal: $$y\left(i\right)=\left|U_{\mathit{HB}1}\left(i+240\right)0\right|-\mathit{lev}\left(i\right),i=0,\dots L-1.$$

   
5. The method according to claim 4, comprising a step of detecting the tonal components based on a detection condition on the residual signal y(i).
6. The method according to claim 5, wherein the detection condition is y(i)>0.
7. The method according to claim 6, wherein the energy of the dominant tonal components is defined by $${\mathit{ener}}_{\mathit{tonal}}={\displaystyle \sum _{i=0\dots 7\left|y\left(i\right)\right|>0}y{\left(i\right)}^2}$$

   
8. A device for extending the frequency band of an audio frequency signal, the signal having been decoded in a first frequency band called the low band, the device being characterized in that it comprises:

   - an extension module (511) on at least a second frequency band higher than the first frequency band implemented on the decoded low band signal to form an extended decoded low band signal U_HB1 (k), k representing the samples covering the U_HB1(k) spectrum,

   - a module for extracting (512) tonal components and an ambient signal from a signal coming from the extended decoded low band signal;

   - a module for combining (513) the tonal components and the ambient signal by adaptive mixing using energy level control factors to obtain a combined audio signal;

   - a module for synthesizing (502) an audio signal to bring the combined signal back into the time domain;

   and wherein the module for extracting (512) the tonal components and the ambient signal is adapted to carry out the following operations:

   (a) calculating the total energy of the extended decoded low band signal;

   (b) calculating the ambience in absolute value corresponding to the average level of the spectrum line by line and calculating the energy of the dominant tonal parts in the high frequency spectrum.
9. An audio-frequency signal decoder characterized in that it comprises a frequency band extension device in accordance with claim 8.

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