EP3103116A1

EP3103116A1 - Improved frequency band extension in an audio signal decoder

Info

Publication number: EP3103116A1
Application number: EP15705687.0A
Authority: EP
Inventors: Magdalena KANIEWSKA; Stéphane RAGOT
Original assignee: Orange SA
Current assignee: Koninklijke Philips NV
Priority date: 2014-02-07
Filing date: 2015-02-04
Publication date: 2016-12-14
Anticipated expiration: 2035-02-04
Also published as: BR122017027991B1; KR20220035271A; CN107993667A; JP6775065B2; FR3017484A1; FI3330966T3; RU2016136008A; CN108109632B; KR20160119150A; BR112016017616A2; DK3103116T3; LT3330966T; EP3330966B1; EP3330967A1; RU2763481C2; HUE062979T2; US20200338917A1; US11325407B2; JP6775063B2; RU2763848C2

Abstract

The invention relates to a method for extending the frequency band of an audio signal during a decoding or improvement process, comprising a step of obtaining the decoded signal in a first frequency band, referred to as a low band. The method is such that it comprises the steps of: extracting (E402) tonal components and a surround signal from a signal from the low-band signal, combining (E403) tonal components and the surround signal by adaptive mixing using energy-level control factors to obtain an audio signal, referred to as a combined signal, extending (E401a) the low-band decoded signal before the extraction step or the combined signal after the combination step over at least one second frequency band which is higher than the first frequency band. The invention also relates to a frequency-band extension device which implements the described method and to a decoder comprising a device of this type.

Description

Improved frequency band extension in an audio-frequency decoder

The present invention relates to the field of coding / decoding and audio-frequency signal processing (such as speech, music or other signals) for their transmission or storage.

More particularly, the invention relates to a method and a device for extending the frequency band in a decoder or a processor performing an audio-frequency signal improvement.

Many techniques exist to compress (with loss) an audiofrequency signal such as speech or music.

Conventional methods of coding for conversational applications are generally classified in waveform coding (MIC for "pulse modulation and coding", ADPCM for "Pulse Modulation and Adaptive Differential Coding, transform coding, etc.), parametric coding (LPC for "Linear Predictive Coding" in English, sinus coding ...) and parametric hybrid coding with a quantification of the parameters by "analysis by synthesis" whose coding CELP (for "Code Excited Linear Prediction" in English) is the best known example.

For non-conversational applications, the state of the art audio signal coding (mono) consists of perceptual encoding by transform or subband, with parametric high frequency band replication coding (SBR for Spectral).

Band Replication in English).

A review of conventional speech and audio coding methods can be found in W. B. Kleijn and K. K. Paliwal (Eds.), Speech Coding and Synthesis, Elsevier, 1995; M. Bosi, R. E. Goldberg, Introduction to Digital Audio Coding and Standards, Springer 2002; J. Benesty, M. Sondhi M., Y. Huang (Eds.), Handbook of Speech Processing, Springer 2008.

Of particular interest here is the 3GPP AMR-WB ("Adaptive Multi-Rate Wideband") codec and decoder, which operates at an input / output frequency of 16 kHz and in which the signal is divided into two sub-bands, the low band (0-6.4 kHz) which is sampled at 12.8 kHz and coded by CELP model and the high band (6.4-7 kHz) which is parametrically reconstructed by "band extension" ( or BWE for "Bandwidth Extension" with or without additional information depending on the mode of the current frame. It can be noted here that the limitation of the coded band of the AMR-WB codec at 7 kHz is essentially related to the fact that the transmission frequency response of the broadband terminals has been approximated at the time of standardization (ETSI / 3GPP then ITU-T T) according to the frequency mask defined in the ITU-T P.341 standard and more precisely by using a so-called "P341" filter defined in the ITU-T G.191 standard. which cuts frequencies above 7 kHz (this filter respects the mask defined in P.341). However, in theory, it is well known that a signal sampled at 16 kHz may have a defined audio band of 0 to 8000 Hz; the AMR-WB codec thus introduces a limitation of the high band in comparison with the theoretical bandwidth of 8 kHz.

The 3GPP AMR-WB speech codec was standardized in 2001 mainly for circuit-mode (CS) telephony applications on GSM (2G) and UMTS (3G). This same codec was also standardized in 2003 in ITU-T as Recommendation G.722.2 "Wideband coding speech at around 16kbit / s using Adaptive Multi-Rate Wideband (AMR-WB)".

It includes nine speeds, called modes, from 6.6 to 23.85 kbit / s, and includes continuous transmission mechanisms (DTX for "Discontinuous Transmission") with voice activity detection (VAD for "Voice Activity Detection") and noise generation of comfort (CNG for "Comfort Noise Generation") from frames of silence description (SID for "Silence Insertion Descriptor"), as well as mechanisms of correction of lost frames (FEC for "Frame Erasure Concealment", sometimes called PLC for "Packet Loss Concealment").

The details of the AMR-WB coding and decoding algorithm are not repeated here, a detailed description of this codec is found in the 3GPP specifications (TS 26.190, 26.191, 26.192, 26.193, 26.194, 26.204) and ITU-TG .722.2 (and the corresponding Appendices and Appendix) and in the article by B. Bessette et al. entitled "The adaptive multirate broadband speech coded (AMR-WB)", IEEE Transactions on Speech and Audio Processing, vol. 10, no. 8, 2002, pp. 620-636 and associated 3GPP and ITU-T standard source codes.

The principle of band extension in the AMR-WB codec is rather rudimentary. Indeed, the high band (6.4-7 kHz) is generated by formatting a white noise through a temporal envelope (applied in the form of gains per subframe) and frequency (by the application of a linear prediction synthesis filter or LPC for "Linear Predictive Coding"). This band extension technique is illustrated in Figure 1.

A white noise, u _HB1 (n), n = 0, · · ·, 79, is generated at 16 kHz per 5 ms subframe per linear congruent generator (block 100). This noise u _HB1 (n) is shaped in time by applying gains per subframe; this operation is broken down into two processing steps (blocks 102, 106 or 109):

• A first factor is calculated (block 101) to set the white noise u _HB1 (n) (block

102) at a level similar to that of the excitation, u (n), "= (), · · -, 63, decoded at 12.8 kHz in the low band:

It can be noted here that the normalization of the energies is done by comparing blocks of different size (64 for u (n) and 80 for u _HB1 (n)), without compensation of the differences of sampling frequencies (12.8 or 16 kHz) .

• The excitation in the high band is then obtained (block 106 or 109) in the form:

u _HB {n) = g _HB u _HB2 {n)

where the gain g _HB is obtained differently depending on the flow rate. If the rate of the current frame is <23.85 kbit / s, the gain g _HB is estimated to be "blind" (ie without additional information); in this case, the block 103 filters the low-band decoded signal by a high-pass filter having a cut-off frequency at 400 Hz to obtain a signal s _hp (n), "= (), · -, - 63 - high-pass filter eliminates the influence of the very low frequencies that can skew the estimate made in the block 104 - then the "tilt" (spectral slope indicator) noted e _tilt of the signal s _hp (n) by standard autocorrelation is calculated (block 104):

and finally we calculate g _HB in the form:

S m = ^w spSsp + ^ - w _SP ) g _BG

where g _sp = l - e _tilt is the gain applied in active speech frames (SP for speech), g _BG = l .25g _sp is the gain applied in inactive speech frames associated with background noise (BG for Background) and w _sp is a weighting function that depends on Voice Activity Detection (VAD). It is understood that the estimate of the tilt (e _tilt ) makes it possible to adapt the level of the high band according to the spectral nature of the signal; this estimate is particularly important when the spectral slope of the decoded signal CELP is such that the average energy decreases when the frequency increases (case of a voiced signal where e _tilt is close to 1, so g _sp = l - e _tilt is thus reduced). Note also that the factor g _HB in the decoding AMR-WB is bounded to take values in the interval [0.1, 1.0]. In fact, for the signals whose spectrum has more energy at high frequencies {e _tiU close to -1, g _sp close to 2), the gain g _HB is usually underestimated.

At 23.85 kbit / s, correction information is transmitted by the encoder AMR-WB and decoded (blocks 107, 108) in order to refine the estimated gain per subframe (4 bits every 5ms, ie 0.8 kbit / s) .

Artificial _HB excitation u (n) is then filtered (block 111) by a LPC synthesis filter transfer function \ l A _HB (z) and operating at the sampling frequency of 16 kHz. The realization of this filter depends on the rate of the current frame:

• At 6.6 kbit / s, the filter \ l A _HB (z) is obtained by weighting by a factor ^ = 0.9 an LPC filter of order 20, l / À ^ex '(z) which "extrapolates" the LPC filter of order 16, I l A (z), decoded in the low band (at 12.8 kHz) - the details of the extrapolation in the domain of the ISF parameters (for "Imittance Spectral Frequency") are described in standard G .722.2 in section 6.3.2.1; in that case,

• At rates> 6.6 kbit / s, the filter 1 / A _HB (z) is of order 16 and simply corresponds to:

where ^ = 0.6. Note that in this case the filter 1 / λ {ζ Ι) is used at 16 kHz, which results in a spread (by homothety) of the frequency response of this filter from [0, 6.4 kHz] to [0, 8 kHz].

The result, s _HB (n), is finally processed by a band-pass filter (block 112) of type FIR ("Finite

Impulse Response "), to keep only the band 6 - 7 kHz, at 23.85 kbit / s, a low-pass filter also FIR type (block 113) is added to the treatment to further attenuate frequencies above 7 kHz. The synthesis at high frequencies (HF) is finally added (block 130) to the low frequency synthesis (BF) obtained with the blocks 120 to 123 and resampled at 16 kHz (block 123). theoretically extends from 6.4 to 7 kHz in the AMR-WB coding, the HF synthesis is rather in the band 6-7 kHz before addition with the synthesis BF.

Several disadvantages can be identified with the AMR-WB coding extension technique:

• The signal in the high band is white noise formatted (by temporal gains per subframe, filtering by 1 / A _HB (z) and bandpass filtering), which is not a good general pattern of the signal in the 6.4-7 kHz band. For example, there are very harmonic music signals for which the 6.4-7 kHz band contains sinusoidal components (or tones) and no noise (or little noise), for these signals the band extension of the AMR-WB coding degrades. strongly the quality.

• The 7 kHz low-pass filter (block 113) introduces an offset of nearly 1 ms between the low and high bands, which can potentially degrade the quality of some signals by slightly desynchronizing the two bands at 23.85 kbit / s - this desynchronization can also be problematic when switching from 23.85 kbit / s to other modes.

• The estimate of gains per subframe (block 101, 103 to 105) is not optimal. In part, it is based on an equalization of the "absolute" energy per sub-frame (block 101) between signals at different frequencies: the artificial excitation at 16 kHz (white noise) and a signal at 12.8 kHz ( ACELP excitation decoded). It can be noted in particular that this approach implicitly induces an attenuation of the high band excitation (by a ratio 12.8 / 16 = 0.8); in fact, it will also be noted that no deemphasis is performed on the high band in the AMR-WB coding, which implicitly induces a relative amplification close to 0.6 (which corresponds to the value of the frequency response s / (l - 0.68z ^_1) to 6400 Hz).

In fact, the factors of 1 / 0.8 and 0.6 compensate each other approximately.

• On speech, the 3GPP AMR-WB coding characterization tests documented in the 3GPP TR 26.976 report showed that the 23.85 kbit / sa mode is not as good as 23.05 kbit / s, its quality is in fact similar to that of the 15.85 kbit / s mode. This shows in particular that the level of artificial RF signal must be controlled very carefully, because the quality is degraded to 23.85 kbit / s while the 4 bits per frame are supposed to better approach the energy of the original high frequencies.

• The limitation of the 7 kHz coded band results from the application of a strict model of the emission response of acoustic terminals (filter P.341 in ITU-T G.191). However, for a sampling frequency of 16 kHz, frequencies in the band 7-8 kHz remain important, especially for music signals, to ensure a good level of quality.

The AMR-WB decoding algorithm has been improved in part with the development of the ITU-T G.718 scalable codec which was standardized in 2008.

ITU-T G.718 includes an interoperable mode, for which core coding is compatible with 12.65 kbit / s G.722.2 (AMR-WB) coding; in addition, the G.718 decoder has the particularity of being able to decode a bit stream AMR-WB / G.722.2 at all possible bit rates of the AMR-WB codec (from 6.6 to 23.85 kbit / s). The G.718 interoperable decoder in low delay mode (G.718-LD) is illustrated in FIG. 2. Below are the improvements made to the bit-stream decoding functionality AMR- WB in the G.718 decoder, with references to Figure 1 when necessary:

The band extension (described for example in clause 7.13.1 of Recommendation G.718, block 206) is identical to that of the AMR-WB decoder, except that the 6-7 kHz band-pass filter and the Synthesis 1 / A _H B (Z) (blocks 111 and 112) are in reverse order. In addition, at 23.85 kbit / s the 4 bits transmitted by AMR-WB encoder subframes are not used in the interoperable G.718 decoder; the synthesis of high frequencies (HF) at 23.85 kbit / s is therefore identical to 23.05 kbit / s which avoids the known problem of quality of AMR-WB decoding at 23.85 kbit / s. A fortiori, the low-pass filter at 7 kHz (block 113) is not used, and the specific decoding mode 23.85 kbit / s is omitted (blocks 107 to 109).

A post-processing of the 16 kHz synthesis (see clause 7.14 of G.718) is implemented in G.718 by "noise gate" in block 208 (to "improve" the quality of silences by reducing the level) , high-pass filtering (block 209), low-frequency post filter (so-called "bass posfilter") in block 210 attenuating inter-harmonic noise at low frequencies and conversion to 16-bit integers with saturation control (with control Gain or AGC) in block 211. However, the band extension in AMR-WB and / or G.718 codecs (interoperable mode) is still limited in several respects.

In particular, the synthesis of high frequencies by shaped white noise (by a temporal approach of the LPC source-filter type) is a very limited model of the signal in the frequency band above 6.4 kHz.

Only the 6.4-7 kHz band is artificially re-synthesized, whereas in practice a wider band (up to 8 kHz) is theoretically possible at the sampling frequency of 16 kHz, which can potentially improve the quality of the signals, if they are not pretreated by a P.341 (50-7000 Hz) filter as defined in the ITU-T Software Tool Library (Standard G.191).

There is therefore a need to improve the band extension in an AMR-WB type codec or an interoperable version of this codec or more generally to improve the band extension of an audio signal, in particular to improve the frequency content of the band extension.

The present invention improves the situation.

To this end, the invention proposes a method of extending the frequency band of an audiofrequency signal during a decoding or improvement process comprising a step of obtaining the decoded signal in a first low band frequency band. The method is such that it comprises the following steps:

extracting tonal components and a room signal from a signal derived from the decoded low band signal;

- combining the tonal components and the ambient signal by adaptive mixing using energy level control factors to obtain an audio signal, said combined signal;

extension on at least one second frequency band higher than the first frequency band of the low band decoded signal before the extraction step or the combined signal after the combining step.

It will be noted that subsequently "band extension" will be taken in the broad sense and will include not only the case of the extension of a subband at high frequencies but also the case of a replacement of subbands set to zero (of type "noise filling" in transform coding).

Thus, both the taking into account of tonal components and a surround signal extracted from the signal resulting from the decoding of the low band makes it possible to perform the band extension with a signal model adapted to the true nature of the band. signal contrary to the use of artificial noise. The quality of the band extension is thus improved and in particular for certain types of signals such as music signals.

Indeed, the signal decoded in the low band has a part corresponding to the sound environment that can be transposed into high frequency so that a mix of harmonic components and the existing environment ensures a high band reconstructed consistent.

It will be noted that even if the invention is motivated by the improvement of the quality of the band extension in the context of the interoperable AMR-WB coding, the different embodiments apply to the more general case of the extension of band of an audio signal, in particular in an enhancement device performing an analysis of the audio signal to extract the parameters necessary for the band extension.

The various particular embodiments mentioned below may be added independently or in combination with each other, to the steps of the extension method defined above.

In one embodiment, the band extension is performed in the field of excitation and the decoded low band signal is a decoded low band excitation signal.

The advantage of this embodiment is that a transformation without windowing (or equivalently with an implicit rectangular window of the length of the frame) is possible in the field of excitation. In this case no artifact (block effects) is audible. In a first embodiment, the extraction of the tonal components and the ambient signal is carried out according to the following steps:

detection of the dominant tone components of the decoded or decoded and extended bass band signal, in the frequency domain;

calculation of a residual signal by extraction of the dominant tonal components to obtain the ambient signal.

This embodiment allows accurate detection of tonal components.

In a second embodiment, of low complexity, the extraction of the tonal components and the ambient signal is carried out according to the following steps:

obtaining the ambient signal by calculating an average value of the spectrum of the decoded or decoded and extended low band signal;

obtaining the tonal components by subtracting the calculated ambient signal from the decoded or decoded and extended low band signal.

In one embodiment of the combining step, a power level control factor used for adaptive mixing is calculated based on the total energy of the decoded or decoded and extended low band signal and the tonal components.

The application of this control factor allows the combining step to adapt to the characteristics of the signal to optimize the relative proportion of the ambient signal in the mixture. The energy level is thus controlled to avoid audible artifacts.

In a preferred embodiment, the decoded low band signal undergoes a step of subband decomposition by transform or filter bank, the extraction and combining steps then taking place in the frequency domain or in sub-bands. .

The implementation of the band extension in the frequency domain makes it possible to obtain a fineness of frequency analysis which is not available with a temporal approach, and also makes it possible to have a frequency resolution sufficient to detect the tonal components. .

In a detailed embodiment, the decoded and extended low band signal is obtained according to the following equation: U _HBl (k) =

with k the index of the sample, U (k) the spectrum of the signal obtained after a transform step U _HB1 (k) the spectrum of the extended signal, and start_band a predefined variable.

Thus, this function includes a re-sampling of the signal by adding samples to the spectrum of this signal. Other ways of extending the signal are however possible, for example by translation in a sub-band processing. The present invention also relates to a frequency band extension device of an audiofrequency signal, the signal having been decoded in a first so-called low band frequency band. The device is such that it comprises:

a module for extracting tonal components and a room signal from a signal derived from the decoded low band signal;

a combination module of the tonal components and the ambient signal by adaptive mixing using energy level control factors to obtain an audio signal, called a combined signal;

an extension module on at least a second frequency band greater than the first frequency band implemented on the decoded low band signal before the extraction module or on the combined signal after the combination module.

This device has the same advantages as the method described above, which it implements.

The invention relates to a decoder comprising a device as described.

It is directed to a computer program comprising code instructions for implementing the steps of the tape extension method as described, when these instructions are executed by a processor.

Finally, the invention relates to a storage medium, readable by a processor, integrated or not to the band expansion device, possibly removable, storing a computer program implementing a band extension method as described above.

Other features and advantages of the invention will appear more clearly on reading the following description, given solely by way of nonlimiting example, and with reference to the appended drawings, in which:

FIG. 1 illustrates a part of an AMR-WB decoder implementing frequency band extension steps of the state of the art and as previously described;

FIG. 2 illustrates a decoder of the interoperable type G.718-LD at 16 kHz according to the state of the art and as described previously;

FIG. 3 illustrates an interoperable decoder with the AMR-WB coding and integrating a band extension device according to one embodiment of the invention; FIG. 4 illustrates in flowchart form the main steps of a band extension method according to one embodiment of the invention;

FIG. 5 illustrates an embodiment in the frequency domain of a band extension device according to the invention integrated in a decoder; and Figure 6 illustrates a hardware embodiment of a tape extender according to the invention. FIG. 3 illustrates an exemplary decoder, compatible with the AMR-WB / G.722.2 standard, in which there is a postprocessing similar to that introduced in G.718 and described with reference to FIG. 2 and an improved band extension according to the extension method of the invention, implemented by the band extension device illustrated by block 309.

Unlike AMR-WB decoding which operates with an output sampling frequency of 16 kHz and G.718 decoding which operates at 8 or 16 kHz, a decoder is considered here which can operate with an output signal (synthesis) at the frequency fs = 8, 16, 32 or 48 kHz. Note that it is assumed here that the coding was performed according to the AMR-WB algorithm with an internal frequency of 12.8 kHz for the low-band CELP coding and at 23.85 kbit / s a sub-frame gain coding at the frequency of 16 kHz, but interoperable variants of the AMR-WB encoder are also possible; even if the invention is described here at the decoding level, it is assumed here that the coding can also operate with an input signal at the frequency fs = 8, 16, 32 or 48 kHz and appropriate resampling operations, beyond the scope of the invention, are implemented at the coding according to the value of fs. It can be noted that when fs = 8 kHz at the decoder, in the case of a decoding compatible with AMR-WB, it is not necessary to extend the low band 0-6.4 kHz, because the audio band reconstructed at the frequency / s is limited to 0-4000 Hz.

In Figure 3, the CELP decoding (BF for low frequencies) always operates at the internal frequency of 12.8 kHz, as in AMR-WB and G.718, and the band extension (HF for high frequencies) being the subject of the invention operates at the frequency of 16 kHz, the synthesis BF and HF are combined (block 312) at the frequency fs after adequate resampling (blocks 307 and 311). In variants of the invention, the combination of the low and high bands can be done at 16 kHz, after resampling the low band of 12.8 to 16 kHz, before resampling the combined signal at the frequency fs.

The decoding according to FIG. 3 depends on the mode (or bit rate) AMR-WB associated with the current frame received. As an indication and without this having an effect on block 309, the decoding of the low band CELP part comprises the following steps:

• Demultiplexing the coded parameters (block 300) in case of a correctly received frame (= 0 where is the "bad frame indicator" worth 0 for a received frame and 1 for a lost frame).

• Decoding of ISF parameters with interpolation and conversion to LPC coefficients (block 301) as described in clause 6.1 of G.722.2.

• Decoding of the CELP excitation (block 302), with an adaptive and fixed part to reconstruct the excitation (exc or u n)) in each subframe of length 64 to

12.8 kHz:

u '(n) = g _p v (n) + g _c c (n), η = 0, · · ·, 63 by following the notations of clause 7.1.2.1 of G.718 concerning CELP decoding, where v (n) and c (n) are respectively the code words of the adaptive and fixed dictionaries, and g and g _c are the decoded gains associates. This excitation a) is used in the adaptive dictionary of the following subframe; it is then post-processed and one discerns as in G.718 the excitement a) (also noted exc) of its modified post-processed version u (n) (also noted exc2) which serves as input to the synthesis filter, 1 / ((z), in block 303. In variants that can be implemented for the invention, the post-treatments applied to the excitation can be modified (for example, the phase dispersion can be improved) or such post-treatments may be extended (for example, interharmonic noise reduction may be implemented) without affecting the nature of the band extension method of the invention.

• Synthetic filtering by 1 / Â (z) (block 303) where the decoded LPC filter Â (z) is of order 16

· Narrowband post-processing (block 304) according to clause 7.3 of G.718 if fs = 8 kHz.

• Deactivation (block 305) by the filter 1 / (l - 0.68z ^_1 )

• Low frequency post-processing (block 306) as described in clause 7.14.1.1 of G.718. This processing introduces a delay which is taken into account in the decoding of the high band (> 6.4 kHz).

· Resampling of the internal frequency of 12.8 kHz to the output frequency fs

(block 307) Several achievements are possible. Without loss of generality, we consider here as an example that if fs = 8 or 16 kHz, the resampling described in clause 7.6 of G.718 is repeated here, and if fs = 32 or 48 kHz, filters Finite Impulse Response (FIR) are used.

· Calculation of "noise gate" parameters (block 308) which is preferably performed as described in clause 7.14.3 of G.718.

In variants that can be implemented for the invention, the post-treatments applied to the excitation can be modified (for example, the phase dispersion can be improved) or these post-treatments can be extended (for example, inter-harmonic noise reduction can be implemented), without affecting the nature of the band extension. The case of the decoding of the low band when the current frame is lost (bfi = 1) which is informative in the 3GPP standard AMR-WB is not described here; in general, whether it is the AMR-WB decoder or a general decoder based on the source-filter model, it is typically to best estimate the LPC excitation and the coefficients of the LPC filter synthesis to reconstruct the lost signal by keeping the source-filter model. When bfi = l is considered here that the band extension (block 309) can function as in the case = 0 and a bit rate <23.85 kbit / s; thus, the description of the invention will assume later and without loss of generality that = 0.

It can be noted that the use of blocks 306, 308, 314 is optional.

Note also that the decoding of the low band described above assumes a current frame called "active" with a rate between 6.6 and 23.85 kbit / s. In fact, when the DTX (Continuous Transmission in French) mode is activated, some frames can be coded as "inactive" and in this case you can either transmit a silence descriptor (on 35 bits) or not transmit anything. In particular, it is recalled that the SID frame of the AMR-WB encoder describes several parameters: ISF parameters averaged over 8 frames, average energy over 8 frames, "dithering flag" for the non-stationary noise reconstruction. In all cases, at the decoder, we find the same decoding model as for an active frame, with a reconstruction of the excitation and an LPC filter for the current frame, which makes it possible to apply the invention itself to inactive frames. The same applies for the decoding of "lost frames" (or FEC, PLC) in which the LPC model is applied.

This example decoder operates in the field of excitation and therefore comprises a step of decoding the low band excitation signal. The band extension device and the band extension method within the meaning of the invention also operates in a field different from the field of excitation and in particular with a low band decoded direct signal or a filter-weighted signal. perceptual.

Unlike the AMR-WB or G.718 decoding, the decoder described makes it possible to extend the decoded low band (50-6400 Hz by taking into account the high-pass filtering at 50 Hz at the decoder, 0-6400 Hz in the general case ) to an extended band whose width varies, ranging from approximately 50-6900 Hz to 50-7700 Hz depending on the mode implemented in the current frame. We can talk about a first frequency band from 0 to 6400Hz and a second frequency band from 6400 to 8000Hz. In fact, in the preferred embodiment, the excitation for the high frequencies and generated in the frequency domain in a band of 5000 to 8000 Hz, to allow bandpass filtering of width 6000 to 6900 or 7700 Hz whose slope is not too stiff in the upper band rejected.

The high band synthesis part is realized in the block 309 representing the band extension device according to the invention and which is detailed in FIG. 5 in one embodiment.

In order to align the decoded low and high bands, a delay (block 310) is introduced to synchronize the outputs of the blocks 306 and 309 and the high band synthesized at 16 kHz is resampled from 16 kHz to the frequency fs (output of block 311). The delay value should be adapted for other cases {fs = 32, 48 kHz) depending on the treatments used. Remember that when fs = 8 kHz, it is not necessary to apply the blocks 309 to 311 because the output signal band of the decoder is limited to 0-4000 Hz. It should be noted that the extension method of the invention implemented in block 309 according to the first embodiment introduces preferentially no additional delay with respect to the low band reconstructed at 12.8 kHz; however, in variants of the invention (for example using a time / frequency transformation with overlap), a delay may be introduced. Thus, in general the value of in block 310 will have to be adjusted according to the specific implementation. For example, in the case where the post-processing of low frequencies (block 306) is not used, the delay to be introduced for fs = 16 kHz can be set at Γ = 15.

The low and high bands are then combined (added) in block 312 and the resulting synthesis is post-processed by high-order 50 Hz (type IIR) high-pass filtering whose coefficients depend on the frequency fs (block 313) and output post-processing with optional noise gate application similar to G.718 (block 314).

The band extension device according to the invention, illustrated by the block 309 according to the embodiment of the decoder of FIG. 5, implements a band extension method (in the broad sense) now described with reference to FIG. figure 4.

This extension device may also be independent of the decoder and may implement the method described in FIG. 4 to perform a band extension of an existing audio signal stored or transmitted to the device, with an analysis of the audio signal to extract it for example an excitation and an LPC filter.

This device receives as input a decoded signal in a first so-called low-band frequency band u (ri) which may be in the field of excitation or that of the signal. In the embodiment described here, a step of subband decomposition (E401b) by time frequency transform or filter bank is applied to the low band decoded signal to obtain the spectrum of the decoded low band signal U (k) for a implemented in the frequency domain.

A step E401a for extending the decoded low band signal into a second frequency band greater than the first frequency band, to obtain an extended low band decoded signal U _HB1 (k), can be performed on this decoded low band signal before or after the analysis step (subband decomposition). This extension step may comprise both a resampling step and an extension step or simply a translation step or frequency transposition as a function of the signal obtained at the input. It will be noted that in variants, step E401a may be performed at the end of the processing described in FIG. 4, that is to say on the combined signal, this processing then being mainly performed on the low band signal before expansion. , the result being equivalent.

This step is detailed later in the embodiment described with reference to FIG. A step E402 for extracting a room signal (U _HBA (k)) and tonal components (y (k)) is performed from the decoded (U (k)) or decoded and extended ( U _HB1 (k)). Ambience is defined here as the residual signal that is obtained by suppressing in the existing signal the main (or dominant) harmonics (or tonal components).

In most broadband signals (sampled at 16 kHz), the high band (> 6 kHz) contains ambient information that is generally similar to that in the low band.

The step of extracting the tonal components and the ambient signal comprises, for example, the following steps:

detection of the dominant tone components of the decoded (or decoded and extended) low band signal, in the frequency domain; and

This step can also be obtained by:

obtaining the ambient signal by calculating an average of the decoded (or decoded and extended) low band signal; and

obtaining tonal components by subtracting the calculated ambient signal from the decoded (or decoded and extended) low band signal.

The tonal components and the surround signal are then adaptively combined using energy level control factors in step E403 to obtain a so-called combined signal (U _HB2 (k)). The extension step E401a can then be implemented if it has not already been performed on the decoded low band signal.

Thus, the combination of these two types of signals makes it possible to obtain a combined signal with characteristics more adapted to certain types of signals, such as musical signals, and richer in frequency content and in the extended frequency band corresponding to the entire band of signals. frequency including the first and the second frequency band.

The band extension according to the method improves the quality for this type of signals compared to the extension described in the AMR-WB standard.

Using a combination of surround signal and tonal components enriches this extension signal to make it closer to the characteristics of the real signal and not to an artificial signal.

This combining step will be detailed later with reference to FIG. 5. A synthesis step, which corresponds to the analysis at 401b, is performed at E404b to bring the signal back into the time domain. Optionally, an energy level adjustment step of the high band signal can be performed at E404a, before and / or after the synthesis step, by applying gain and / or adequate filtering. This step will be explained in more detail in the embodiment described in FIG. 5 for blocks 501 to 507.

In an exemplary embodiment, the band extension device 500 is described now with reference to FIG. 5 illustrating both this device and also processing modules suitable for implementation in a decoder of interoperable type with a coding AMR-WB. This device 500 implements the band extension method described above with reference to FIG. 4.

Thus, the processing block 510 receives a decoded low band signal (u (n)). In a particular embodiment, the band extension uses the decoded 12.8 kHz excitation (exc2 or u (n)) at the output of the block 302 of FIG. 3.

This signal is broken down into frequency subbands by the subband decomposition module 510 (which implements step E401b of FIG. 4) which generally performs a transform or applies a filter bank, to obtain a sub-band decomposition U (k) of the signal u (n).

In a particular embodiment, a DCT-IV type transform (for "Discrete Cosine Transform" - Type IV in English) (block 510) is applied to the current frame of 20 ms (256 samples), without windowing, which is to directly transform a) with η = 0, · · ·, 255 according to the following formula:

where N = 256 and fc = 0, - - -, 255.

A transformation without windowing (or equivalently with an implicit rectangular window of the length of the frame) is possible when the processing is performed in the field of excitation, and not the domain of the signal. In this case no artefact (block effects) is audible, which is an important advantage of this embodiment of the invention.

In this embodiment, the DCT-IV transformation is implemented by FFT according to the "Evolved ZXT (EDCT)" algorithm described in the article by D. M. Zhang, HT. Li, A Low Complexity Transform - Evolved DCT, IEEE 14th International Conference on Computational Science and Engineering (CSE), Aug. 2011, pp. 144-149, and implemented in ITU-T G.718 Annex B and G.729.1 Annex E.

In variants of the invention and without loss of generality, the DCT-IV transformation may be replaced by other short-term time-frequency transformations of the same length and in the field of excitation or in the domain of the signal. as an FFT (for "Fast Fourier Transform" in English) or a DCT-II (Discrete Cosine Transform - Type II). Alternatively, the DCT-IV can be replaced on the frame by a recovery-addition and windowing transformation of length greater than the length of the current frame, for example using an MDCT (for "Modified Discrete Cosine Tranform" in English). In this case the delay in the block 310 of Figure 3, should be adjusted (reduced) adequately depending on the additional delay due to analysis / synthesis by this transform.

In another embodiment, the subband decomposition is performed by the application of a real or complex filter bank, for example of the PQMF (Pseudo-QMF) type. For some filterbanks, for each subband in a given frame, not a spectral value but a series of time values associated with the subband are obtained; in this case, the preferred embodiment in the invention can be applied by producing for example a transform of each subband and calculating the ambient signal in the range of absolute values, the tonal components always being obtained by difference between the signal (in absolute value) and the ambient signal. In the case of a complex filter bank, the complex module of the samples will replace the absolute value.

In other embodiments, the invention will be applied in a system using two subbands, the low band being analyzed by transform or filterbank.

In the case of a DCT, the DCT spectrum, U (k), of 256 samples covering the band

0-6400 Hz (at 12.8 kHz), is then extended (block 511) into a spectrum of 320 samples covering the band 0-8000 Hz (at 16 kHz) in the following form:

where preferentially start_band = 160 is used.

Block 511 implements step E401a of FIG. 4, that is to say the extension of the decoded low band signal. This step can also include a resampling of 12.8 to 16 kHz in the frequency domain, adding ¼ samples (k = 240, · ^■ -, 319) to the spectrum, the ratio between 16 and 12.8 being 5/4 .

In the frequency band corresponding to samples ranging from indices 200 to

239, the original spectrum is preserved, in order to be able to apply a gradual attenuation response of the high-pass filter in this frequency band and also not to introduce audible defects during the addition step of the low frequency synthesis at high frequency synthesis. It should be noted that in this embodiment, the generation of the extended over-sampled spectrum is carried out in a frequency band ranging from 5 to 8 kHz, thus including a second frequency band (6.4-8 kHz) greater than the first band of frequency (0-6.4 kHz).

Thus, the extension of the decoded low band signal is performed at least on the second frequency band but also on a part of the first frequency band.

Of course, the values defining these frequency bands may be different depending on the decoder or the processing device in which the invention applies. In addition, block 511 performs high pass filtering implicit in the 0-5000 Hz band since the first 200 samples of U _HB1 (k) are set to zero; as explained later, this high-pass filtering can also be completed by a progressive attenuation part of the spectral values of indices k = 200, · · -, 255 in the band 5000-6400 Hz, this progressive attenuation is set implemented in the block 501 but could be carried out separately outside the block 501. Equivalently and in variants of the invention, the implementation of high-pass filtering separated into blocks of coefficients of index = = 0, · · ·, Set to zero, with coefficients k = 200, · · -, 255 attenuated, in the transformed domain, can therefore be performed in a single step.

In this exemplary embodiment and according to the definition of U _HB1 (k), it is noted that the 5000-6000 Hz band of U _HB1 (k) (which corresponds to the indices k = 200, · - -, 239) is copied from of the 5000-6000 Hz U (k) band. This approach preserves the original spectrum in this band and avoids introducing distortions in the 5000-6000 Hz band during the addition of HF synthesis with BF synthesis - particularly the signal phase (implicitly represented in the DCT-IV domain) in this band is preserved.

The 6000-8000 Hz band of U _HB1 (k) is here defined by copying the 4000-6000 Hz band of U (k) since the value of start_band is preferably fixed at 160.

In a variant of the embodiment, the value of start_band can be made adaptive around the value of 160, without changing the nature of the invention. The details of the adaptation of the value start_band are not described here because they go beyond the scope of the invention without changing the scope.

In most wideband signals (sampled at 16 kHz), the high band (> 6 kHz) contains background information that is naturally similar to that in the low band. Ambience is defined here as the residual signal that is obtained by suppressing in the existing signal the main (or dominant) harmonics. The level of harmonicity in the 6000-8000 Hz band is generally correlated to that of the lower frequency bands.

This decoded and extended low band signal is provided at the input of the extension device 500 and in particular at the input of the module 512. Thus the block 512 for extracting tonal components and a room signal, implements the step E402 of Figure 4 in the frequency domain. The ambient signal, ½ _BA () for £ = 240, · · ·, 319 (80 samples) is thus obtained for a second so-called high frequency frequency band in order to then adaptively combine it with the extracted tonal components y ( k), in the combination block 513.

In a particular embodiment, the extraction of the tonal components and the ambient signal (in the 6000-8000 Hz band) is carried out according to the following operations: • Calculation of the total energy of the extended decoded low band signal ener _HB : ^ener HB U _HB1 (k) ² + e

where £ = 0.1 (this value may be different, it is fixed here as an example).

Calculation of the atmosphere (in absolute value) which corresponds here to the average level of the high spectrum (i) (line by line) and calculation of the ener _tonal energy of the dominant _tonal parts (in the high frequency spectrum)

= 0 ... L - 1, we obtain this average level by the following equation:

)

l ^{ev (Î) =} _Μ - 1 Σ l ^ ™ - + 240) |

This corresponds to the average level (in absolute value) and therefore represents a sort of envelope of the spectrum. In this embodiment, L = 80 and represents the length of the spectrum and the index i from 0 to L - 1 corresponds to the indices 240 + 31 of 240 to 319, ie the spectrum of 6 to 8 kHz.

In general β {ϊ) = ί - Ί and fn {ï) = i + l, however the first and last 7 indices (ζ ^' = 0, · · ·, 6 and i = L - 7, - - -, Ll ) request special treatment and without loss of generality then define:

fl i) = 0 and fa (i) = i + 7 for ι = 0, · · ·, 6

β {ϊ) = i-1 and jh {ï) = L-1 for i = L-7, - - -, L-1

In variants of the invention, the average of | £ / _{/ FFLL} ( _I / + 240) | , j = ft> (i), ..., fn (ï), can be replaced by a median value on the same set of values, either lev (i) = median _{j = ft? (ik (i)} (| L / _{/ ffll} ( _i / + 240) |) This variant has the defect of being more complex

(in terms of the number of calculations) than a sliding average. In other variants, a non-uniform weighting may be applied to the averaged terms, or the median filtering may be replaced for example by other nonlinear filters of "stack filter" type.

The residual signal is also calculated:

Y (= _HB1 d + 240) | - lev (i), i = 0, ..., L - 1

which corresponds (approximately) to the tonal components if the value y (z) at a given line i is positive (y (z)> 0).

This calculation therefore involves an implicit detection of the tonal components. The tonal parts are thus implicitly detected using the intermediate term y (i) representing an adaptive threshold. The detection condition being y (z)> 0. In variants of the invention this condition can be changed for example by defining an adaptive threshold depending on the local envelope of the signal or in the form y (i)> lev (i) + xdB where x has a predefined value (for example = 10 dB).

The energy of the dominant tonal parts is defined by the following equation:

^ tonal =

i = 0 ... Σ y

7LY (i)> 0

Other methods of extracting the ambient signal can of course be envisaged. For example, this ambient signal can be extracted from a low frequency signal or possibly another frequency band (or several frequency bands).

The detection of peaks or tonal components can be done differently.

The extraction of this ambient signal could also be done on the decoded but not extended excitation, that is to say before the extension or spectral translation step, that is to say, for example on a portion of the low frequency signal rather than directly on the high frequency signal.

In an alternative embodiment, the extraction of the tonal components and the ambient signal is performed in a different order and according to the following steps:

detection of the dominant tonal components of the decoded low band signal

(or decoded and extended), in the frequency domain;

calculating a residual signal by extracting the dominant tonal components to obtain the ambient signal.

This variant can for example be made in the following way: A peak (or tonal component) is detected at a line of index i in the amplitude spectrum * (i + 240 + 1) | ,

for Ï = 0, ..., L - 1. As soon as a peak is detected at the line of index i, a sinusoidal model is applied in order to estimate the amplitude, frequency and possibly phase parameters of a tonal component associated with this peak. The details of this estimate are not presented here, but the estimate of the frequency can typically use a parabolic interpolation on 3 points to locate the maximum of the parabola approximating the 3 points of amplitude | t / _iffil (z ^' + 240) | (reduced in dB), the amplitude estimate being obtained by means of this same interpolation. Since the transform domain used here (DCT-IV) does not make it possible to obtain the phase directly, it will be possible, in one embodiment, to neglect this term, but in variants it will be possible to apply a quadrature transform of the DST type to estimate a phase term. The initial value of y (z) is set to zero for i = 0,. . . , L -1. Since the sinusoidal parameters (frequency, amplitude, and possibly phase) of each tonal component are estimated, the term y (z) is calculated as the sum of predefined prototypes (spectra) of pure sinusoids transformed in the DCT-IV domain (or other if another subband decomposition is used) according to the estimated sinusoidal parameters. Finally, we apply an absolute value to the terms y (z) to reduce to the domain of the amplitude spectrum in absolute values. Other methods for determining the tonal components are possible, for example it would also be possible to calculate an envelope of the signal env (i) by interpolation by splines of the local maximum values (detected peaks) of | t / _{/ ffll} (z ^' + 240) | , to lower this envelope by a certain level in dB to detect the tonal components as the peaks exceeding this envelope and to define y (z) as

y (z ^' ) = max (i + 240) | - env (i), θ)

In this variant the atmosphere is thus obtained by the equation:

lev (ï) = 240) | - y (z), i = {), ..., L -l

In other variants of the invention, the absolute value of the spectral values will be replaced for example by the square of the spectral values, without changing the principle of the invention; in this case a square root will be needed to return to the signal domain, which is more complex to achieve.

The combination module 513 performs a step of combining by adaptive mixing of the ambient signal and the tonal components. For this, a control factor Γ of the ambient level is defined by the following equation: γ ₌ β ^ener HB - _tonal ener

ener _HB - ener _tonal

β being a factor of which an example of calculation is given below.

To obtain the extended signal, we first obtain the combined signal in absolute values for / = 0 ... - 1:

to which the signs of U _HB1 (k) are applied:

f (Î) = sgn {U _HBl (i + 240)) (Î) where the function sgn.) gives the sign:

By definition the factor Γ is> 1. The tonal components, detected line by line by the condition y (i)> 0, are reduced by the factor Γ; the average level is amplified by the factor / Γ.

In the adaptive mixing block 513, a power level control factor is calculated based on the total energy of the decoded (or decoded and extended) low band signal and the tonal components.

In a preferred embodiment of the adaptive mixing, the energy adjustment is performed as follows:

U _HB2 (k) = fac.y "(k-240), k = 240, ^••• , 319

U _HB2 (k) being the combined band extension signal.

The adjustment factor is defined by the following equation:

Where γ avoids over-estimation of energy. In an exemplary embodiment, we calculate /? so as to maintain the same ambient signal level with respect to the energy of the tonal components in the consecutive bands of the signal. The energy of the tonal components is calculated in three bands: 2000-4000 Hz, 4000-6000 Hz and 6000-8000 Hz, with

V ₄ = Σ u ^a (k)

i eN (80,159) ^J IV4-6 Σ U ' ² (k)

N (160.239)

^J IV4-6 Σ U (k)

N (240.319)

or

U k) U (k) = = 160, ..., 239

Σ U ² (k)

fc = 160

319 -u _HB1 (k) £ = 240, ..., 319

Σ U _HB1 ² (k)

fc = 240

And where N (£ ₁ , £ ₂ ) is the set of indices k for which the index coefficient k is classified as being associated with the tonal components. This set can be obtained for example by detecting the local peaks in U > lev (k) or lev {k) is calculated as the average level of the spectrum line by line.

It may be noted that other methods of calculating the energy of the tonal components are possible, for example by taking the median value of the spectrum on the band considered. Β is fixed so that the ratio between the energy of the tonal components in the 4-6 kHz and 6-8 kHz bands is the same as between the 2-4 kHz and 4-6 kHz bands: o _ PE _N6 _ _&

239

ΣU ² (k) - E _m _,

i = 160

or

E ²

E = MZX _N4 _ ₆ (E _{N 4} _ ₆ _ E _{N 2} _4), p = -J! , = max (/ ?, E _A

^ IV2-4 and max (.,.) Is the function that gives the maximum of both arguments.

In variants of the invention, the calculation of β may be replaced by other methods. For example, in one variant, it will be possible to extract (calculate) different parameters (or "features" in English) characterizing the low band signal, including a "tilt" parameter similar to that calculated in the AMR-WB codec, and it will be estimated the postman β as a function of a linear regression from these different parameters by limiting its value between 0 and 1. The linear regression could for example be estimated in a supervised manner by estimating the factor β by giving the original high band in a base d 'learning. It will be noted that the method of calculating β does not limit the nature of the invention.

Then, the parameter β can be used to calculate γ, taking into account that a signal with a surround signal added in a given band is generally perceived as stronger than a harmonic signal at the same energy in the same direction. bandaged. If we define a as the quantity of ambient signal added to the harmonic signal:

we can calculate γ as a decreasing function of, for example = b - a,

£> = 1.1, a = 1.2 and γ limited from 0.3 to 1. Here again other definitions of a and γ are possible within the scope of the invention.

At the output of the band extension device 500, the block 501, in a particular embodiment, optionally carries out a dual operation of application of bandpass filter frequency response and deemphasis filtering (or deemphasis filtering). ) in the frequency domain.

In a variant of the invention, the deemphasis filtering may be performed in the time domain, after the block 502 or even before the block 510; however, in this case, the bandpass filtering performed in the block 501 may leave some low frequency components of very low levels which are amplified by de-emphasis, which may slightly discern the decoded low band. For this reason, it is preferred here to perform the deemphasis in the frequency domain. In the preferred embodiment, the index coefficients = = (), ^{■ ■} , 199 are set to zero, so the deemphasis is limited to the higher coefficients.

The excitation is first de-emphasized according to the following equation:

0 £ = 0, · · ·, 199

U _m2 '(*) = G _deemph (k - 200) U _HB2 (k) k = 200, -, 255

G _deemph (55) U _HB2 (k) = 256, · · ·, 319 where G _deemph (k) is the frequency response of the filter l / (l - 0.68z ^_1 ) over a restricted discrete frequency band. Taking into account the discrete (odd) frequencies of the DCT-IV, we define here G _{deem h} (k) as:

or

In the case where a transformation other than the DCT-IV is used, the definition of 0 _k can be adjusted (for example for even frequencies).

It is noted that the deemphasis is applied in two phases for £ = 200, · · -, 255 corresponding to the frequency band 5000-6400 Hz, where the response l / (l - 0.68z ^_1 ) is applied as at 12.8 kHz, and for k = 256, · · -, 319 corresponding to the 6400-8000 Hz frequency band, where the response is extended from 16 kHz here to a constant value in the 6.4-8 kHz band.

It may be noted that in the AMR-WB coding the HF synthesis is not de-emphasized. In the embodiment presented here, the high-frequency signal is on the contrary de-emphasized so as to bring it back to a domain coherent with the low-frequency signal (0-6.4 kHz) which leaves block 305 of FIG. 3. This is important for the estimation and subsequent adjustment of the energy of HF synthesis.

In a variant of the embodiment, in order to reduce the complexity, it will be possible to fix G _deemph (k) at a constant value independent of, for example, taking

_Deemph G (k) = 0.6 which corresponds approximately to the mean value of _deemph G (k) for k = 200, · · -, 319 under the conditions of the embodiment described above.

In another variant of the embodiment of the decoder, the de-emphasis can be performed in an equivalent manner in the time domain after inverse DCT.

In addition to de-emphasis, band-pass filtering is applied with two separate parts: one fixed high-pass, the other adaptive low-pass (flow-rate function).

This filtering is performed in the frequency domain.

In the preferred embodiment, the partial low-pass filter response in the frequency domain is calculated as follows:

G k) = 1 - 0.999 - ^ -

N _lp - l

where N _lp = 60 to 6.6 kbit / s, 40 to 8.85 kbit / s, 20 at rates> 8.85 bit / s.

Then we apply a bandpass filter in the form:

0 £ = 0, · · ·, 199

G _hp (k-200) U _HB2 \ k) £ = 200, · · -, 255

U _HB3 (k)

U _HB2 \ k) £ = 256, - - -, 319 - N _lp

G _lp (k - 320 - N _p) U _HB2 \ k) fc = 320 - N _lp, -, 319

The definition of G _h (k), k = 0, · · ·, 55, is given for example in Table 1 below. 9 K _f kk) KK 9 k) k 9 _f kk)

0 0.001622428 14 0.114057967 28 0.403990611 42 0.776551214

1 0.004717458 15 0.128865425 29 0.430149896 43 0.800503267

2 0.008410494 16 0.144662643 30 0.456722014 44 0.823611104

3 0.012747280 17 0.161445005 31 0.483628433 45 0.845788355

4 0.017772424 18 0.179202219 32 0.510787115 46 0.866951597

5 0.023528982 19 0.197918220 33 0.538112915 47 0.887020781

6 0.030058032 20 0.217571104 34 0.565518011 48 0.905919644

0.037398264 21 0.238133114 35 0.592912340 49 0.923576092

8 0.045585564 22 0.259570657 36 0.620204057 50 0.939922577

0.054652620 23 0.281844373 37 0.647300005 51 0.954896429

10 0.064628539 24 0.304909235 38 0.674106188 52 0.968440179

11 0.075538482 25 0.328714699 39 0.700528260 53 0.980501849

12 0.087403328 26 0.353204886 40 0.726472003 54 0.991035206

13 0.100239356 27 0.378318805 41 0.751843820 55 1.000000000

Table 1

It will be noted that in variants of the invention the values of G _hp (k) may be modified while keeping a gradual attenuation. Similarly, the low-pass filtering with variable bandwidth, G _lp {k), may be adjusted with different values or frequency support, without changing the principle of this filtering step.

Note also that the bandpass filtering can be adapted by defining a single filtering step combining the high-pass and low-pass filtering.

In another embodiment, the bandpass filtering may be performed equivalently in the time domain (as in block 112 of FIG. 1) with different filter coefficients according to the bit rate, after an inverse DCT step. However, it will be noted that it is advantageous to carry out this step directly in the frequency domain because the filtering is carried out in the field of LPC excitation and therefore the problems of circular convolution and edge effects are very limited in this field. .

The inverse transform block 502 performs an inverse DCT on 320 samples to find the high frequency signal sampled at 16 kHz. Its implementation is identical to block 510 because the DCT-IV is orthonormed, except that the length of the transform is 320 instead of 256, and we obtain:

where _M = 320 and fc = 0, - - -, 319 In the case where the block 510 is not a DCT, but another transformation or decomposition into subbands, the block 502 performs the synthesis corresponding to the analysis carried out in the block 510.

The signal sampled at 16 kHz is then optionally scaled by gains defined by subframe of 80 samples (block 504).

In a preferred embodiment, one calculates first (block 503) a gain g _H Bi (m) per subframe by energy ratios of the subframes such that in each subframe of index 77 = 0, 1, 2 or 3 of the current frame:

e ₃ (m)

e ₂ (m)

or

63

e ₁ (m) = Σ u (n + 64m) ² + ε

319

Σu _HB (n † +

e ₃ (m) = e ₁ (m) 255

+ £ with ε = 0.01. The gain was written by sub-frame g _HB1 (m) in the form

which shows that the signal u _{HB has} the same ratio between energy per subframe and energy per frame as in the signal u (n).

Block 504 scales the combined signal (included in step E404a of FIG. 4) according to the following equation:

^U _HB '(") = S _HBI ( ^m ) u _HB (n), n = 80m, · · ·, 80 (m + 1) -1

Note that the realization of the block 503 differs from that of the block 101 of Figure 1, because the energy at the current frame is taken into account in addition to that of the subframe. This makes it possible to have the ratio of the energy of each subframe with respect to the energy of the frame. Energy ratios (or relative energies) are compared rather than the absolute energies between low band and high band.

Thus, this scaling step makes it possible to keep in the high band the energy ratio between the subframe and the frame in the same way as in the low band.

Optionally, block 506 then scales the signal (included in step E404a of FIG. 4) according to the following equation:

^U HB "(") = 8HB2 ( ^m ) ^U HB X ⁿ ) / ^H = ^80m ^"""> ⁸⁰ ( ^m + 1) ^-1

where the gain g _HB2 (m) is obtained from the block 505 by executing the blocks 103, 104 and 105 of the AMR-WB coding (the input of the block 103 being the decoded excitation in the low band, u (n)) . Blocks 505 and 506 are useful for adjusting the level of the LPC synthesis filter (block 507), here depending on the tilt of the signal. Other methods of calculating the gain g _HB2 (m) are possible without changing the nature of the invention.

Finally, the signal, u _HB '(ri) or u _HB "(ri), is filtered by the filtering module 507 which can be realized here by taking as transfer function 1 / Α (ζ / γ), where γ = 0.9 to 6.6 kbit / s and ^ = 0.6 to other data rates, which limits the order of the filter to 16.

In a variant, this filtering can be done in the same way as described for the block 111 of FIG. 1 of the AMR-WB decoder, however the order of the filter goes to 20 at the rate of 6.6, which does not change. not significantly the quality of the synthesized signal. In another variant, it will be possible to carry out LPC synthesis filtering in the frequency domain, after having calculated the frequency response of the filter implemented in block 507.

In alternative embodiments of the invention, the coding of the low band (0-6.4 kHz) may be replaced by a CELP coder other than that used in AMR-WB, for example the CELP coder in G.718 to 8. kbit / s. Without loss of generality other encoders in wide band or operating at frequencies higher than 16 kHz, in which the coding of the low band operates at an internal frequency at 12.8 kHz could be used. Moreover, the invention can be obviously adapted to other sampling frequencies than 12.8 kHz, when a low frequency encoder operates at a sampling frequency lower than that of the original or reconstructed signal. When the low band decoding does not use a linear prediction, it does not have an excitation signal to be extended, in this case it will be possible to carry out an LPC analysis of the reconstructed signal in the current frame and calculate an LPC excitation. so as to be able to apply the invention.

Finally, in another variant of the invention, the excitation or the low band signal (u (n)) is resampled, for example by linear interpolation or cubic "spline", of 12.8 to 16 kHz before transformation (for example DCT-IV) of length 320. This variant has the defect of being more complex, because the transform (DCT-IV) of the excitation or the signal is then calculated on a larger length and resampling is not performed in the transform domain.

Moreover, in variants of the invention, all the calculations necessary for estimating the gains (G _HBN , g _HB1 (m), g _HB2 (m), g _HBN , ...) can be carried out in a field logarithmic.

FIG. 6 represents an exemplary hardware embodiment of a band extension device 600 according to the invention. This may be an integral part of an audio-frequency signal decoder or equipment receiving decoded or non-decoded audio signals.

This type of device comprises a PROC processor cooperating with a memory block BM having a memory storage and / or work MEM.

Such a device comprises an input module E adapted to receive a decoded audio signal or extracted in a first frequency band said low band brought into the frequency domain (U (k)). It comprises an output module S adapted to transmit the extension signal in a second frequency band (U _HB2 (k)) for example to a filtering module 501 of FIG. 5.

The memory block may advantageously comprise a computer program comprising code instructions for implementing the steps of the band extension method in the sense of the invention, when these instructions are executed by the processor PROC, and in particular the steps for extracting (E402) tonal components and a surround signal from a signal derived from the decoded low band signal (U (k)), combining (E403) the tonal components (y (k)) and the ambient signal (U _HBA k)) by adaptive mixing using energy level control factors to obtain an audio signal, said combined signal (U _HB2 (k)), of extension (E401a) on at minus a second frequency band higher than the first frequency band of the low band decoded signal before the extraction step or the combined signal after the combining step.

Typically, the description of FIG. 4 repeats the steps of an algorithm of such a computer program. The computer program can also be stored on a memory medium readable by a reader of the device or downloadable in the memory space thereof.

The memory MEM generally records all the data necessary for the implementation of the method.

In a possible embodiment, the device thus described may also include the low band decoding functions and other processing functions described for example in FIGS. 5 and 3 in addition to the band extension functions according to the invention.

Claims

A method of extending the frequency band of an audiofrequency signal during a decoding or improvement process comprising a step of obtaining the decoded signal in a first so-called low band frequency band, the method being characterized in that it includes the following steps:

- extracting (E402) tonal components and a room signal from a signal from the decoded low band signal;

- combining (E403) tonal components and the ambient signal by adaptive mixing using energy level control factors to obtain an audio signal, said combined signal;

extension (E401a) on at least one second frequency band higher than the first frequency band of the low band decoded signal before the extraction step or the combined signal after the combining step.

Method according to claim 1, characterized in that the decoded low band signal is a decoded low band excitation signal.

Method according to one of claims 1 or 2, characterized in that the extraction of the tonal components and the ambient signal is performed according to the following steps: detection of the dominant tonal components of the decoded or decoded and extended low band signal, in the frequency domain;

Method according to one of claims 1 or 2, characterized in that the extraction of the tonal components and the ambient signal is carried out according to the following steps: obtaining the ambient signal by calculating a mean value of the spectrum of the decoded or decoded and extended bass band signal;

A method according to claim 1 characterized in that a power level control factor used for adaptive mixing is calculated as a function of the total energy of the decoded or decoded and extended low band signal and the tonal components.

6. Method according to one of the preceding claims, characterized in that the decoded low band signal undergoes a subband decomposition step by transform or by filter bank, the extraction and combining steps then taking place in the frequency domain or in subbands.

7. Method according to one of the preceding claims, characterized in that the step of extending the decoded low band signal is performed according to the following equation:

U _HBl (k) =

with k the index of the sample, U (k) the spectrum of the decoded low band signal obtained after a transform step, U _H BI (k) the spectrum of the extended signal, and start_band a predefined variable.

8. Device for frequency band extension of an audiofrequency signal, the signal having been decoded in a first so-called low band frequency band, the device being characterized in that it comprises:

- an extraction module (512) of tonal components and a room signal from a signal from the decoded low band signal;

a combination module (513) of the tonal components and the adaptive mixing ambient signal using energy level control factors to obtain an audio signal, called a combined signal;

an extension module (511) on at least a second frequency band greater than the first frequency band implemented on the decoded low band signal before the extraction module or on the combined signal after the combination module.

9. Audio frequency signal decoder characterized in that it comprises a frequency band extension device according to claim 8.

Computer program comprising code instructions for implementing the steps of the frequency band extension method according to one of claims 1 to 7, when these instructions are executed by a processor.

11. Storage medium readable by a frequency band extension device on which is recorded a computer program comprising code instructions for performing the steps of the frequency band extension method according to one of claims 1 to 1. at 7.