EP3198820B1 - Linear equalization for use in low latency high speed communication systems - Google Patents
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- EP3198820B1 EP3198820B1 EP15845356.3A EP15845356A EP3198820B1 EP 3198820 B1 EP3198820 B1 EP 3198820B1 EP 15845356 A EP15845356 A EP 15845356A EP 3198820 B1 EP3198820 B1 EP 3198820B1
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- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2602—Signal structure
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- H—ELECTRICITY
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- H04L25/00—Baseband systems
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- H04L25/03019—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception
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- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
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- H04L25/03019—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception
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- H04L25/03828—Arrangements for spectral shaping; Arrangements for providing signals with specified spectral properties
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- H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
- H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
- H04L25/03878—Line equalisers; line build-out devices
- H04L25/03885—Line equalisers; line build-out devices adaptive
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- H—ELECTRICITY
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- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
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- H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
- H04L27/38—Demodulator circuits; Receiver circuits
- H04L27/3845—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier
- H04L27/3854—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier using a non - coherent carrier, including systems with baseband correction for phase or frequency offset
- H04L27/3863—Compensation for quadrature error in the received signal
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Description
- The present invention relates generally to digital communication over a channel and, in particular, to linear equalizers used in a low latency high speed communication system.
- Wireless communications in millimetre wavelength frequency bands, such as the E-band (71-76 GHz and 81-86 GHz), typically have data rates in the order of Giga bits per second (Gbps). At such high data rates, mitigating Intersymbol Interference (ISI) caused by radio signal multipath propagation in a wireless channel, and signal reflection caused by connecting cables, is always a significant technical challenge.
- Orthogonal frequency division multiplexing (OFDM) and its variants, such as single carrier with frequency domain equalization (SC-FDE), typically cope with large multipath delay spreads in broadband communications. However, equalization of OFDM signals in the frequency domain introduces large processing delays, and the spectrum efficiency is also reduced due to the use of guard intervals.
- A single carrier system with advanced equalizers, such as a decision-feedback equalizer, is another option for coping with ISI. However, for high speed systems which demand very high clock rates in firmware implementations, such equalization cannot be performed at sufficiently high speeds to satisfy the data rate requirements. Therefore, a single carrier system with linear equalization becomes the only viable solution to ISI mitigation for high speed systems, when low processing delay is demanded.
- Transmitter side equalization, which is referred to as pre-equalization hereafter, is efficient in reducing the implementation complexity and noise enhancement effect associated with receiver side linear equalization. Generally, a linear equalizer needs to have a long impulse response to equalize a linear channel with even a short delay spread, which implies that the equalization complexity will generally be very high if the equalization is implemented at the receiver. Such equalization can also cause a significant noise enhancement effect. Shifting such equalization from the receiver to the transmitter (i.e., pre-equalization) can significantly reduce the implementation complexity and latency by using predefined lookup tables created based on a pre-defined signal constellation. The noise enhancement effect can also be mitigated as the signal to noise ratio at the transmitter is much larger compared to that at the receiver.
- More generally, both pre-equalization at the transmitter side and equalization at the receiver side are implemented at the same time. In one approach, the impulse response of the communications channel is factorized as a product of two impulse responses, and each is compensated for by either transmitter or receiver equalization. However, very complex computations are required for such factorization. In another aspect, channel equalization is mainly implemented at the transmitter, and receiver side equalization is only used to deal with residual channel effects after pre-equalization at the transmitter.
- Coefficients used in the pre-equalization at the transmitter need to be generated using, for example, the impulse response of the channel estimated at the receiver. However, when channels are time varying, a mechanism is required to track the channel variation and update the equalization coefficients. Typically, pre-equalized or non- pre-equalized training sequences are used for estimating the impulse response of the channel, and generating the equalization coefficients at both transmitter and receiver.
- The impulse response of the channel is typically estimated in the frequency domain due to its low complexity. When a pre-equalized training sequence is used, in the frequency domain, the received signal at one frequency point can be represented as y=hpx+n, where y is the received signal, h is the channel response, p is the pre-equalizer coefficient, x is the training signal, and n is the noise. The receiver equalizer coefficient can be generated by treating hp as a combined channel response, while the transmitter pre-equalization coefficient needs to be determined through the channel response h, which can be obtained by removing the pre-equalizer coefficient p from the estimate of the combined channel response hp. When non- pre-equalized training sequences are used, the received signal is y = hx+n, and the estimation of the receiver side equalization coefficient needs to combine the pre-equalizer p with the estimate of the channel response h. In either case, the receiver needs to know when the pre-equalization coefficients are updated. Estimation performance is also affected by using non-constant magnitude training signals in the frequency domain in the case of using pre-equalized training and by a doubled noise effect by combining the estimate of the impulse response of a noisy channel and the pre-equalizer in the case of using the non- pre-equalized training sequence.
- In existing systems, these training sequences in every frame are generally identical. If multiple training sequences are required, they are concatenated in the preamble of a frame. However, long preamble causes long delay.
- In-phase and quadrature (I/Q) imbalance is another significant concern for a wireless system with I/Q modulation architecture, i.e., the baseband signal is modulated onto (or demodulated from) an intermediate frequency (IF) or a radio frequency (RF) carrier through two separate in-phase (1) and quadrature (Q) channels. Due to the difference between the I and Q channel transmission characteristics (therefore termed I/Q imbalance or mismatch), the signal will be distorted if such impairment exists at the transmitter and/or receiver side(s). If the signal bandwidth is large, the I/Q imbalance is also frequency dependent (i.e., the I/Q imbalance is different at different frequencies throughout the bandwidth).
- There are a number of techniques found in the prior art for I/Q imbalance compensation. Most of those techniques deal with I/Q imbalance compensation at the receiver side only, whereas both transmitter and receiver side imbalances exist at the same time in real systems. Estimating and compensating for both transmitter and receiver side imbalance are very challenging as the imbalance signals are entangled and therefore generally complex to separate them to achieve good estimation. Existing approaches typically require offline calibration to obtain the estimate for the transmitter side mismatch, and then estimate the receiver side mismatch using the received signal. However, this calibration will interrupt the normal operation, and is infeasible in continuous transmission systems such as backhaul systems. A limited number of approaches propose to jointly estimate the transmitter and receiver side mismatches, however, their complexity is very high which makes them impractical for implementation in real hardware.
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EP 2 378 730 A1 -
EP 1 953 982 B1 - A need therefore exists for alternative equalizers for use in a low latency high speed communication system.
- It is an object of the present invention to substantially overcome, or at least ameliorate, one or more disadvantages of existing arrangements.
- According to a first aspect of the present disclosure, there is provided the communication system of
claim 1. - According to a second aspect of the present disclosure, there is provided the method of
claim 8. - Other aspects of the invention are also disclosed.
- One or more embodiments of the present invention will now be described with reference to the drawings, in which:
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Figs. 1A and1B show schematic block diagrams of baseband communication systems according to the present disclosure; -
Fig. 2A illustrates a structure of a variable linear equalizer with I/Q imbalance compensation; -
Fig. 2B illustrates a structure of a linear equalizer with I/Q imbalance compensation; -
Figs. 3A and 3B illustrate a sequence of odd and even data frames, and the contents of those data frames; -
Fig. 4 shows a schematic flow diagram of a method for initial estimation of the channel shown inFig. 1 ; -
Fig. 5 illustrates the effect down-sampling on the spectrum of a training signal; -
Fig. 6A shows a schematic block diagram of a structure of an RxFilter for an example case; -
Fig. 6B shows a schematic block diagram of a structure of a polyphase filter; -
Fig. 7 shows a schematic block diagram of a structure of a precoder with I/Q imbalance compensation and equalization corresponding to the example case ofFig. 6A ; -
Fig. 8 shows a packet fed back from the receiver to the transmitter; -
Fig. 9 illustrates modulating feedback bits with a preamble of data frames; -
Fig. 10 illustrates implementing baseband processing over the period of a current frame, and applying the results of the processing to the next two frames, instead of the current frame; -
Fig. 11 illustrates a special precoding structure used in the processing illustrated inFig. 10 ; -
Fig. 12 shows a schematic flow diagram of a method, performed by a receiver of the system shown inFig. 1 , of estimating I/Q imbalance parameters and an impulse response of the channel; -
Fig. 13 shows a schematic flow diagram of a method, performed by a transmitter of the system shown inFig. 1 , of estimating I/Q imbalance parameters and refining the impulse response of the channel; and -
Fig. 14 shows a configuration in which the system disclosed herein is implemented both in a combined field-programmable gate array and a personal computer. - Disclosed herein is a method and apparatus for realizing a high speed low latency full duplex wireless point-to-point link. This includes transmitting one or more training sequences, performing channel estimation and equalization, channel feedback, transmitter and receiver filters, and applying transmitter and receiver filtering to a data payload in a particularly configured frame structure. I/Q imbalance estimation and compensation are also optionally performed.
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Fig. 1A shows a schematic block diagram of a basicbaseband communication system 100. The basicbaseband communication system 100 includes atransmitter 110 which communicates via achannel 120 with areceiver 130. - In the
system 100 linear equalization is performed in both thetransmitter 110 and thereceiver 130. More particularly, thetransmitter 110 includes a variablelinear equalizer 112 as well as a fixedlinear equalizer 114, whereas thereceiver 130 includes alinear equalizer 138 only. - The fixed
linear equalizer 114 uses equalization coefficients that are fixed over time. More particularly, the equalization coefficients of the fixedlinear equalizer 114 are predetermined during installation and calibration of thesystem 100. In the simplest case, the impulse response of the fixedlinear equalizer 114 is a delta function in the time domain (i.e. a constant 1 at each subcarrier in the frequency domain), when either calibration is omitted or the calibration result suggests a single coefficient of 1. - The variable
linear equalizer 112 and thelinear equalizer 138 use equalization coefficients that vary over time. The values of the varying equalization coefficients of the variablelinear equalizer 112 are determined by thetransmitter 110 using a feedback channel (not illustrated) from thereceiver 130. - In the
systems 100, a training sequence together withdata symbols 151 are forward error coded (FEC) and modulated bymodule 105. The resulting data symbols x(n) are then input to the variablelinear equalizer 112 to generate output symbols y(n). The output symbols y(n) are next provided to apulse shaping filter 118 before the signal is transmitted over thechannel 120. - At the
receiver 130, the signal received from thechannel 120 is first passed to a matchedfilter 132, which corresponds to thepulse shaping filter 118 of thetransmitter 110. The output r(n) of the matchedfilter 132 is then input to thelinear equalizer 138 to generate output symbols z(n). The output symbols z(n) are then provided tomodule 140 which applies FEC decoding and demodulation to provideoutput data bits 155. Theoutput data bits 155 are processed for packet synchronization, channel estimation and other functions such as carrier frequency offset (CFO) estimation. - In a similar manner, a
different training sequence 152 is input to the fixedlinear equalizer 114, followed by pulse shaping by thepulse shaping filter 118. The resulting signal is then transmitted over thechannel 120. In thereceiver 130, the signal received from thechannel 120 is also passed to the matchedfilter 132, before the resulting data symbols are input to achannel estimation module 139 which outputsdata 156. - In the manner described in detail below, the varying equalization coefficients of the variable
linear equalizer 112 and thelinear equalizer 138 are determined using the training sequence added to thedata receiver 130 using thetraining sequence 152, is fed back from thereceiver 130 to thetransmitter 110, and then used for computing the coefficients of the variablelinear equalizer 112. -
Fig. 1B shows a schematic block diagram of an extended baseband communication system 100'. The extended baseband communication system 100' has many elements in common with the basicbaseband communication system 100 described above with reference toFig. 1A . Accordingly, where reference is made to elements inFig. 1B which have the same reference numerals used inFig. 1A , those elements have for the purposes of this description the same function(s) or operation(s), unless the contrary intention appears. - Similar to the
system 100 described above, the transmitter 110' of the system 100' includes a variable linear equalizer 112' as well as a fixedlinear equalizer 114, whereas the receiver 130' includes a linear equalizer 138' only. However, the variable linear equalizer 112' of the transmitter 110' and the linear equalizer 138' of the receiver 130' combine channel equalization with in-phase (I) and quadrature (Q) channel (I/Q) imbalance compensation. - Another difference between the
systems 100 and 100' is that the transmitter 110' of the system 100' includes a sampling rate conversion (SRC)module 115 which converts the symbols, output from the variable linear equalizer 112' or the fixedlinear equalizer 114, from the symbol rate to a desired chip rate. Similarly, the receiver 130' of the system 100' includes anotherSRC module 135 which receives the output of the matchedfilter 132 and converts the symbols back from the chip rate to the symbol rate before the resulting data symbols are input to the linear equalizer 138', which also performs I/Q imbalance compensation, or thechannel estimation module 139. - In the manner described in detail below, the I/Q imbalance parameters, as well as the the varying equalization coefficients of the variable linear equalizer 112' and the linear equalizer 138', are determined using the training sequence added to the
data - Referring again to
data Fig. 3A illustrates a sequence of odd and even data frames.Fig. 3B illustrates the odd and even data frames in more detail. A physical-layer data frame typically consists of preamble, PHY header and data payload. The preamble of each data frame includes the training sequence. In principle, the training sequences may appear in any location of the data frames. In the disclosedsystems 100 and 100', two different training sequences are used as the preambles of alternating frames. Accordingly, odd and even frames, odd and even preambles, and odd and even training sequences, are defined. The odd data frame, as well as the PHY header and data payload of the even data frame, are processed in the manner described for the training sequence anddata symbols 151 with reference toFig. 1A . The even preamble is processed in the manner described fortraining sequence 152. Hence, the data and preamble of the even frames usedifferent equalizers - The preferred training sequence described below is specifically designed to achieve the best I/Q imbalance estimation performance. The essential property required for such training sequences is that the frequency domain responses of the real and imaginary parts of the time domain signal are orthogonal. The training sequence in the discrete time domain at the symbol rate is denoted as:
wherein - One example of constructing such a training sequence x(n) is as follows: Let X(k), k = 0,1,..., Ns - 1 be real and only takes on a value +1 or -1 (for computational simplicity). Let Xe (k) = X(k) for any even k and Xe (k) = 0 for any odd k, and Xo (k) = X(k) for any odd k and Xo (k) = 0 for any even k. Let x1 (n) and xQ (n) be the Ns -point inverse discrete Fourier transform (IDFT) of Xe (k) and Xo (k) respectively. X(k)=Xe (k)+Xo (k) is then the DFT of x(n) = xI (n) + jxQ (n).
- The same training sequence as designed above can be used in both even and odd frames. For other purposes, such as identification of odd and even frames, different training sequences can also be used. For simplicity, it is assumed hereafter that the same sequence is used in both odd and even frames. However, it is noted that the processing applied to the sequence in the even and odd frames are different, as has been described before.
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Fig. 2A illustrates the structure of the variable linear equalizer 112' of the transmitter 110' with I/Q imbalance compensation, both operating at the symbol rate. The variable linear equalizer receives as input the real xI (n) and imaginary parts xQ (n) of data symbols x(n), which are separately equalized. The I/Q imbalance compensation, which follows the equalization, is represented by parameters tg(θ) wherein θ is the phase imbalance and tg( ) is the tangent function, and hQ / I (n) which is the frequency-dependent amplitude imbalance as is described in detail below. -
Fig. 2B illustrates the structure of the linear equalizer of the receiver 130' with I/Q imbalance compensation, also operating at symbol rate. The output r(n) ofSRC module 135 is the input denoted as real rI (n) and imaginary parts rQ (n). In the receiver the I/Q imbalance is represented by parameters tg(Θ) where Θ is the phase imbalance and h I/Q (n) which is the frequency-dependent amplitude imbalance. The real and imaginary components of the received signal r(n) is firstly passed to the I/Q mismatch compensation module based on the estimates of tg(Θ) and h I/Q (n). The compensated signal is then processed by the receiver equalizer RxEqz in the manner described in detail below. - Estimating the I/Q imbalance parameters and the impulse response of the
channel 120 is next described. As will be described, the odd preamble is used for estimating receiver side I/Q imbalance parameters (tg(Θ) and h I/Q (n)) and the coefficients of the linear equalization performed by linear equalizer 138'. The even preamble on the other hand is used for estimating the transmitter side I/Q imbalance parameters (tg(θ) and h Q/I (n)) and the coefficients of the variable linear equalization performed in variable linear equalizer 112' based on the estimation of the impulse response of thechannel 120 which is fed back from the receiver 130' to the transmitter 110'. - As was described with reference to
Fig. 1B , the odd preamble is pre-equalized with I/Q imbalance compensation by the transmitter 110', and more particularly the variable linear equalizer 112'. After the signal is passed through thechannel 120, an initial channel estimation is obtained in the frequency domain.Fig. 4 shows a schematic flow diagram of amethod 400 for initial estimation of thechannel 120 performed in the receiver 130'. Thesame method 400 is also applied to even frames. Hardware (processing) resources may thus be shared in an efficient implementation. The initial channel estimation obtained from even frames is fed back to the transmitter 110' for estimating the transmitter side I/Q imbalance parameters (tg(θ) and h Q/I (n)) and the coefficients of the variable linear equalization at a pre-defined time interval. - Before describing the initial estimation of the impulse response of the
channel 120 performed inmethod 400, the sampling rate conversion performed inmodule 115 is first described by way of an example. Assume a symbol rate of 3.75 Gsps and a chip rate of 5 Gsps. In the example K training sequences are used in each preamble. One training sequence serves as a cyclic prefix and will therefore absorb multipath interference from the previous frame at the receiver 130'. The training sequence at symbol rate has a length Ns , whereas the training sequence at chip rate has a length - The matched
filter 132 is assumed to be a root raised cosine (RRC) filter. For SRC from 5 Gsps to 3.75 Gsps, a polyphase filter bank consisting of three filters is required. Each filter is sampled at thechip rate 5 Gsps. - Referring to
Fig. 4 , after determining the synchronization point, a block of samples, averaged over the remaining K-1 training sequences, is input instep 410 to an N-point DFT. The output ofstep 410 is next multiplied instep 420 by a pre-computed RRC waveform in the frequency domain to apply the matched filtering ofmodule 132, resulting in a N-ppoint frequency domain received training signal S(k). - To convert the received training signal S(k) from the 5 Gsps chip rate to the 3.75 Gsps symbol rate, as is performed in the
SRC module 135 of the receiver 130', the received training signal S(k) is down-sampled instep 430 to obtain a frequency domain training signal R(k) corresponding to an Ns point time domain signal. -
Fig. 5 illustrates the effect the down-sampling has on the spectrum of the training signal. More particularly, the downsampling of the received training signal S(k) at the 5 Gsps chip rate to obtain the training signal R(k) at the 3.75 Gsps symbol rate causes overlap of the spectrum of the training signal R(k). - Note that the spectrum overlapping procedure in the channel estimation above is applicable to any other sampling rate conversion problem where the conversion ratio is a rational number. By changing the parameters of the sampling rate conversion, the width of the overlapped spectrum will change.
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- The inverse of the initial channel impulse response H̃(k) can be directly used for equalization if no I/Q mismatch exists. Note that instead of using 1/H̃(k), which is essentially a least square equalization approach, the minimal ratio combining (MRC) approach may be used. Let C(k) = S(k)X5 (k), k=0, 1, ..., N-1, where X 5(k) is the signal X(k) resampled at 5Gsps. The MRC equalizer coefficients may then be represented as:
- Hereafter, the least square equalization will be used as an example, but extension to MRC equalization would be obvious to a person with general expertise in related areas.
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- Step 210 follows where the even samples of the initial channel impulse response H̃(k), k = 0,2,..., Ns - 2, are interpolated to obtain Heven (k). Similarly, in
step 215, the odd samples of the initial channel impulse response H̃(k), k = 1,3,..., Ns - 1, are interpolated to obtain Hodd (k). Next, instep 220, Heven (k) is divided by Hodd (k) to obtain: - The receiver side I/Q imbalance parameters are then in
step 225 estimated as:
wherein (·) e and (·) o denote the conjugate symmetric part and conjugate antisymmetric part of the function in brackets respectively, and H I/O (k) represents the DFT of the frequency-dependent amplitude imbalance h I/Q (n). -
- The even training sequence is pre-equalized by the fixed
linear equalizer 114 of thetransmitter 110, without any I/Q imbalance compensation. After processing the equalized even training sequence bysteps Fig. 4 ), the received even training sequence in the frequency domain after SRC instep 430 is still denoted R(k) for convenience. Instep 440 the initial channel impulse response is estimated as by the channel estimation module 139: - The initial channel impulse response H̃(k) is then fed back to the
transmitter 110 for estimating the I/Q imbalance parameters.Fig. 13 shows a schematic flow diagram of amethod 250 of estimating the transmitter-side I/Q imbalance parameters and refining the channel impulse response H(k). -
Method 250 starts instep 255 where the even samples of the initial channel impulse response H̃(k), k = 0,2,..., Ns - 2, are interpolated to obtain Heven (k). In a similar manner, the odd samples of the initial channel impulse response H̃(k), k = 1,3,..., Ns - 1, are interpolated instep 260 to obtain Hodd (k). Instep 265 which follows Hodd (k) is divided by Heven (k) to obtain: -
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- In the preferred implementation post processing is performed in order to improve the accuracy of the I/Q imbalance parameters estimated at both the receiver 130' and the transmitter 110'. The post processing includes low-pass filtering the frequency-dependent amplitude imbalance H I/Q (k) (or H Q/I (k)) to reduce the impact of unknown channels and averaging both the imbalance parameters tg(Θ) and H I/Q (k) (or both tg(θ) and H Q/I (k)) over time to reduce the impact of noise.
- Below is another I/Q imbalance estimation and compensation approach, which is different to that described above but also takes advantages of the multiple training sequences structure.
- This approach treats I/Q imbalance as a 2x2 MIMO problem, where the I and Q channels at the transmitter 110' and receiver 130' are analysed independently. Thus there is a channel from I-transmit to I-receive, from I-transmit to Q-receive, from Q-transmit to Q-receive, and from Q-transmit to I-receive.
- In one embodiment, each preamble consists of a sequence of 64 samples (with a 32 sample cyclic prefix). The I/Q imbalance is calculated in the frequency domain. A 64 point FFT converts the 64 transmitted or received samples into 64 frequencies. The 2x2 MIMO channel is calculated at each frequency.
- As there is only a single received data point at a given frequency from a single preamble, a single preamble does not provide enough information to calculate the 2 channels I-I and Q-I channels. To overcome this without extending the preamble (which would be bad for latency), several successive preambles are used. The preambles must be different so that they do not provide redundant information. For n used preambles, at each frequency we have:
- The use of more preambles (i.e. larger value for n) gives a more robust estimate of the channel, at the expense of a longer lag time between the data used for the calculation and the new I/Q imbalance being applied.
- Note that since I and Q are real signals, the channel being calculated is conjugate symmetric in the frequency domain.
- The 2x2 channel matrix is inverted and used in the equalization. It contains both I/Q imbalance and channel information, and hence replaces the separate channel estimation and I/Q imbalance in the first approach.
- In order to reduce processing latency, a more efficient implementation may be achieved by combining the pre-equalization (with or without I/Q imbalance compensation), SRC, and pulse shaping in the transmitter 110' into one integrated transmitter filter, termed a
precoder 111. More particularly, two precoders result, the first without I/Q imbalance compensation which includes the equalization performed by the fixedlinear equalizer 114, and the second with I/Q imbalance compensation and equalization as performed by the variablelinear equalizer 112. Similarly, the matched filtering, SRC, and equalization with I/Q imbalance compensation in the receiver 130' may be combined into a single integrated receiver filter, termed aRxFilter 131. - The
RxFilter 131 andprecoder 111 are implemented as respective filter banks, each with a number of polyphase filters depending on the SRC ratio. For the example case described above, with 3.75Gsps symbol rate and 5Gsps chip rate, the number of polyphase filters is 3, each being a time-shifted version of a base filter. -
Fig. 6A shows a schematic block diagram of the structure of theRxFilter 131 for the example case. In this case, the length of eachpolyphase filter receiver 130 cannot afford to use anRxFilter 131 that is too long. -
Fig. 6B shows a schematic block diagram of the structure of a polyphase filter. The polyphase filter has twoparts adder 613 which adds the outputs fromparts -
- The Ns -point impulse responses A(k) and B(k) are then expanded to N-point vectors as:
which are composed of the 1st toFig. 5 ; -
- The two parts of each polyphase filter for p =0,1, or 2 , denoted as a p and b p , are then obtained by converting vectors A p and B p to the time domain by applying N-point IDFT and then circularly shifting the IDFT outputs by a pre-designed number P, which is the length of the precursor part of the filters, such that the maximum tap is at P+1.
- Finally, the filter length is truncated to the desired length, if required.
-
Fig. 7 shows a schematic block diagram of the structure of theprecoder 111 with I/Q imbalance compensation and equalization as performed by the variable linear equalizer 112' according to the example case described above. Theprecoder 111 includes 3polyphase filters polyphase filter Fig. 6B . For simplicity, the two filter parts are represented by vectors a p and b p with filter index p = 0,1, or 2 as before, but now a p and j b p are used for filtering the real part and imaginary part of the data symbols after serial-to-parallel conversion 710 respectively. The precursor length of thefilters - Firstly, the time domain channel impulse h(n) and the frequency-dependent amplitude I/Q imbalance h Q/I (n) represented in time domain are calculated by performing Ns -point IDFT to the estimated channel impulse response H(k) and frequency-dependent amplitude I/Q imbalance H Q/I (n) in the frequency domain.
-
- Next, h'(n) and h' Q/I (n) are converted to the frequency-domain by applying Ls - point FFT as H'(k) and H' Q/I (n) respectively.
-
-
-
- The two parts of each polyphase filter for p =0,1, or 2 , denoted as a p and b p are obtained by converting the vectors A p and B p to the time domain by applying L-point IDFT and then circularly shifting the IDFT outputs by the precursor length P.
- It is noted that the value of precursor length P is chosen such that no significant power in the variable linear equalization precoder and receiver side linear equalization is truncated.
- The filters for the fixed linear equalization precoder may be computed similarly to those of the variable linear equalization precoder described above, but using a fixed channel impulse response obtained by initial system calibration and without considering I/Q imbalance compensation. The initial value of the variable linear equalization precoder may be set similarly to the fixed linear equalization precoder.
- As is described above with reference to
Fig. 1A , CSI, estimated at thereceiver 130 using thetraining sequence 152, is fed back from thereceiver 130 to thetransmitter 110, and then used for computing the coefficients and I/Q imbalance parameters of the variablelinear equalizer 112. The CSI, together with other information that needs to be sent back, forms apacket 800 as shown inFig. 8 . Thepacket 800 consists of a stream start delimiter (SSD) 801, theCSI 802,other information bits 803, and a 16 bit CRC check 804. - The
SSD 801 is used by the transmitter 110 (which is the receiver of the packet 800) for detecting the start of the stream, and is a fixed 8-bit sequence. Before transmitting, thepacket 800 is encoded by an 8b/10b line code that maps 8-bit symbols to 10-bit symbols to achieve DC-balance and bounded disparity, and to allow packet synchronization and reasonable clock recovery. TheSSD 801 is treated as a control symbol, and encoded following, e.g., the K. 28.5 rule in the 8b/10b code. Packet synchronization can then be referred to the encoded SSD symbol because of its uniqueness in the coded stream. - The
CSI packet 800 is typically sent to thetransmitter 110 by using dedicated feedback channels. However, in a full-duplex system having two transceivers in communication with each other, according to the present disclosure feedback bits are modulated with the preamble of the data frames being communicated to the other transceiver for use in itstransmitter 110, and hence does not require overhead for feedback. The speed of feedback is slower, but it has zero overhead. - Each bit in the
feedback packet 800 is binary phase shift key (BPSK) modulated to be eithersymbol 1 or -1, and is multiplied to the preamble in each frame. As is illustrated inFig. 9 , the symbol multiplied to the n-th odd frame is also multiplied to the coded and modulated data payload in the n-th even frame and the (n+1)-th odd frame. This is to avoid the {1,-1} -ambiguity in equalization. - In very high speed communications, some baseband processing, such as synchronization, CFO estimation, channel estimation and generating receiver equalization coefficients, may not be able to be completed in time due to the processing requirements at the
receiver 130. For applications where thechannel 120 remains unchanged (or the change is small) over a few frames, this baseband processing is implemented over the period of the current frame. The results are then applied to the next frame. In particular, for the proposed frame structure described with reference toFig. 3 where odd and even frames are transmitted alternately, as is illustrated inFig. 10 , the processing is done over the entire odd frame. The results of the processing are applied to the next two frames, instead of the current frame. - Since the RxFilter computed from the channel estimates obtained from an odd preamble will be used for the next two frames, a specific precoding structure is used to avoid these coefficients become incorrect when the precoder in the transmitter is changed. That precoding structure is illustrated in
Fig. 11 . When a precoder is to be updated, precoder (t+1) is first applied to the odd preamble of the current frame, while precoder (t) is still applied to the rest of this frame. For the following frames, only precoder (t+1) is applied. - At the
receiver 130, thechannel 120 over consecutive frames generally changes insignificantly, and hence equalization coefficients over consecutive frames derived from channel estimates are also similar. However, there will always be minor clock differences between thetransmitter 110 andreceivers 130 at different nodes. The minor clock difference over a long period will cause the change of synchronization point. When the change is small, no adjustment is necessary, only if the same synchronization point and equalization coefficients are used with the previous frame. However, when such an accumulated timing drift is significantly large, the synchronization point needs to be adjusted to make sure at least the signals used for the channel estimation is always part of the preamble. When it is decided to adjust the synchronization by K samples, the channel estimates are multiplied by a phase shifting sequence exp(j2π K [0, 1,..., Ns -1]/Ns ), where K may be positive or negative, depending on how the phase is shifted. - All the processing described above, including channel estimation, I/Q imbalance parameter estimation and compensation, computation of the variable linear equalization precoder, the fixed linear equalization precoder and RxFilter can be implemented flexibly in a combined field-programmable gate array (FPGA) and general purpose processors (such as a personal computer (PC)).
Fig. 14 shows onepossible configuration 900, which considers both the fast processing capability of FPGA and the powerful computational capability of PCs. In thatconfiguration 900 the channel estimation and I/Q imbalance parameter estimation are performed in amodule 901 implemented in aFPGA 903. The computation of the variable linear equalization precoder, the fixed linear equalization precoder and RxFilter is then performed inmodule 905 implemented in a PC. Finally, precoding, equalization and I/Q imbalance compensation are performed inmodule 902, also implemented in theFPGA 903.
Claims (11)
- A communication system (100) comprising:a transmitter (110) adapted to transmit at least a first frame and a second consecutive frame, wherein the first frame contains a first training sequence and the second frame contains a second training sequence,and wherein the first training sequence and the second training sequence are different training sequences;the transmitter (110) comprising a time variable linear equalizer (112) adapted to apply equalization to at least the first training sequence to produce an equalized first training sequence; andthe transmitter comprising a fixed linear equalizer (114) adapted to apply equalization to the second training sequence to produce an equalized second training sequence;a receiver (130) adapted to receive the first frame and the second frame transmitted from the transmitter (110) over a channel (120),the receiver comprising a receiver linear equalizer (138) adapted to receive the equalized first training sequence and to apply equalization thereto, and to estimate equalization coefficients to be used by the receiver linear equalizer (138) in future equalization;the receiver (130) further comprising a channel estimator (139) adapted to receive the equalized second training sequence, and to estimate a channel state of the channel (120) from the equalized second training sequence; andwherein the receiver is adapted to feed back the channel state to the transmitter (110), which is adapted to estimate from the channel state equalization coefficients to be used for generating the coefficients of the time variable linear equalizer (112) in future equalization.
- A communication system according to claim 1, wherein the time variable linear equalizer (112) of the transmitter (110) is further adapted to apply equalization to a data payload to produce an equalized data payload and, upon receipt of the equalized data payload by the receiver (130), the receiver linear equalizer (138) is further adapted to apply equalization to the equalized data payload.
- A communication system according to claim 1 wherein:the time variable linear equalizer (112) and the receiver linear equalizer (138) further are adapted to perform I/Q mismatch compensation;the receiver linear equalizer (138) is further adapted to estimate parameters to be used for the I/Q mismatch compensation in the receiver linear equalizer (138) from the equalized first training sequence; andthe transmitter (110) is further adapted to estimate parameters to be used for the I/Q mismatch compensation in the time variable linear equalizer (112) from the channel state fed back from the receiver (130).
- A communication system according to claim 3, wherein the I/Q mismatch compensation is performed using linear filters.
- A communication system according to claim 1, wherein the equalization applied to the second training sequence by the fixed linear equalizer (114) has a constant impulse response in the frequency domain.
- A communication system according to claim 1, wherein the first and second training sequences are orthogonal in the frequency domain.
- A communication system according to claim 1, wherein the channel state is fed back to the transmitter (110) over a return channel.
- A method for performing equalization in a communication system, the method comprising the steps of
applying equalization to at least a first training sequence by a time variable linear equalizer (112) of a transmitter to produce an equalized first training sequence; wherein the first training sequence is contained in a first frame and
applying equalization to a second training sequence by a fixed linear equalizer (114) of the transmitter to produce an equalized second training sequence; wherein the second training sequence is contained in a second frame consecutive to the first frame,
transmitting by the transmitter the first and consecutive second frame containing the equalized first and second training sequence to the receiver (130) over the channel (120);
receiving by a receiver the first frame and the second frame transmitted from the transmitter (110) over a channel (120),
receiving the equalized first training sequence by a receiver linear equalizer (138) and applying equalization thereto, and estimating equalization coefficients to be used by the receiver linear equalizer (138) in future equalization;
receiving by a channel estimator (139) the equalized second training sequence, and estimating a channel state of the channel (120) from the equalized second training sequence; and
feeding back the channel state to the transmitter (110) from the receiver; and
estimating by the transmitter from the channel state equalization coefficients to be used for generating the coefficients of the time variable linear equalizer (112) in future equalization. - A method according to claim 8, further comprising the steps of:applying, by the time variable linear equalizer (112) of the transmitter (110), equalization to a data payload to produce an equalized data payload;transmitting the equalized data payload by the transmitter to the receiver (130) over the channel (120); andapplying, by the receiver linear equalizer (138), equalization to the equalized data payload.
- A method according to claim 8, wherein the time variable linear equalizer (112) and the receiver linear equalizer (138) further perform I/Q mismatch compensation, and the method further comprises the steps of:estimating by the receiver linear equalizer (138) parameters to be used for the I/Q mismatch compensation in the receiver linear equalizer (138) from the equalized first training sequence; andestimating by the transmitter (110) further parameters to be used for the I/Q mismatch compensation in the time variable linear equalizer (112) from the channel state fed back from the receiver (130).
- A method according to claim 8, wherein the channel state is fed back to the transmitter (110) from the receiver (130) over a return channel.
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