EP2811644B1 - Motor control device - Google Patents
Motor control device Download PDFInfo
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- EP2811644B1 EP2811644B1 EP13743139.1A EP13743139A EP2811644B1 EP 2811644 B1 EP2811644 B1 EP 2811644B1 EP 13743139 A EP13743139 A EP 13743139A EP 2811644 B1 EP2811644 B1 EP 2811644B1
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- Prior art keywords
- command
- current
- magnetic flux
- motor
- calculating unit
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P23/00—Arrangements or methods for the control of AC motors characterised by a control method other than vector control
- H02P23/03—Arrangements or methods for the control of AC motors characterised by a control method other than vector control specially adapted for very low speeds
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/02—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for optimising the efficiency at low load
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/04—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for very low speeds
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P23/00—Arrangements or methods for the control of AC motors characterised by a control method other than vector control
- H02P23/02—Arrangements or methods for the control of AC motors characterised by a control method other than vector control specially adapted for optimising the efficiency at low load
Definitions
- the present invention relates to a motor control apparatus.
- Patent Literature 1 Japanese Patent Application Laid-Open No. H9-191700 ("0043", Formula (16))
- Patent Literature 1 a division of square roots and a multiplication and power calculation of a trigonometric function are included in a calculation formula of a square root. Therefore, there is a problem in that control calculation cannot be easily performed.
- a motor control apparatus controls rotational speed of a control target motor
- the motor control apparatus performs V/F fixing control for fixing a voltage-frequency ratio from a start to a low-speed region or a medium-speed region.
- V/F fixing control for fixing a voltage-frequency ratio from a start to a low-speed region or a medium-speed region.
- arithmetic processing using the numerical formula is performed in all speed regions from a start to a high-speed region. There is a problem in that effective control corresponding to a control form is not always performed.
- a voltage-frequency ratio corresponding to a rotational speed command for the motor rather than a value corresponding to a load is determined.
- An output voltage characteristic is given such that an optimum characteristic is obtained at rating time. Therefore, in the low-speed region or a light-load driving region, an excessive voltage is applied to the motor. There is a problem in that a motor loss is larger than an optimum value and highly efficient operation is not always performed.
- the present invention has been devised in view of the above and it is an object of the present invention to provide a motor control apparatus that, in particular, reduces a motor loss in a low-speed region or a light-load driving region to enable further improvement of efficiency in a motor control apparatus that uses a V/F fixing control system.
- a motor control apparatus that divides an electric current flowing into and out of a motor driven by an inverter into a torque current and an excitation current and individually controls the torque current and the excitation current
- the motor control apparatus is constructed to include: a secondary-magnetic-flux-command calculating unit including a first calculating unit that calculates a minimum current secondary magnetic flux command for minimizing a current root-mean-square value due to the torque current and the excitation current; and a PWM-signal generating unit that generates a torque current command for outputting a torque command and an excitation current command for outputting the secondary magnetic flux command, performs vector control such that a detection value of the torque current and a detection value of the excitation current respectively coincide with the torque current command and the excitation current command, and generates a control signal for turning on and off a switching element included in the inverter.
- FIG. 1 is a block diagram of a configuration example of a motor control apparatus according to a first embodiment.
- a motor control apparatus 1 according to the first embodiment is a control apparatus by a vector control system that divides an electric current (a primary current) flowing into and out of a motor 12 driven by an inverter 11 into a torque current and an excitation current and individually controls the torque current and the excitation current.
- the motor control apparatus 1 includes a torque-command calculating unit 4, a secondary-magnetic-flux-command calculating unit 5, a motor-constant calculating unit 6, a coordinate converting unit 7, a speed control unit 8, and a PWM-signal generating unit 9 that generates a PWM signal as a voltage command.
- the motor control apparatus 1 generates, on the basis of the notch command, the variable load signal, the motor currents IU, IV, and IW, and the PG pulse signal, PWM signals (U, V, W, X, Y, and Z) serving as voltage commands output from the PWM-signal generating unit 9 located in a last stage for controlling the inverter 11 to be,
- the operation of the motor control apparatus 1 according to the first embodiment is explained.
- the notch command and the variable load signal and an internally-generated inverter frequency FINV are input to the torque-command calculating unit 4.
- the torque-command calculating unit 4 has a pattern of a torque command (a torque pattern) for fixing torque when speed is equal to or lower than a threshold as shown in the figure, and for reducing the torque when the speed exceeds the threshold.
- the torque-command calculating unit 4 generates a torque command PTR corresponding to the notch command, the variable load signal, and the inverter frequency FINV and outputs the torque command PTR to the secondary-magnetic-flux-command calculating unit 5 and the PWM-signal generating unit 9.
- the inverter frequency FINV can be an input signal from the outside.
- the motor-constant calculating unit 6 calculates a motor constant of the motor 12 represented as an equivalent circuit model.
- the motor constant primary resistance, secondary resistance, primary inductance, secondary inductance, mutual inductance, the number of pole pairs, and the like are representative.
- the motor-constant calculating unit 6 outputs at least values of secondary inductance L2R and the number of pole pairs Pm to the secondary-magnetic-flux-command calculating unit 5.
- the motor 12 cannot change the number of pole pairs Pm because of the structure of the motor 12, the number of pole pairs Pm output to the secondary-magnetic-flux-command calculating unit 5 is a fixed value.
- the motor 12 can change the number of pole pairs Pm, a value corresponding to a change in the number of pole pairs Pm is output to the secondary-magnetic-flux-command calculating unit 5.
- the phase motor currents IU, IV, and IW detected by the current detectors 14 are input to the coordinate converting unit 7.
- the coordinate converting unit 7 generates a d-axis current I1DF and a q-axis current I1QF obtained by converting the phase motor currents IU, IV, and IW, which are current detection values of a three-phase coordinate system, into current detection values of a dq-axis coordinate system.
- the d-axis current I1DF is input to both of the PWM-signal generating unit 9 and the secondary-magnetic-flux-command calculating unit 5 and the q-axis current I1QF is input to the PWM-signal generating unit 9.
- phase motor currents IU, IV, and IW are input to the coordinate converting unit 7.
- the d-axis current I1DF and the q-axis current I1QF can be calculated as long as any two kinds of information among the phase motor currents IU, IV, and IW are present.
- the inverter frequency FINV, the torque command PTR, the secondary inductance L2R, the number of pole pairs Pm, and the d-axis current I1DF are input to the secondary-magnetic-flux-command calculating unit 5.
- the secondary-magnetic-flux-command calculating unit 5 generates a secondary magnetic flux command F2R on the basis of the inverter frequency FINV, the torque command PTR, the secondary inductance L2R, the number of pole pairs Pm and the d-axis current I1DF, and outputs the secondary magnetic flux command F2R to the PWM-signal generating unit 9. Note that the internal configuration and a more detailed operation of the secondary-magnetic-flux-command calculating unit 5 are explained below.
- the PG pulse signal detected by the pulse generator (PG) 13 is input to the speed control unit 8.
- the speed control unit 8 generates a motor frequency FM, which is a rotation frequency of the motor 12, on the basis of information such as a cycle of the PG pulse signal and the number of pulses per one cycle included in the PG pulse signal and outputs the motor frequency FM to the PWM-signal generating unit 9.
- the torque command PTR, the secondary magnetic flux command F2R, the d-axis current I1DF, the q-axis current I1QF, and the motor frequency FM are input to the PWM-signal generating unit 9.
- the inverter frequency FINV and a filter capacitor voltage EFC, which is a voltage of a not-shown filter capacitor provided on a direct-current section side of the inverter 11, are also input to the PWM-signal generating unit 9.
- the PWM-signal generating unit 9 internally generates a torque current command and an excitation current command.
- the PWM-signal generating unit 9 performs vector control such that the q-axis current I1QF, which is a detection value of a torque current, and the d-axis current I1DF, which is a detection value of an excitation current, respectively coincide with the torque current command and the excitation current command.
- the PWM-signal generating unit 9 generates PWM signals U, V, W, X, Y, and Z for controlling a switching element 16 included in the inverter 11 to be turned on and off and outputs those signals to the inverter 11.
- the PWM signals U, V, W, X, y, and Z are an example obtained when the inverter 11 is a three-phase inverter. Switching signals for switching elements forming an upper arm correspond to U, V, and W. Switching signals for switching elements forming a lower arm correspond to X, Y, and Z.
- FIG. 2 is a diagram for explaining a relation between a motor constant and d and q axis currents.
- FIG. 3 is a diagram for explaining a relation between d and q axis currents on a d-q plane and torque.
- a torque current command I1QR and an excitation current command I1DR generally used in the control apparatus, which performs the vector control, can be represented as indicated by the following formulas using the torque command PTR, the secondary inductance L2R, the secondary magnetic flux command F2R, the number of pole pairs Pm, a mutual inductance MR, and a secondary resistance R2R as shown in FIG. 2 :
- I 1 ⁇ QR PTR / F 2 R ⁇ 1 / Pm ⁇ L 2 R / MR
- I 1 ⁇ DR F 2 R / MR + L 2 R / MR ⁇ 1 / R 2 R ⁇ d F 2 R / dt
- the torque command PTR has a magnitude proportional to a product of the torque current command I1QR and the excitation current command I1DR, that is, an area of a portion of a rectangle indicated by hatching in FIG. 3 . Therefore, when the torque command PTR is given, it is possible to select I1QR and I1DR having any values satisfying a condition that I1QR ⁇ I1DR is fixed (a condition that the area is fixed (i.e., the torque is fixed).
- the motor control apparatus makes use of the idea explained above.
- the excitation current command I1DR can be represented as indicated by the following formula using the torque command PTR.
- I 1 ⁇ DR ⁇ L 2 R / Pm ⁇ PTR / MR
- the secondary magnetic flux command F2R can be calculated according to the torque command PTR, the secondary inductance L2R, and the number of pole pairs Pm. Therefore, the secondary-magnetic-flux-command calculating unit 5 shown in FIG. 1 is configured as shown in FIG. 4.
- FIG. 4 is a block diagram of a configuration example of the secondary-magnetic-flux-command calculating unit 5 according to the first embodiment.
- the secondary-magnetic-flux-command calculating unit 5 includes a minimum-current-secondary-magnetic-flux-command calculating unit 21 functioning as a first calculating unit, a magnetic-flux-command-compensation calculating unit 22 functioning as a second calculating unit, a multiplier 23, and a subtracter 24.
- the torque command PTR, the secondary inductance L2R, and the number of pole pairs Pm are input to the minimum-current-secondary-magnetic-flux-command calculating unit 21.
- the minimum-current-secondary-magnetic-flux-command calculating unit 21 performs arithmetic processing indicated by Formula (6) on the basis of the torque command PTR, the secondary inductance L2R, and the number of pole pairs Pm and outputs a result of the arithmetic processing as a minimum current secondary magnetic flux command F2R1.
- FIG. 5 is a diagram of a loss curve with respect to the secondary magnetic flux command F2R.
- a solid line represents a copper loss and a broken line represents an iron loss.
- the copper loss and the iron loss are predominant in a loss that occurs in a motor. Therefore, if a sum of the copper loss and the iron loss can be minimized, it is possible to substantially minimize the motor loss.
- the minimum current secondary magnetic flux command value F2R1 for minimizing the copper loss is not a magnetic flux condition for a minimum loss when the iron loss is taken into account as well.
- the inverter frequency FINV is small and the voltage V applied to the motor is small, the iron loss is small, so that it can be ignored.
- the iron loss is innegligibly large.
- the secondary magnetic flux command value F2R is slightly reduced from the minimum current secondary magnetic flux command value F2R1
- a decrease of the iron loss is larger than an increase in the copper loss, and the total loss of the copper loss and the iron loss also decreases. That is, the secondary magnetic flux command value F2R for minimizing the total loss of the copper loss and the iron loss becomes smaller than F2R1.
- the magnetic-flux-command-compensation calculating unit 22 for calculating, on the basis of the frequency of the motor, compensation for a decrease from the minimum current secondary magnetic flux command F2R1 of the secondary magnetic flux command value for minimizing the total loss of the copper loss and the iron loss of the motor is provided.
- An output of the magnetic-flux-command-compensation calculating unit 22 and the d-axis current I1DF are multiplied together in the multiplier 23.
- a multiplied value obtained by the multiplication is subtracted from an output of the minimum-current-secondary-magnetic-flux-command calculating unit 21.
- an iron loss consideration table for calculating an iron loss according to a motor characteristic is prepared in advance at its designing stage.
- the magnetic-flux-command-compensation calculating unit 22 generates, on the basis of the input inverter frequency FINV, an optimum compensation coefficient for reducing a loss due to the iron loss with respect to the d-axis current I1DF and outputs the optimum compensation coefficient to the multiplier 23.
- the minimum current secondary magnetic flux command F2R1 generated by the minimum-current-secondary-magnetic-flux-command calculating unit 21 is output to the PWM-signal generating unit 9 directly as the secondary magnetic flux command F2R.
- the voltage-frequency ratio is fixed, the secondary magnetic flux command for minimizing the current root-mean-square value due to the torque current command and the excitation current command in the driving region for driving at fixed torque is calculated, and the vector control is performed such that a detection value of the torque current and the excitation current respectively coincide with the torque current command and of the excitation current command. Therefore, it is possible to reduce the motor loss including the copper loss and the iron loss. It is possible to realize the motor control apparatus that enables more highly efficient operation control.
- the switching element 16 included n the inverter 11 is explained.
- the switching element 16 used in the inverter 11 is a semiconductor switching element made of silicon (Si) (IGBT, MOSFET, etc.; hereinafter abbreviated as "Si-SW").
- Si-SW semiconductor switching element made of silicon
- the technology explained in the first embodiment can be configured using the general Si-SW.
- the technology in the first embodiment is not limited to the Si-SW. It is naturally possible to use a semiconductor switching element made of silicon carbide (SiC), which attracts attention in recent years (hereinafter abbreviated as "SiC-SW"), instead of silicon (Si) as the switching element 16.
- SiC-SW silicon carbide
- a loss in the inverter 11 is mainly a switching loss and a conduction loss of the switching element 16.
- the SiC-SW is formed in a MOSFET structure, it is expected that the switching loss can be greatly reduced.
- the SiC-WS is formed in the MOSFET structure, a conduction loss of the MOSFET increase in proportion to a square of an electric current. Therefore, it is possible to reduce the conduction loss by reducing a current value flowing to the SiC-SW.
- the motor control apparatus in the first embodiment it is possible to minimize an electric current for generating the same torque. Therefore, by using the SiC-SW as the switching element 16 included in the inverter 11 in the first embodiment, it is possible to greatly reduce the conduction loss. Consequently, it is possible to reduce the loss in the inverter 11. It is possible to realize the motor control apparatus that enables more highly efficient motor control.
- an output frequency of the inverter 11 is controlled by sequentially switching a plurality of control modes including a multi-pulse mode and a one-pulse mode.
- the switching element 16 formed by a wide band gap semiconductor such as SiC it is possible to perform asynchronous PWM control in all control regions. Therefore, a loss reduction effect by the motor control apparatus in this embodiment extends over all the regions. It is possible to perform highly efficient motor control in all the regions. In particular, when a current value is set high to perform the asynchronous PWM control in all the regions, the loss reduction effect for the motor is extremely large.
- the SiC is an example of a semiconductor called wide band gap semiconductor (on the other hand, Si is called narrow band gap semiconductor).
- Si is an example of a semiconductor called wide band gap semiconductor
- a semiconductor formed using a gallium nitride material or diamond also belongs to the wide band gap semiconductor. Characteristics of the gallium nitride material and the diamond have many similarities to the characteristics of the silicon carbide.
- the switching elements formed by such wide band gap semiconductors have high voltage resistance and high allowable current density. Therefore, it is possible to reduce the size of the switching elements. By using the switching elements reduced in the size, it is possible to reduce the size of the semiconductor module incorporating these elements.
- the switching elements formed by the wide band gap semiconductors also have high heat resistance. Therefore, in the case of the switching element that requires a cooling mechanism such as a heat sink, it is possible to reduce the size of the cooling mechanism. It is possible to further reduce the size of the switching element module.
- the present invention is a motor control apparatus that reduces a motor loss in a low-speed region or a low-load driving region.
Description
- The present invention relates to a motor control apparatus.
- There has been disclosed a method of analytically deriving a numerical formula representing an excitation current command value for minimizing a total loss that occurs in an induction electric motor (motor) and controlling the induction motor with current control using the derived excitation current command value (e.g., Patent Literature 1).
- Patent Literature 1: Japanese Patent Application Laid-Open No.
H9-191700 - However, in the numerical formula derived in
Patent Literature 1, a division of square roots and a multiplication and power calculation of a trigonometric function are included in a calculation formula of a square root. Therefore, there is a problem in that control calculation cannot be easily performed. - When a motor control apparatus controls rotational speed of a control target motor, for example, in general, the motor control apparatus performs V/F fixing control for fixing a voltage-frequency ratio from a start to a low-speed region or a medium-speed region. On the other hand, in the conventional technology, arithmetic processing using the numerical formula is performed in all speed regions from a start to a high-speed region. There is a problem in that effective control corresponding to a control form is not always performed.
- In the V/F fixing control system, a voltage-frequency ratio corresponding to a rotational speed command for the motor rather than a value corresponding to a load is determined. An output voltage characteristic is given such that an optimum characteristic is obtained at rating time. Therefore, in the low-speed region or a light-load driving region, an excessive voltage is applied to the motor. There is a problem in that a motor loss is larger than an optimum value and highly efficient operation is not always performed.
- The present invention has been devised in view of the above and it is an object of the present invention to provide a motor control apparatus that, in particular, reduces a motor loss in a low-speed region or a light-load driving region to enable further improvement of efficiency in a motor control apparatus that uses a V/F fixing control system.
- In order to solve the aforementioned problems, a motor control apparatus that divides an electric current flowing into and out of a motor driven by an inverter into a torque current and an excitation current and individually controls the torque current and the excitation current, the motor control apparatus is constructed to include: a secondary-magnetic-flux-command calculating unit including a first calculating unit that calculates a minimum current secondary magnetic flux command for minimizing a current root-mean-square value due to the torque current and the excitation current; and a PWM-signal generating unit that generates a torque current command for outputting a torque command and an excitation current command for outputting the secondary magnetic flux command, performs vector control such that a detection value of the torque current and a detection value of the excitation current respectively coincide with the torque current command and the excitation current command, and generates a control signal for turning on and off a switching element included in the inverter.
- According to the present invention, there is an effect that, in particular, a motor loss in a low-speed region or a light-load driving region is further reduced to enable a more highly efficient operation in a motor control apparatus that uses a V/F fixing control system.
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FIG. 1 is a block diagram of a configuration example of a motor control apparatus according to a first embodiment. -
FIG. 2 is a diagram for explaining a relation between a motor constant and d and q axis currents. -
FIG. 3 is a diagram for explaining a relation between d and q axis currents and torque. -
FIG. 4 is a block diagram of a configuration example of a secondary-magnetic-flux-command calculating unit according to the first embodiment. -
FIG. 5 is a diagram of a loss curve with respect to a secondary magnetic flux command. - A motor control apparatus according to embodiments of the present invention is explained with reference to the accompanying drawings. Note that the present invention is not limited by the embodiments explained below.
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FIG. 1 is a block diagram of a configuration example of a motor control apparatus according to a first embodiment. Amotor control apparatus 1 according to the first embodiment is a control apparatus by a vector control system that divides an electric current (a primary current) flowing into and out of amotor 12 driven by aninverter 11 into a torque current and an excitation current and individually controls the torque current and the excitation current. As shown in the figure, themotor control apparatus 1 includes a torque-command calculating unit 4, a secondary-magnetic-flux-command calculating unit 5, a motor-constant calculatingunit 6, a coordinate converting unit 7, aspeed control unit 8, and a PWM-signal generatingunit 9 that generates a PWM signal as a voltage command. - A notch command and a variable load signal output from a not-shown external control apparatus, phase (U phase, V phase, and W phase) motor currents IU, IV, and IW detected by current detectors 14 (14U, 14V, and 14W) provided between the
inverter 11 and the electric motor (the motor) 12, a PG pulse signal detected by a pulse generator (PG) 13 provided in themotor 12, and the like are input to themotor control apparatus 1. Themotor control apparatus 1 generates, on the basis of the notch command, the variable load signal, the motor currents IU, IV, and IW, and the PG pulse signal, PWM signals (U, V, W, X, Y, and Z) serving as voltage commands output from the PWM-signal generatingunit 9 located in a last stage for controlling theinverter 11 to be, - The operation of the
motor control apparatus 1 according to the first embodiment is explained. First, the notch command and the variable load signal and an internally-generated inverter frequency FINV are input to the torque-command calculating unit 4. The torque-command calculating unit 4 has a pattern of a torque command (a torque pattern) for fixing torque when speed is equal to or lower than a threshold as shown in the figure, and for reducing the torque when the speed exceeds the threshold. The torque-command calculating unit 4 generates a torque command PTR corresponding to the notch command, the variable load signal, and the inverter frequency FINV and outputs the torque command PTR to the secondary-magnetic-flux-command calculating unit 5 and the PWM-signal generating unit 9. Note that the inverter frequency FINV can be an input signal from the outside. - The motor-constant calculating
unit 6 calculates a motor constant of themotor 12 represented as an equivalent circuit model. As the motor constant, primary resistance, secondary resistance, primary inductance, secondary inductance, mutual inductance, the number of pole pairs, and the like are representative. In the first embodiment, the motor-constant calculatingunit 6 outputs at least values of secondary inductance L2R and the number of pole pairs Pm to the secondary-magnetic-flux-command calculating unit 5. When themotor 12 cannot change the number of pole pairs Pm because of the structure of themotor 12, the number of pole pairs Pm output to the secondary-magnetic-flux-command calculating unit 5 is a fixed value. On the other hand, when themotor 12 can change the number of pole pairs Pm, a value corresponding to a change in the number of pole pairs Pm is output to the secondary-magnetic-flux-command calculating unit 5. - The phase motor currents IU, IV, and IW detected by the current detectors 14 (14U, 14V, and 14W) are input to the coordinate converting unit 7. The coordinate converting unit 7 generates a d-axis current I1DF and a q-axis current I1QF obtained by converting the phase motor currents IU, IV, and IW, which are current detection values of a three-phase coordinate system, into current detection values of a dq-axis coordinate system. Of these electric currents, the d-axis current I1DF is input to both of the PWM-signal generating
unit 9 and the secondary-magnetic-flux-command calculating unit 5 and the q-axis current I1QF is input to the PWM-signal generating unit 9. Note that, inFIG. 1 , all the phase motor currents IU, IV, and IW are input to the coordinate converting unit 7. However, the d-axis current I1DF and the q-axis current I1QF can be calculated as long as any two kinds of information among the phase motor currents IU, IV, and IW are present. - The inverter frequency FINV, the torque command PTR, the secondary inductance L2R, the number of pole pairs Pm, and the d-axis current I1DF are input to the secondary-magnetic-flux-
command calculating unit 5. The secondary-magnetic-flux-command calculating unit 5 generates a secondary magnetic flux command F2R on the basis of the inverter frequency FINV, the torque command PTR, the secondary inductance L2R, the number of pole pairs Pm and the d-axis current I1DF, and outputs the secondary magnetic flux command F2R to the PWM-signal generating unit 9. Note that the internal configuration and a more detailed operation of the secondary-magnetic-flux-command calculating unit 5 are explained below. - The PG pulse signal detected by the pulse generator (PG) 13 is input to the
speed control unit 8.
Thespeed control unit 8 generates a motor frequency FM, which is a rotation frequency of themotor 12, on the basis of information such as a cycle of the PG pulse signal and the number of pulses per one cycle included in the PG pulse signal and outputs the motor frequency FM to the PWM-signal generating unit 9. - In this way, the torque command PTR, the secondary magnetic flux command F2R, the d-axis current I1DF, the q-axis current I1QF, and the motor frequency FM are input to the PWM-
signal generating unit 9. In addition to these signals (information), the inverter frequency FINV and a filter capacitor voltage EFC, which is a voltage of a not-shown filter capacitor provided on a direct-current section side of theinverter 11, are also input to the PWM-signal generating unit 9. The PWM-signal generatingunit 9 internally generates a torque current command and an excitation current command. The PWM-signal generatingunit 9 performs vector control such that the q-axis current I1QF, which is a detection value of a torque current, and the d-axis current I1DF, which is a detection value of an excitation current, respectively coincide with the torque current command and the excitation current command. The PWM-signal generatingunit 9 generates PWM signals U, V, W, X, Y, and Z for controlling aswitching element 16 included in theinverter 11 to be turned on and off and outputs those signals to theinverter 11. Note that the PWM signals U, V, W, X, y, and Z are an example obtained when theinverter 11 is a three-phase inverter. Switching signals for switching elements forming an upper arm correspond to U, V, and W. Switching signals for switching elements forming a lower arm correspond to X, Y, and Z. - An arithmetic formula applied to the
motor control apparatus 1 in the first embodiment is explained with reference toFIG. 2 and FIG. 3 and the like.FIG. 2 is a diagram for explaining a relation between a motor constant and d and q axis currents.FIG. 3 is a diagram for explaining a relation between d and q axis currents on a d-q plane and torque. - A torque current command I1QR and an excitation current command I1DR generally used in the control apparatus, which performs the vector control, can be represented as indicated by the following formulas using the torque command PTR, the secondary inductance L2R, the secondary magnetic flux command F2R, the number of pole pairs Pm, a mutual inductance MR, and a secondary resistance R2R as shown in
FIG. 2 : - In a V/F fixing control region, the torque command PTR is controlled to a substantially fixed value. Therefore, a second term of Formula (2) representing a temporal change of the secondary magnetic flux command F2R can be put as zero. The secondary magnetic flux command F2R can be represented by the following formula using an excitation current command 11DR:
FIG. 3 . Therefore, when the torque command PTR is given, it is possible to select I1QR and I1DR having any values satisfying a condition that I1QR×I1DR is fixed (a condition that the area is fixed (i.e., the torque is fixed). On the other hand, a current root-mean-square value I1=√(I1DR2+I1QR2)/√3 is minimized when I1QR=I1DR among the I1QR and I1DR having any values, that is, a hatching portion shown inFIG. 4 is a square. - The motor control apparatus according to the first embodiment makes use of the idea explained above. When the condition I1QR=I1DR and the condition of Formula (3) are applied to Formula (4), the excitation current command I1DR can be represented as indicated by the following formula using the torque command PTR.
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- According to Formula (6), the secondary magnetic flux command F2R can be calculated according to the torque command PTR, the secondary inductance L2R, and the number of pole pairs Pm. Therefore, the secondary-magnetic-flux-
command calculating unit 5 shown inFIG. 1 is configured as shown inFIG. 4. FIG. 4 is a block diagram of a configuration example of the secondary-magnetic-flux-command calculating unit 5 according to the first embodiment. As shown in the figure, the secondary-magnetic-flux-command calculating unit 5 includes a minimum-current-secondary-magnetic-flux-command calculating unit 21 functioning as a first calculating unit, a magnetic-flux-command-compensation calculating unit 22 functioning as a second calculating unit, a multiplier 23, and asubtracter 24. - In the secondary-magnetic-flux-
command calculating unit 5, the torque command PTR, the secondary inductance L2R, and the number of pole pairs Pm are input to the minimum-current-secondary-magnetic-flux-command calculating unit 21. The minimum-current-secondary-magnetic-flux-command calculating unit 21 performs arithmetic processing indicated by Formula (6) on the basis of the torque command PTR, the secondary inductance L2R, and the number of pole pairs Pm and outputs a result of the arithmetic processing as a minimum current secondary magnetic flux command F2R1. - When the minimum-current-secondary-magnetic-flux-
command calculating unit 21 generates the minimum current secondary magnetic flux command F2R1 in this way, current adjustment of the torque current command I1QR = the excitation current command I1DR is performed. It is possible to minimize an electric current necessary for generating the same torque. A copper loss in themotor 12 depends on the magnitude of an electric current. The loss increases as the electric current increases. Therefore, when the electric current is reduced, the copper loss also decreases. A loss in the switchingelement 16 of theinverter 11 also depends on the magnitude of the electric current. Therefore, it is possible to reduce the loss in theinverter 11 according to control for minimizing the electric current. - Subsequently, a significance of providing the magnetic-flux-command-
compensation calculating unit 22 is explained.FIG. 5 is a diagram of a loss curve with respect to the secondary magnetic flux command F2R. InFIG. 5 , a solid line represents a copper loss and a broken line represents an iron loss. The copper loss and the iron loss are predominant in a loss that occurs in a motor.
Therefore, if a sum of the copper loss and the iron loss can be minimized, it is possible to substantially minimize the motor loss. - However, when the minimum-current-secondary-magnetic-flux-
command calculating unit 21 generates the minimum current secondary magnetic flux command F2R1, the current adjustment of I1QR=I1DR is performed as explained above to perform adjustment for minimizing the copper loss. However, the minimum current secondary magnetic flux command value F2R1 for minimizing the copper loss is not a magnetic flux condition for a minimum loss when the iron loss is taken into account as well. When the inverter frequency FINV is small and the voltage V applied to the motor is small, the iron loss is small, so that it can be ignored. - On the other hand, when the inverter frequency FINV increases and the voltage V applied to the motor increases, the iron loss is innegligibly large. In that case, as it is seen from
FIG. 5 , when the secondary magnetic flux command value F2R is slightly reduced from the minimum current secondary magnetic flux command value F2R1, a decrease of the iron loss is larger than an increase in the copper loss, and the total loss of the copper loss and the iron loss also decreases. That is, the secondary magnetic flux command value F2R for minimizing the total loss of the copper loss and the iron loss becomes smaller than F2R1. Therefore, in the secondary-magnetic-flux-command calculating unit 5 in the first embodiment, as shown in the figure, the magnetic-flux-command-compensation calculating unit 22 for calculating, on the basis of the frequency of the motor, compensation for a decrease from the minimum current secondary magnetic flux command F2R1 of the secondary magnetic flux command value for minimizing the total loss of the copper loss and the iron loss of the motor is provided. An output of the magnetic-flux-command-compensation calculating unit 22 and the d-axis current I1DF are multiplied together in the multiplier 23. A multiplied value obtained by the multiplication is subtracted from an output of the minimum-current-secondary-magnetic-flux-command calculating unit 21. - In the magnetic-flux-command-
compensation calculating unit 22, an iron loss consideration table for calculating an iron loss according to a motor characteristic is prepared in advance at its designing stage. The magnetic-flux-command-compensation calculating unit 22 generates, on the basis of the input inverter frequency FINV, an optimum compensation coefficient for reducing a loss due to the iron loss with respect to the d-axis current I1DF and outputs the optimum compensation coefficient to the multiplier 23. - When an output of the multiplier 23 is not zero, a command obtained by subtracting compensation from the minimum current secondary magnetic flux command F2R1 taking into account the iron loss, which is the output of the multiplier 23, is generated as the final secondary magnetic flux command F2R.
- Note that, when the output of the multiplier 23 is zero or small compared with the minimum current secondary magnetic flux command F2R1 (e.g., when the d-axis current I1DF is small or when the compensation coefficient output by the magnetic-flux-command-
compensation calculating unit 22 is zero or small), the minimum current secondary magnetic flux command F2R1 generated by the minimum-current-secondary-magnetic-flux-command calculating unit 21 is output to the PWM-signal generating unit 9 directly as the secondary magnetic flux command F2R. - As explained above, with the motor control apparatus in the first embodiment, the voltage-frequency ratio is fixed, the secondary magnetic flux command for minimizing the current root-mean-square value due to the torque current command and the excitation current command in the driving region for driving at fixed torque is calculated, and the vector control is performed such that a detection value of the torque current and the excitation current respectively coincide with the torque current command and of the excitation current command. Therefore, it is possible to reduce the motor loss including the copper loss and the iron loss. It is possible to realize the motor control apparatus that enables more highly efficient operation control.
- In a second embodiment, the switching
element 16 included n theinverter 11 is explained. In general, the switchingelement 16 used in theinverter 11 is a semiconductor switching element made of silicon (Si) (IGBT, MOSFET, etc.; hereinafter abbreviated as "Si-SW"). The technology explained in the first embodiment can be configured using the general Si-SW. - On the other hand, the technology in the first embodiment is not limited to the Si-SW. It is naturally possible to use a semiconductor switching element made of silicon carbide (SiC), which attracts attention in recent years (hereinafter abbreviated as "SiC-SW"), instead of silicon (Si) as the switching
element 16. - A loss in the
inverter 11 is mainly a switching loss and a conduction loss of the switchingelement 16. In particular, when the SiC-SW is formed in a MOSFET structure, it is expected that the switching loss can be greatly reduced. When the SiC-WS is formed in the MOSFET structure, a conduction loss of the MOSFET increase in proportion to a square of an electric current. Therefore, it is possible to reduce the conduction loss by reducing a current value flowing to the SiC-SW. - In the motor control apparatus in the first embodiment, it is possible to minimize an electric current for generating the same torque. Therefore, by using the SiC-SW as the switching
element 16 included in theinverter 11 in the first embodiment, it is possible to greatly reduce the conduction loss. Consequently, it is possible to reduce the loss in theinverter 11. It is possible to realize the motor control apparatus that enables more highly efficient motor control. - Conventionally, when a large-capacity electric motor of an electric vehicle or the like is driven, an output frequency of the
inverter 11 is controlled by sequentially switching a plurality of control modes including a multi-pulse mode and a one-pulse mode. However, in the switchingelement 16 formed by a wide band gap semiconductor such as SiC, it is possible to perform asynchronous PWM control in all control regions. Therefore, a loss reduction effect by the motor control apparatus in this embodiment extends over all the regions. It is possible to perform highly efficient motor control in all the regions. In particular, when a current value is set high to perform the asynchronous PWM control in all the regions, the loss reduction effect for the motor is extremely large. - Note that, taking notice of a characteristic that a band gap of the SiC is larger than a band gap of Si, the SiC is an example of a semiconductor called wide band gap semiconductor (on the other hand, Si is called narrow band gap semiconductor). Apart from the SiC, for example, a semiconductor formed using a gallium nitride material or diamond also belongs to the wide band gap semiconductor. Characteristics of the gallium nitride material and the diamond have many similarities to the characteristics of the silicon carbide.
- The switching elements formed by such wide band gap semiconductors have high voltage resistance and high allowable current density. Therefore, it is possible to reduce the size of the switching elements. By using the switching elements reduced in the size, it is possible to reduce the size of the semiconductor module incorporating these elements.
- The switching elements formed by the wide band gap semiconductors also have high heat resistance. Therefore, in the case of the switching element that requires a cooling mechanism such as a heat sink, it is possible to reduce the size of the cooling mechanism. It is possible to further reduce the size of the switching element module.
- As explained above, the present invention is a motor control apparatus that reduces a motor loss in a low-speed region or a low-load driving region.
-
- 1 Motor control apparatus
- 4 Torque-command calculating unit
- 5 Secondary-magnetic-flux-command calculating unit
- 6 Motor-constant calculating unit
- 7 Coordinate converting unit
- 8 Speed control unit
- 9 PWM-signal generating unit
- 11 Inverter
- 12 Motor
- 13 Pulse generator (PG)
- 14 Current detectors
- 16 Switching element
- 21 Minimum-current-secondary-magnetic-flux-command calculating unit
- 22 Magnetic-flux-command-compensation calculating unit
- 23 Multiplier
- 24 Subtracter
Claims (7)
- A motor control apparatus (1) that divides an electric current flowing into and out of a motor (12) driven by an inverter (11) into a torque current and an excitation current and individually controls the torque current and the excitation current, characterised in that the motor control apparatus comprises:a secondary-magnetic-flux-command calculating unit (5) including a first calculating unit (21) that calculates a minimum current secondary magnetic flux command (F2R1) for minimizing a current root-mean-square value (I1) by using a torque current command (I1QR) and an excitation current command (I1DR), and outputs a secondary magnetic flux command (F2R) based on the minimum current secondary magnetic flux command (F2R1), wherein the current root-mean-square value (I1) is I1=√(I1DR2+I1QR2)/√3 and is minimized when the torque current command (I1QR) equals the excitation current command (I1DR); anda PWM-signal generation unit (9) that generates, on the basis of the secondary magnetic flux command (F2R), the torque current command (I1QR) and the excitation current command (I1DR), performs vector control such that a detection value of the torque current (I1QF) and a detection value of the excitation current (I1DF) respectively coincide with the torque current command (I1QR) and the excitation current command (I1DR), and generates a control signal for turning on and off a switching element (16) included in the inverter (11).
- The motor control apparatus (1) according to claim 1, wherein the first calculating unit (21) further calculates the minimum current secondary magnetic flux command (F2R1) on the basis of secondary inductance (L2R) and a number of pole pairs (Pm) of the motor (12) and a torque command (PTR).
- The motor control apparatus (1) according to claim 2, wherein the secondary-magnetic-flux-command calculating unit (5) further comprises a second calculating unit (22) providing, on the basis of a frequency of the inverter(FINV), compensation for a decrease from the minimum current secondary magnetic flux command (F2R1) of the secondary magnetic flux command (F2R) for minimizing the total loss of a copper loss and an iron loss of the motor (12).
- The motor control apparatus (1) according to claim 1 or 2, wherein the secondary-magnetic-flux-command calculating unit (5) further calculates, on the basis of an inverter frequency (FINV) and the minimum current secondary magnetic flux command (F2R1), a final secondary magnetic flux command (F2R) with which a total loss of a copper loss and an iron loss of the motor (12) decreases to be smaller than the total loss that occurs when the minimum current secondary magnetic flux command (F2R1) is used.
- The motor control apparatus (1) according to claim 4, wherein the secondary-magnetic-flux-command calculating unit (5) further generates compensation for reducing a loss due to the iron loss on the basis of the inverter frequency (FINV) and subtracts the compensation from the minimum current secondary magnetic flux command (F2R1) to calculate the final secondary magnetic flux command (F2R) .
- The motor control apparatus (1) according to any of claims 1 to 5, wherein the switching element (16) is formed of a wide band gap semiconductor.
- The motor control apparatus (1) according to claim 6, wherein the wide band gap semiconductor is a semiconductor in which silicon carbide, a gallium nitride material, or diamond is used.
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JP2012017212 | 2012-01-30 | ||
PCT/JP2013/052026 WO2013115240A1 (en) | 2012-01-30 | 2013-01-30 | Motor control device |
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EP2811644A1 EP2811644A1 (en) | 2014-12-10 |
EP2811644A4 EP2811644A4 (en) | 2016-01-20 |
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EP (1) | EP2811644B1 (en) |
JP (1) | JP5586798B2 (en) |
CN (1) | CN104081653B (en) |
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WO (1) | WO2013115240A1 (en) |
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CN104852655B (en) * | 2015-04-08 | 2017-08-01 | 华中科技大学 | A kind of Control of Induction Motors method based on vector controlled |
KR20180109351A (en) * | 2017-03-28 | 2018-10-08 | 엘에스산전 주식회사 | Proportional and resonant current controller |
EP3462600A1 (en) * | 2017-09-29 | 2019-04-03 | Siemens Aktiengesellschaft | Energy efficient asynchronous machine |
DE102018202854B4 (en) * | 2018-02-26 | 2020-01-02 | Audi Ag | Method for operating an on-board network of a hybrid motor vehicle and hybrid motor vehicle |
JP6939693B2 (en) * | 2018-04-27 | 2021-09-22 | 株式会社豊田自動織機 | Pulse pattern generator |
US11008014B2 (en) * | 2018-08-14 | 2021-05-18 | Ford Global Technologies, Llc | Methods and apparatus to determine vehicle weight information based on ride height |
JP2021005922A (en) * | 2019-06-25 | 2021-01-14 | 株式会社日立産機システム | Power conversion device |
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JPH0669307B2 (en) | 1987-09-21 | 1994-08-31 | 富士電機株式会社 | Induction motor control method |
JP2778753B2 (en) | 1989-09-01 | 1998-07-23 | 株式会社東芝 | Vector controller for induction motor |
JPH07118960B2 (en) * | 1990-01-24 | 1995-12-18 | 三菱電機株式会社 | Induction motor controller |
JPH0687596A (en) | 1991-04-15 | 1994-03-29 | Ain Tec:Kk | Winding device for double sheave drum |
CN1161933A (en) * | 1996-01-27 | 1997-10-15 | Lg产电株式会社 | Device for regulating stop-level of elevator |
JP3266790B2 (en) * | 1996-03-26 | 2002-03-18 | 三菱電機株式会社 | Induction motor control device |
JP3067660B2 (en) | 1996-11-11 | 2000-07-17 | 株式会社日立製作所 | Control method of induction motor |
US5949210A (en) * | 1998-03-16 | 1999-09-07 | Lockheed Martin Corp. | Two-dimensional variable limit proportional internal regulator for the current controller in synchronous frame |
JP4455248B2 (en) * | 2004-09-24 | 2010-04-21 | 三菱電機株式会社 | Vector control device for induction motor |
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WO2011111175A1 (en) | 2010-03-09 | 2011-09-15 | 三菱電機株式会社 | Power semiconductor module, power conversion device, and railway vehicles |
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US20150028793A1 (en) | 2015-01-29 |
JP5586798B2 (en) | 2014-09-10 |
EP2811644A1 (en) | 2014-12-10 |
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US9667187B2 (en) | 2017-05-30 |
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