EP2315505A1

EP2315505A1 - Method of controlling a half-bridge circuit, and a half-bridge circuit controlled thereby

Info

Publication number: EP2315505A1
Application number: EP09173168A
Authority: EP
Inventors: Arjan Van Den Berg; Wilhelmus Hinderlikus Maria Langeslag
Original assignee: NXP BV
Current assignee: NXP BV
Priority date: 2009-10-15
Filing date: 2009-10-15
Publication date: 2011-04-27

Abstract

A method of controlling a half-bridge circuit, in particular for use with lamps such as compact fluorescent lamps, is disclosed. The method is particularly useful for preventing excess power dissipation in the half-bridge transistors under mains over-voltage supply conditions. Whereas conventional methods rely on adjusting the switching frequency in response to the - measured - input voltage, the method disclosed herein relies on measurement of the mean square voltage of the half-bridge node. By controlling the switching frequency to maintain a constant mean square half-bridge voltage, the power dissipated by the half-bridge transistors is held constant independent of variations in mains input voltage.

In one embodiment, the control is effected by charging a capacitor with a current corresponding to the instantaneous square of the half-bridge node voltage, and then discharging it with a current corresponding to the instantaneous square of the reference voltage. The voltage across residual charge is used to control a voltage-controlled oscillator, to modify the half-bridge switching frequency and maintain the mean square value.

Description

Field of the Invention

This invention relates to methods of controlling half-bridge circuits, and to half-bridge circuits controlled thereby.

Background of the Invention

Switched power supplies comprising a half-bridge circuit are used in many applications, and one particularly interesting field is that of lamps such as compact fluorescent lamps (CFLs). Relative to mains voltages, compact fluorescent lamps typically operate at different voltages, and a high-frequency switched mode circuit such as a half-bridge circuit, operating at typically several tens of kilohertz, provides a convenient means for matching CFLs to a mains supply.
In order to maximise the efficiency of the overall system, it is important to minimise losses such as the power dissipated in the half-bridge's power transistors. Moreover, CFLs typically operate at elevated temperatures, typically up to 150°C: variation in the power dissipated in the half-bridge power transistors can result in variation or fluctuation in the operating temperature as a result of which it is necessary to specify the power transistors for even higher temperatures. Furthermore, the power dissipated in the transistors varies with the square of the operating voltage, rather than linearly. Thus a 15% increase in the mains voltage could result in a 32% dissipation increase, such that the power transistor would have to be 32% bigger to withstand the variation. This leads to increased costs which is undesirable commercially, and in the case where power transistors are integrated into a controller IC the problem is particularly acute, since integrated power transistors can typically dominate the cost of such a controller IC,
In order to reduce or eliminate the variation in power dissipation with mains voltage variation, it is known to provide feed-forward control: here, the mains voltage is measured, and the oscillating frequency of the half-bridge is adjusted. In this way, a first order compensation can be established, by scaling the operating frequency linearly with mains voltage above a certain threshold voltage value. However, a mains voltage dependency still remains. Moreover, such compensation is different for different types of CFL applications. Furthermore, feed-forward control requires additional components to sense the mains voltage.
Thus there is an ongoing need to provide a cost-effective method of reducing or eliminating the variation in dissipated power with mains voltage.

Summary of the invention

It is an object of the present invention to provide a means of controlling a half-bridge circuit which effectively reduces the effect of mains variation on dissipation in the half-bridge power transistors.
According to the present invention there is provided a method of controlling a half-bridge circuit for a lamp such as a compact fluorescent lamp, which circuit switches with a high-frequency period, the method comprising, not necessarily in sequence: determining a half-bridge current; determining a mean square value from the determined half-bridge current over a first part the high-frequency period; determining a reference signal, and adjusting the high-frequency period in dependence on the difference between the mean square value and the reference signal.
In embodiments, the step of adjusting the high-frequency period in dependence on the difference between the mean square value and the reference signal comprises adjusting the high-frequency period in proportion to the difference between the mean square value and the reference signal.
In embodiments the half-bridge current is determined by measuring a voltage across a sense resistor connected in series with the half-bridge. Such a measurement is particularly convenient and avoids the need for additional components.
In embodiments the step of determining a mean square value from the determined half-bridge current over a first part the high-frequency period comprises deriving a current from the square of the determined half-bridge current, and a first one of charging and discharging a capacitor with the derived current over the first part the high-frequency period.
In embodiments the step of determining a reference signal comprises sensing a reference voltage, and determining a further mean square value from the sensed reference voltage over a second part of the high-frequency period.
In embodiments the step of determining a further mean square value from the sensed reference voltage over a second part of the high-frequency period comprises deriving a further current from the square of the sensed reference voltage, and the other one of charging and discharging the capacitor with the further current over the second part of the high-frequency period.
In embodiments the step of adjusting the high-frequency period in dependence on the difference between the mean square value and the reference signal comprises determining a voltage across the capacitor, using the voltage so determined as a control input to a voltage controlled oscillator, and using the output of the voltage controlled oscillator to set the high-frequency period.
In embodiments the voltage across the capacitor is determined at the end of the second part of the high-frequency period.
In embodiments the high frequency period consists of consecutive first, second, third and fourth intervals, the controller comprises a high-side switch and a low-side switch which are respectively in an on-state only during the first and third intervals of the high-frequency period, the first part of the high-frequency period consists of the second and third intervals, and the second part of the high-frequency period consists of the fourth and first intervals.
In embodiments, the total duration of the first and second intervals is equal to the total duration of the third and fourth intervals. Thus, the controller operates at 50% duty cycle.
According to another aspect of the invention there is provided a half-bridge circuit configured to operate according to any of the above methods and there may be provided a power supply for a fluorescent lamp comprising such a half-bridge circuit, of a fluorescent lamp comprising such a power supply.
According to yet another aspect of the invention there is provided an integrated circuit configured to drive a half-bridge circuit operating according to any of the above methods.
These and other aspects of the invention will be apparent from, and elucidated with reference to, the embodiments described hereinafter.

Brief description of Drawings

Embodiments of the invention will be described, by way of example only, with reference to the drawings, in which

Fig. 1 is a schematic of a CFL lamp assembly arrangement including half-bridge controller in an arrangement according to an embodiment of the invention;
Fig. 2 shows a graph of known feedforward control, and
Fig. 3 shows a schematic of a controller arrangement according to an embodiment of the invention.

It should be noted that the Figures are diagrammatic and not drawn to scale. Relative dimensions and proportions of parts of these Figures have been shown exaggerated or reduced in size, for the sake of clarity and convenience in the drawings. The same reference signs are generally used to refer to corresponding or similar feature in modified and different embodiments

Detailed description of embodiments

In figure 1 there is shown a schematic of a CFL lamp assembly arrangement. A lamp 10, which in this case is a fluorescent lamp and may be a CFL, has a capacitor Clamp connected across its electrodes. One end electrode is connected between a rectified mains supply and ground by means of respective capacitors C_HB1 and C_HB2. The other electrode is connected, via a series ballast which in this case is an inductor L_lamp, to the half-bridge node HB of a half-bridge circuit.
The half-bridge circuit comprises two power transistors: a first, high side, power transistor HS is connected between the half-bridge node and the rectified mains voltage; the other, low side, power transistor LS is connected between the half-bridge node and ground PGND, via a sense a resistor Rsense 50. The node between the low side power transistor and the sense resister is designated 'sense'. The high side power transistor HS and low side power transistor LS are driven by respective high side driver 12 and low side driver 14. A clock signal CLK to control the timing of drivers 12 and 14 is generated by a voltage controlled oscillator (VCO) 16. The clock signal is directed to the low side driver 14 via a non-overlap circuit 18 which outputs a signal Lson to the low side driver 14 and a signal HSon to a pulse generator 20, and thence via a latch 22 to the high side driver 12.
Also connected to the sense node 'sense', is a RMS control unit 52. The control unit has as one input the voltage at the sense node 'sense' , and as another input a reference voltage Vref 54.
As will be explained in more detail herebelow RMS current control unit 52 outputs a current which is switchably connected to a capacitor Csw. The capacitor is also connected to the VCO 16.
As will be further described below, during the second and third intervals (which will be described below) the output current of the RMS control block is determined by Vsense. During the fourth and first intervals, the output current (which is then in an opposite direction) is determined by Vref. Thus the switching element is integral within the RMS control block. It should be noted that, advantageously, any offset in the RMS block is thereby cancelled as both the Vsense signal as the Vref goes through this block and offset is automatically substracted.
Normal operation of the half-bridge circuit comprising a high side switch HS and low side switch LS will be well-known to the skilled person: switches HS and LS are alternately closed by means of drivers 14 and 12 respectively. Non-overlap circuit 18, in conjunction with pulse generator 20 and latch 22, ensures that the switches are never both closed at the same time. Thus the high-frequency period comprises a first interval, during which only the HS switch is on; a second interval, being a non-overlap interval during which neither switch is on; a third interval, during which only the LS switch is on; and a fourth interval, being another non-overlap interval during which neither switch is on. In normal operation the switching frequency is held constant under the control of the VCO 16.
It is well-known that the power dissipated by the power transistors in a half-bridge circuit is proportional to the square of the mains voltage: since the current through the power switches is linear with the input voltage, the so-called "I²R", or (I*I*Ron) losses in the power switches scale quadraticaly with the input voltage.
It is also known that the power dissipation can be reduced by decreasing the operating frequency of the half-bridge circuit. This is because the operating current is set by the input voltage and the impedance of the inductor. The latter is 2*Π*f*L. Thus a higher frequency f will result in a larger impedance of the inductor and in a lower inductor and power switch current. However due to the presence of a frequency clamp and the lamp resistance itself, the overall frequency dependency is more complex and not linear with the frequency.
In known arrangements, the mains voltage is sensed, and supplied as an input to the VCO. If the mains voltage exceeds a threshold value, the VCO is used to increase the frequency of the half-bridge circuit, in proportion to the sense the voltage. In other circuits, instead of sensing the mains voltage directly, a current derived from the mains voltage is sensed and used to control a current controlled oscillator (CCO) rather than a VCO. Figure 2 shows a graph of such a control mechanism. At relatively lower values of the mains voltage Vm the half-bridge switching frequency is fixed at a value fb; beyond the threshold voltage, the frequency increases linearly with the voltage.
In contrast, according to embodiments of the invention, frequency control is provided by means of the RMS control unit 52, as will now be described with reference to Figure 3, which shows part of the assembly of Figure 1, and in particular the controller:
During a first part of the period of the switching cycle of the half-bridge power transistors, (which will hereinafter also be referred to as the high frequency, or HF, period), the output current for the lamp flows through the closed LS transistor. The current is sensed by measuring the voltage across the sense resister 50, which is in series with the LS transistor. In the RMS control unit 52, this voltage is squared and converted to a current: various arrangement to carry this out will be familiar to the skilled person: for example a translinear loop can be used to provide a squaring function. However since this function is in the current domain, first an internal voltage-to-current converter is used to convert the Vsense voltage to a proportional current. The current (for example the output current of the translinear loop) is used to charge the capacitor Csw which also acts as the VCO input capacitor, and thus at the end of the first part of the HF period the capacitor Csw stores a charge which is representative of the integral of the square of the instantaneous voltage across the sense resistor and thus the half-bridge current.
At this point it should be noted that "RMS" control can be considered to be equivalent to "MS", or mean square control, since taking the "root" is a time independent function and can be omitted.
Generally, the capacitor value is very large compared to the charge and discharge current i.e. a difference in charge / discharge current over one HF cycle will result in only a very minor voltage increase / decrease. So the averaging function is spread out over multiple HF cycles. Thus, advantageously, no sample-and-hold circuit is needed. Of course, the invention is not limited to this implementation, and in other embodiments, a sample-and-hold circuit can be used in place of the large capacitor.
Further, in the embodiment described above the capacitor is charged during the first part of the high frequency period, and discharged during the second part. It will be immediately apparent that the converse arrangement, during which the capacitor is discharged during the first part and charged during the second part of the high frequency period, is also within the scope the invention.
As will be shown below, the lamp power level can be shown to be dependent on the RMS value of the current, and thus the value of the charge stored in the integrating capacitor Csw. According to embodiments of the invention the frequency of the HF period is adjusted to maintain this at a constant level. To do this, a reference charge is subtracted from the integrating capacitor, and the difference used as an output to adjust the frequency by means of the voltage controlled oscillator.
The first part of the high frequency period concludes at the end of the non-overlap period, or in other words, at the moment when the HS switch is closed. Thus the first part of the high frequency period consists of the second and third intervals discussed above.
To provide the reference charge to the integrating capacitor Csw, during the second part of the HF period, whilst the HS transistor is closed (and including the preceding non-overlap interval), a reference voltage of Vref 54 is input to the RMS control unit 52. The reference voltage is squared and converted into a current by the RMS control unit 52, and the result used to discharge the integrating capacitor Csw. At the end of the second period, the residual charge in the integrating capacitor Csw thus corresponds to a perturbation of the voltage across sense resistor sense, assuming, for the present, a 50% duty cycle of the LS transistor. This perturbation is fed as the differential signal to the VCO 16, and used to adjust the frequency of the HS period.
The control method just described depends on the fact that the RMS half-bridge current (or equivalently, the dissipated power, since control of the
RMS half-bridge current to a fixed value results in control of the dissipation of the power switches (lrms*lrms*Rdson) to a fixed value) can be determined from Vref and Rsense. To show this, it should be noted that, in steady state conditions such that the residual charge in the integrating capacitor is zero: $\int_{0}^{d * T} Ihb {(t)}^{2} * {Rsense}^{2} dt = \int_{d * T}^{T} Vref^{2} dt$
Then, provided that the lamp burner current is much greater than that of the current through the lamp capacitor , and since Ihb ≅ Ila , $\int_{0}^{d * T} Iha (t) {()}^{2} * {Rsense}^{2} dt = \int_{d * T}^{T} Vref^{2} dt$
Rearranging, first by dividing by d*T $\frac{1}{d * T} \int_{0}^{d * T} Ila (t) {()}^{2} dt = \frac{1}{d * T} \int_{d * T}^{T} Vref \frac{^{2}}{{Rsense}^{2}} dt$

and then by taking the square root: $\sqrt{\frac{1}{d * T} \int_{0}^{d * T} Ila (t) {()}^{2} dt} = \sqrt{\frac{1}{d * T} \int_{d * T}^{T} Vref \frac{^{2}}{{Rsense}^{2}} dt}$

yields an equation for the root mean square of the lamp current, which for a 50% duty cycle (d=0.5), results in: ${Ila}_{rms} = \frac{Vref}{Rsense}$
The skilled person will appreciate that, absent a 50% duty, equation 5 no longer directly follows from equation 4; the simplicity of equation 5 results in 50% duty cycle being a preferred embodiment; however, the invention is not limited thereto, since, with other duty cycles, a similar relationship (albeit with more complex scaling), holds between Ila_rms and V_ref.
Finally, it will be apparent to the skilled person, that the power in the lamp is dependant on the burner current: since the lamp can be considered as a resistive load, higher current means quadratically more lamp power, and the power dissipated in the half-bridge transistors is similarly dependant on the RMS of the half-bridge current. Thus by maintaining the burner current (and the half-bridge current) constant as described above, both the power in the lamp and that dissipated in the half-bridge power transistors are maintained constant independent of changes in the mains voltage
Between these two effects, the more important is that the half-bridge RMS current is constant as this means a constant IC temperature independent of the mains voltage. However, an advantageous secondary effect is that the RMS lamp current is also nearly constant, as a result of which the observer experiences no variation in illumination when the mains voltage changes.
Of course, it will be appreciated that the RMS burner current will still vary a little as due to the control, the frequency will change. As result the current through the clamp will change and given the fact that the total rms current is fixed (controlled), the lamp current needs to change a little.
The embodiment above has been described in relation to a half-bridge control circuit for a CFL. However, the invention is not so limited, and is equally applicable to other types of lamps such as cold cathode fluorescent lamps or conventional fluorescent lamps. Moreover, the invention extends to other low impedance applications which may be driven from a half-bridge circuit, such as inductive motor drives, half bridge power supplies / adapters.
For the sake of completeness, it is mentioned that the term "high frequency", as used herein, takes its normal meaning, as will be immediately apparent to the skilled person. That is to say, "high frequency" means a frequency which is high in relation to any normal mains frequency including 50, 60 and 110 Hz. In particular, "high frequency" encompasses frequencies in the kilohertz (that is, < 1,000Hz), and the megahertsz (that is, < 1,000,000 Hz) ranges.
Thus in summary, a method of controlling a half-bridge circuit, in particular for use with lamps such as compact fluorescent lamps, has been disclosed. The method is particularly useful for preventing excess power dissipation in the half-bridge transistors under mains over-voltage supply conditions. Whereas conventional methods rely on adjusting the switching frequency in response to the - measured - input voltage, the method disclosed herein relies on measurement of the mean square voltage of the half-bridge node. By controlling the switching frequency to maintain a constant mean square half-bridge voltage, the power dissipated by the half-bridge transistors is held constant independent of variations in mains input voltage.
In one embodiment, the control is effected by charging a capacitor with a current corresponding to the instantaneous square of the half-bridge node voltage, and then discharging it with a current corresponding to the instantaneous square of the reference voltage. The voltage across residual charge is used to control a voltage-controlled oscillator, to modify the half-bridge switching frequency and maintain the mean square value.
From reading the present disclosure, other variations and modifications will be apparent to the skilled person. Such variations and modifications may involve equivalent and other features which are already known in the art of half-bridge controllers, and which may be used instead of, or in addition to, features already described herein.
Although the appended claims are directed to particular combinations of features, it should be understood that the scope of the disclosure of the present invention also includes any novel feature or any novel combination of features disclosed herein either explicitly or implicitly or any generalisation thereof, whether or not it relates to the same invention as presently claimed in any claim and whether or not it mitigates any or all of the same technical problems as does the present invention.
Features which are described in the context of separate embodiments may also be provided in combination in a single embodiment. Conversely, various features which are, for brevity, described in the context of a single embodiment, may also be provided separately or in any suitable sub-combination.
The applicant hereby gives notice that new claims may be formulated to such features and/or combinations of such features during the prosecution of the present application or of any further application derived therefrom.
For the sake of completeness it is also stated that the term "comprising" does not exclude other elements or steps, the term "a" or "an" does not exclude a plurality, a single processor or other unit may fulfil the functions of several means recited in the claims and reference signs in the claims shall not be construed as limiting the scope of the claims.

Claims

A method of controlling a half-bridge circuit for a lamp such as a compact fluorescent lamp, which circuit switches with a high-frequency period,
the method comprising the steps, not necessarily in sequence, of:
- determining a half-bridge current;

- determining a mean square value from the determined half-bridge current over a first part the high-frequency period;

- determining a reference signal, and

- adjusting the high-frequency period in dependence on the difference between the mean square value and the reference signal.
A method as claimed in claim 1, wherein the step of adjusting the high-frequency period in dependence on the difference between the mean square value and the reference signal comprises:
adjusting the high-frequency period in proportion to the difference between the mean square value and the reference signal.
A method as claimed in claim 1, wherein the half-bridge current is determined by measuring a voltage across a sense resistor connected in series with the half-bridge.
A method as claimed in claim 1 or 2, wherein the step of determining a mean square value from the determined half-bridge current over a first part the high-frequency period comprises:
- deriving a current from the square of the determined half-bridge current; and

- a first one of charging and discharging a capacitor with the derived current over the first part the high-frequency period
A method as claimed in claim 3, wherein the step of determining a reference signal comprises:
- sensing a reference voltage, and

- determining a further mean square value from the sensed reference voltage over a second part of the high-frequency period.
A method as claimed in claim 4, wherein the step of determining a further mean square value from the sensed reference voltage over a second part of the high-frequency period comprises:
- deriving a further current from the square of the sensed reference voltage, and

- the other one of charging and discharging the capacitor with the further current over the second part of the high-frequency period.
A method as claimed in claim 5, wherein the step of adjusting the high-frequency period in dependence on the difference between the mean square value and the reference signal comprises:
- determining a voltage across the capacitor;

- using the voltage so determined as a control input to a voltage controlled oscillator, and

- using the output of the voltage controlled oscillator to set the high-frequency period.
A method as claimed in claim 6, wherein the voltage across the capacitor is determined at the end of the second part of the high-frequency period.
A method as claimed in any proceeding claim, wherein the high frequency period consists of consecutive first, second, third and fourth intervals, the controller comprises a high-side switch and a low-side switch which are respectively in an on-state only during the first and third intervals of the high-frequency period, the first part of the high-frequency period consists of the second and third intervals, and the second part of the high-frequency period consists of the fourth and first intervals.
A method as claimed in any proceeding claim, wherein the total duration of the first and second intervals is equal to the total duration of the third and fourth intervals.
A half-bridge circuit configured to operate according to a method as claimed in any preceding claim.
An integrated circuit configured to drive a half-bridge circuit operating according to a method as claimed in any of claims 1 to 10.
A power supply for a fluorescent lamp, comprising a half-bridge circuit as claimed in claim 11.
A fluorescent lamp comprising a power supply as claimed in claim 13.