EP1842262B1 - Aperture-coupled antenna - Google Patents

Abstract

An aperture-coupled antenna has a first radiation electrode, a ground area and a wave guide which is implemented to supply energy to the antenna. The wave guide is arranged spaced apart from the ground area on a first side of the ground area, and the first radiation electrode is arranged spaced apart from the ground area on a second side of the ground area. The ground area has an aperture including a first slot in the ground area, a second slot in the ground area and a third slot in the ground area. The first slot and the second slot together form a slot in the shape of a cross. The third slot passes through an intersection of the first slot and the second slot. The wave guide and the radiation electrode are arranged such that energy can be coupled from the wave guide through the aperture to the patch.

Description

The present invention generally relates to an aperture-coupled antenna, more particularly to an aperture-coupled circularly polarized planar antenna.

Currently, more and more wireless systems are being developed that need to work in multiple frequency bands. For this purpose, compact antennas are often necessary to keep the volume of the antennas small and to allow use in portable devices.

It is possible to provide a separate antenna for each frequency band to be used. The disadvantage of using separate antennas, however, is that a multiplexer must be used. Furthermore, when using separate antennas, the area required for the antennas increases.

The reception of a plurality of different wireless transmission systems with a single broadband antenna is problematic, since broadband antennas are conventionally not produced in a small design at low cost. So if you wanted to receive all relevant systems with only a single broadband antenna, so this is not possible with a small low-cost antenna.

For receiving a plurality of frequency bands, a multi-element antenna may be used which has its own radiator for each frequency range. Most known antenna concepts that are suitable for receiving two or more frequency bands (dual band concepts or multiband concepts), such as As integrated inverted-F antennas (inverted-F antennas, IFA) and planar inverted-F antennas (planar inverted-F antennas, PIFA) have only a linear polarization. Such known antenna forms are described, for example, in the book " Planar Antennas for Wireless Communications "by Kin-Lu Wong (John Wiley & Sons, Inc., Hoboken, New Jersey, 2003 ).

Especially for mobile applications, it is desirable to use a circular polarization, since in this case the alignment of the transmitting and receiving antenna is not critical, while using linear polarization, the orientation of the antennas must be selected appropriately.

While a number of integrable antennas having circular polarization are known, many of the integrable geometries for generating circular polarization have significant disadvantages. For example, almost square patches (planar conductive surfaces) with coaxial feed have a low impedance bandwidth, as for example in the thesis " Investigation and construction of multiband antennas for the reception of circularly polarized signals "by U. Wiesman, published in 2002 at the Fraunhofer Institute for Integrated Circuits in Erlangen. The same applies to aperture-locked patch antennas with a cross-slit, which are described in the master's thesis "Investigation of circularly polarized patch antenna with aperture coupling" produced in 2004 at the Fraunhofer Institute for Integrated Circuits in Erlangen by A. Popugaev. Overall, it can be said that the polarization purity in known broadband circularly polarized patch antennas with only one feed point is low. On the other hand, spiral antennas have great losses.

An overview of aperture-coupled microstrip antennas can be found in the article " A review of aperture coupled microstrip antennas: history, operation, developement and applications ", by DM Pozar, published in May 1996 at the University of Massachusetts at Amherst and available on the Internet at www.ecs.umass.edu/ece/pozar/aperture.pdf . Further information on broadband patch antennas can also be found in the book "Broadband Patch Antennas" by J.-F. Zuercher, which was published in 1995 by Artech-House Verlag.

In summary, it can be stated that no technically advantageous design of an antenna is found in the prior art which, with good radiation efficiency and sufficient impedance bandwidth, enables radiation of circularly polarized waves with high suppression of orthogonal polarization. Furthermore, no technically simple and inexpensive realizable antenna design is known, which allows for good efficiency and sufficient bandwidth radiation of a circularly polarized electromagnetic wave in two different frequency bands.

It is thus the object of the present invention to provide an aperture-coupled patch antenna which enables the radiation of a circularly polarized electromagnetic wave and which has both a good suppression of orthogonal polarization and a large impedance bandwidth compared to conventional antennas.

This object is achieved by an antenna according to claim 1.

The present invention provides an aperture-coupled antenna having a first radiation electrode, a ground plane, and a waveguide configured to energize the antenna. The waveguide is spaced from the ground plane on a first side of the ground plane and the radiation electrode is, spaced from the ground plane, disposed on a second side of the ground plane. The ground plane has an aperture comprising a first slot in the ground plane, a second slot in the ground plane, and a third slot in the ground plane, wherein the first slot and the second slot together form a slot of the shape of a cross, and wherein the third slot extends through an intersection of the first slot and the second slot. The geometrical shape of the radiation electrode is designed to allow radiation of a circularly polarized electromagnetic wave. For this purpose, the radiation electrode preferably has a faulty geometry. For example, the radiation electrode can be almost square with slightly different dimensions or edge lengths. Likewise, the radiation electrode may be rectangular or almost square, wherein at least one corner is chamfered. Finally, the radiation electrode may also have slots designed to allow the radiation of a circularly polarized wave. However, any other geometry of the radiation electrode is possible as long as it allows a circular polarization. Furthermore, in an antenna according to the invention, the waveguide and the radiation electrode are arranged so that energy from the waveguide can be coupled via the aperture to the radiation electrode.

It is the gist of the present invention that it is possible to provide an aperture-coupled antenna having particularly advantageous properties by coupling energy from a waveguide through an aperture to a radiation electrode, the aperture having a combination of three slits. In this case, in conjunction with a suitably designed radiation electrode, a circularity of a radiated electromagnetic wave can be improved (ie, a suppression of an undesired orthogonal polarization in the emission of a circularly polarized wave can be improved) by two of the slits forming the aperture together form a slit of the shape of a cross. The radiation electrode is in this case designed so that it allows the emission of a circularly polarized wave. For example, the radiation electrode may have a rectangular or square shape, wherein at least one of the corners is chamfered. An almost square radiation electrode with slightly different dimensions or edge lengths can also be used. Furthermore, the radiation electrode may have one or more slots, which are preferably arranged in the middle of the radiation electrode. In addition to the aforementioned embodiments, however, any type of radiation electrode is suitable, which allows the emission of a circularly polarized wave. In addition, the impedance bandwidth of the antenna according to the invention may be increased by providing a third slot passing through an intersection in which the first and second slots form the center of a cross in which the first and second slots intersect ,

The introduction of a third slot has created a new degree of design freedom that allows the antenna to be designed so that the widest possible impedance bandwidth can be achieved. Impedance bandwidth here is to be understood as a bandwidth within which the adaptation of the antenna is so good that a predefined standing wave ratio (SWR) is not exceeded.

It is particularly surprising that the introduction of a third slot does not significantly degrade the polarization characteristics of the aperture-coupled antenna. Namely, according to the results known from the prior art, it would be expected that a circular polarization excited by the presence of two slots, which together form a cross shape, would be greatly affected by the addition of another slot becomes so that the orthogonal polarization increases strongly. However, unlike the knowledge of the prior art, it has been found that even with the use of three slots, a very high suppression of undesired polarization can be achieved. This is all the more surprising since, according to the conventional conception, two mutually orthogonal modes with suitable phase shift must be excited in order to achieve a circular polarization with a small proportion of orthogonal polarization. Thus, it is surprising to those skilled in the art that in the presence of three slots forming an aperture, but of course not all may be orthogonal to one another, a circular polarization can still be achieved with a small amount of orthogonal polarization.

The advantage of the present invention is thus that a planar antenna is provided which has a circular polarization, provides a good suppression of orthogonal polarization and at the same time has a large impedance bandwidth. Furthermore, the antenna according to the invention can be constructed in a completely planar manner, which results in a design that is smaller in comparison with conventional antennas and lower costs. The structure of the antenna can be made by conventional techniques, wherein only electrically conductive layers forming a radiation electrode and a ground plane have to be produced. These conductive structures may, for example, be arranged on dielectric carrier materials, wherein a structuring of metallizations by means of conventional etching technologies is offered. The supply of energy to the antenna can be done with any waveguide structure capable of coupling electromagnetic energy through the aperture to the radiation electrode. Thus, a very flexible power supply of the antenna according to the invention is possible. Another advantage of an antenna structure according to the invention is that dual-band and multi-band concepts are implemented where a circularly polarized electromagnetic wave can be generated in multiple frequency bands, and the overall size does not exceed the size of the antenna structure needed for the lowest operating frequency. This is made possible by the fact that the electromagnetic energy from the back of the antenna is coupled via an aperture. The size of the radiation electrode is determined by the operating frequency. Feed structures and other active and passive elements (eg, amplifiers, phase shifters or mixers) can be placed behind the aperture-coupled antenna and do not increase the area requirement of the overall arrangement. It should also be noted that the antenna structure according to the invention makes it possible to minimize losses by using dielectric materials only to a limited extent. It is sufficient to mechanically support the radiation electrode, the ground plane and optionally the waveguide with dielectric support materials. Furthermore, in the case of an antenna structure according to the invention, there are no very long and narrow conductor structures, as are usual, for example, in spiral antennas. This also reduces the losses of an antenna according to the invention.

For the sake of clarity, it should also be pointed out here that the radiation electrode is preferably a planar structure, as is customary in aperture-coupled antennas. Such a radiation electrode is typically referred to in the relevant literature as a "patch". The overall structure of the aperture-coupled antenna according to the invention thus represents a special case of a patch antenna.

It should also be noted that in the aperture-coupled antennas, the ground plane is preferably parallel or approximately parallel to the radiation electrode, wherein a deviation from a parallelism up to about 20 degrees may occur. It should also be noted that a Aperture-coupled antenna is preferably constructed as a planar antenna, wherein both the radiation electrode and the ground plane are flat. Similarly, the waveguide is preferably flat. However, a curvature of the radiation electrode and the ground plane is possible.

In a preferred embodiment of the present invention, the third slot is longer than the first slot and also longer than the second slot. This is particularly advantageous because the bandwidth of the antenna can be increased by a third slot that is longer than the first and second slots. This is understandable, since the third slot can then particularly effectively improve the bandwidth of the antenna if it influences the electromagnetic field distribution as much as possible without causing a deterioration in the separation of mutually orthogonal polarizations.

It is further preferred that the first slot and the second slot are orthogonal to each other and together form a slot of the shape of a right-angled cross with equal length arms. In this case, therefore, the lengths of the two slots are the same, and the slots are arranged so that they intersect orthogonally in the middle. An orthogonal arrangement of the first and second slot is particularly advantageous, since an optimal excitation of a circular polarization can be achieved thereby. An orthogonal arrangement of the slots thus results in that only one right-handed or one left-handed circularly polarized wave is excited by the first and the second slot. In order to produce an optimally pure polarization, however, the acute angle between the first and the second slot can be varied between 70 ° and 90 °. Thus, an optimization of the antenna structure in the presence of the third slot is possible.

Furthermore, it is preferable that the center of the third slot coincides with a center of the cross-shaped slot formed by the first and second slots. In other words, the first, second and third slots intersect in a common space area. There is thus only one area in the center of the aperture in which the three slots intersect. The three slots thus form the shape of a star. Moreover, it is achieved by the described arrangement in an advantageous manner that the third slot is arranged symmetrically, in the sense that the length of the third slot on both sides of the intersection with the first and the second slot is the same. This prevents asymmetries in the emission of the antenna according to the invention from arising.

Furthermore, a highly symmetrical arrangement is preferred in which a geometric center of the first slot, a geometric center of the second slot and a geometric center of the third slot coincide, and wherein the aperture is axisymmetric with respect to an axis of the third slot. The axis of the third slot is defined along a largest dimension of the third slot. For the rectangular third slot, the axis is defined as a centerline of the rectangle that is parallel to the two longer edges of the rectangle. Such a geometry allows a very high symmetry, which is reflected in the radiation behavior of the antenna, in particular in the purity of the polarization.

Further, it is preferred that the third slot is orthogonal to the feed line. This arrangement leads to a further increase in the symmetry, which in turn can improve the radiation properties and the polarization purity.

In another preferred embodiment, the first slot and the second slot are configured such that the first slot and the second slot are not resonated in an operating frequency range for which the aperture-coupled antenna is designed. This can be achieved for example by a suitable choice of the length of the first and the second slot. To avoid a resonance behavior of the first and second slots, they are preferably designed shorter than a predetermined length, wherein the predetermined length is of the order of a half free-space wavelength at an operating frequency. Such a measure is advantageous because the first slot and the second slot essentially serve to enable the excitation of the radiation electrode in such a way that a radiated wave has a circular polarization. Therefore, it is not desirable for the first and second slots to be operated close to resonance. Namely, a resonance that would occur in the first and second slots would involve steep changes in phase, which would greatly change the polarization over the frequency. Moreover, a resonance of the first and the second slot also causes a strong radiation to the rear, d. H. from the ground plane in the direction in which the feed line is located takes place. This should be avoided.

Furthermore, it is preferred that the third slot is designed such that an operating frequency for which the aperture-coupled antenna is designed deviates by at most 30% from a resonance frequency of the third slot. It is therefore required that the resonance frequency of the slot differs at most by 30% from a permissible operating frequency. Thus, at least one operating frequency for which the antenna is designed, the third slot is operated near the resonance. But just a resonance-like behavior of the third slot leads to the fact that the impedance bandwidth of the invention Antenna improved. Namely, when the third slot is resonated, a large amount of electromagnetic energy is stored in the space surrounding the third slot, thereby creating an energy reservoir by which reactive impedance components of the input impedance of the antenna of the present invention can be compensated. Therefore, operation of the third slot in the vicinity of its resonance provides improved impedance matching of the entire aperture-coupled antenna structure of the present invention.

In another preferred embodiment, the third slot is configured so that a resonant frequency of the third slot is within an operating frequency range for which the aperture-coupled antenna is designed. With such a design, a maximum improvement of the bandwidth of the antenna according to the invention can be achieved. At the resonant frequency, the area around the third slot stores maximum electromagnetic energy and thus can maximize the impedance impact.

Furthermore, it is preferred that the waveguide through which the antenna is fed is a microstrip line, a coplanar waveguide, a stripline, a dielectric waveguide or a cavity waveguide. A microstrip line is particularly advantageous, since this is easy to implement and can be combined well with active circuits. A coplanar waveguide offers the advantage that no connection (vias) is necessary for coupling to a reference potential. A stripline that is completely embedded in a dielectric offers a particularly advantageous dispersion behavior. The use of a dielectric waveguide is recommended, for example, at very high frequencies, as in a dielectric waveguide metallic losses are avoided. A cavity waveguide can also serve as a low-loss feed line.

Preferably, the aperture and the radiation electrode are designed so that the aperture-coupled antenna, apart from parasitic effects, radiates a circularly polarized electromagnetic wave. With regard to the design of the radiation electrode, it is preferred to use a rectangular patch. A particularly good circular radiation results when the patch is almost square, ie the lengths of the longer and shorter sides differ by a maximum of 20%. Moreover, it is advantageous to cut off corners of the rectangular or almost square patch, as this allows the polarization can be determined. It is excited a suitable mode that allows the emission of a circularly polarized electromagnetic wave. In this case, it is preferable to cut off two opposite corners. The purity of the polarization can be influenced by changing the geometrical details of the slit aperture while maintaining the principal shape of the aperture, which has three slits.

In a further preferred embodiment, the antenna according to the invention further comprises a second planar radiation electrode and a third planar radiation electrode. The second planar radiation electrode is arranged substantially parallel to the first radiation electrode, wherein the first radiation electrode is located between the second radiation electrode and the ground plane. A substantially parallel arrangement here means that a maximum tilt between the second planar radiation electrode and the first radiation electrode does not exceed 20 degrees. The geometric arrangement is such that in a sequence from bottom to top first the waveguide, then the ground plane, then the first radiation electrode and then the second radiation electrode are arranged. The first radiation electrode thus lies in the order of the layers between the second radiation electrode and the ground surface. However, the term "between" is not a restriction for planar electrodes, the spatial arrangement is to be understood such that a plane in which the first radiation electrode lies lies between a plane in which the second radiation electrode lies and a plane in which the ground surface lies. is arranged. If the electrodes are not completely flat, the corresponding definition should be applied mutatis mutandis, where instead of the planes sufficiently smooth surfaces occur, in which the respective electrodes are located.

Furthermore, in a preferred embodiment of the present invention, the third radiation electrode is arranged such that in a projection along an axis normal to the second radiation electrode, the third radiation electrode encloses the second radiation electrode. A corresponding definition is analogously applicable to cases in which the second and the third radiation electrode are not completely flat but have a slight curvature. It should be defined here that in a plan view, in which the viewing direction corresponds to a mean surface normal of the second radiation electrode, the third radiation electrode encloses the second radiation electrode. Such an arrangement, which thus has a first radiation electrode and a second and a third radiation electrode, is suitable for enabling a multi-band operation of the antenna according to the invention. At very high frequencies, the first radiation electrode acts as an essential radiating element. Although the third radiation electrode encloses the second radiation electrode, there is a slot or gap between the two, through which a radiation can take place starting from the first radiation electrode. For a better understanding, it should be pointed out once again that the second radiation electrode and the third radiation electrode together are typically larger than the first radiation electrode and lie in the direction of the main radiation in front of the first radiation electrode. Therefore, it is by a The arrangement according to the invention, in which a second radiation electrode and a third radiation electrode are separated, allows the first radiation electrode to still radiate effectively, despite the presence of a second or third radiation electrode.

In a further preferred embodiment, the second radiation electrode and the third radiation electrode lie in one plane, wherein in turn the third radiation electrode encloses the second radiation electrode. This arrangement enables a particularly advantageous joint production of the second and the third radiation electrode, which can be carried, for example, by a common substrate. Incidentally, therefore, the second and the third radiation electrode can strongly interact with each other, thereby effectively forming a radiation electrode having approximately the size of the third radiation electrode.

Preferably, the inventive antenna is designed so that an impedance matching is achieved with a VSWR of less than 2 in at least two frequency bands. Thus, a dual band operation or multi-band operation of the antenna according to the invention is possible, with a good adaptation is achieved. However, a good adaptation enables an effective coupling of energy into the antenna.

The antenna according to the invention can preferably be constructed in several layers. In a preferred embodiment, the antenna according to the invention comprises a first dielectric layer, a first low-dielectric constant layer, and a second dielectric layer. The first dielectric layer carries on its first surface the waveguide and on its second surface the ground plane. The second dielectric layer carries on one side the first radiation electrode. The low-dielectric-constant layer is disposed between the first dielectric layer and the second dielectric layer. The dielectric constant of the first low-dielectric-constant layer is lower than the dielectric constant of the first dielectric layer and lower than the dielectric constant of the second dielectric layer. Such an embodiment of an antenna enables a particularly simple production, whereby the radiation properties of the antenna are improved by the layers with a low dielectric constant. A layer with a very low dielectric constant reduces the dielectric losses and also reduces the occurrence of surface waves.

A multi-band structure may preferably be achieved by further introducing a second low-dielectric-constant layer and a third dielectric layer. The third dielectric layer in this case carries the second radiation electrode and the third radiation electrode. The second low-dielectric-constant layer is disposed between the second dielectric layer and the third dielectric layer. The dielectric constant of the second low-dielectric-constant layer is lower than the dielectric constant of the first, second and third dielectric layers.

A particularly simple and inexpensive production can be achieved by the first, second and third dielectric layer of FR4 material (conventional printed circuit board material) are produced. The low-dielectric-constant layer may preferably be formed by air. It has been found that an antenna according to the invention can be produced very inexpensively with a corresponding design, wherein the radiation properties are not influenced in a negative manner despite the inexpensive materials used.

Fig. 1

ein Schrägbild einer erfindungsgemäßen Antennenstruktur gemäß einem ersten Ausführungsbeispiel der vorliegenden Erfindung;

Fig. 2

ein Schrägbild einer erfindungsgemäßen Strahlergeometrie gemäß einem zweiten Ausführungsbeispiel der vorliegenden Erfindung;

Fig. 3

ein Schrägbild einer erfindungsgemäßen Antennenstruktur gemäß einem dritten Ausführungsbeispiel der vorliegenden Erfindung;

Fig. 4

ein Schrägbild einer erfindungsgemäßen Antennenstruktur gemäß einem vierten Ausführungsbeispiel der vorliegenden Erfindung;

Fig. 5

eine Fotografie eines Prototypen einer erfindungsgemäßen Antennenstruktur gemäß dem dritten Ausführungsbeispiel der vorliegenden Erfindung;

Fig. 6

eine Fotografie eines Prototypen einer erfindungsgemäßen Antennenstruktur gemäß dem vierten Ausführungsbeispiel der vorliegenden Erfindung;

Fig. 7

eine grafische Darstellung des Verlaufs des Reflexionskoeffizienten S11 für einen Prototypen einer erfindungsgemäßen Antenne gemäß dem dritten Ausführungsbeispiel der vorliegenden Erfindung;

Fig. 8

eine grafische Darstellung des Verlaufs der Pola-risationsentkopplung für einen Prototypen einer erfindungsgemäßen Antenne gemäß dem dritten Ausführungsbeispiel der vorliegenden Erfindung; und

Fig. 9

eine grafische Darstellung des Verlaufs des Reflexionskoeffizienten S11 für einen Prototypen einer erfindungsgemäßen Antenne gemäß dem vierten Ausführungsbeispiel der vorliegenden Erfindung.

Preferred embodiments of the present invention will be explained in more detail below with reference to the accompanying drawings. Show it:

Fig. 1

an oblique view of an antenna structure according to the invention according to a first embodiment of the present invention;

Fig. 2

an oblique view of a radiator geometry according to the invention according to a second embodiment of the present invention;

Fig. 3

an oblique view of an antenna structure according to the invention according to a third embodiment of the present invention;

Fig. 4

an oblique view of an antenna structure according to the invention according to a fourth embodiment of the present invention;

Fig. 5

a photograph of a prototype of an antenna structure according to the invention according to the third embodiment of the present invention;

Fig. 6

a photograph of a prototype of an antenna structure according to the invention according to the fourth embodiment of the present invention;

Fig. 7

a graphical representation of the course of the reflection coefficient S11 for a prototype of an antenna according to the invention according to the third embodiment of the present invention;

Fig. 8

a graphical representation of the course of polarization decoupling for a prototype of an antenna according to the invention according to the third embodiment of the present invention; and

Fig. 9

a graphical representation of the course of the reflection coefficient S11 for a prototype of an antenna according to the invention according to the fourth embodiment of the present invention.

Fig. 1 shows an oblique view of an antenna structure according to the invention according to a first embodiment of the present invention. The antenna structure is designated 100 in its entirety. The antenna structure 100 includes a ground plane 110 having an aperture 120. Furthermore, the antenna structure according to the invention comprises a radiation electrode 130, which is arranged above the ground surface 110. A feed line 140, shown here as a conductive strip, is disposed below the ground plane 110. The aperture 120 includes a first slot 150, a second slot 152 and a third slot 154. The first, second and third slots 150, 152, 154 each have a rectangular shape and constitute an opening of the ground plane 110. The first slot 150 and the second slot 152 are arranged so as to form a cross. The lengths of the first slot 150 and the second slot 152 are the same in the embodiment shown. The third slot 154 is longer than the first slot 150 and the second slot 152, and intersects the first and second slots 150, 152 in the area where the first and second slots 150, 152 also intersect, that is in the area Center of the cross formed by the first and the second slot. It should also be noted that the third slot 154 is perpendicular to the feed line 140 in a plan view, along a direction indicated by an arrow 170. The aperture 120 also has a high symmetry. The geometric center of the first, second and third slot 150, 152, 154 coincide, apart from manufacturing tolerances together. Furthermore, there is an axial symmetry of the aperture with respect to an axis 158 of the third slot 154. Furthermore, the aperture 120 is arranged with respect to the feed line 140 such that in a plan view the feed line 140 passes through the area where the first, second and third slots 150, 152, 154 intersect.

The radiation electrode 130 is a planar conductive electrode, which may also be referred to as a patch. It is arranged above the aperture 120 in the embodiment shown. The radiation electrode 130 shown is substantially rectangular. The radiation electrode 130 is designed to allow the radiation of a circularly polarized electromagnetic wave. In the embodiment shown, the radiation electrode is nearly square. However, it is also possible to use a rectangular radiation electrode in which at least one corner is beveled or cut off. Also, a radiation electrode having a slit in the center which allows circular polarization can be used. Finally, other geometries can be used as long as it is ensured that they allow a circular polarization. The radiation electrode 130 is arranged so that the aperture 120 is symmetrically located below the radiation electrode 130 in a plan view along a direction indicated by the arrow 170.

It should also be noted that overall, the waveguide and the radiation electrode are arranged so that energy from the waveguide can be coupled via the aperture to the radiation electrode (patch).

The operation of the present antenna structure can be easily described. The aperture 120 forms a resonant cross aperture according to the invention. The first slot 150 and the second slot 152 form a slot of the shape of a cross. The slots are sized so that no resonance of the cross-shaped slot occurs in an operating frequency range of the antenna. Thus it is achieved that on the radiation electrode, a vibration is excited, the radiation of a circularly polarized electromagnetic Wave entails. The cross-shaped configuration of the first and second slots 150, 152 of the aperture 120 assists in exciting a suitable mixed mode of vibration which enables such circular polarization of the radiated waves. The third slot 154 is operated in the vicinity of its resonance, so that it contributes to improving the adaptation of the antenna according to the invention. As shown, the third slot 154 is typically longer than the first and second slots 150, 152, which drives the slot 154 closer to resonance than the first and second slots. It is further noted that it is surprising that the third slot 154 does not interfere with the circular polarization of the radiated electromagnetic wave as would be expected according to conventional theories.

The geometry shown can be varied in a wide range without departing from the spirit of the present invention. For example, lengths of the three slots 150, 152, 154 that form the aperture 120 may be changed. For example, the length of the third slot 154 may be increased or decreased. Likewise, it is not necessary that the first slot 150 and the second slot 152 have the same length. Rather, the length of the slots 150, 152, 154 can be changed from each other to allow fine adjustments of the antenna structure according to the invention. Furthermore, it is possible to deviate from the strict symmetry of the aperture. This may be helpful, for example, even if the radiation electrode 130 does not have complete symmetry. Also with regard to the angle between the slots and between a slot and the feed line changes can be made. Twisting the slots by up to 20 degrees is possible to allow fine tuning of the antenna structure. Thus, the angle between the first slot and the second slot may deviate from a right angle by up to 20 degrees. The same applies to the angle between the third slot and the feed line.

Also, the radiation electrode 130 can be changed in a wide range. This can for example be rectangular or almost rectangular. It is preferable to use a radiation electrode which is almost square, with the dimensions or edge lengths slightly different. Such a radiation electrode allows the radiation of a circularly polarized electromagnetic wave. Likewise, it is preferably possible to use a radiation electrode having a nearly rectangular or square shape, wherein at least one corner is chamfered. It is further preferred in this case for symmetry reasons, to skew two opposite corners. Finally, a radiation electrode can be used which has a slot in the middle, wherein the slot is designed so that a circularly polarized wave can be radiated. Common extensions are possible, for example, the coupling of additional metallic elements to the radiation electrode 130. Also parasitic elements, for example, capacitive, inductive or resistive nature, can be coupled to the radiation electrode 130. This can be forced to form a desired mode. In addition, the bandwidth of the antenna can be further improved by parasitic elements. Finally, it is possible to cut corners of the radiation electrode 130 or beveled. This results in a coupling of different vibration modes that may exist between the radiation electrode 130 and the ground plane 110. As a result, a proper phase relationship is established between the various modes so that right-handed or left-handed circular polarization can be set. Moreover, the radiation electrode can also be changed in another form, for example by adding slots into the radiation electrode, suppress the unwanted modes or provide a suitable phase relationship between the desired modes.

The feeding of the antenna structure shown can be done in various ways. The metallic stripline 140 shown here may be replaced by various waveguides. For example, these waveguides may be a microstrip line. A coplanar waveguide can also be used. Furthermore, the supply of electrical energy can be effected by a stripline, a dielectric waveguide or a cavity waveguide.

It should also be noted that the Fig. 1 only represents a schematic representation of the basic structure of an antenna according to the invention. Features that are not essential to the antenna are not shown here. It is therefore to be noted that the illustrated metallic structures, in particular the ground plane 110, the radiation electrode 130 and the stripline 140 are typically supported by dielectric materials. Namely, it is possible to incorporate into the illustrated antenna structure 100 almost arbitrarily layers or structures of dielectric materials. Such structures may be, for example, layers that run parallel to the ground plane 110. The conductive structures may be deposited on these dielectric layers and patterned by a suitable method, such as an etching process. All that is required here is that the dielectric constant of a dielectric layer is not too large, since this increases the losses occurring in the antenna structure and worsens the radiation. Furthermore, care must be taken when introducing dielectric structures that no surface waves are excited, since these also significantly impair the radiation efficiency of an antenna structure.

For example, a dielectric layer may be present between ground plane 110 and stripline 140 to form a microstrip line. Such a microstrip line is particularly advantageous for the coupling of an antenna structure according to the invention. Moreover, a microstrip line can also be combined particularly well with active and passive circuit structures.

In addition to planar dielectric structures, differently shaped dielectric structures are also possible. For example, the radiation electrode 130 may be supported by a spacer made of a dielectric material. Such a design improves the mechanical stability of the antenna according to the invention and enables a cost-effective production.

The combination of dielectric layers and layers with very low dielectric constant, such as air layers, is also possible. Air layers reduce the electrical losses and may possibly reduce the excitation of surface waves.

Fig. 2 shows an oblique view of a radiator geometry according to the invention according to a second embodiment of the present invention. The radiator geometry is designated in its entirety by 200. It should be noted that in the Fig. 1 and 2 as well as in the other figures, like reference numerals designate like devices. Shown here is a ground plane 110 having an aperture 120. Details of the aperture are not shown for reasons of clarity, but the aperture corresponds to the basis of the Fig. 1 shown and described. Furthermore, the radiator geometry 200 according to the invention comprises a first radiation electrode 130. The aperture 120 represents an opening in the ground plane 110 which lies below the first radiation electrode 130 in a plan view along a direction which is indicated by the arrow 210. Above the first radiation electrode is a second radiation electrode 220. It is enclosed by the third radiation electrode 230, wherein a gap 240 exists between the second radiation electrode 220 and the third radiation electrode 230. The second radiation electrode 220 is connected to the third radiation electrode 230 via four conductive bars 250, 252, 254, 256. In the embodiment shown, these webs are arranged approximately in the middle of the edges of the second radiation electrode 220. The second radiation electrode 220 is thus arranged such that the first radiation electrode 130 lies between the second radiation electrode 220 and the ground surface 110. Furthermore, in the embodiment shown, the second radiation electrode 220 and the third radiation electrode 230 lie in a common plane. Furthermore, the dimensions of the second radiation electrode 220 deviate only slightly from the dimensions of the first radiation electrode 130. Preferably, the deviation is less than 20%.

Based on the structural description, the operation of a radiator geometry according to the invention will be described in more detail below. It should be noted that such a geometry enables the construction of circularly polarized dual or multi-band antennas. The individual layers can be carried by different boards. For example, a first board made of a dielectric material may carry the ground plane 110, while a second board carries the first radiation electrode 130 and a third board carries the second radiation electrode 220 and the third radiation electrode 230. The boards are not shown here for the sake of clarity but can be arranged so that the respective radiation electrodes are supported by any surface of the board. On the underside of a circuit board, which carries the ground plane 110, there may be a microstrip line, from which power is transmitted via the aperture 120 in the ground plane only to a smaller patch, which is formed by the first radiation electrode 130 become. The smaller patch formed by the first radiation electrode 130 is designed for the upper frequency band of two frequency bands. The power coupled through the aperture may subsequently be overcoupled to a larger patch designed for the lower of two frequency bands. The larger patch effectively consists of two patches, which in the embodiment shown are formed by the second radiation electrode 220 and the third radiation electrode 230. The larger patch can be interpreted as two nested patches with a short key. The inner minor patch formed by the second radiation electrode 220 is approximately the same size as the lower minor patch formed by the first radiation electrode 130. Conductive connecting webs 250, 252, 254, 256 connect the second radiation electrode 220 and the third radiation electrode 230. The connecting webs 250, 252, 254, 256 act on the second radiation electrode and the third radiation electrode depending on their position as a capacitive or inductive load or coupling whereby exerting an influence on the resonant frequency of the upper radiator formed by the second radiating electrode 220 and the third radiating electrode 230. A change in the position of a connecting web 250, 252, 254, 256 (with respect to the second and third radiation electrodes 220, 230 and with respect to the remaining connecting webs) can thus be used for a fine tuning of the antenna structure. For example, it is possible to move the connecting webs 250, 252, 254, 256 away from the center of the edges of the second radiation electrode 220 toward the corners of the second radiation electrode 220. In the case where two corners of the second radiation electrode 220 are chamfered, it has been found advantageous to move the connecting webs 250, 252, 254, 256 towards these chamfered corners. For the rest, it should be noted that the connecting webs need not be arranged in a strictly symmetrical manner. Rather, it is appropriate the connecting webs 250, 252, 254, 256 to be arranged slightly offset at opposite edges of the second radiation electrode, so that a connecting line between two opposite connecting webs 250, 252, 254, 256 is not parallel to an edge of the second radiation electrode. Such an asymmetrical arrangement results in a particularly large freedom in the fine tuning of the upper radiator. Finally, it should also be pointed out that the connecting webs can also be omitted if there is sufficient near-field coupling between the second radiation electrode 220 and the third radiation electrode 230.

The structure according to the invention thus effectively comprises two radiation-capable structures, namely a so-called lower patch, which is formed by the first radiation electrode 130, and which is effective in particular at higher frequencies, and an upper, larger patch, which by the second radiation electrode 220 and the third radiation electrode 230 is formed.

It should further be noted that the distance between the small patch formed by the first radiation electrode 130 and the ground surface is smaller than the distance between the second larger patch formed by the second radiation electrode 220 and the third radiation electrode 230 and the ground plane 110.

A structure according to the invention offers significant advantages over known structures, whereby a circularly polarized radiation in two frequency bands can be achieved without substantially influencing the purity of the polarization or by exciting surface waves to a greater extent.

It should be noted that in general an increase of an electrical substrate thickness to a Formation of surface waves of higher order leads. If such surface waves occur, the antenna gain is greatly reduced. In order to avoid or minimize the formation of surface waves, the two antenna structures, which are contained in a geometry according to the invention, have different effective substrate thicknesses for different frequency ranges. At low frequencies, the upper major patch formed by second radiation electrode 220 and third radiation electrode 230 is effective. The effective substrate thickness is equal to the distance of the second and third radiation electrodes from the ground plane 110. This distance is denoted by D here. At higher frequencies, on the other hand, the lower small patch formed by the first radiation electrode 130 is effective. The effective substrate thickness is equal to the distance between the first radiation electrode 130 and the ground plane 110, which is designated here by d.

Thus, it can be seen that the effective substrate thickness for low frequencies, denoted by D, is greater than the effective substrate thickness for high frequencies, denoted by d. This corresponds to the requirement that antennas for different frequencies must have different substrate thicknesses. Thus, by having the radiators operating at different frequencies at different planes and at different distances from the ground plane 110, the generation of surface waves can be effectively reduced. In fact, it is precisely the requirement that the effective substrate thickness should be lower for high frequencies than for low frequencies.

Likewise, the geometry according to the invention satisfies the requirement that the antenna for the upper frequency band (formed by the first radiation electrode 130) must be closer to the ground plane 110 and to the aperture 120 than the antenna for the lower frequency band (formed by the second Radiation electrode 220 and the third radiation electrode 230). That is, if the larger patch down (ie near the aperture) and the smaller patch at the top (away from the aperture) would result in poor polarization characteristics in the upper frequency range because the aperture would be shielded by the larger patch. In such a case, an effective coupling of the small patch over the aperture would no longer be possible. Similarly, a smaller patch that would be separated from the aperture by a larger patch could not radiate a circularly polarized wave with a small amount of orthogonal polarization.

Furthermore, the geometry according to the invention, in which the larger patch is composed of two parts, namely the second radiation electrode 220 and the third radiation electrode 230, avoids that the radiation of the lower patch lying below is too much shielded by the larger patch lying on top , Namely, if the antenna for the upper frequency band is closer to the ground plane 110 than the antenna for the lower frequency band, the strong shielding of the small radiator with the large one is to be avoided.

Reduced shielding of the radiation of the lower patch 130 by the overhead patch 220, 230 is achieved by the gap 140 between the second radiation electrode 220 and the third radiation electrode 230.

The radiator geometry 200 according to the invention can also be substantially changed. Thus, all the previously described changes can be applied to the individual radiation electrodes 130, 220, 230. For example, it is advantageous to trim the corners of the corresponding radiation electrodes. As a result, several modes required for circular emission can be coupled, while undesired modes can be suppressed.

Fig. 3 shows an oblique view of an antenna structure according to the invention according to a third embodiment of the present invention. The antenna structure is designated in its entirety by 300. It essentially corresponds to the basis of Fig. 1 shown antenna structure 100, so that the same facilities and geometry features are provided here with the same reference numerals. Unchanged features will not be described separately here. It should be noted, however, that in the antenna arrangement 300, a first corner 310 and a second corner 320 of the first radiation electrode 130 are cut off or bevelled. This geometric change helps to radiate a circularly polarized electromagnetic wave. Furthermore, the antenna arrangement 300 has a stub 330, which is attached to the strip line 140. This stub 330 is for further impedance matching of the present antenna structure. The dimensioning of such a stub for adaptation is well known to a person skilled in the art.

Furthermore, the shows Fig. 3 an enclosing box 340 that encloses the entire antenna structure. Such an enclosing cuboid can be used, for example, to limit a simulation area in an electromagnetic simulation of an antenna structure.

The Fig. 4 shows an oblique view of an antenna structure according to the invention according to a fourth embodiment of the present invention. The antenna structure is designated 400 in its entirety. The antenna structure 400 comprises a feed line 140, a ground plane 110 with an aperture 120 and a first radiation electrode 130, a second radiation electrode 220 and a third radiation electrode 230. The geometry of the first radiation electrode 130 corresponds substantially to the geometry of FIG Fig. 3 The first and second radiation electrodes 220, 230 are arranged in much the same way as they are based on Fig. 2 is described. However, in the antenna structure 400, two opposite corners 410, 420 of the second radiation electrode 220 are chamfered. The third radiation electrode 230 in turn encloses the second radiation electrode 220, wherein between the second radiation electrode 220 and the third radiation electrode 230, a slot or gap 240 is present. It should also be noted that the third radiation electrode 230 is adapted in shape to the second radiation electrode 220. That is, the third radiation electrode 230 is fitted to the chamfered corners 410, 420 of the second radiation electrode 220 such that the gap 240 between the second radiation electrode 220 and the third radiation electrode 230 substantially coincides also in the region of the chamfered corners 410, 420 remains constant width. The inner edges of the third radiation electrode 230 are thus substantially parallel to the outer edges of the second radiation electrode 220. Also, the third radiation electrode 230 has two outer bevelled corners 430, 440 adjacent the chamfered corners 410, 420 of the second radiation electrode 220. Thus, each of the first, second, and third radiation electrodes 130, 220, 230 has beveled corners 310, 320, 410, 420, 430, 440, with each of the adjacent corners of the various radiation electrodes being chamfered. The second and third radiation electrodes 220, 230 are coupled via connecting webs 250, 252, 254, 256, wherein the connecting webs 250, 252, 254, 256 approximately in the middle of edges of a rectangle, the second radiation electrode 220, apart from the bevelled Corners, describes, are arranged.

It should also be noted that the size of the second radiation electrode 220 is equal to the size of the first radiation electrode 130, except for a deviation of at most 20%. Also differ in the shape of the first and second radiation electrodes 130, 220 are not essential. They are therefore almost parallel electrodes of almost the same shape and with almost the same dimensions.

It is also referred to here again explicitly on the layer order. The feed line 140 forms the lowermost conductive layer. In addition, a ground plane 110 is arranged, which has an aperture 120. Once again, the first radiation electrode 130 lies in one plane. In a further, further above, plane, the second radiation electrode 220 and the third radiation electrode 230 are arranged. The respective metallizations, d. H. the feed line 140, the ground plane 110 and the first, second and third radiation electrodes 130, 220, 230 are each supported by dielectric layers.

It should also be noted here that the width of the feed line 140 is changed for adaptation purposes. Remote from the aperture, the feedline 140 has a wide section 450 while the feedline 140 is narrower near the aperture. A narrow feed line is advantageous because it causes a greater concentration of the electric field. This allows a stronger coupling of the radiation electrodes to the feed line through the aperture 120. Incidentally, the change in the width of the feed line also serves to match the impedance, and the adaptation can be influenced by a suitable choice of the length of the thin piece 460.

In addition, an enclosing rectangle 470 is shown, which delimits a simulation area in which the antenna structure is simulated. The enclosing rectangle also indicates the thickness of the respective layers.

Fig. 5 shows a photograph of a prototype of an antenna structure according to the invention according to the third embodiment of the present invention. Shown here is a constructed monoband antenna, the frequency range from 2.40GHz to 2.48GHz. The antenna is designated 500 in its entirety. It has a first plate 510 of a dielectric material and a second plate 520 of a dielectric material. The two plates are separated or fixed by four spacers 530 made of a dielectric material. The first dielectric plate 510 carries a first radiation electrode 130. The second dielectric plate 520 carries on a top surface the ground plane 110 having an aperture 120. The lower side of the dielectric plate 530 carries a feed line through which electrical power is supplied to the antenna from an SMA jack 550.

The antenna assembly 500 has a first dimension 570, which can be considered as a width, of 75 mm. A second dimension 572, which can also be considered as a length, is also 75 mm. Finally, a third dimension 574, which can be understood as height, is 10 mm. Only for size comparison here is a one-euro coin 576 shown.

Fig. 6 shows a photograph of a prototype of an antenna structure according to the invention according to the fourth embodiment of the present invention. The antenna structure is designated in its entirety by 600. It comprises a first dielectric layer 610, a second dielectric layer 620 and a third dielectric layer 630.

The 3 dielectric layers or plates 610, 620, 630 are held by dielectric spacers 640. The first dielectric plate 610 carries here a second radiation electrode 220 and a third radiation electrode 30. The second dielectric plate carries a first radiation electrode 130. The third dielectric plate 630 carries on one side a ground plane 110 and on the other side a feed line 140. The feed line is incidentally, led out to an SMA socket 650. The entire antenna structure 600 forms a dual band antenna.

The antenna 600 has a first dimension 670, which may also be considered as a length. This first dimension is 75 mm. Furthermore, the antenna 600 has a second dimension 672, which can be considered as a width, and which is also 75 mm. A third dimension 674 of the antenna 600 may be considered as a height. This height is 10.5 mm.

The dual-band antenna 600 shown is based on the monoband antenna 500, whereby the monoband antenna has been improved into a dual-band antenna. The antenna 600, the in its basic structure of the in Fig. 4 shown antenna 400 is constructed of several layers, which will be explained in more detail below. The lowermost position of the antenna is formed by a structured conductive layer, for example a metallization layer or metal layer, which as a whole forms a microstrip line. This microstrip line is deposited on the underside of a first FR4 type substrate, with the first substrate having a thickness of 0.5 mm. The first substrate corresponds to the third dielectric layer 630. On top of the first substrate is applied a ground plane having a total extension of 75 mm x 75 mm. The ground plane further includes an aperture 120. Above the ground plane is a layer that is not filled with a dielectric material. Accordingly, the antenna thus comprises an air layer having a thickness of 5 mm. Above this layer of air is another conductive layer on which the first radiation electrode is formed as a patch. The further conductive layer is supported by a second dielectric layer of FR4, again having a thickness of 0.5 mm. The second dielectric FR4 layer is the same as in FIG Fig. 6 shown second dielectric layer 620. Above the second dielectric FR4 layer is again a layer in which there is no solid dielectric. This creates a second layer of air whose thickness is 4 mm. Again above it is a third dielectric FR4 layer having a thickness of 0.5 mm. The third FR4 dielectric layer carries another conductive layer on which the second radiation electrode and the third radiation electrode are formed in the form of patches by structuring. Conductive connecting webs between the second radiation electrode and the third radiation electrode have a width of 1 mm. The entire antenna structure thus comprises the following layers in the order shown: microstrip line; FR4 (0.5 mm); Ground area (75 mm x 75 mm, with aperture); Air (5 mm); Patch 1 (first radiation electrode); FR4 (0.5 mm); Air (4 mm); FR4 (0.5 mm) and patch 2 (second radiation electrode and third radiation electrode). All layers and dimensions can vary by up to 30%. However, it is preferred that the deviation from the preferred dimensions is not more than 15%.

Fig. 7 FIG. 12 is a graph showing the course of the reflection coefficient S11 for a prototype 500 of an antenna according to the third embodiment of the present invention. The graphical representation is designated in its entirety by 700. The input reflection factor S11 was measured for a patch antenna designed for a frequency range of 2.40 to 2.48 GHz. A photograph of such an antenna 500 is shown in FIG Fig. 5 shown.

The abscissa 710 has the frequency of 2.15 GHz to 2.85 GHz. The ordinate 712 shows in logarithm form the amount of the input reflection factor S11. Here, the input reflection factor is plotted in a range of -50 dB to 0 dB. A first curve 720 shows a simulated input reflection factor. A second curve 730 shows the measured value for the input reflection factor.

According to the measurement, the input reflection factor is below -10 dB in the entire frequency range shown from 2.15 GHz to 2.85 GHz. The simulation also shows a similar broadband characteristic of the antenna.

Fig. 8 shows a graphical representation of the polarization decoupling for a prototype 500 of an antenna according to the invention according to the third embodiment of the present invention. The graphical representation is designated in its entirety by 800. At abscissa 810, the frequency is plotted in a range of 2.3 GHz to 2.55 GHz. The ordinate 812 shows the polarization decoupling in decibels in a range between 0 and 25 dB. A first curve 820 shows a simulated history of polarization decoupling, while a second curve 830 represents measured values. In the required bandwidth from 2.40 GHz to 2.48 GHz, the cross polarization is suppressed by more than 15.5 dB with a sufficient matching factor.

Fig. 9 FIG. 12 is a graph showing the course of the reflection coefficient S11 for a prototype 600 of an antenna according to the fourth embodiment of the present invention. The graph is designated 900 in its entirety. Shown here are measurement results for the reflection coefficient of a dual-band antenna according to the invention, as described on the basis of FIG Fig. 4 and 6 has been described. The abscissa 910 shows the frequency range between 2 GHz and 6 GHz. At the ordinate 912, the amount of the input reflection factor S11 is plotted in logarithmic form from -40 dB to + 40 dB. A curve 920 shows the variation of the input reflection factor versus frequency. Also shown are a first marker 930, a second marker 932, a third marker 934 and a fourth marker 936. The first marker indicates that the input reflection factor at 2.40 GHz is -13.618 dB. The second marker shows an input reflection factor of -16.147 dB at 2.48 GHz. The third Marker shows an input reflection factor of -9.457 dB at 5.15 GHz, and the fourth marker shows an input reflection factor of -10.011 dB at 5.35 GHz. Finally, the fifth marker shows an input reflection factor of -0.748 dB at 4.0008 GHz.

Thus, it can be seen that the input reflection factor in the ISM band between 2.40 GHz and 2.48 GHz is less than -13 dB, and that the input reflection factor in the ISM band between 5.15 GHz and 5.35 GHz is less than -13 dB -9.4 dB.

In addition to the input reflection factor, the radiation characteristics of the dual-band antenna were also measured. In the ISM band between 2.40 GHz and 2.48 GHz, the antenna gain of a prototype dual band antenna is between 8.9 dBic and 9.3 dBic. The half-value width is 70 °, and the polarization decoupling is between 11 dB and 22 dB.

In the ISM band between 5.15 GHz and 5.35 GHz, the antenna gain is between 5.9 dBic and 7.3 dBic. The half width is 35 °, the polarization decoupling between 5 dB and 7 dB.

The required matching properties and radiation properties can thus be achieved with a dual-band antenna according to the invention. It should also be noted that the polarization purity for the upper frequency range can still be optimized. For this example, geometric details can be changed.

In summary, therefore, the present invention provides a planar circularly polarized antenna that can be used in the ISM bands from 2.40 GHz to 2.48 GHz and 5.15 GHz to 5.35 GHz. The proposed shape of the slot for an aperture-coupled patch antenna allows the radiation almost purely circular polarized waves at a relatively large bandwidth of the reflection coefficient S11. This is especially possible for multiband antennas. With an antenna according to the invention, a radio link can be achieved in which the strength of the signal received by an antenna according to the invention in a linear polarization of a transmitter is independent of the installation position of the receiving antenna. In other words, by a circularly polarized antenna, a linearly polarized signal can be received regardless of the orientation of the antenna.

The antenna according to the invention was developed in several steps. A first sub-task was to develop an aperture-coupled antenna for a frequency range of 2.40 to 2.48 GHz with right-handed circular polarization (RHCP). In the simulation, particular care was taken to achieve a strong suppression of the orthogonal polarization within the required bandwidth. It has been found that when a patch is fed via a non-resonant cross-aperture, the cross-polarization is very strongly suppressed. However, in such a non-resonant cross-aperture, the bandwidth of the reflection coefficient is narrow. A resonant rectangular aperture (so-called SSFIP principle) has a larger bandwidth, but the polarization decoupling is weaker. Finally, an earlier unknown combination of the two slot geometries has proven to be advantageous, which is referred to here as a resonant cross aperture. A corresponding antenna geometry was in the Fig. 1 . 3 and 5 shown.

Furthermore, it has been shown that a geometry of the aperture or slot according to the invention also enables the construction of circularly polarized dual or multi-band antennas. For this purpose, the concept described below can be used. In the case of two bands, the antenna consists of three boards. Corresponding arrangements are for example in the Fig. 4 and 6 shown. On the bottom On the lower circuit board there is a microstrip line whose power is coupled via an aperture in the ground plane first to a small patch (for the upper frequency band) and then to a larger patch (for the lower frequency band), consisting of two patches. The larger patch can be interpreted as "nested patches with short circuits". The inner smaller patch is preferably the same size as the lower patch.

By such a structure or such a dual-band concept, a number of problems that occur in conventional antennas can be solved. Thus, the increase in the electrical substrate thickness conventionally leads to the formation of surface waves of higher order, which greatly reduces the antenna gain. Therefore, the two antennas must have different substrate thicknesses for different frequency ranges. The antennas must therefore be in different planes. This can be achieved with an antenna geometry according to the invention.

A conventional variant with a larger patch at the bottom and a smaller patch at the top has poor polarization properties because the aperture is shielded with the larger patch. The antenna for the upper frequency band must therefore be closer to ground than the antenna for the lower frequency band, which can be achieved with a geometry according to the invention.

Since the antenna for the upper frequency band must therefore be closer to the ground plane than the antenna for the lower frequency band, a strong shielding of the small radiator for the upper frequency band by the large radiator for the lower frequency band is to be avoided. This can be achieved by forming the radiator for the lower frequency band by two radiation electrodes between which there is a gap.

The adaptation of an antenna according to the invention can be done by a transformer or by a stub.

An antenna according to the invention has a number of advantages over conventional antennas. The feeding of an antenna via a resonant Phillips allows the construction of completely planar relatively small and inexpensive antennas. At the same time, high polarization purity and a large impedance bandwidth can be achieved. It is also possible to construct planar circularly polarized multiband antennas. In this case, the area requirement of the entire antenna is determined only by the size of the antenna element for the lowest frequency. In comparison to broadband antennas, an antenna according to the invention furthermore offers a better prefiltering.

Claims (19)

1. An aperture-coupled antenna (100; 300; 400; 500; 600) comprising:

   a first radiation electrode (130) the geometrical shape of which is implemented to allow radiation of a circularly polarized electromagnetic wave;

   a ground area (110); and

   a wave guide (140) which is implemented to supply energy to the antenna,

   wherein the wave guide (140) is arranged spaced apart from the ground area (110) on a first side of the ground area (110), and wherein the first radiation electrode (130) is arranged spaced apart from the ground area (110) on a second side of the ground area (110);

   wherein the ground area (110) comprises an aperture (120) including a first slot (150) in the ground area (110), a second slot (152) in the ground area (110) and a third slot (154) in the ground area (110), wherein the first slot (150) and the second slot (152) together form a slot in the shape of a cross, wherein the third slot (154) passes through the intersection of the first slot (150) and the second slot (152);

   wherein additionally the wave guide (140) and the first radiation electrode (130) are arranged such that energy can be coupled from the wave guide (140) through the aperture (120) to the first radiation electrode (130); characterized in that

   the third slot (154) is implemented such that an operating frequency for which the aperture-coupled antenna is designed deviates by at most 30% from the resonant frequency of the third slot (154); and that

   the length of the first slot (150) and the length of the second slot (152) differ from the length of the third slot (154) so that the third slot (154) at the operating frequency is operated nearer to its resonance than the first slot (150) and the second slot (152).
2. The aperture-coupled antenna (100; 300; 400; 500; 600) according to claim 1, wherein the third slot (154) is longer than the first slot (150), and wherein the third slot (154) is longer than the second slot (152).
3. The aperture-coupled antenna (100; 300; 400; 500; 600) according to claim 1 or 2, wherein the first slot (150) and the second slot (152) are orthogonal to each other and together form a slot in the shape of a rectangular cross having arms of equal lengths.
4. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 1 to 3, wherein a midpoint of the third slot (154) coincides with a midpoint of the cross-shaped slot formed by the first slot (150) and the second slot (152).
5. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 1 to 4, wherein a geometrical midpoint of the first slot (150), a geometrical midpoint of the second slot (152) and a geometrical midpoint of the third slot (154) coincide, and wherein the aperture (120) is axisymmetrical relative to an axis (158) of the third slot (154), wherein the axis (158) of the third slot (154) passes along a greatest dimension of the third slot (154).
6. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 1 to 5, wherein the first slot (150) and the second slot (152) are implemented such that the first slot (150) and the second slot (152) do not comprise resonance in an operating frequency range for which the aperture-coupled antenna is designed.
7. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 1 to 6, wherein the third slot (154) is implemented such that a resonant frequency of the third slot (154) is within an operating frequency range for which the aperture-coupled antenna is designed.
8. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 1 to 7, wherein the aperture-coupled antenna is a planar antenna.
9. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 1 to 8, wherein the wave guide (140) is a microstrip line, a coplanar wave guide, a strip line, a dielectric wave guide or a cavity wave guide.
10. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 1 to 9, wherein the aperture (120) and the first radiation electrode (130) are implemented such that the aperture-coupled antenna, except for parasitic effects, radiates a circularly polarized electromagnetic wave.
11. The aperture-coupled antenna (400; 600) according to one of claims 1 to 10, further comprising a second radiation electrode (220) and a third radiation electrode (230), wherein the second radiation electrode (220) is basically parallel to the first radiation electrode (130) and arranged such that the first radiation electrode (130) is arranged between the second radiation electrode (220) and the ground area (110), and wherein the third radiation electrode (230) encloses the second radiation electrode (220) in a projection along an axis normal to the second radiation electrode (220).
12. The aperture-coupled antenna (400; 600) according to claim 11, wherein the second radiation electrode (220) and the third radiation electrode (230) are in one plane, and wherein the third radiation electrode (230) encloses the second radiation electrode (220) in the plane.
13. The aperture-coupled antenna (400; 600) according to claim 11 or 12, wherein the second radiation electrode (220) and the third radiation electrode (230) are coupled to each other via at least one conductive connective land (250, 252, 254, 256).
14. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 1 to 10, comprising a first dielectric layer, a first layer of low dielectric constant, and a second dielectric layer,
    wherein the first dielectric layer supports the wave guide (140) on its first surface and supports the ground area (110) on its second surface,
    wherein the second dielectric layer supports the first radiation electrode (130) on a surface;
    wherein the second layer of low dielectric constant is arranged between the first dielectric layer and the second dielectric layer;
    wherein a dielectric constant of the first layer of low dielectric constant is smaller than a dielectric constant of the first dielectric layer, and wherein the dielectric constant of the first layer of low dielectric constant is smaller than a dielectric constant of the second dielectric layer.
15. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 11 to 13, comprising a first dielectric layer, a first layer of low dielectric constant, and a second dielectric layer,
    wherein the first dielectric layer supports the wave guide (140) on its first surface and supports the ground area (110) on its second surface,
    wherein the second dielectric layer supports the first radiation electrode (130) on a surface;
    wherein the first layer of low dielectric constant is arranged between the first dielectric layer and the second dielectric layer;
    wherein a dielectric constant of the first layer of low dielectric constant is smaller than a dielectric constant of the first dielectric layer, and wherein the dielectric constant of the first layer of low dielectric constant is smaller than a dielectric constant of the second dielectric layer.
16. The aperture-coupled antenna (400; 600) according to claim 15, further comprising a second layer of low dielectric constant and a third dielectric layer,
    wherein the third dielectric layer supports the second radiation electrode (220) and the third radiation electrode (230);
    wherein the second layer of low dielectric constant is arranged between the second dielectric layer and the third dielectric layer;
    wherein a dielectric constant of the second layer of low dielectric constant is smaller than the dielectric constant of the first dielectric layer, wherein the dielectric constant of the second layer of low dielectric constant is smaller than the dielectric constant of the second dielectric layer, and wherein the dielectric constant of the second layer of low dielectric constant is smaller than a dielectric constant of the third dielectric layer.
17. The aperture-coupled antenna (100; 300; 400; 500; 600) according to claim 14, 15 or 16, wherein the first, the second or the third dielectric layer is made of FR4 material.
18. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 14 to 17, wherein the first layer of low dielectric constant or the second layer of low dielectric constant is an air layer.
19. The aperture-coupled antenna (100; 300; 400; 500; 600) according to one of claims 1 to 18, which is implemented such that impedance matching can be achieved with a standing wave ratio of smaller than 2 in at least two frequency bands.

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