EP1540565B1

EP1540565B1 - Switched capacitor system, method, and use

Info

Publication number: EP1540565B1
Application number: EP03791719A
Authority: EP
Inventors: Patrick J. Quinn
Original assignee: Xilinx Inc
Current assignee: Xilinx Inc
Priority date: 2002-08-29
Filing date: 2003-08-20
Publication date: 2012-01-18
Anticipated expiration: 2023-08-20
Also published as: CA2494264A1; JP2005537749A; WO2004021251A2; EP1540565A2; JP4454498B2; WO2004021251A3; CA2494264C

Abstract

An apparatus and method for adding input voltage signals. First 206 and second 208 input voltage signals are respectively sampled onto first 218 and second 228 capacitors during a first clock phase 202. In response to a second clock phase 204, the first sampled input voltage 206 that is held on the first capacitor 218 is coupled to the negative input terminal 236 of an amplifier 230, and the second sampled voltage 208 held on the second capacitor 228 is coupled to the positive terminal 240 of the amplifier 230. A feedback voltage is provided from the amplifier output 216 to the negative amplifier input 236 via the first capacitor 218 during the second clock phase 204. The first 206 and second 208 input voltage signals are added at the amplifier 230 during the second clock phase 204 to output 216 the sum in response to the sampled input voltage signals and the output feedback, whereby the resulting transfer function is independent of capacitor mismatch and non-linearity.

Description

FIELD OF THE INVENTION

The present invention generally relates to switched capacitor circuits, and more particularly to a switched capacitor summing circuit and its use in an analog-to-digital converter.

BACKGROUND

The ubiquitous switched capacitor charge transfer circuit has long been used in a wide range of signal processing applications. Switched capacitor circuits are a class of discrete-time systems that are often used in connection with filters, analog-to-digital converters (ADCs), digital-to-analog converters (DACs), and other analog/mixed signal applications. Conventional switched capacitor circuits are based on creating coefficients of a transfer function by transferring charge from one input capacitor C₁ to a second capacitor C₂ in the feedback loop of an amplifier via the virtual node of that amplifier so as to create a transfer of C₁ /C₂.
However, finite amplifier DC gain and bandwidth results in incomplete charge transfer from C₁ to C₂. This, together with inaccuracies in the matching of the capacitors C₁ and C₂, results in the creation of an inaccurate transfer function. Many applications, such as ADCs, accurate high-Q filters, etc. require very high accuracies in the transfer function, such as accuracies exceeding 0.1%. This kind of accuracy is virtually impossible using conventional circuits in modern day CMOS processes. Often, the values of the capacitors are trimmed at manufacture, or some active calibration routines are executed, switching in and out small value capacitors in order to create an accurate transfer. Such schemes are expensive for high volume manufacture. To reduce capacitor mismatch problems, special capacitors such as double poly or Metal-Insulator-Metal (MiM) capacitors may be used, but the capacitor mismatch problem is not eliminated. Further, such circuits that employ voltage-to-charge and charge-to-voltage translations via the virtual earth node have limited immunity to extraneous noise sources, as the virtual earth node is a well known pick-up point for unwanted noise.
US 6 362 770 describes an example of a switched capacitor gain stage having a first and second input and an output and includes an amplifier having an output connected to the output of the gain stage and further including a first, second and third capacitor pair. A switch network is operable to switch selected ones of the first second and/or third capacitor pairs to receive varying voltage from the first or second inputs and/or switch a capacitor pair across the amplifier and a reference voltage. The capacitors of the capacitor pairs are connectable via switches to an inverting input of the amplifier.
Prior art switched capacitor circuits such as those described above are often used in the design of analog-to-digital converters (ADCs), such as pipelined and algorithmic ADCs. The transfer characteristic of such ADCs is affected by nonlinearities in the analog hardware. While offsets in the amplifier and comparators may be corrected through the use of digital error correction (DEC) logic, other sources of error remain. These include the inaccuracies in the creation of a multiply-by-two (MX2) gain function (including subtraction of sub-DAC levels), and variations in the reference levels. Variations in the reference levels is only an issue in pipelines ADCs, in which separate hardware in each stage samples +Vref and -Vref. Static errors in the reference levels are not an issue for algorithmic ADCs, since each rotation of the ADC sample the same references in the same way with the same hardware. The absolute accuracy of the reference levels is not important in a differential implementation, as long as they are stable and do not vary from conversion to conversion. Thus, the remaining sources of error that limit the accuracy of the complete ADC are the accuracy of the MX2 function, and the accuracy of the sub-DAC through the accuracy with which the DAC levels can be generated. In actual state of the art implementations, these errors are predominantly caused by the capacitor mismatch problems described above.
The present invention addresses these and other shortcomings of the prior art, and provides a solution to the problems exhibited by prior art switch capacitor circuits and ADCs.

SUMMARY OF THE INVENTION

In various embodiments, the present invention provides a method and apparatus for summing a plurality of input voltage signals and providing optional level shifting,
where the resulting transfer function is independent of capacitor mismatch and non-linearity.
In accordance with one embodiment of the invention, a circuit is provided for adding a plurality of input signals. The circuit includes an amplifier having first and second input terminals and an output terminal. A first capacitance is coupled to receive a first input signal and to store a corresponding first voltage in response to a first clock phase, and a second capacitance is coupled to receive a second input signal and to store a corresponding second voltage in response to the first clock phase. In response to a second clock phase, a first switch circuit is coupled to the first capacitance to provide the first voltage to the first input terminal of the amplifier, and to couple the output terminal of the amplifier to the first capacitance via a feedback loop. A second switch circuit is coupled to the second capacitance to provide the second voltage to the second input terminal of the amplifier in response to the second clock phase. In this manner, the amplifier outputs a voltage signal corresponding to a sum of the first and second input signals that is independent of a ratio of the first and second capacitances.
In accordance with another embodiment of the invention, a method is provided for adding input voltage signals. First and second input voltage signals are respectively sampled onto first and second capacitors during a first clock phase. In response to a second clock phase, the first sampled input voltage that is held on the first capacitor is coupled to the negative input terminal of an amplifier, and the second sampled voltage held on the second capacitor is coupled to the positive terminal of the amplifier. A feedback voltage is provided from the amplifier output to the negative amplifier input via the first capacitor during the second clock phase. The first and second input voltage signals are added at the amplifier during the second clock phase to output the sum in response to the sampled input voltage signals and the output feedback, whereby the resulting transfer function is independent of capacitor mismatch and non-linearity.
In various other embodiments, the present invention provides a method, apparatus, and system for providing accurate level shifting, residue multiplication, and sample-and-hold functions for analog-to-digital conversions, without requiring charge transfer between capacitors in a switched capacitor arrangement, thereby eliminating capacitor mismatch as a source of ADC errors.
In accordance with one embodiment of the invention, an ADC stage is provided for use in analog-to-digital conversions. The ADC stage includes an amplifier having first and second input terminals, and an output terminal to provide an analog ADC residue signal. First and second capacitances sample an input voltage signal and a complemented input voltage signal respectively, in response to a first clock phase. A first switch circuit is coupled to the first capacitance to provide the sampled input voltage signal to the first input terminal of the amplifier, and to couple the output terminal of the amplifier to the first capacitance via a feedback loop, in response to a second clock phase. A second switch circuit is coupled to the second capacitance to provide an inverted version of the sampled complemented input voltage signal to the second input terminal of the amplifier in response to the second clock phase. A level shifting circuit is coupled to receive the input voltage signal, and in response, to select one of a plurality of reference voltages. The amplifier adds the input signal to the inverted version of the complemented input signal as shifted by the level shifting circuit, to create the analog ADC residue signal for use in a subsequent ADC stage. Differential and/or double-sampling versions are also provided in accordance with an embodiment of the present invention. Further, an embodiment of the present invention may be used in a number of ADC configurations, including algorithmic and pipelined ADC configurations.
In accordance with another embodiment of the invention, a method is provided for converting an analog input signal to a digital signal using an amplifier. The method includes sampling the analog input signal onto a first capacitor, and the complement of the analog input signal onto a second capacitor. The sampled analog input signal is provided to a first input terminal of the amplifier by controllably connecting the first capacitor between the amplifier output and the first input terminal in a unity gain feedback configuration. An inverted version of the sampled complemented analog input signal, level shifted by one of a plurality of selectable reference voltages, is provided at a second input terminal of the amplifier by controllably coupling the second capacitor between a selected reference voltage and the second input terminal of the amplifier. The sampled analog input signal is added to the inverted version of the sampled complemented analog input signal, and the selected reference voltage is subtracted therefrom to provide a residue signal available for use in subsequent conversion stages.
It will be appreciated that various other embodiments are set forth in the Detailed Description and Claims which follow.

BRIEF DESCRIPTION OF THE DRAWINGS

Various aspects and advantages of the invention will become apparent upon review of the following detailed description and upon reference to the drawings in which:

FIG. 1A illustrates a conventional switched capacitor circuit that exhibits inherent capacitor mismatch and non-linearity problems addressed by an embodiment of the preset invention;
FIG. 1B illustrates another conventional switched capacitor circuit having an inverting charge transfer stage with no delay;
FIG. 2A illustrates a representative single-sampling circuit implementing the principles of the present invention;
FIG. 2B illustrates a representative single-sampling circuit implementing the principles of the present invention and referenced to a common reference voltage;
FIG. 3 illustrates a representative double-sampling circuit implementing the principles of the present invention;
FIG. 4 illustrates an example of an N-path sum-delay-shift circuit in accordance with one embodiment of the present invention;
FIG. 5 is a flow diagram illustrating a method for adding at least two input voltage signals in accordance with the principles of the present invention;
FIG. 6 is a block diagram illustrating a typical 1.5-bit ADC stage;
FIG. 7 is a block diagram of an N-bit algorithmic ADC;
FIG. 8 is a block diagram of a representative pipelined ADC;
FIG. 9 illustrates an example of a residue transfer characteristic of a complete 1.5-bit ADC stage;
FIG. 10A is a graph illustrating the effects on the transfer function of an ADC,exhibiting a gain error greater than two in the multiply-by-two function;
FIG. 10B is a graph illustrating the effects on the transfer function of an ADC exhibiting a gain error less than two in the multiply-by-two function;
FIG. 10C is a graph illustrating the effect of sub-DAC errors in the first stage of the ADC on the total transfer function;
FIG. 11A illustrates a switched capacitor implementation of a 1.5-bit stage for a single-ended application;
FIG. 11B illustrates a differential switched capacitor implementation of a 1.5-bit stage;
FIGs. 12A and 12B illustrate two halves of a representative differential 1.5-bit ADC stage in accordance with the principles of the present invention;
FIG. 13 illustrates an implementation of a differential ADC stage in accordance with the principles of the present invention;
FIG. 14 illustrates a representative waveform diagram corresponding to an algorithmic ADC in accordance with an embodiment of the present invention;
FIGS. 15A and 15B illustrate representative examples of an ADC stage corresponding to a first half of a differential, algorithmic ADC implementation in accordance with an embodiment of the present invention;
FIG. 16 illustrates a representative portion of an algorithmic ADC stage 1100 which implements such a reset circuit in accordance with one embodiment of the invention;
FIG. 17 illustrates a non-differential, single-sampling ADC stage in accordance with the principles of the present invention; and
FIG. 18 is a flow diagram of a method for converting an analog input signal to a digital input signal in accordance with one embodiment of the present invention.

DETAILED DESCRIPTION

In the following description of the exemplary embodiment, reference is made to the accompanying drawings which form a part hereof, and in which is shown by way of illustration various manners in which the invention may be practiced. It is to be understood that other embodiments may be utilized.

Switched Capacitor

An exemplary embodiment of the present invention is directed to an apparatus and methodology that provides highly accurate, scalable addition and subtraction functions with optional output voltage level shifting, without requiring special circuit or calibration options. The exemplary embodiment of the present invention can serve as a replacement for existing switched capacitor circuits that inherently exhibit capacitance mismatch and non-linearity characteristics. In accordance with the exemplary embodiment of the present invention, input signals are sampled onto corresponding capacitor circuits, and the resulting voltages stored thereon are subsequently coupled to a buffering amplifier to determine the sum/difference of the input signals. No transfer of charge occurs between the capacitor circuits, which provides a transfer function that is independent of capacitor mismatch concerns. A voltage level shift can also be implemented, by providing a level shifting voltage as a reference voltage to one of the capacitor circuits during the summing operation.
FIG. 1A illustrates a conventional switched capacitor that exhibits inherent capacitor mismatch and non-linearity problems addressed by the exemplary embodiment of the present invention. A conventional manner for creating analog sampled data signal processing functions is based on the charge transfer stage 100 shown in FIG. 1A. The charge transfer stage 100 is a non-inverting charge transfer stage with a half clock period delay.
The circuit 100 includes three input signals, labeled Vin_1 102, Vin_2 104, and Vin_3 106. Vin_2 104 is the voltage to which the positive terminal of the amplifier 108 is connected, and thus is the virtual earth voltage between the positive and negative terminals of the amplifier 108. Generally, Vin_2 104 at the positive terminal of the amplifier 108 is the voltage to which the top plate of capacitor C ₁ 110 is connected to on the first clock phase, clk1 112. If this were not the case, the negative input of the amplifier 108 would have to be returned to voltage Vin_2 on a second clock phase, clk2 114, which would considerably reduce the settling speed_of the amplifier 108. Furthermore, Vin_2 104 is generally a fixed reference voltage. The voltage Vin_3 106 does not necessarily have to be equivalent to Vin_2 104, but it generally is in conventional designs.
On the first clock phase, clkl 112, the signal voltage Vin_1 102 is sampled on to C ₁ 110 with respect to Vin_2 104. This occurs due to switches 116, 118 closing on the clk1 112 clock phase, thereby placing the capacitor C ₁ 110 between the signal voltage Vin_1 102 and the reference voltage Vin_2 104. On the subsequent clock phase clk2 114, switches 116, 118, and 120 open, and switches 122, 124, and 126 close. This coupled the top plates of capacitors C₁ 110 and C ₂ 128, and the charge on C ₁ 110 from the sampling phase is transferred to C ₂ 128 via the virtual earth node of the amplifier 108 between the positive and negative input terminals. More particularly, in response to assertion of the clk2 114 phase, the negative feedback through C₂ drives the amplifier 108 input differential voltage and thus the voltage across C₁ to zero (assuming for purposes of discussion that Vin_2 = Vin_3) via the virtual earth node. The charge stored on C₁ is must then be transferred to C₂, producing an output voltage equal to the signal voltage Vin_1 102 times the ratio of C₁/C₂. Taking into consideration clock phase delays, the net effect (assuming Vin_3 106=Vin_2 104) is that a voltage Vout 130 is available at the output with the value shown in Equation 1 below (where T is the clock period): $V_{out} [nT] = C_{1} / C_{2} \times V_{in_1} [(n - \frac{1}{2}) T]$

As stated above, the extra voltage Vin_3 106 does not have to be the same as Vin_2 104, such that the circuit 100 would have a transfer function given by Equation 2 below: $V_{out} [nT] = (C_{1} / C_{2}) \times \{V_{in_1} [(n - \frac{1}{2}) T] - V_{in_3} [nT]\}$
Alternatively, a negative transfer function may be created as shown in FIG. 1B, which illustrates an inverting charge transfer stage 150 with no delay. The charge transfer stage 150 is analogous to the charge transfer stage 100 of FIG. 1A, but the clock phases are switched on the top plate of the capacitor 110. In this charge transfer stage 150, there is a direct feedthrough path between input and output on clock phase clk1 112. There is no delay in this circuit, with the output voltage given by Equation 3 below, assuming that Vin_2 104 is equivalent to Vin_3 106: $V_{out} [nT] = - C_{1} / C_{2} \times V_{in_1} [nT]$
The amplifier 108 in FIGs. 1A and 1B has the dual function of providing charge transport via its virtual earth node (i.e., active charge redistribution), and buffering so as to allow the following stage to read the output voltage without affecting the charge on the capacitors. However, finite amplifier DC gain and bandwidth cause incomplete charge redistribution, resulting in incomplete charge transfer from C₁ to C₂. This, together with inaccuracies in the matching of the capacitors C₁ and C₂, results in the creation of an inaccurate transfer function. Many applications, such as ADCs, accurate narrowband filters including FIR and IIR filters, etc. require very high accuracies in the transfer function, such as accuracies exceeding 0.1%. This kind of accuracy is virtually impossible using the standard circuits of FIGs. 1A and 1B in current Complementary Metal-Oxide Semiconductor (CMOS) processes. Often, the values of the capacitors are trimmed at manufacture, or some active calibration routines are executed, switching in and out small value capacitors in order to create an accurate transfer. Such schemes are expensive for high volume manufacture. The exemplary embodiment of the present invention solves these problems, and provides the requisite transfer function accuracy by design.
FIG. 2A illustrates a representative single-sampling circuit 200 implementing the principles of an embodiment of the present invention. The transfer function of circuit 200 is independent of capacitor mismatch, and can be realized in a standard digital CMOS process requiring no special options such as double poly or Metal-Insulator-Metal (MiM) capacitors, expensive trimming or calibrations, etc. It is based on delta-charge redistribution where the only charge transfer (other than to an external load capacitor) is to the parasitic capacitors at the amplifier inputs. No charge transfer takes place via the virtual earth node of the amplifier, making the circuit inherently accurate and second order independent of both the mismatch and non-linearity of the signal capacitors. The circuit is faster than prior art solutions due at least in part to the buffer-type configuration used. Further, it has better immunity to extraneous noise sources due to the fact that there is primarily voltage processing with no voltage-charge-voltage translations via the virtual earth node which is a well-known pick-up point for unwanted noise.
The representative single-sampling circuit 200 of FIG. 2A includes two opposite phased clock signals, namely clock phases clk1 202 and clk2 204. The analog sampled data input signals are shown as input signals Vin_1 206 and Vin_2 208, and may be either direct current (DC) or time varying signals. The signals Vin_4 210 and Vin_5 212 may be either DC or time-varying signals. The signal Vin_3 214 may be used, for example, as a variable DC shift in order to level shift the output signal Vout 216.
In operation, the input signal Vin_1 is sampled onto capacitance C ₁ 218 with respect to the reference voltage Vin_5 212 on clock phase clk1 202 by closing switches 220 and 222. During clock phase clk1 of the illustrated embodiment, switches 224 and 226 are also closed to sample the input signal Vin_2 208 onto capacitance C ₂ 228. In one embodiment of the invention, bottom plate sampling is used, where the input signals Vin_1 206 and Vin_2 208 are sampled on to the bottom plate of capacitances C₁ 218 and C ₂ 228 respectively. The top plates of capacitances C₁ 218 and C ₂ 228 are coupled to reference voltages Vin_5 212 and Vin_4 210 respectively during the clk1 202 phase.
On the next clock phase, clk2 204, C ₁ 218 is coupled across the amplifier 230 due to switches 232 and 234 closing, and switches 220 and 222 opening. Thus, the top plate of capacitance C ₁ 218 is coupled to the negative input 236 of the amplifier 230, and the bottom plate of capacitance C ₁ 218 is coupled to the output Vout 216 of the amplifier 230. In one embodiment of the invention, capacitance C ₂ 228 may be coupled at its bottom plate to Vin_3 214 by closing switch 238 on the clk2 204 clock phase. Further, the top plate of capacitance C ₂ 228 may be coupled to the positive input terminal 240 of the amplifier 230 on clk2 204 by closing switch 242. In this manner, the voltage Vin_3 214 is coupled to the positive terminal 240 of the amplifier 230 through the capacitor C ₂ 228, in order to provide voltage level shifting at the output Vout 216.
The transfer function for the single-sampling circuit 200 realization depicted in FIG. 2A can be determined using voltage superposition, resulting in the transfer function shown in Equation 4A: $V_{out} [nT] = V_{in_3} (nT) - (V_{in_2} [(n - \frac{1}{2}) T] - V_{in_4} [(n - \frac{1}{2}) T]) + (V_{in_1} [(n - \frac{1}{2}) T] - V_{in_5} [(n - \frac{1}{2}) T])$

or alternatively written in Equation 4B: $V_{out} [nT] = V_{in_1} [(n - \frac{1}{2}) T] - V_{in_2} [(n - \frac{1}{2}) T] + V_{in_3} [nT] + V_{in_4} [(n - \frac{1}{2}) T] - V_{in_5} [(n - \frac{1}{2}) T]$
Typically, but not necessarily, the analog sampled data input signals Vin_1 and Vin_2 are sampled with respect to AC ground set at a reference voltage Vref. With this AC ground 252 shown in FIG. 2B, and all signals referenced to AC ground, the relationship between Vin_5 212 and Vin_4 210 of FIG. 2A becomes that shown in Equation 5 below: $V_{in_4} [(n - \frac{1}{2}) T] = V_{in_5} [(n - \frac{1}{2}) T] = 0$

which in turn provides the simplified transfer function shown in Equation 6 below: $V_{out} [nT] = V_{in_1} [(n - \frac{1}{2}) T] - V_{in_2} [(n - \frac{1}{2}) T] + V_{in_3} [nT]$
As can be seen, Equations 4A, 4B, and 6 are independent of the capacitances C₁ and C₂, illustrating that the circuits 200, 250 can provide a summing function independent of capacitor mismatch that is inherently exhibited in prior art solutions. No charge transfer takes place via the virtual earth node of the amplifier, making the design inherently accurate and second order independent of both the mismatch and non-linearity of the signal capacitors. Further, because the circuit configuration primarily utilizes voltage processing with no voltage-to-charge and charge-to-voltage translations via a virtual earth node, the circuit configuration exhibits much better noise immunity than prior art solutions. This makes the circuit configuration suitable for use in standard digital CMOS processes that are uncharacterized for analog performance and have no special analog options.
Due to the accurate transfer function created by the circuit configuration of an embodiment of the present invention, it can be adapted to a double-sampling version that is free of the typical, inherent problems of double-sampling switched capacitor circuits that arise from mismatch of capacitors. An example of such a double-sampling circuit is shown in FIG. 3.
The representative double-sampling circuit 300 of FIG. 3 again includes two opposite phased clock signals, clkl and clk2. The analog sampled data input signals are shown as input signals Vin_1 302 and Vin_2 304, and the signal Vin_3 306 may again be used as a variable DC shift in order to level shift the output signal Vout 308. In this example, the data input signals Vin_1 302 and Vin_2 304 are sampled with respect to an AC ground.
In operation, the input signals Vin_1 302 and Vin_2 304 are sampled onto capacitances C₂ 310 and C ₄ 312 respectively on clock phase clk1 by closing the appropriate switches 314, 316, 318, and 320. The top plates of capacitances C₂ 310 and C ₄ 312 are coupled to ground during the clk1 phase. On the next clock phase, clk2, C ₂ 310 is coupled across the amplifier 322 due to switches 324, 326 closing, and switches 314, 316 opening. Thus, the top plate of capacitance C ₂ 310 is coupled to the negative input 328 of the amplifier 322, and the bottom plate of capacitance C ₂ 310 is coupled to the output Vout 308 of the amplifier 322. In one embodiment of the invention, capacitance C ₄ 312 may be coupled at its bottom plate to Vin_3 306 by closing switch 330 on the clk2 clock phase. Further, the top plate of capacitance C ₄ 312 may be coupled to the positive input terminal 332 of the amplifier 322 on clk2 by closing switch 334. In this manner, the voltage Vin_3 306 is coupled to the positive terminal 332 of the amplifier 322 through the capacitor C ₄ 312, in order to provide voltage level shifting at the output Vout 308. As can be seen, the operation is analogous to that described in connection with FIGs. 2B.
The embodiment of FIG. 3 allows for the sampling of the inputs Vin_1 302 and Vin_2 304 on a first clock phase (e.g., clk1) and delivery of the output on a subsequent clock phase (e.g., clk2) as described above. Further, in accordance with the double-sampled embodiment shown in FIG. 3, inputs Vin_1 302 and Vin_2 304 can also be sampled and delivered on alternate clock phases through the use of an additional set of capacitors, whereby the input signals are sampled on the second clock phase (e.g., clk2) and the output delivered on the first clock phase (e.g., clk1). By doubling the capacitors and making use of the alternate clock phases in this way, it is possible to double the processing rate of the circuit for the same analog power dissipation.
More particularly, in the double-sampled embodiment of FIG. 3, C ₁ 336 and C ₃ 338 perform similar functions to those described in connection with C ₂ 310 and C ₄ 312, but perform these functions on opposite phased clock signals. Thus, input signal Vin_1 302 is sampled onto capacitance C ₁ 336 with respect to ground when switches 338 and 340 close, which will occur on the opposite clock phase as when C ₂ 310 is sampled. On the same clock phase that Vin_2 302 is sampled onto C ₁ 336, Vin_2 302 is also sampled onto capacitance C ₃ 338 due to switches 342 and 344 being closed. In this manner, Vin_2 302 is sampled onto capacitors C ₁ 336 and C ₃ 338 on the clock phase opposite to that in which Vin_2 302 is sampled onto C ₂ 310 and C ₄ 312.
On the following clock phase, C ₁ 336 is connected across the amplifier 322 due to switches 346 and 348 closing. Thus, the top plate of capacitance C ₁ 336 is coupled to the negative input 328 of the amplifier 322, and the bottom plate of capacitance C ₁ 336 is coupled to the output Vout 308 of the amplifier 322. On this same clock phase, the bottom plate of capacitance C ₃ 338 is coupled at its bottom plate to Vin_3 306 by closing switch 350. Further, the top plate of capacitance C ₃ 338 may be coupled to the positive input terminal 332 of the amplifier 322 on this clock phase by closing switch 352. In this manner, the voltage Vin_3 306 is coupled to the positive terminal 332 of the amplifier 322 through the capacitor C ₃ 338, in order to provide voltage level shifting at the output Vout 308.
Using the additional circuitry in such a double-sampled embodiment, the inputs Vin_1 302 and Vin_2 304 can be processed at double the rate of a single-sampling implementation, thereby doubling the processing speed of the circuit (assuming the same amplifier hardware is being used).
The example circuit 300 of FIG. 3 has a transfer function shown by Equation 7 below: $V_{out} [nT] = V_{in_1} [(n - 1) T] - V_{in_2} [(n - 1) T] + V_{in_3} [nT]$
The double-sampling circuit that can operate independent of capacitor matching has a number of advantages compared to the single-sampling version. For example, the double-sampling circuit can operate at double the speed of the single-sampling circuit for the same frequency of non-overlapping clocks (e.g., clk1 and clk2), since the input can be processed on both clk1 and clk2 phases. Even with this increased speed of operation, the double-sampling circuit consumes the same analog power as the single-sampling circuit. Further, the double-sampling circuit offers a full period delay, which is a requirement for any sampled data system operating at a sampling rate of 1/T. Furthermore, a full period (T) hold signal is possible when used as an interface from analog sampled data to continuous time data. Since the single-sampling circuit only has a delay of T/2, an extra delay of T/2 must be found in order that all analog sampled data samples are available at time intervals of T only.
The representative circuits described in connection with FIGs. 2A, 2B, and 3 present balanced impedances from the capacitors and accompanying switches at the two sensitive input terminals of the single-ended amplifier. This ensures accurate settling between clock edges. As previously noted, the transfer functions associated with these circuits do not contain any capacitor ratios so that the processing of the signals occurs independent of the mismatch of the two signal capacitors with nominal value C. Only errors of a second order nature occur due to the presence of parasitic capacitances at the input nodes of the amplifier. Any imbalances either between the capacitors of nominal value C, or the input parasitic capacitors, will give rise to an error that is second order with respect to the absolute imbalance itself.
In accordance with one embodiment of the present invention, various combinations of clock phase control may be utilized. In the previously described examples, two clock phases were described (e.g., clk1 and clk2). However, any number of desired clock phases may be used. For example, using three clock phases clk1, clk2, and clk3, a first of the voltage signals may be added at one clock delay, where another voltage signal may be added at, for example, two clock delays. This provides additional variability and flexibility in the choice of delays. This may be beneficial for circuit applications benefiting from extended and/or variable clock delays. For example, delays may be required in the case of filter design, such as with Finite and Infinite Impulse Response (FIR/IIR) filters. More particularly, such filters may be of an nth order where a plurality of previous inputs (in the case of non-recursive filters) and/or a plurality of previous outputs (in the case of recursive filters) are utilized to perform the desired filtering function. Flexibility in delay lines in the switched capacitor summer/level shifter in accordance with an embodiment of the present invention is highly advantageous. Therefore, where the transfer function requires the addition of signals separated by one or more delays, the addition of additional clock phases in accordance with an embodiment of the present invention provides this ability.
FIG. 4 illustrates an example of an N-path sum-delay-shift circuit 400 in accordance with one embodiment of the present invention. Thus, where the additional clock phase was used to facilitate double-sampling in the embodiment illustrated in FIG. 3, additional clock phases may be used for circuits requiring delays. The circuit of FIG. 4 operates similarly to the circuit described in connection with FIG. 3, however additional switched capacitor circuits are provided, as well as N clock phases. For example, N switched capacitor circuits 402, 404, 406 are coupled to the negative input 408 of the amplifier 410, and N switched capacitor circuits 412, 414, 416 are coupled to the positive input 418 of the amplifier 410.
The analog sampled data input signals are shown as input signals Vin_1 420 and Vin_2 422, and the signal Vin_3 424 may again be used as a variable DC shift in order to level shift the output signal Vout 426. In this example, the data input signals Vin_1 420 and Vin_2 422 are sampled with respect to an AC ground. In operation, the input signals Vin_1 420 and Vin_2 422 are sampled onto capacitances C within their respective N switched capacitor circuit 402, 404, 406, 412, 414, 416. For example, sampling for first switched capacitor circuits 402, 412 occurs on clk1, sampling for N-1 switched capacitor circuits 404, 414 occurs on clkN-1, sampling for N switched capacitor circuits 406, 416 occurs on clkN, and so forth. On different clock phases, each of the switched capacitor circuits can then be coupled across the amplifier 426 to perform the summing/level shifting function previously described. In this manner, input signals may be added at any desired delay, thereby facilitating realization of a wide variety of different circuit implementations, such as, for example, FIR and IIR filter circuits.
FIG. 5 is a flow diagram illustrating a method for adding at least two input voltage signals in accordance with the principles of an embodiment of the present invention. A first input voltage signal is sampled 500 onto a first capacitor during a first clock phase. Analogously, a second input voltage signal is sampled 502 onto a second capacitor during the first clock phase. On the second clock phase, the first capacitor is switched 504 in order to connect to the negative input terminal of the amplifier, and the second capacitor is switched 506 to connect to the positive input terminal of the amplifier. Also during the second clock phase, the output voltage is fed back from the amplifier output to the negative input of the amplifier by way of the first capacitor, as shown at block 508. The sum of the first and second input voltage signals is output 510 from the amplifier in response to the feedback voltage, and in response to the first and second sampled input voltages, during the second clock phase.
The signal processing capability of the method and architecture in accordance with an embodiment of the present invention enables its use in a wide variety of applications where accurate addition and subtraction of analog sampled data signals can be performed independent of capacitor mismatch. The transfer function is also independent of non-linearity of the capacitors, since there is only voltage sampling and no charge transfer takes place from signal capacitor to signal capacitor. The only significant charge transfer (other than that to the load capacitance) is to the parasitic capacitors at the amplifier inputs, which is only a small fraction of the total charge held on the signal capacitors with nominal values C. This, however, does not affect the accuracy of the transfer function. This is referred to herein as delta-charge redistribution, since the only main charge transfer is that to charge parasitic capacitance.
The principles of an exemplary embodiment of the present invention may be used in a wide variety of applications, such as Finite and Infinite Impulse response Filters (FIR and IIR filters), N-path filters, delay lines, comb filters, integrators, differentiators, voltage multipliers to any level, accurate inverters, level shifters, voltage multipliers, single-to-differential and differential-to-single ended converters, etc. These functions can be realized with an order of magnitude improved accuracy, and at least twice the speed than previous circuits in standard CMOS processes (assuming the use of similar hardware components).
It should be noted that any known circuit components may be used to provide the operations in accordance with an exemplary embodiment of the present invention. For example, a capacitor may be used where capacitors are indicated, however groups of series and/or parallel capacitors may also be used. Further, other components exhibiting capacitive properties and capable of storing a charge thereon may be used. As another example, the switches employed may be any component capable of performing a switching function. For example, the principles of an exemplary embodiment of the present invention may be implemented using field-effect transistors (FETs) and variations such as metal-oxide-semiconductor field-effect transistor (MOSFETs), JFETs, VMOS, CMOS, etc. Other transistor technologies may also be employed, such as bipolar technologies. The switches may also be implemented using electrically-controlled mechanical switches and/or relays. Speed, efficiency, power consumption, and other factors will determine the type of switches to be employed, and in one particularly beneficial embodiment CMOS switches are implemented to provide the desired speed and power consumption characteristics. The amplifier components may be any of a wide variety of operational amplifiers facilitating single-ended operation.

Use of Switched Capacitor in ADC

Another exemplary embodiment of the present invention is directed to an analog-to-digital converter (ADC) for use in various ADC architectures, such as algorithmic and pipelined ADC architectures. The ADC circuit in accordance with the another exemplary embodiment of the present invention provides a very accurate manner of subtracting/level shifting, residue multiplication, and sample-and-hold (S&H) functions, all within a single clock cycle. In accordance with the invention, these functions are performed using a switched capacitor technique that is first order independent of capacitor matching. This enables its use in new digital technology processes, such as Complementary Metal-Oxide Semiconductor (CMOS) processes, that are uncharacterized for capacitor matching and analog performance.
In prior art ADC circuits such as 1.5-bit ADC stages, charge transfer occurs from one input capacitor to a second capacitor in the feedback look of an amplifier via the virtual earth node of the amplifier. In this manner, the input capacitor discharges to the feedback capacitor, giving rise to an output voltage that is proportional to the capacitor ratio (i.e., input capacitance/feedback capacitance). For example, a gain of "2" may be created by providing an input capacitor having a capacitance value twice that of the feedback capacitor.
The another exemplary embodiment of the present invention, on the other hand, adds capacitor voltages only, with the amplifier serving as a buffer. For example, in one particular embodiment of the invention utilizing 1.5-bit ADC stages, a signal voltage may be sampled onto two capacitors on one clock cycle. On a following clock cycle one of the capacitors is placed in the feedback loop of the amplifier, and the other capacitor is inverted and connected between the amplifier's negative input terminal and any one of a predetermined number of voltages used in the 1.5-bit stage (e.g., +Vref, 0, -Vref), giving rise to an effective doubling of the input sample voltage combined with subtraction of one of the predetermined voltages. The resulting voltage is held at an output on a subsequent clock cycle so that it can be, for example, sampled by a subsequent stage of a pipeline ADC, or sampled in once again by a subsequent set of capacitors in an algorithmic ADC. By summing only capacitor voltages and using the amplifier as a buffer, multiplication by two, for example, does not depend on the absolute values of the capacitors, giving rise to a very robust solution suitable for embedding in digital environments. Chip area and power consumption are consequently reduced, thereby providing enhanced power and area figures-of-merit (FOMs) compared to current ADC designs.
A number of ADC architectures currently exist, and design choices are often made based on parameters including speed, power consumption, required real estate, complexity, etc. For example, a straightforward and fast ADC architecture is the flash architecture, where a number of parallel comparator circuits compare sampled/held analog signals with different reference levels. However, because each reference level should be no further than one least significant bit (LSB) apart, a large number of comparators may be required for such an architecture. For example, an N-bit ADC requires 2ⁿ comparators. Where the full scale input is a relatively small voltage, the LSB size will be relatively small, and the offset of the comparator needs to be very small which may be difficult to achieve with technologies such as CMOS, and special circuit techniques may be required. Flash ADCs are therefore generally limited to smaller resolution converters, such as 8-bit or less resolution.
Two-step flash architectures arose to address some of the problems of flash ADCs, where the two-step flash ADCs first performs a course quantization, the held signal is the subtracted from an analog version of the course quantization, and the residue is then more finely quantized. This significantly reduces the number of comparators required in a standard flash ADC architecture, but additional clock cycles are required to process the signal due to the extra stage. Another enhancement arose, where interstage gain was used to tolerate larger comparator offset for second stage comparators, which ultimately led to the pipelined ADC architecture employing multiple stages. The sampled input at each stage of a pipelined ADC architecture is converted to a particular resolution of the stage, such as n bits.
An ADC architecture resolving 1 bit per stage with one-half bit overlap is referred to as a "1.5-bit" ADC architecture. In order to facilitate an understanding of the invention, various embodiments of the description provided herein are described in terms of such a 1.5-bit architecture. Examples of such architectures are set forth below to provide an appropriate, representative context in which the principles of the another exemplary embodiment of the present invention may be described. However, it will be apparent to those skilled in the art from the description provided herein that the another exemplary embodiment of the present invention is scalable and equally applicable to other analogous ADC architectures.
FIG. 6 is a block diagram illustrating a typical 1.5-bit ADC stage 1100. The circuit 1100 includes a sample-and-hold (S&H) circuit 1102, a 1.5-bit sub-ADC 1104, a 1.5-bit sub-DAC 1106, a subtractor 1108, and a multiplier 1110. Such an architecture is used in pipelined or algorithmic ACS to provide maximum bandwidth and low sensitivity to component mismatches. This is because each stage 1100 requires only two comparators (not shown) having an accuracy of +/-(Vref/4) for the 1.5-bit sub-ADC 1104, and one multiplier (e.g., amplifier) 1110. The associated comparator and amplifier offset can easily be corrected using standard digital error correction (DEC) techniques.
In the circuit of FIG. 6, the input voltage "In" is sampled by the sample-and-hold 1102 and resolved into a 1.5-bit digital code in a course analog-to-digital sub-converter (sub-ADC) 1104. With a 1.5-bit sub-ADC, only three codes are possible, such as 00, 01, 10. The resulting 1.5-bit code 1112 is outputted to a digital error correction circuit. The code is also converted, via a digital-to-analog sub-converter (sub-DAC) 1106, back into a course analog signal with one of three predetermined analog values, such as -Vref/2, 0, +Vref/2. The result is subtracted from the sampled-and-held analog input signal "In" via subtractor 1108. The resulting analog "residue" is gained up by a factor of two using the multiplier 1110 to become the input voltage for the successive conversion.
As can be seen, the analog equivalent of the sub-ADC 1104 output plus the output residue (prior to multiplication) is equal to the analog input voltage. Thus, any perturbation in the residue due to non-idealities can introduce differential nonlinearity (DNL) errors. Effectively, all errors in the gained up analog residue after the first conversion should be less than 1 LSB of the remaining resolution of the ADC (or less than 2 LSBs of the total resolution at N-bit level).
An N-bit algorithmic ADC 200 shown in FIG. 2 is formed by sampling the input signal on the first clock cycle, and sampling the output of the 1.5-bit stage 202 on the next N-1 cycles. The 1.5-bit data 204 from each rotation are added up with 1-bit overlap in the DEC 1206 circuit such that the least significant bit (LSB) from one rotation is added to the most significant bit (MSB) from the next rotation. Each rotation of the ADC resolves one effective bit from the MSB level down to the LSB-1 level. The final LSB bit is often resolved using a simple 1-bit flash 1208, e.g., a comparator with its threshold set to 0V. This bit 1210 is not added, but rather is concatenated to the parallel data 1212 of the DEC 1206.
Alternatively, a series of such stages may be used to create a pipelined ADC, such as the representative pipelined ADC 300 shown in FIG. 8. The pipelined ADC 1300 includes a series of N-2 stages 1300, 1302,...1304, such as those described in connection with FIG. 6, as well as an Nth stage 1306. Stages 1300, 1302,...1304 may be used to resolve N-2 bits, with the final stage 1308 being a 2-bit flash to absolutely resolve the final two bits. The 1.5- bit data 1310, 1312,...1314 and 2-bit data 1316 is provided to the DEC 1318 to create the N-bit parallel output data 1320. The sample rate of the pipeline is approximately N times faster than that of the algorithmic architecture, depending ultimately on what resolution flash converter is used for the final stage 1308.
An example of a residue transfer characteristic of the complete 1.5-bit ADC stage is shown in FIG. 9. In this example, it is assumed that the full signal range is between -Vref and +Vref. The transfer function is defined by Equation 1 below: $V_{out} = 2 \times V_{in} + D \times V_{ref}$

where D can take on any one of the values {-1, 0, +1} depending on whether the analog input voltage falls within corresponding ranges of $\{(+ V_{ref} \to + \frac{V_{ref}}{4}), (+ \frac{V_{ref}}{4} \to - \frac{V_{ref}}{4}), (- \frac{V_{ref}}{4} \to - V_{ref})\} .$

Vout of Equation 1 may either be resampled into the algorithmic ADC on a subsequent rotation, or may become the input voltage for a subsequent stage of a pipelined ADC.
In an actual implementation, the transfer characteristic is affected by non-idealities in the analog hardware. As previously indicated, offsets in the amplifier and comparators can be corrected by the DEC. The two remaining sources of error in an actual implementation include inaccuracies in the creation of the multiply-by-two (MX2) gain function (including subtraction of sub-DAC levels), and variations in the reference levels. Variations in reference levels are issues only in pipelined ADCs, in which separate hardware in each 1.5-bit stage samples +Vref and -Vref, where uncorrelated errors can occur from stage to stage. Static errors in the reference levels are not an issue for algorithmic ADCs, since each rotation of the ADC samples the same references in the same way with the same hardware. The absolute accuracy of the reference levels is not important in a differential implementation, as long as the reference levels are stable, within the usable dynamic range of the active circuitry, and do not vary from conversion to conversion. At most, the gain transfer is affected without affecting DNL/INL. Thus, the only two remaining sources of error that limit accuracy of the complete ADC are the accuracy of the multiply (MX2) function and the DAC levels (sub-DAC). In conventional implementations, this error is predominantly caused by capacitor mismatch.
The combined accuracy of the MX2 and sub-DAC functions must be better than one LSB of the remaining resolution of the ADC in order to guarantee no missing codes. The first stage of the pipeline has the most stringent requirement here, as the MX2/sub-DAC functions for an N-bit ADC must be accurate to at least N-1 bits, which is the number of bits yet to be resolved after the first stage. The required resolution of an N-bit algorithmic ADC is commensurate with the required resolution of the first stage of a pipeline, i.e., N-1 bits. For a robust design - and to account for other sources of error, most notably noise - the accuracy of the MX2 amplifier with sub-DAC, after including all possible contributions of error, should be designed to be at least 0.5 LSBs of the remaining resolution, i.e., N bits accuracy.
The effect of a gain error in the first stage of a pipeline, or the first rotation of an algorithmic, is illustrated in FIG. 9. The comparator levels of the two comparators of the 1.5-bit stage are set to -Vref/4 and +Vref/4 respectively. It can be seen that when the gain of the stage is too high, over-ranging can occur where the slope 400 of the MX2 is greater than the ideal slope 402 of the MX2. This causes the input signal to the next stage to go beyond the maximum allowable range {+Vref and -Vref} for conversion.
The effects on the complete transfer function of the ADC are shown in FIGs. 10A, 10B, and 10C for gain errors and sub-DAC errors in the first stage of a pipeline or in the algorithmic ADC. FIG. 10A shows the effect of a gain error greater than two in the MX2 which produces non-monotonicity and the potential for missing codes. Where the ideal gain is equal to two as shown on dashed line 1500, non-ideal gain error greater than two as shown on lines 1502A, 1502B, 1502C can result in missing digital output codes. Similarly, FIG. 10B shows the effect of a gain error less than two in the MX2 which produces missing codes. Where the ideal gain is again equal to two as shown on dashed line 500 of FIG. 10B, non-ideal gain error less than two as shown on lines 1504A, 1504B, 1504C can result in missing digital output codes. Further, FIG. 10C shows the effect of sub-DAC errors in the first stage of the ADC on the total transfer function. The ideal transfer function is shown on dashed line 1506, and various representative DAC level shift errors are shown on lines 1508A, 1508B, and 1508C, which will result in missing codes. These errors are caused by capacitor mismatch and non-linearity. In practice, all these errors will propagate from the MSB to the LSB level, eventually (and undesirably) producing a jagged transfer function for the complete ADC.
Current 1.5-bit designs exhibit characteristics that are responsible for much of this gain error. A switched capacitor implementation of a 1.5-bit stage for a single-ended application is shown in FIG. 11A, a portion of which includes a prior art switched capacitor (SC) circuit 1600. The switched capacitor circuit 1600 includes an amplifier 1602, two nominally equal capacitors C_f 1604 and C _s 1606, and several switches 1608, 1610, 1612, 1614, 1616, 1618, 1620. Two opposite phased clock signals, clk1 and clk2, are non-overlapping. The switched capacitor circuit 1600 performs the level shifting, residue multiplication by two (MX2), and sample-and-hold buffering as is known in the art. The input signal Vin is applied to the sub-ADC including comparators 1622, 1624, with voltage thresholds set at +Vref/4 and -Vref/4 respectively. Concurrently, the input signal Vin is sampled onto C _s 1606 and C _f 1604. At the end of the first clock phase, clk1, Vin is completely sampled onto C _s 1606 and C _f 1604, while the output of the sub-ADC 1622, 1624 is latched and held by latches associated with the latches and clock generator 1626. During clk2, C _f 1604 is switched via switch 608 and placed across the amplifier 1602, completing its negative feedback loop 1628. At the same time, one of the input switches 1614, 1616, 1618 connected to C _s 1606 is closed by the sub-DAC using only one of the clock signals top, mid, bot. In this manner, the analog residue voltage is produced at the output 1630, such that Vout is provided as shown in Equation 2: $V_{out} = (1 + \frac{C_{s}}{C_{f}}) \times V_{in} + D \times V_{ref}$

where:

V_in D Bot Mid Top

V_in > V_ref /4 -C_s l C_f 1 0 0

-V_ref / 4≤V_in ≤+V_ref /4 0 0 1 0

V_in <-V_ref /4 +C_s l C_f 0 0 1

By choosing capacitors C_s 1606 and C _f 1604 to have the same value, Equation 2 is made to correspond to the ideal transfer function of Equation 1 of a 1.5-bit stage. The reference levels can be generated accurately and is generally not a limitation on the realization of a high resolution ADC (e.g., 12-bit level). The single factor that ultimately determines the maximum resolution of the ADC is the capacitor mismatch. This mismatch has two effects on the performance of current state-of-the-art designs, including 1) it affects the accuracy of the MX2 function, and 2) it affects the accuracy of the sub-DAC through the accuracy with which the DAC levels {-Vref, 0, +Vref} can be generated.
In order to achieve 10-bit performance, a matching of the order of 0.1% is needed between C _s 1606 and C _f 1604. This is currently not possible to achieve in standard CMOS processes without using special capacitor options, such as the use of poly-poly capacitors. Even using such specialized capacitors, very large values for the capacitors are needed (i.e., on the order of many picofarads), to guarantee 0.1% matching across all process corners. Such large value capacitors would be responsible for creating an ADC that requires a great deal of real estate and exhibits significant power consumption. For a pipelined ADC with N-1 stages, such an approach is unacceptable. Alternatively, calibration routines are sometimes used to either trim the values of the capacitors or to digitally calibrate out the gain error in a post-processing routine. Such correction/calibration routines are needed to achieve a resolution better than ten bits due to the limitations of the processing technology on prior art ADC circuit architectures. Complicated calibration routines exist which add area, power consumption, and latency to the conversion. Typically, many (e.g., up to seven) clock cycles per bit are needed to calibrate away capacitor mismatch errors. Still a further point of issue can be capacitor linearity: any non-linearity in C _s 1606 and C _f 1604 of FIG. 11A will cause non-linearity in the MX2 amplifier 602 and cause differential nonlinearity (DNL) and integral non-linearity (INL) errors.
For well known reasons of noise immunity and increased dynamic range, conventional ADC solutions may be realized using a fully differential amplifier. FIG. 11B illustrates a differential switched capacitor implementation of a 1.5-bit ADC stage. The conventional switched capacitor implementation includes a differential amplifier 1650, as well as a differential input signal Vin 1652 and a differential output signal 1654. In such a conventional differential amplifier implementation, the differential amplifier 1650 is used, charge is transferred between capacitors, and a capacitor ratio is still used to establish the gain (multiplication by 2, for example). As previously stated in the single-ended example, all of the charge on one capacitor is transferred to the other capacitor, and any error in the charge transfer results in errors in the total transfer function. The capacitance mismatch and non-linearity problems may be exacerbated where double-sampling techniques are used. A double-sampling ADC stage may be realized that samples the inputs on a first clock phase clk1 and delivers its output on a second clock phase clk2, and can also sample the inputs on clk2 and deliver its output on clk1 through the use of an additional set of capacitors. By doubling the capacitors in this way, it is possible to double the conversion rate of the ADC for the same analog power dissipation. However, in current state-of-the-art designs, double-sampling introduces unwanted characteristics around half the sampling frequency due to the extra mismatch that occurs between both of the double-sampling channels from mutual capacitor mismatch on clk1 and clk2. To reduce such a mismatch, the capacitors would need to be even larger than in the single-sampling version, meaning more power and area consumption which is undesirable. Mainly for these reasons, double-sampling is often not used in current ADC implementations.
The another exemplary embodiment of the present invention addresses a number of shortcomings of prior art ADC technologies, including the aforementioned error situations exhibited by current ADC technologies. The another exemplary embodiment of the present invention significantly reduces errors in the MX2 (or other multiplier) function, as well as errors in the generation of DAC levels, that are present in conventional ADC technologies. The another exemplary embodiment of the present invention is first order independent of capacitor matching, enabling accurate, relatively high bit-width ADCs in CMOS (and other technologies) that are otherwise uncharacterized for matching of analog components. Further, the apparatus and methodology in accordance with the another exemplary embodiment of the present invention allows for use of simple metal layer capacitors as the signal capacitors, while still achieving accurate, high bit-width performance. The another exemplary embodiment of the present invention is also substantially faster than prior art ADCs employing analogous hardware. Thus, with use of similar amplifiers and capacitors in both prior art systems and in the another exemplary embodiment of the present invention, the another exemplary embodiment is substantially faster than the prior art systems by virtue of the fact that the feedback factor (and, consequently the gainbandwidth) for the amplifiers is substantially larger.
Referring to FIG. 12A, a block diagram of a representative 1.5-bit ADC stage 1700 corresponding to a first half of a differential implementation is illustrated. FIG. 12B illustrates a second half of the representative differential implementation. Two opposite phased clock signals are used, namely clock phases clk1 and clk2. First considering the top half of the differential implementation shown in FIG. 12A, In_p 1702 of the differential input signal is sampled onto capacitance C _1a 1704 with respect to ground on clock phase clk1 by closing switches 1706 and 1708. During clock phase clk1 of the illustrated embodiment, a number of other different switches are closed, including switches 1714 and 1716. Thus, In_n 1720 of the differential input signal is also sampled onto capacitance C _3a 1722 due to switches 1714 and 1716 being closed during clock phase clk1. In one embodiment of the invention, bottom plate sampling is used, where the input signals In_p 1702 and In_n 1720 are sampled on to the bottom plate of the capacitances C_1a 1704 and C _3a 1722 respectively. The top plates of capacitances C_1a 1704 and C _3a 1722 are coupled to ground during the clk1 phase.
On the next clock phase, clk2, C _1a 1704 is connected across the amplifier 1724 due to switches 1726 and 1728 closing, and switches 1706 and 1708 opening. Thus, the top plate of capacitance C _1a 1704 is coupled to the negative input 1730 of the amplifier 1724, and the bottom plate of capacitance C _1a 1704 is coupled to the output (Out_p 1732) of the amplifier 1724. Assertion of clock phase clk2 also causes capacitance C _3a 1722 to have its bottom plate connected to any one of the voltages +Vref, 0, -Vref. Such voltages are controllably selected by sub-DAC control signals labeled as the top (top_a), middle (mid_a), or bottom (bot_a). The top plate of capacitance C _3a 1722 is then coupled to the positive input terminal 734 of the amplifier 1724 on clk2. In this manner, one of the output control signals of the sub-DAC (i.e., bot_a, mid_a, top_a) selects a corresponding +Vref, 0, or -Vref voltage, which in turn serves as a reference voltage to the capacitance C _3a 1722 during the second clock phase clk2. The net consequence of these actions is that after one clock period delay, In_p is added to an inverted version of In_n, while at the same time it is level shifted by either +Vref, 0, -Vref. This is accomplished without ever creating a transfer of charge between capacitors.
In a double-sampled embodiment, C _2a 1736 and C _4a 1738 perform similar functions to those described in connection with C _1a 1704 and C _3a 1722, but with opposite phased clock signals. More particularly, In_p 1702 of the differential input signal is sampled onto capacitance C _2a 1736 with respect to ground on clock phase clk2 by closing switches 1740 and 1742. During clock phase clk2 of the illustrated embodiment, In_n 720 of the differential input signal is also sampled onto capacitance C _4a 1738 due to switches 1744 and 1746 being closed during clock phase clk2. In one embodiment of the invention, bottom plate sampling is used, where the input signals In_p 1702 and In_n 1720 are sampled on to the bottom plate of the capacitances C _2a 1736,and C _4a 1738 respectively. The top plates of capacitances C_2a 1736 and C _4a 1738 are coupled to ground during the clk2 phase.
On the next clock phase, clk1, C _2a 1736 is connected across the amplifier 1724 due to switches 1748 and 1750 closing, and switches 1740 and 1742 opening. Thus, the top plate of capacitance C _2a 1736 is coupled to the negative input 1730 of the amplifier 1724, and the bottom plate of capacitance C _2a 1736 is coupled to the output (Out_p 1732) of the amplifier 1724. Assertion of clock phase clk1 also causes capacitance C _4a 1738 to have its bottom plate connected to any one of the voltages +Vref, 0, -Vref, in response to the appropriate control output from the sub-DAC. Such sub-DAC control signals are labeled as the top (top_a), middle (mid_a), or bottom (bot_a). The top plate of capacitance C _4a 1738 is then coupled to the positive input terminal 1734 of the amplifier 1724. In this manner, one of the output control signals of the sub-DAC (i.e., bot_a, mid_a, top_a) selects the corresponding voltage +Vref, 0, or -Vref, which in turn serves as a reference voltage to the capacitance C _4a 1738 during the first clock phase clk1.
Using the additional circuitry in such a double-sampled embodiment, the inputs In_p 1702 and In_n 1720 can be processed at double the rate of a single-sampling implementation, thereby doubling the conversion speed of the ADC using such circuit stages.
FIG. 12B illustrates a representative 1.5-bit ADC stage 1760 corresponding to the second half of the differential implementation described in connection with FIG. 12A. The circuit stage 760 operates in an analogous manner as that described in connection with FIG. 12A, using another set of capacitances C_1b 1762 and C _3b 1764, as well as capacitances C_2b 1766 and C _4b 1768 for the double-sampling implementation. Further, because the circuit 1760 forms a second half of a differential implementation, the input signals In_p 1702 and In_n 1720 are reversed such that the input signal In_n 1720 is ultimately coupled to the negative input 1730 of the amplifier 1724, and the input signal In_p 1702 is ultimately coupled to the positive input 1734 of the amplifier 1724. The amplifier 1724 outputs the other differential signal, shown as output signal Out_n 1770 in FIG. 12B. Otherwise, the operation is analogous to that described in connection with FIG. 12A, ultimately producing differential output signals Out_p 1732 and Out_n 1770.
FIG. 13 illustrates an implementation of the differential ADC stage 1800 described in connection with FIGs. 12A and 12B. The illustrated embodiment represents an implementation of the differential ADC stage in the context of an algorithmic ADC. In this embodiment, circuit stages 1802 and 1804 correspond respectively to the circuits 1700 and 1760 described in connection with FIGs. 12A and 12B. In this embodiment, all voltage levels are shifted by a common mode voltage, refcm, such that the signal range is between refn and refp. Therefore, a single supply voltage may be used (i.e., 0 to Vdd). The illustrated ADC stage 1800 is applied in an algorithmic ADC as previously described in connection with FIG. 7, with non-overlapping clocks ADC_clk and ADC_clk_n such that ADC_clk is high for one clock period and ADC_clk_n is high for the remaining N-2 clock periods as explained in connection with FIG. 7. The differential analog input signal (i.e., In_p 1806; In_n 1808) is sampled at the start of each conversion, using ADC_clk, while the gating with ADC_clk_n ensures that the differential output signal (i.e., Out_p 1810; Out_n 1812) is sampled for the remaining N-2 clock periods. A final instantaneous decision can be made with a 1-bit flash to determine the last bit, giving a total of N-1 clock cycles to resolve N bits.
Matching of the absolute values of reference voltages -Vref/4 and +Vref/4, and consequently refp-refcm and refcm-refn, is not needed in differential algorithmic/pipelined ADCs. Furthermore, refcm may be nominally set halfway between refp and refn, but its exact position is not critical.
FIG. 14 illustrates a representative waveform diagram corresponding to an algorithmic ADC such as described in connection with FIG. 13. A master clock 1900 is provided, where clk1 and clk2 are non-overlapping phases of the clock. For this algorithmic ADC, clocks ADC_clk 1906 and ADC_clk_n 1908 are non-overlapping, such that ADC_clk 1906 is high for one clock period and ADC_clk_n 1908 is high for the remaining N-2 clock periods. The data ready signal (DRDY) 1914 is asserted when the ADC_clk 1906 is asserted, thereby allowing the parallel data 1912 to begin accumulating the associated digital data.
Non-overlapping clocks with early turn-off times, i.e., clk1_e 1914 and clk2_e 1916, may be applied in the implementation of the algorithmic ADC. When the capacitors are sampling the input signals or references, the input switches switching with respect to refcm switch off early in one embodiment of the invention. On the other hand, switches connecting the capacitors to the inputs of the amplifiers should switch off late in accordance with this embodiment of the invention. In this manner, when in cyclic mode, the outputs of the amplifiers can be sampled by the oppositely-phased capacitor networks before any switching occurs around the amplifiers, ensuring clean sampling.
An example of an ADC stage corresponding to a first half of a differential, algorithmic ADC implementation, such as that described in connection with FIG. 13, is illustrated in FIGs. 15A and 15B. The example of FIGs. 15A and 15B is provided as a representative implementation, and those skilled in the art will appreciate that many variations to such an implementation are possible.
FIG. 15A corresponds to the circuitry coupled to the negative input of an amplifier, such as the switches and capacitors coupled to the negative input of the amplifier shown in block 1802 of FIG. 13. As was described in connection with FIGs. 13 and 14, two opposite phased clock signals are used, namely clock phases clkl and clk2. The signal In_p 2000 of the differential input signal is sampled onto capacitance C _1a 2002 with respect to a reference voltage such as refcm, on clock phase clk1 2004. The signal 2000 is sampled onto C _1a 2002 via switch circuit 2006. The ADC_clk 2008 enables the clk1 2004 to be passed for one clock period, via the NAND gate 2010 and associated inverters 2012, 2014 to the CMOS switch 2016. Thus, when the ADC_clk 2008 and clk1 2004 are asserted, the switch 2016 samples the In_p 2000 signal onto C _1a 2002 with respect to the reference voltage through CMOS switch 2018 when switched by the early turn-off clock clk1_e 2020.
On the next clock phase, clk2 2022, C _1a 2002 is coupled to the negative terminal 2024 of the amplifier via switch circuit 2026. As previously indicated, the ADC_clk_n 2028 is high for the remaining N-2 clock periods, thereby gating the appropriate clock phase to the CMOS switch 2030 via the logic components 2032, 2034, 2036, 2038. The output signal Out_p 2040, from the output of the amplifier (not shown), is thus fed back to switch 2030 and coupled to the bottom plate of the capacitor C_1a on clk2 2022.
In a double-sampled embodiment, switch circuits 2042 and 2044 are also provided. These switch circuits 2042, 2044 operate analogously to switch circuits 2006 and 2026 respectively, with the clk1 2004 and clk2 2022 signals reversed with respect to switch circuits 2006 and 2026. In the double-sampled embodiment, In_p 2000 is sampled onto capacitance C _2a 2046, and on the next clock phase C _2a 2046 is coupled to the negative terminal 2024 of the amplifier via switch circuit 2048.
FIG. 15B corresponds to a portion of the circuitry coupled to the positive input of an amplifier, such as the switches and capacitors coupled to the positive input of the amplifier shown in block 802 of FIG. 13. Because the circuits associated with each of the capacitors C_3a and C_4a in a double-sampling implementation of FIG. 13 are analogous, only the circuitry of one such circuit is described in FIG. 15B.
In_n 2050 is sampled onto capacitance C _4a 2052 via switch circuit 2054. This occurs when clk2 2056 is high, and ADC_clk 2008 is asserted on the first clock period of the algorithmic implementation. NAND gate 2056 and inverters 2058, 2060 enable passage of the In_n 2050 signal through the CMOS switch 2062 to be sampled on to C _4a 2052. On all remaining stages, ADC_clk_n 2028 gates the clk2 2022 signal via switch circuit 2064, which includes NAND gate 2066 and inverters 2068, 2070, such that passage of the Out_n 2072 signal from the differential counterpart circuit is enabled through switch 2074 to be sampled on to C _4a 2052, and ultimately switched via switch 2074 to the positive terminal 2076 of the amplifier.
The sub-DAC provides control signals, such as bot_b, mid_b, and top_b, which selectively provide a corresponding voltage refp, refcm, or refn to the bottom plate of the capacitor C_4a via the level-shifting circuit 2078. In this manner, one of the output control signals of the sub-DAC (i.e., bot_b, mid_b, top_b) allows a corresponding voltage to level shift the voltage at the positive terminal 2076 of the amplifier.
A counterpart circuit (not shown) corresponding to the other half of the differential circuit shown in FIGs. 15A and 15B operates analogously.
Amplifiers that may be used in connection with the another exemplary embodiment of the present invention, such as amplifier 2724 described in connection with FIGs. 12A and 7B, can retain a significant amount of residual charge when switching from one N-bit conversion to the next. This is due to the parasitic capacitance at the input to the amplifiers, where this parasitic capacitance includes the oxide input capacitance of the amplifier, wiring capacitance, switch diffusion capacitance, etc. This charge is transferred to the signal capacitors at the start of the next new conversion, giving rise to a substantial degradation in performance when any over-range occurs in the ADC (i.e., an input signal which has an amplitude larger than refp-refn).
In accordance with the another exemplary embodiment of the present invention, a novel amplifier reset methodology is implemented to address this residual charge problem between conversions. In one embodiment, a number of reset switches are timed to remove the residual charge on the amplifier terminals, while performing the N-bit conversion in N clock cycles of the master clock. As previously indicated, by using a final flash stage it is possible to convert an analog signal into a digital signal using N-1 clock periods of the master clock. The final decision is instantaneous and becomes available with the final LSB+1 bit in the DEC. Thus, during the sampling-in period with ADC_clk (described in connection with FIGs. 13, 14, 15A, 15B), the amplifiers may be reset, since their output is no longer necessary for the DEC. In this manner, the N-bit conversion can be performed in only one additional clock cycle, thereby resulting in an N-bit conversion in N clock cycles of the master clock. If no such reset action were performed, and the input were to fall below 0V for example, then the input signal must reach a minimum level of the offset that has been transferred to the signal capacitors before the ADC starts to convert properly again. Therefore, the reset circuit used in connection with the another exemplary embodiment of the present invention dramatically improves the performance of the algorithmic ADC.
FIG. 16 illustrates a representative portion of an algorithmic ADC stage 2100 which implements such a reset circuit. The amplifier 2102, a single-ended amplifier in the illustrated embodiment, includes a negative input 2104, a positive input 2106, and an output 2108. As indicated in connection with previously described embodiments, the derived clock signal ADC_clk 2110 may be used to trigger the initial sampling of the input signal in an algorithmic ADC, and the derived clock signal ADC_clk_n 2112 is used for the remaining N-2 clock periods. During the time that the new input signal is being sampled as enabled by ADC_clk 2110, the amplifier 2102 can be reset. It should be recognized that the amplifier 2102 can be reset using an additional clock cycle rather than during the sampling-in period corresponding to the ADC_clk 2110, however resetting the amplifier during this period allows the total conversion to be minimized.
Thus, when the ADC_clk 2110 is asserted, each of the switches 2114, 2116, 2118, and 2120 close, and discharge any charge to a reference voltage which is refcm in the illustrated embodiment. Reset switch 2114 is coupled between the negative input 2104 of the amplifier 2102 and refcm, and reset switch 2118 is coupled between the positive input 2106 of the amplifier 2102 and refcm. A reset switch 2116 is also coupled between the negative 2104 and positive 2106 inputs of the amplifier, which in turn are coupled to refcm. Finally, reset switch 2120 is coupled between the amplifier 2102 output 2108 and refcm. When the ADC_clk 2110 is asserted (e.g., transitions high), each of the switches 2114, 2116, 2118, 2120 are closed, thereby discharging parasitic capacitances to refcm.
As indicated above, the ADC stage in accordance with the another exemplary embodiment of the present invention may be used in a differential implementation. However, the principles of the another exemplary embodiment of the present invention may also be implemented in a non-differential mode. FIG. 17 is an example of how the another exemplary embodiment of the present invention may be implemented in a non-differential, single-sampling ADC stage 2200. In this example, the input signal, Vin 2202 is sampled onto a first capacitor C ₁ 2204 when switches 2206, 2208 are closed during clk1. A complemented version of the Vin 2202 signal is generated in any known manner, represented by the inverter 2210. Thus, this inverted signal, Vin' 2212 is sampled onto C ₂ 2214 during clk1 when switches 2216, 2218 are closed.
During the clk1 phase, the Vin 2202 signal is also received at the sub-ADC circuit 2220 of a level shifting circuit 2230, where the sub-ACD circuit 2220 provides the 1.5-bit (or other) data 2221, the value of which depends on the Vin 2202 analog voltage level. This 1.5-bit digital output is received by the decoder/clock generator (clkgen) circuit 2222. On the next clock phase clk2, the decoder/clkgen 2222 asserts one of a plurality of control signals based on the 1.5-bit data 2221, such as the "bottom," "middle," or top" signals. The asserted one of the bottom, middle, or top signals closes a corresponding one of the switches 2224, 2226, 2228 of the level shift circuit 2230. Depending on which of the switches 2224, 2226, 2228 is closed, the corresponding reference voltage -Vref, 0, +Vref is used to shift the output signal RESIDUE 2232 of the amplifier 2234, by providing the selected reference voltage to the positive input 2235 of the amplifier 2234.
The RESIDUE 2232 signal 2232 is generated during the clk2 phase, where the sampled voltage on C ₁ 2204 is coupled between the output 2236 and the negative terminal 2238 of the amplifier 2234, due to switches 2240 and 2242 closing and switches 2206 and 2208 opening. Further, the sampled voltage on C ₂ 2214 is coupled to the positive input 2235 of the amplifier 2234 when switch 2244 closes in response to clk2.
The Vin 2202 signal is therefore inverted, and the complementary signals Vin 2202 and Vin' 2212 are sampled, and provided to the amplifier 2234 as the Vin 2202 signal and an inverted version of the complemented Vin signal, to provide the MX2 function by adding these signals. The RESIDUE 2232 is provided as a result of the subtraction of the voltage provided by the level shift circuit 2230 and the MX2 function performed at the amplifier 2234. As can be seen, the subtraction/level shifting, residue multiplication by two, and sample/hold functions are all performed in one clock cycle, independent of any capacitor mismatch that may occur between the signal capacitors C₁ 2204 and C ₂ 2214.
It is noted that the sub-ADC 2220, decoder/clkgen 2222, and level shift circuit 2230 are representative of the circuit (or equivalent thereof) that may be used to provide the coarse analog-to-digital conversion, decoding, and level shift functions for any of the embodiments of the present invention described herein.
FIG. 18 is a flow diagram of a method for converting an analog input signal to a digital input signal in accordance with one embodiment of the present invention. The analog input signal is sampled 2300 onto a first capacitor, or group of capacitors or capacitive elements collectively providing a capacitance in which the input signal may be stored. A complemented analog input signal, i.e., an inversion of the analog input signal, is similarly sampled 2302 onto a second capacitor(s). One or more switches are actuated 2304 in order to couple the first capacitor between the amplifier output and a first amplifier input, in a unity gain feedback arrangement. The sampled input signal is thus provided to the first amplifier input, such as the inverting/negative amplifier input. One or more switches are also actuated 2306 in order to couple the second capacitor between a selected reference voltage and a second amplifier input, in order to provide an inverted version of the sampled complemented input signal to the first amplifier input as level-shifted by the selected reference voltage. The sampled input signal is added 2308 to the inverted version of the complemented input signal using the amplifier, and the selected reference voltage is effectively subtracted from the output in order to provide a residue signal available for use in subsequent conversion stages.
If there are more ADC residue stages in the ADC as determined at decision block 2310, then the next stage 2312 is considered, and the process is repeated for that stage. When there are no further stages, such as when N-1 stages have been processed in an algorithmic or pipelined ADC configuration, then the final flash stage can be processed 2314 as previously described.
Each of the illustrated embodiments (as well as other embodiments of the present invention not illustrated herein) not only provide a significantly more accurate conversion, the resulting ADC is substantially faster than prior art ADCs employing analogous hardware. In other words, the use of amplifiers and capacitors in both prior art systems and in the another exemplary embodiment of the present invention, the another exemplary is substantially faster than the prior art systems by virtue of the fact that the feedback factor (and, consequently the gainbandwidth) for the amplifiers is substantially larger.
The foregoing description of various exemplary embodiments of the invention has been presented for the purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise form disclosed. Many modifications and variations are possible in light of the above teaching. It is intended that the scope of the invention be limited not with this detailed description, but rather by the claims appended hereto.

Claims

A circuit (200, 300) for adding a plurality of input signals, comprising:
an amplifier (230, 322) having inverting (236, 328) and non-inverting (240,332) input terminals and an output terminal (216, 308);

a first sampling circuit coupled between a first input (206, 302) signal and a first reference signal (212) to store a first voltage across a first capacitor (218, 310) in response to a first clock phase (202);

a second sampling circuit coupled between a second input signal (208, 304) and a second reference signal (210, 252) to store a second voltage across a second capacitor (228, 312) in response to the first clock phase (202); and

a switching circuit (220, 222, 224, 226, 232, 234, 238, 242, 314, 316, 318, 320, 324, 326, 330, 334, 338, 340, 342, 344, 346, 348, 350, 352) coupled to the amplifier (230, 322) and the first and second sampling circuits, wherein, in response to a second clock phase (204), the switching circuit switches the first capacitor (218, 310) storing the first voltage between the inverting input terminal (236, 328) and the output terminal (216, 308) of the amplifier (230, 322), and further switches the second capacitor (228, 312) storing the second voltage between the non-inverting input terminal (240, 332) and a third input signal (214, 306).
The circuit of Claim 1, further comprising an N-phase clock signal comprising the first (202) and second (204) clock phases and remaining clock phases of the N-phase clock signal, and wherein the switching circuit switches the first capacitor (218, 310) between the inverting input terminal (236, 328) and the output terminal (216, 308) of the amplifier (230, 322), and switches the second capacitor(228, 312) between the non-inverting input terminal (236, 332) and a third input signal (214, 306), in response to selected ones of the second (204) and remaining clock phases of the N-phase clock signal.
The circuit of Claim 1, wherein the first reference signal (206) comprises a DC reference voltage or a time-varying signal.
The circuit of Claim 1, wherein the first (202) and second (204) reference signals comprises a common DC reference voltage.
The circuit of Claim 1:
(a) further comprising:
(i) a third sampling circuit coupled between the first input signal (302) and the first reference signal to store a third voltage across a third capacitor (336) in response to the second clock phase (204);

(ii) a fourth sampling circuit coupled between the second input signal (304) and the second reference signal to store a fourth voltage across a fourth capacitor (338) in response to the second clock phase (204); and

(b) wherein the switching circuit is further coupled to the third and fourth sampling circuits, wherein, in response to the first clock phase, the switching circuit switches the third capacitor (336) storing the third voltage between inverting input terminal (328) and the output terminal of the amplifier (308), and further switches the fourth capacitor (338) storing the fourth voltage between the non-inverting input terminal (332) and the third input signal (306).
A method for adding at least two input voltage signals (206, 208, 302, 304), comprising:
Sampling first (206, 302) and second (208, 304) input voltage signals onto first (218, 310) and second (228, 312) capacitor circuits respectively during a first clock phase (202);

coupling the first sampled input voltage (206, 302) held on the first capacitor circuit (218, 310) to a negative input terminal (236, 328) of an amplifier (230, 322) and coupling the second sampled input voltage (208, 304) held on the second capacitor circuit (228, 312) to a positive input terminal (240, 332) of the amplifier (230, 322) during a second clock phase (204);

providing a feedback voltage from an output (216, 308) of the amplifier (230, 322) to the negative input (236, 328) of the amplifier (230, 322) via the first capacitor circuit (218, 310) during the second clock phase (204); and

outputting a sum of the first (206, 302) and second (208, 304) input voltage signals in response to the feedback voltage and the first (206, 302) and second (208, 304) sampled input voltages during the second clock phase (204).
The method of Claim 6, further comprising shifting the voltage level at the output (216, 308) during the second clock phase (204) by applying a shift level voltage (214, 306) to the second capacitor circuit (228, 312) to algebraically modify the second sampled input voltage (208, 304) present at the positive input terminal (240, 332) of the amplifier (230, 322).
The method of Claim 6, further comprising activating at least one switch to create an electrical connection between the second capacitor circuit (228, 312) and the shift level voltage (214, 306) in response to the second clock phase (204).
The method of Claim 6, further comprising:
sampling the first (302) and second (304) input voltage signals onto third (236) and fourth (338) capacitor circuits respectively during the second clock phase (204);

coupling the first sampled input voltage (302) held on the third capacitor circuit (336) to the negative input terminal (328) of the amplifier (322), and coupling the second sampled input voltage (304) held on the fourth capacitor circuit (338) to the positive input terminal (332) of the amplifier (322), during the first clock phase (302);

providing a second feedback voltage from an output (308) of the amplifier (322) to the negative input (328) of the amplifier (322) via the third capacitor circuit (336) during the first clock phase (202); and

outputting a sum of the first (302) and second (304) input voltage signals in response to the feedback voltage and the first (302) and second (304) sampled input voltages during the first clock phase (202).
The method of Claim 9, further comprising shifting the voltage level at the output (308) during the second clock phase by applying a shift level voltage (306) to the second capacitor circuit (312) to algebraically modify the second sampled input voltage present at the positive input terminal (332) of the amplifier (322).
The method of Claim 9, further comprising shifting the voltage level at the output (308) during the first clock phase (202) by applying a shift level voltage (306) to the fourth capacitor circuit (338) to algebraically modify the second sampled input voltage present at the positive input terminal (332) of the amplifier (322).
The method of Claim 6, wherein coupling the first sampled input voltage (206, 302) held on the first capacitor circuit (218, 310) to the negative input terminal (236, 328) of the amplifier (230, 322) comprises activating at least one switch (232, 234, 324, 326) in response to the second clock phase (204) to create an electrical connection between the first capacitor circuit (218, 310) and the negative input terminal (236, 328) of the amplifier (230, 322).
The method of claim 6, wherein coupling the second sampled input voltage (208, 304) held on the second capacitor circuit (228, 312) to the positive input terminal (240, 332) of the amplifier (230, 322) comprises activating at least one switch (238, 242, 330, 334) in response to the second clock phase (204) to create an electrical connection between the second capacitor circuit (228, 312) and the positive input terminal (240, 332) of the amplifier (230, 322).
The circuit of Claim 1, further comprising an analog-to-digital converter (ADC) stage for use in ADCs comprising:
an amplifier (1724, 2234) having first (1730, 2238) and second (1734, 2235) input terminals, and an output terminal (1732, 1770, 2232) to provide an analog ADC residue signal;

first (1704, 2204) and second (1722, 2214) capacitances coupled to sample an input voltage signal (1702, 2202) and a complemented input voltage signal (1720, 2212) respectively in response to a first clock phase (clk1);

a level shifting circuit (2230) coupled to receive the input voltage signal, and to select one of a plurality of reference voltages in response to a second clock signal;

a first switch circuit (1706, 1708, 1726, 1728, 2206, 2208, 2240) coupled to the first capacitance (1704, 2204) to provide the sampled input voltage signal (1702, 2202) to the first input terminal (1730, 2238) of the amplifier (1724, 2234) and to couple the output terminal (1732, 2232) of the amplifier (1724, 2234) to the first capacitance (1704, 2204) via a feedback loop, in response to the second clock phase (clk2); and

a second switch circuit (1714, 1716, 2218, 2244) coupled to the second capacitance (1722, 2214) to provide an inverted version of the sampled complemented input voltage signal (1720, 2212) to the second input terminal (1734, 2235) of the amplifier (1724, 2234) and to reference the second capacitance to the selected reference voltage in response to the second clock phase (clk2); and

wherein the amplifier (1724, 2234) adds the input signal (1702, 2202) to the inverted version of the complemented input signal (1720, 2212) as shifted by the selected reference voltage to create the analog ADC residue signal for use in a subsequent ADC stage.
The circuit of Claim 14, wherein the level shifting circuit comprises:
a sub-ADC (2220) coupled to receive the input voltage signal (2202) and to provide a digital code (2221) based on a voltage of the input voltage signal (2202);

a decoder circuit (2222) coupled to the sub-ADC (2220) to receive the digital code (2221) and to assert one of a plurality of switch signals in response thereto; and

a plurality of switches (2224, 2226, 2228), each coupled to a different one of the plurality of reference voltages; and

wherein the asserted one of the switch signals closes a corresponding one of the plurality of switches (2224, 2226, 2228) to couple a corresponding one of the plurality of reference voltages to the second capacitance (2214) to add to the inverted version (2212) of the sampled complemented input voltage.
The circuit of Claim 15, wherein the digital code is an n-bit binary code having 2ⁿ possible values, and wherein each of the 2ⁿ possible values enables a different one of the plurality of switch signals to be asserted by the decoder circuit (2222).
The circuit of Claim 15, wherein the digital code is a 1.5-bit binary code having three possible values, and wherein each of the three possible values enables a different one of the plurality of switch signals to be asserted by the decoder circuit (2222).
The circuit of Claim 14, wherein;
the first capacitance (1704, 2204) comprises at least one capacitor having a top plate and a bottom plate;

the top plate of the capacitor is coupled to a first reference voltage via the first switch circuit (1706, 1708, 1726, 1728, 2206, 2208, 2240) during the first clock phase (clk1) and to the first input terminal (1730, 2238) of the amplifier (1724, 2234) via the first switch circuit during the second clock phase (clk2); and

the bottom plate of the capacitor is coupled to the input voltage signal (1702, 2202) through the first switch circuit during the first clock phase (clk1); and to the output terminal (1732, 2232) of the amplifier (1724, 2232) via the first switch circuit during the second clock phase (clk2).
The circuit of Claim 14, wherein:
the second capacitance (1722, 2214) comprises at least one capacitor having a top plate and a bottom plate;

the top plate of the capacitor is coupled to a second reference voltage via the second switch circuit (1714, 1716, 2218, 2244) during the first clock phase (clk1), and to the second input terminal (1734, 2235) of the amplifier (1724, 2234) via the second switch circuit during the second clock pahse (clk2); and

the bottom plate of the capacitor is coupled to the complemented input voltage signal (1720, 2212) through the second switch circuit during the first clock phase (clk1), and to the reference voltage selected by the level shifting circuit (2230) via the second switch circuit during the second clock phase (clk2).
The circuit of Claim 14, further comprising a reset circuit (2110, 2114, 2116, 2118) coupled to the amplifier (1724, 2234, 2102) to discharge residual charge present at one or more of the first (1730, 2104, 2238) and second (1734, 2106, 2235) input terminals and output terminal (1732, 2108, 2232) of the amplifier (1724, 2102, 2234) to clear a current analog ADC residue signal in preparation for output of a subsequent analog ADC residue signal.
The circuit of Claim 14, wherein the input voltage signal and complemented input voltage signal comprise complementary input voltage signals of a differential input voltage signal.
The circuit of Claim 14, further comprising:
third (1736, 1766) and fourth (1738, 1768) capacitances coupled to sample the input voltage signal (1702) and the complemented input voltage signal (1720) respectively in response to the second clock phase (clk2);

a second level shifting circuit coupled to receive the input voltage signal, and to select one of a plurality of second reference voltages in response to the first clock phase (clk1);

a third switch circuit (1740, 1742, 1748, 1750) coupled to the third capacitance (1736, 1766) to provide the sampled input voltage signal (1702, 1720) to the first input terminal (1730) of the amplifier (1724), and to couple the output terminal (1732, 1770) of the amplifier (1724) to the third capacitance (1736, 1766) via a second feedback loop, in response to the first clock phase (clk1); and

a fourth switch circuit (1744, 1746) coupled to the fourth capacitance (1738, 1768) to provide an inverted version of the sampled complemented input voltage signal (1720, 1702) to the second input terminal (1734) of the amplifier (1724) and to reference the fourth capacitance (1738, 1768) to the selected second reference voltage in response to the first clock phase (clk1); and

wherein the amplifier (1724) adds the input signal (1702, 1720) to the inverted version of the complemented input signal (1720, 1702) as shifted by the selected second reference voltage to create a second analog ADC residue signal for use in a subsequent ADC stage.
The method of Claim 6, comprising converting an analog input signal (1704, 2202) to a digital signal using an amplifier (1724, 2234) comprising:
(a) sampling the analog input signal (1704, 2202) onto a first capacitor (1704, 2204) and a complement (1720, 2212) of the analog input signal onto a second capacitor (1722, 2214).

(b) providing the sampled analog input signal (1704, 2202) at a first input terminal (1730, 2238) of the amplifier (1714, 2234) by controllably coupling the first capacitor (1704, 2204) between the amplifier output (1732, 2232) and the first input terminal (1730, 2238) in a unity gain feedback configuration;

(c) providing the sampled complemented analog input signal (1720, 2212), level shifted by one of a plurality of selectable reference voltages, at a second input terminal (1734, 2235) of the amplifier (1714, 2234) by controllably coupling the second capacitor (1722, 2214) between a selected one of the reference voltages and the second input terminal (1734, 2235) of the amplifier; and

(d) adding the sampled analog input signal (1704, 2202) to an inverted version of the sampled complemented analog input signal (1720, 2212) and subtracting the selected one of the reference voltages to provide a residue signal available for use in subsequent conversion stages
The method of Claim 23, further comprising repeating steps (a) - (d) for each of the first M-1 stages of an M-stage analog-to-digital conversion having an N-bit resolution.
The method of Claim 24, further comprising resolving least significant bits of the digital signal in an M-th flash stage of the analog to digital conversion, by comparing the residue signal from the M-1 stage to a set of predetermined reference voltages.
The method of Claim 25, wherein the set of predetermined reference voltages comprises 2ⁿ-1 reference voltages, wherein n corresponds to a resolution of the M-th stage.
The method of Claim 26, further comprising resolving N-M bits at the M-th stage of the analog-to-digital conversion having the N-bit resolution
The method of Claim 23, further comprising providing a multi-phase clock signal including a first clock phase (clk1) and a second clock phase (clk2), and wherein step (a) is performed during the first clock phase (clk1) and steps (b), (c), and (d) are performed during the second clock phase (clk2)
The method of Claim 28, wherein controllably coupling the first capacitor (1704, 2202) between the amplifier output and the first input terminal comprises activating one or more switches (1726, 1728, 2240) coupled between the amplifier output (1732, 2232) and the first input terminal (1730, 2238) to complete a circuit path therebetween in response to a transition of the second clock phase (clk2).
The method of Claim 29, further comprising activating one or more sampling switches coupled between the analog input signal (1702, 1720, 2202) and a reference voltage in response to a first transition of the first clock phase (clk1), and deactivating the sampling switches in response to a second transition of the first clock phase (clk1)