EP1177618A1 - Moyens et procede permettant d'ameliorer la performance des recepteurs a antiparasitage - Google Patents

Moyens et procede permettant d'ameliorer la performance des recepteurs a antiparasitage

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Publication number
EP1177618A1
EP1177618A1 EP00916379A EP00916379A EP1177618A1 EP 1177618 A1 EP1177618 A1 EP 1177618A1 EP 00916379 A EP00916379 A EP 00916379A EP 00916379 A EP00916379 A EP 00916379A EP 1177618 A1 EP1177618 A1 EP 1177618A1
Authority
EP
European Patent Office
Prior art keywords
interference
fsle
dfe
receiver
signal
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP00916379A
Other languages
German (de)
English (en)
Inventor
François TRANS
Tho Le-Ngoc
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Sapphire Communications Inc
Original Assignee
Sapphire Communications Inc
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from US09/417,528 external-priority patent/US6553085B1/en
Application filed by Sapphire Communications Inc filed Critical Sapphire Communications Inc
Publication of EP1177618A1 publication Critical patent/EP1177618A1/fr
Withdrawn legal-status Critical Current

Links

Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B3/00—Line transmission systems
    • H04B3/02—Details
    • H04B3/32—Reducing cross-talk, e.g. by compensating
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03012—Arrangements for removing intersymbol interference operating in the time domain
    • H04L25/03019—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception
    • H04L25/03038—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception with a non-recursive structure
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L7/00—Arrangements for synchronising receiver with transmitter
    • H04L7/0008—Synchronisation information channels, e.g. clock distribution lines
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J3/00—Time-division multiplex systems
    • H04J3/02—Details
    • H04J3/06—Synchronising arrangements
    • H04J3/0635—Clock or time synchronisation in a network
    • H04J3/0638—Clock or time synchronisation among nodes; Internode synchronisation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/0335—Arrangements for removing intersymbol interference characterised by the type of transmission
    • H04L2025/03356—Baseband transmission
    • H04L2025/03363—Multilevel
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/03433—Arrangements for removing intersymbol interference characterised by equaliser structure
    • H04L2025/03439—Fixed structures
    • H04L2025/03445—Time domain
    • H04L2025/03471—Tapped delay lines
    • H04L2025/03484—Tapped delay lines time-recursive
    • H04L2025/0349—Tapped delay lines time-recursive as a feedback filter
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/03433—Arrangements for removing intersymbol interference characterised by equaliser structure
    • H04L2025/03439—Fixed structures
    • H04L2025/03445—Time domain
    • H04L2025/03471—Tapped delay lines
    • H04L2025/03509—Tapped delay lines fractionally spaced

Definitions

  • This invention is related to maximizing the transmission of data over communication networks. More specifically, this invention is related to suppression of cross-talk and interference over communications channels.
  • C. BACKGROUND Interference (echo and crosstalk) is one of the major performance-limiting impairments on UTP cables.
  • various receiver structures suitable to the transmission of Gigabit Ethernet over 4 pairs of UTP cables are described.
  • the performance of the invention is disclosed through use of a receiver structure that uses a cascade of FSLE and DFE for interference suppression.
  • the present invention provides a method for equalizing interference over a synchronized packet or frame based baseband transmission system wherein the crosstalk on the system is cyclostationary or periodic with a period equal to a symbol interval.
  • the method comprises the steps of : synchronizing the transmitters and receivers using the uncorrelated transmit signals; generating the cyclostationary NEXT and FEXT interference along with ISI using the uncorrelated symbols at the synchronized transmitters at one or more remote stations and the centrally station ;using cascaded Fractionally Spaced Linear Equalizer (FSLE) and Decision Feed back Equalizer (DFE) for both interference suppression and equalization to minimize excess bandwidth at central receivers at the central station; increasing the receiver's FSLE filter taps (NT) to maximize Signal to noise ration; combining FSLE/DFE and proper phase sampling adjustments, enabling use of the spectral correlation properties peculiar to the modified signals.
  • FSLE Fractionally Spaced Linear Equalizer
  • DFE Decision Feed back Equalizer
  • Figure 2a illustrates the worst-case insertion loss associated with data transmitted over a 100 meter, Cat-5 cable
  • Figure 2b illustrates the cable impulse response over a 100 meter, Cat-5 cable
  • Figure 3 shows the plot of the measured return loss and the return loss limit
  • Figure 4a illustrates the worst-case return loss between pairs of Cat-5 cable pairs
  • Figure 4b illustrates the worst-case NEXT loss between pairs of Cat-5 cable pairs
  • Figure 5 illustrates the worst case FEXT loss between pairs of Cat-5 cable pairs
  • Figure 6 illustrates the channel model including the effects of partial response, DAC and hybrid filtering in the transmitter, the main and coupling channel characteristics, and the filtering in the receiver front-end;
  • FIG. 7 illustrates the received pulse responses including both the mail pulse response and the
  • Figure 8 illustrates the Echo pulse response using the present method
  • Figure 9a illustrates the NEXT pulse response achieved using the present method
  • Figure 9b illustrates the FEXT pulse response using the present method
  • Figure 10a provides a sample receiver structure using Interference cancellers prior to equalisation
  • Figure 10b provides a sample receiver structure using Interference cancellers after equalisation
  • Figure 10c provides a sample receiver structure using cascaded FSLE/DFE for both interference suppression and equalisation
  • Figure lOd illustrates a C17 Cable Impulse Response Graph
  • Figure 20 provides a System Model illustrating the described system
  • FIG. 50A Intersymbol Interference (ISI) At High Transmission Rate Over MIL-C17 Cable
  • Figure 50B Decision Feedback Equalization To Remove Postcursor ISI
  • Figure 60 Proposed High Level MDI-1553+ Transceiver Structure High Level
  • Figure 70 Proposed Detailed MDI-1553+ Transceiver Structure using DPIC
  • Each UTP supports a 250Mb/s full-duplex channel using a 5-level 125Mbaud transmission scheme.
  • the transmitter #1L sends a signal to the Receiver #1R, but also generates spurious signal (called echo) to its own Receiver#lL.
  • the interference signals generated by Transmitters 2L-4L appear at the input of the Receiver #1L are called near-end crosstalk (NEXT) interferers, NEXT_21 to NEXT 41.
  • the interference signals generated by Transmitters 2R-4R on the right appear at the input of the Receiver #1L are called far-end crosstalk (FEXT) interferers, FEXT_21 to FEXT_41.
  • NEXT near-end crosstalk
  • FEXT far-end crosstalk
  • ⁇ (f) is the propagation constant
  • a (f) is the attenuation constant
  • ⁇ (f) is the phase constant.
  • the propagation loss (or insertion loss) limit Lp (f) for category 5 (cat-5) 100m cable is a positive quantity expressed in dB
  • ECHO Loss The Echo loss is indicated by the return loss.
  • Figure 3 shows the plot of the measured return loss and the return loss limit which is 15dB for frequency from 1 to 20M ⁇ z and 15- 101og(_720) for frequency from 20 to 100MHz.
  • the wavy curves in Fig. 4 give the measured pair-to-pair NEXT loss characteristics for three different combinations of twisted pairs in 100m cat-5 cables.
  • the existence of the minima (small loss) and maxima (large loss) in these curves is due to the fact that the frequencies considered here correspond to wavelengths that are in the same length range as the distance between points of unbalance in the NEXT coupling path. Notice that the minima and maxima usually occur at different frequencies for the three pair combinations.
  • NEXT loss corresponding to the minima decreases with increasing frequency and tends to follow the smooth dotted curve on the bottom in the figure, which is defined as the worst-case pair-to-pair NEXT loss (or NEXT loss limit) as a function of frequency.
  • the worst-case TIA/EIA-568-A NEXT loss model shown in Figure 4 is 27.1-16.81og(f/100) in dB.
  • D. Channel Modeling Figure 6 shows the channel model including the effects of partial response, DAC and hybrid filtering in the transmitter, the main and coupling channel characteristics, and the filtering in the receiver front-end.
  • the DAC and hybrid filtering is represented by the cascade of two identical first-order Butterworth sections with a corner frequency of 180MHz. This introduces a 4ns rise/fall time.
  • the receiver front-end is modelled as a fifth-order Butterworth filter with a corner frequency of 80MHz.
  • the main channel, echo coupling and NEXT coupling channels are represented by C( ⁇ ), E( ⁇ ), N 2 ( ⁇ ), N 3 ( ⁇ ), and N 4 ( ⁇ ), respectively.
  • the models for the FEXT's are similar to those of the NEXT's except the coupling channels will be F,( ⁇ ), F 3 ( ⁇ ), and F 4 ( ⁇ ), instead of N 2 ( ⁇ ), N 3 ( ⁇ ), and N 4 ( ⁇ ).
  • the pulse responses of the main, echo, NEXT's and FEXT's at the input of the RECEIVER shown in Figure 6 are shown in Figures 7, 8, and 9, respectively.
  • a NEXT canceller synthesizes, in an adaptive fashion, a replica of the NEXT interferer. The interferer is then cancelled out by subtracting the output of the canceller from the signal appearing at the receiver.
  • a NEXT canceller has the same principle of operation as an echo canceller, and all the familiar structures used for echo cancellers can also be used for NEXT cancellers.
  • the cancellers needs to have access to the local transmitters from which they get their input signals. Typically, this input signal is the stream of symbols generated by the transmitter's encoder.
  • the output signal of the canceler is subtracted from the received signal immediately after the AID. With such an approach, the canceler generates outputs at the same rate as the sampling rate of the A/D.
  • An alternative embodiment is to make the subtraction at the input of the sheer as shown in Fig. b. In this case, the outputs of the canceller need only be generated at the symbol rate.
  • the FFE (feed-forward equalizer) in Figures 10a and b can be a symbol-spaced (SS) or fractionally spaced (FS) FFE or an analog equalizer. It is used to equalize the precursor ISI.
  • the DFE is used to remove the post cursor ISI. Note that the performance of the DFE is also dependent on the reliability of the symbols detected by the slicer and influenced by the error propagation. For this, one may replace the simple slicer by a sequence detector (such as Viterbi decoder) for a better performance. In that case, the long processing delay of the decoder can be an issue.
  • Figure 6 shows the overall system that is used to study the performance of the receiver structure using a FSLE cascaded with a DFE in the presence of interference (echo and NEXT's), ISI, and additive white noise (AWN).
  • the AWN has power spectral density of NJ2.
  • the waveform received by the receiver is:
  • ⁇ the first term of r(t) is the desired signal (i.e., sequence to be detected), while the second term represent N interferers, and n(t) is the AWN at the input of the FFE.
  • ⁇ 0 ⁇ , ⁇ T is the 1th interferer's delay.
  • ⁇ 0 (t) is the overall end-to-end pulse response (e.g., Figure 7)
  • ⁇ ,(t) is the pair-to-pair pulse response of the 1th interferer (e.g., Figures 8-9).
  • ⁇ a k is the transmitted symbol
  • b kI is the interfering symbol. It is assumed that: 1) all a k and b H are uncorrelated;
  • w m ' s and ⁇ beaut are the tap settings of the FF ⁇ and DF ⁇ , respectively, and p is the delay in the receiver's decision relative to the receiver's input.
  • the FF ⁇ and DF ⁇ coefficients are optimized to minimize the mean squared error (MS ⁇ ), where the error is:
  • Equation (2) for the output of the slicer can be expressed as:
  • R n T [r(nT - ⁇ )r(nT -D - ⁇ ) -- r(nT - N w D
  • A E(R_ 1 R n T )
  • A2 E(R-a ⁇ n . ⁇ . p )
  • I is the identity matrix
  • R(t) is the autocorrelation function oFrhe- ower spectral density of AWN at the output of the receiver filter. Note that for stationary interference with power spectrum equal to that of the cyclostationary interference, the results are the same except the q(i, j) term becomes:
  • SNR 10 * log 10 (1 /MSE) where the mean squared error (MSE) expression is shown by Equation (7) above.
  • MSE mean squared error
  • ⁇ NT is the span of the FFE in terms of the number of symbol intervals, and D is the delay 5 element used in the FFE.
  • the number of taps of the FFE is given by the product of
  • ⁇ NF is the number of DFE taps.
  • the AS1553 standard commonly referred to as MIL-STD-1553, was introduced in the early 1970's to define a digital communications bus for the interconnection of different subsystems that were required to share or exchange information in a multi-drop configuration. Since its introduction, the AS 1553 standard has been evolving to incorporate functional and user community enhancements. However, the basic communications and architectural characteristics of the bus have not varied from its original release. Message-based communications over the multi-drop bus make use of the Manchester II bi-phase coding for IMb/s transmission in a half-duplex mode over Twisted-Shielded Pair (TSP) with 90% shield coverage. The largest message is 32 word long where a word has 20 bits.
  • TTP Twisted-Shielded Pair
  • Transmission performance is specified for a word error rate (WER) of 10 "7 or better for an equivalent worst- case Additive White Noise (AWG) of 140mVrms in a bandwidth from 1kHz to 4MHz, and a signal level of 2.1 Vpp .
  • WER word error rate
  • AVG Additive White Noise
  • lOOft-cable shows an insertion loss of 2dB or less for frequency range from 100kHz to 8MHz and the insertion loss increases rapidly beyond 8MHz.
  • An insertion loss of 8dB was measured at 100MHz.
  • a group delay variation within 3ns was measured for frequencies from 75kHz to 100MHz.
  • High speed data transmission of digital data over C-17 cables requires adaptive equalization to equalize channel distortion and adaptive interference cancellation to remove both echo and crosstalk interference (NEXT's and FEXT's).
  • Channel distortion includes mainly amplitude distortion and delay dispersion. It causes the smearing and elongation of the duration of each symbol.
  • High speed 1553 network communications where the data symbols closely follow each other, particularly at multiple hundred megabit speeds, time dispersion results in an overlap of successive symbols, an effect known as inter-symbol interference (ISI).
  • ISI inter-symbol interference
  • An Equalization system in concert with a synchronous communication environment, alleviates the relative phase dispersion of the interfered and interfering signals that greatly reduces ISI.
  • MDI-1553+ which utilizes the Com2000TM Signal Equalization of Decision Precursor ISI Canceller (DPIC) (described above), and Com2000TM Signal Coding of Coded Synchronous M-PAM with the emphasis of backward compatibility with existing Mil-STD-1553 standard.
  • DPIC Decision Precursor ISI Canceller
  • the above discussions indicate that it is desired to find advanced signaling techniques for high-speed data transmissions over the multi-drop bus using the existing MIL-C-17 Cable.
  • the present invention provides a method and system, hereinafter referred to as the MDI-1553+, that create an enhanced 1553 System for supporting new terminals with data rate up to lOOMb/s using enhanced coupler.
  • the invention also provides interoperability with existing low-speed AS 1553 terminals at rate IMb/s using the existing AS 1553 transformer assemblies.
  • the cable channel has a severe frequency-selective attenuation at frequencies beyond 1MHz, which limits the transmission at higher rate.
  • the transmission using Manchester coding is limited by the bandwidth of IMHz in which the attenuation is relatively flat.
  • the present invention further provides equalization techniques and advanced combined coding and modulation schemes enabling transmissions at lOOMb/s or above.
  • the system uses a baseband bandwidth up to 30MHz.
  • the insertion loss variation is about 2dB, i.e., the frequency-selective attenuation has a depth of 2dB.
  • Our DPIC equalization technique will be used to remove the inter-symbol interference and crosstalk due to such frequency-selective attenuation.
  • adaptive equalization will be applied to adapt to a particular bus in use.
  • Multilevel modulation such as baseband Synchronous Pulse Amplitude Modulation (SPAM) will increase the bandwidth efficiency required to support transmission of lOOMb/s over a bandlimited channel of up to 30MHz. However, it will require higher signal level to maintain the WER of 10 "7 for the specified noise floor.
  • SAM baseband Synchronous Pulse Amplitude Modulation
  • a specified noise floor of 140mVrms in a frequency range from 1kHz to 4MHz is equivalent to 383mVrms in a frequency range from 1kHz to 30MHz.
  • the use of multi-level modulation scheme for high bandwidth efficiency alone will require a much larger signal level to maintain the same WER of 10 "7 , especially when the bandwidth is also increased.
  • the combined coding and modulation technique takes into account the frequency-selective attenuation of the cable. This is achieved by using a new signaling scheme that combines modulation, coding and advanced equalization for noise suppression to achieve a high performance and high capacity suitable to support lOOMb/s over the existing MIL-C-17-Cable.
  • MDI-1553+ signaling scheme for lOOMb/s speed does not require new coupler.
  • the current legacy passive coupler supports the new transceiver chip operations.
  • an active coupler which is provided power by the new MDI-1553+ node via a new stub wire, is optional.
  • Section II we describe the channel characte ⁇ stics and modeling used to evaluate the performance of various receiver structures.
  • Section III we present the receiver structures currently for the IMb/s transmission of M ⁇ l-STD-1553 over single Twisted Pairs cable and their limited performance.
  • Section IV describes the MDI-1553+ Interference and Noise suppression via DPIC techniques and its applications in the design of various receiver structures using both Echo and NEXT cancellers Their performance and complexity as compared to the existing schemes are discussed.
  • Section V desc ⁇ bes an embodiment of the MDI-1553+ Receiver Architecture
  • the propagation loss (or insertion loss) limit Lp (f) for C-17 100m cable is a positive quantity expressed in dB
  • Figure 20 shows the channel model including the effects of partial response, DAC and hybrid filtering in the transmitter, the main and coupling channel characteristics, and the filtering in the receiver front-end.
  • the DAC and hybrid filtering is represented by the cascade of two identical first-order Butterworth sections with a corner frequency of 180MHz. This introduces a 4ns rise/fall time.
  • the receiver front-end is modelled as a fifth-order Butterworth filter with a corner frequency of 80MHz.
  • the main channel, echo coupling and NEXT coupling channels are represented by C( ⁇ ), E( ⁇ ), N 2 ( ⁇ ) respectively.
  • the models for the FEXT's are similar to those of the NEXT's except the coupling channels will be F 2 ( ⁇ ) instead ofN 2 ( ⁇ ).
  • Figure 30 shows the data coding scheme and bus cabling architecture for the 1553 .
  • the symbol timing recovery is shown in this figure.
  • the receiver is a standard Bi-Phase Manchester Signaling Receiver.
  • the chart below illustrates the difference between the traditional approach and the Com2000TM approach.
  • Reliable duplex operation at 300Mb/s over single pair of a C-17 Twisted Pairs cable requires the usage of some kind of technique to remove inter-symbol interference (ISI) and to combat interference including echo, NEXT and FEXT.
  • ISI inter-symbol interference
  • a NEXT canceller synthesizes, in an adaptive fashion, a replica of the NEXT interferer. The interferer is then cancelled out by subtracting the output of the canceller from the signal appearing at the receiver.
  • a NEXT canceller has the same principle of operation as an echo canceller, and all the familiar structures used for echo cancellers can also be used for NEXT cancellers.
  • the cancellers preferably have access to the local transmitters from which they get their input signals. Typically, this input signal is the stream of symbols generated by the transmitter's encoder.
  • the FFE feed-forward equalizer
  • the FFE can be a symbol-spaced (SS) or fractionally spaced (FS) FFE or an analog equalizer. It is used to equalize the precursor ISI.
  • the DFE is also used to remove the post cursor ISI. Note that the performance of the DFE is also dependent on the reliability of the symbols detected by the slicer and influenced by the error propagation. One may optionally replace the simple slicer by a sequence detector (such as Viterbi decoder) for a better performance.
  • Figure 50A provides a graph demonstrating INTERS YMBOL INTERFERENCE (ISI) at high transmission rates over C-17 cable without modification.
  • Figure 50B demonstrates the results of using Feedforward filtering and Equalization to suppress the precursor intersymbol interference in the signal.
  • the MDI- 1553+ transceiver includes a 20-tap fractionally-spaced (T/2) equalizer, an 128-tap DFE and an 165-tap symbol-spaced echo canceller.
  • T/2 20-tap fractionally-spaced
  • 128-tap DFE 128-tap DFE
  • 165-tap symbol-spaced echo canceller 165-tap symbol-spaced echo canceller.
  • the performance margin is very tight and a rate 3 ⁇ 512-state Trellis Code is used in order to provide 6dB of coding gain required for a proper operation. It is therefore desired to enhance the transceiver
  • Short command/signaling messages use 1 Mb/s so that all low- speed and high-speed devices can "understand”.
  • New highspeed devices can exchange data at rate of 100 b/s.
  • Figure 70 shows the proposed transceiver structure using DPIC.
  • the critical issues were that the required performance would include a BER of 1E-7 and that the margin used in theoretical and simulation studies would be 12dB, while the margin on a measured piece of equipment need only be 6dB.
  • the crosstalk model has a NEXT loss of about 57dB at 80kHz and decreases at about 15dB per decade for frequencies above about 20kHz.
  • An embodiment of the system using a single-pair Advanced 1553 achieved a 17dB margin using coded Synchronous 16PAM on existing 1553 test loops for 300ft.
  • the basics of the proposed MDI- 1553+ standard include the following recommendations:
  • CONFIDENTIAL use of a programmable encoder for rate-3/4, 512-state trellis codes for extra 6dB of coding gain. While the present system and method have been described with reference to specific embodiments, those skilled in the art will recognize that these procedures may be applied to all kinds communications channels and filtering mechanisms. Thus, the scope of this invention and claims should not be limited by the described implementations.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Power Engineering (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
  • Mobile Radio Communication Systems (AREA)

Abstract

L'invention porte sur un canal de transmission de données bidirectionnel placé entre une station centrale et une pluralité de terminaux distants, et concerne notamment un procédé permettant d'équilibrer les parasites dans un système d'émission en bande de base par paquets ou trames synchronisé, un écho magnétique du système étant cyclostationnaire ou périodique et la période étant égale à un intervalle entre les symboles. Ledit procédé comprend les étapes suivantes : synchronisation des émetteurs et des récepteurs au moyen des signaux d'émission non corrélés; génération des parasites NEXT et FEXT cyclostationnaires avec des brouillages inter-symboles (ISI) au moyen des symboles non corrélés au niveau des émetteurs synchronisés d'un ou de plusieurs terminaux distants et de la station centrale; utilisation d'un égaliseur linéaire à faible espacement (FSLE) et d'un égaliseur à rétroaction de décision (DFE) en cascade pour à la fois supprimer et équilibrer les parasites, afin de réduire au minimum une bande passante excessive au niveau des récepteurs centraux de la station centrale ; augmentation des prises des filtres (NT) à FSLE des récepteurs pour augmenter au maximum le rapport signal/brut; et combinaison FSLE/DFE et réglage de l'échantillonnage de phase adéquat, ce qui permet l'utilisation des propriétés de corrélation spectrale particulières aux signaux modifiés.
EP00916379A 1999-04-14 2000-03-15 Moyens et procede permettant d'ameliorer la performance des recepteurs a antiparasitage Withdrawn EP1177618A1 (fr)

Applications Claiming Priority (9)

Application Number Priority Date Filing Date Title
US444007 1982-11-23
US12931499P 1999-04-14 1999-04-14
US129314P 1999-04-14
US09/417,528 US6553085B1 (en) 1997-07-31 1999-10-13 Means and method for increasing performance of interference-suppression based receivers
US417528 1999-10-13
US44400799A 1999-11-19 1999-11-19
US17045599P 1999-12-13 1999-12-13
US170455P 1999-12-13
PCT/US2000/006842 WO2000062415A1 (fr) 1999-04-14 2000-03-15 Moyens et procede permettant d'ameliorer la performance des recepteurs a antiparasitage

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EP1177618A1 true EP1177618A1 (fr) 2002-02-06

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EP00930112A Withdrawn EP1173949A1 (fr) 1999-04-14 2000-04-14 Systeme de reseau synchrone universel pour processeur internet et environnement de fonctionnement internet

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AU (1) AU3748900A (fr)
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WO2002058315A1 (fr) * 2001-01-17 2002-07-25 Telcordia Technologies, Inc. Identification de la diaphonie pour la gestion du spectre dans des systemes de telecommunications large bande
EP1413121A4 (fr) * 2001-07-31 2006-01-11 Telcordia Tech Inc Identification amelioree de la diaphonie pour la gestion du spectre dans des systemes de telecommunications large bande
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