Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04S—STEREOPHONIC SYSTEMS 
  • H04S3/00—Systems employing more than two channels, e.g. quadraphonic
  • H04S3/02—Systems employing more than two channels, e.g. quadraphonic of the matrix type, i.e. in which input signals are combined algebraically, e.g. after having been phase shifted with respect to each other

Definitions

• This invention relates to sound reproduction systems involving the decoding of a stereophonic pair of input audio signals into a multiplicity of output signals for reproduction after suitable amplification through a like plurality of loudspeakers arranged to surround a listener.
• the invention concerns an improved set of design criteria and their solution to create a decoding matrix having optimum psychoacoustic performance, with high separation between left and right components of the stereo signals while maintaining non-directionally encoded components at a constant acoustic level regardless of the direction of directionally encoded components of the input audio signals.
• this invention relates to the encoding of multi-channel sound onto two channels for reproduction by decoders according to the invention.
• it relates to improved matrixing coefficients for a 5-2-5 matrix encoder and decoder system.
• Apparatus for decoding a stereophonic pair of left and right input audio signals into a multiplicity of output signals is commonly referred to as a surround sound decoder or processor.
• Surround sound decoders work by combining the left and right input audio signals in different proportions to produce the multiplicity N of output signals.
• the various combinations of the input audio signals may be mathematically described in terms of a N row by 2 column matrix, in which there are 2N coefficients each relating the proportion of either left or right input audio signals contained in a particular output signal.
• the matrix coefficients may be fixed, in which case the matrix is called passive, or they may vary in time in a manner defined by one or more control signals, in which case the matrix is described as active.
• the coefficients in a decoding matrix may be real or complex.
• a passive matrix which is defined as a matrix in which the coefficients are constant, such as the Dolby Surround matrix
• properties include the following:
• Signals encoded with a standard encoder will be reproduced by a passive matrix decoder with equal loudness regardless of their encoded direction.
• the input signals are a combination of a directionally encoded component and a decorrelated component there is no change in either the loudness or the apparent separation of the decorrelated component as the encoded direction of the directionally encoded component changes.
• a disadvantage of passive decoders is that the separation of both directional and decorrelated components of the input signals is not optimal. For example, a signal intended to come from front center is also reproduced in the left and right front output channels usually with a level difference of only 3dB. Therefore, most modern decoders employ some variation of the matrix coefficients with the apparent direction of the predominant sound source, that is, they are active rather than passive.
• these directional control signals can be possibly derived from directional information recorded on a subchannel of a digital audio signal.
• This invention concerns the use to which these directional control signals are put in controlling an active matrix which takes the signals on the two inputs and distributes them to a number of output channels in appropriately varying proportions dependent upon the directional control signals.
• each of these matrices is constructed somewhat differently, but in each case each output is formed by a sum of the two input signals, each input signal having been first multiplied by a coefficient.
• each matrix in the prior art can be completely specified by knowing the value of two coefficients for each output and how these coefficients vary as a function of the directional control signals which provide directional information as described above.
• These two coefficients are the matrix elements of a N by 2 matrix, where N is the number of output channels, which completely specifies the character of the decoder.
• these matrix elements are not explicitly stated, but can be inferred from the descriptions given. In a particular embodiment they can also be easily measured.
• a boost is applied to the front channels when a strongly steered signal such as dialog is present. This upsets the balance between such signals and background effects or music, relative to the balance between such signals in the discrete 5 channel movie theater system.
• An improved active encoder described herein is needed to correct the balance between the strongly steered front signals and music.
• a further improvement in the decoder is to limit the effects of abrupt changes in the directional control signals to provide better dynamic response to rapid changes therein.
• the present invention is concerned with realization of the active matrix having certain properties which optimize its psychoacoustic performance.
• the invention is a surround sound decoder having variable matrix values so constructed as to reduce directionally encoded audio components in outputs which are not directly involved in reproducing them in the intended direction; enhance directionally encoded audio components in the outputs which are directly involved in reproducing them in the intended direction so as to maintain constant total power for such signals; while preserving high separation between the left and right channel components of non-directional signals regardless of the steering signals; and maintaining the loudness defined as the total audio power level of non-directional signals effectively constant whether or not directionally encoded signals are present and regardless of their intended direction if present.
• a surround sound decoder for redistributing a pair of left and right audio input signals including directionally encoded and non-directional components into a plurality of output channels for reproduction through loudspeakers surrounding a listening area, and incorporating circuitry for determining the directional content of the left and right audio signals and generating therefrom at least a left-right steering signal and center-surround steering signal.
• the decoder includes delay circuitry for delaying each of the left and right audio input signals to provide delayed left and right audio signals; a plurality of multipliers equal to twice the number of output channels, organized in pairs, a first element of each pair receiving the delayed left audio signal and a second element receiving the delayed right audio signal, each of the multipliers multiplying its input audio signal by a variable matrix coefficient to provide an output signal; the variable matrix coefficient being controlled by one or both of the steering signals.
• a plurality of summing devices are provided, one for each of the plurality of output channels, with each of the summers receiving the output signals of a pair of the multipliers and producing at its output one of the plurality of output signals.
• the decoder has the variable matrix values so constructed as to reduce directionally encoded audio components in outputs which are not directly involved in reproducing them in the intended direction; and so constructed to enhance directionally encoded audio components in the outputs which are directly involved in reproducing them in the intended direction so as to maintain constant total power for such signals; while preserving high separation between the left and right channel components of non- directional signals regardless of the steering signals; and so constructed to maintain the loudness defined as the total audio power level of non-directional signals effectively constant whether or not directionally encoded signals are present and regardless of their intended direction if present.
• This invention also includes improved active encoder embodiments which correct the balance between strongly steered front signals and decorrelated music signals due to the boost of front signals which occurs in a standard film decoder, and which also increase the separation between encoder outputs when uncorrelated left and right side inputs are presented to the encoder. It also encompasses modified performance in the film decoder specifications with regard to left or right side encoded signals.
• a further improvement in the decoder relates to the effects of abrupt changes in the directional control signals and limits the more slowly changing signal to provide better dynamic response to the rapidly changing signal.
• an advantage of the invention is that it can be implemented as a digital signal processor.
• An advantage of the present invention is that the design of the decoding matrix provides high left to right separation in all output channels.
• a further advantage of the invention is that it maintains this high separation regardless of the direction of the dominant encoded signal.
• Another advantage of the invention is that the total output energy level of any non-encoded decorrelated signal remains constant regardless of the direction of the dominant encoded signal.
• Another advantage of the invention is that it provides an active encoder which has better performance in respect to the left and right surround inputs than that achievable with a passive five channel encoder.
• the decoder of the invention operates optimally when the active five channel encoder
• another advantage of the invention is that with an added phase correction network it can also optimally reproduce movie soundtracks encoded with either the standard four channel passive encoder of the prior art or the five channel passive matrix encoder which is an aspect of the present invention.
• An advantage of the active matrix encoder of the invention is that it provides dynamic control of the balance between strongly steered front signals and non-directional music to compensate for the boost applied to such steered signals in standard film decoders.
• a further advantage of the encoder is that it provides improved separation of simultaneous left side and right side signals when decoded with a standard film decoder.
• An advantage of the decoder of the invention is that it provides more of a level change in the front loudspeakers relative to the rear when a signal ispanned on either side of the listener, improving the apparent motion of such signal sources.
• Another advantage of the decoder according to this invention is to limit the absolute value of one of the two steering signals when the other is rapidly changing, so that dynamic effects are better reproduced.
• the present invention is concerned with improvements to the derivation of suitable variable matrix coefficients as previously disclosed in Griesinger's U.S. Patent Nos. 4,862,502 (1989), 5,136,650 (1992), the July 1996 Griesinger U.S. Patent Application No. 08/684,948, and the November 1996 Griesinger U.S. Patent Application No. 08/742,460, as disclosed in the Provisional Patent Application filed September 1997 .
• the previously used coefficients were implemented in a decoder referred to here as version 1.11.
• the present invention includes two principal changes to the coefficients derived in the previous patent application No. 08/684,948 of July 19, 1996. The first is a change to the "TV matrix" correction in the rear channels.
• an advantage is that the center boost function has been chosen carefully on the basis of listening tests to give a minimal sense of motion of vocals or dialog between the left and right main speakers and the center speaker, while maximizing the left - right separation of instruments which are present along with the vocals.
• adding a special function CF which replaces the previous boost in LFR along the / 0 axis with a cut, the cut designed to preserve in the sum of the powers from the outputs of the decoder the ratio of the power of the center component of signals to the encoder to the total power of signals to the encoder.
• An advantage of this aspect of the invention is that this procedure makes vocals in music, and dialog in films, have the identical balance in the decoded environment that they did in the material before encoding. This procedure also preserves the balance in recordings which were originally mixed for two channel playback.
• the new function CF remains close to zero - that is there is no subtraction of the right input to the decoder from the left input of the decoder when forming the left front output, and this low value is maintained until cs reaches about 30 degrees toward the front. As the control signal cs increases over this range the center channel level rises rapidly at first to a value about 3dB lower than the value for Dolby Pro-Logic, and then holds constant.
• the center level rises rapidly to the same maximum used for Dolby Pro-Logic.
• the CF function also decreases rapidly over this range, increasing the subtraction, and removing the center component from the left and right front outputs.
• the value of CF also drops rapidly to the previous value when the absolute value of control signal Ir approaches the boundary.
• the principal advantage of this invention is a reduction in the variations of various directional signals in the presence of strong steering, especially in the rear signals when steering is to the front, and in center signals when steering is in other directions. This is seen particularly in the corrections to TV matrix decoding.
• An additional advantage of this invention is that it provides a smoother and more transparent reproduction of the surround sound effects without unwanted variations of the total acoustic output of center front signals due to steering activity.
• Another advantage of this invention is to more accurately balance the levels of vocals in music and dialog in films with respect to the non-directional sounds so that the balance is identical in the decoded environment to that in the material before encoding.
• Another advantage of the invention is to preserve the balance in recordings that were originally mixed for two channel playback.
• FIG. 1 is a block schematic of a passive matrix Dolby surround decoder according to the prior art
• FIG. 2 is a block schematic of a standard Dolby matrix encoder according to the prior art
• FIG. 3 is a block schematic of a five channel encoder for producing Dolby matrix compatible encoding of discrete five channel soundtracks according to the present invention
• FIG. 4 is a block schematic of a five channel embodiment of the decoder according to the invention.
• FIGs. 5a and 5b show detailed schematics for a typical phase shifter that may be used in the circuit of FIG. 4;
• FIGs. 6a-6e show the relationships between various signals in the decoder of FIG. 4;
• FIG. 7 shows a block schematic of an active encoder according to the invention.
• FIG. 8 shows a phase sensitive detection circuit for generation of an ls/rs signal for use with the phase correction circuit of FIG. 9;
• FIG. 9 shows an input phase correction circuit to be applied ahead of the decoder of FIG. 4 for optimal decoding of passively encoded movie soundtracksincluding a graph showing the relationship between the control signal ls/rs and the steering angle ⁇ S ;
• FIG. 10 shows a block schematic of a simplified active encoder according to the invention, also including a graph of the steering angle ⁇ RS against the control signal rs/ls;
• FIG. 11 shows a block schematic of an active matrix encoder having amplitude compensation for strongly steered front signals and better separation or simultaneous side inputs, according to the invention
• FIGs. 12a- 12c show graphically the variation of the GL, GC and GR signals for front quadrant steering and of the left-left (LL) and left-right (LR) matrix elements as steering goes from left to left side in the encoder of FIG. 11; and FIG. 13 shows graphically the maximum permissible values of each of control signals 1/r and c/s as the other changes, for signals steered between left and center, as applied to the decoder of FIG. 4 or the seven channel variant thereof.
• FIG. 14 is a perspective graphical view showing the value of the left rear left (LRL) matrix element in decoder version 1.11 of the general type shown in FIG. 4, illustrating a discontinuity near the left vertex;
• LNL left rear left
• FIG. 16 shows in perspective graphical view the LRL matrix element as it was intended to be in decoder version 1.11, as contrasted with the flawed matrix element in FIG. 14 which was actually implemented;
• FIG. 17 similarly shows the left front left (LFL) matrix element from U.S. Patent No. 4,862,502 and Dolby Pro-Logic, scaled so the maximum value is one;
• FIG. 18 similarly shows the left front right (LFR) matrix element from U.S. Patent No. 4,862,502 and Dolby Pro-Logic, scaled so the maximum value is one;
• FIG. 18 similarly shows the left front right (LFR) matrix element from U.S. Patent No. 4,862,502 and Dolby Pro-Logic, scaled so the maximum value is one;
• FIG. 18 similarly shows the left front right (LFR) matrix element from
• FIG. 19 graphically represents in perspective the square root of the sum of the squares of LFL and LFR from U.S. Patent No. 4,862,502, scaled so the maximum value is one, showing that the value is constant at 0.71 along the axis from unsteered to right, while the unsteered to left rises 3dB to the value 1, and the unsteered to center or to rear falls by 3dB to the value 0.5, in which graph the rear direction profile is identical to that of the center direction;
• FIG. 20 similarly represents the square root of the sum of the LFL and LFR matrix elements from the previous U.S. Patent Application No. 08/742,460, scaled so the maximum value is 1, illustrating the constant value of .71 in the entire right half of the plane, and the gentle rise to one toward the left vertex;
• FIG. 22 similarly shows the left front left matrix (LFL) element having the correct amplitude along the left to center boundary, as well as along the center to right boundary;
• FIG. 23 is a graph showing the behavior of LFL and LFR along the rear boundary between left and full rear, where the slight glitch is due to the absence of a point at 22.5 degrees.
• LFL left front left
• FIG. 26 illustrates the root mean squared sum of LFL and LFR, according to the present invention
• FIG. 27 shows the square root of the sum of the squares of LFL and LFR including the correction to the rear level, viewed from the left rear, showing that the unsteered (middle) to right axis has the value one, the center vertex has the value 0.71, the rear vertex has the value 0.5, and the left vertex has the value 1.41, and showing the peak along the middle to center axis;
• FIG. 27 shows the square root of the sum of the squares of LFL and LFR including the correction to the rear level, viewed from the left rear, showing that the unsteered (middle) to right axis has the value one, the center vertex has the value 0.71, the rear vertex has the value 0.5, and the left vertex has the value 1.41, and showing the peak along the middle to center axis;
• FIG. 27 shows the square root of the sum of the squares of LFL and LFR including the correction to the rear level, viewed from the left rear, showing that the unsteered (middle) to right axi
• FIG. 28 is a graph showing as a solid curve the center matrix value as a function of CS in dB, assuming sound power ratios identical to stereo, and using Dolby matrix elements with 3dB less power in the rear than typically used, and as a dotted curve the actual value of the center matrix elements in Pro-Logic, illustrating that while the actual values give reasonable results for an unsteered signal and a fully steered signal, they are about 1.5dB too low in the middle;
• FIG. 30 shows the square root of the sum of the squares of LRL and LRR, using the elements implemented in decoder version 1.11 illustrating that in the front left quadrant there is a 3dB dip along the line from the middle to the left vertex, and nearly a 3dB boost in the level along the boundary between left and center, also showing the "mountain range” in the rear quadrant and including the "TV matrix” dip of 3dB at the center of the plane, which is hard to see in this projection;
• FIG. 32 shows in perspective graphically the square root of the sum of the squares of LRL and LRR using the values for GR and GS according to the present invention, illustrating that except for the valley created by the "TV matrix” correction, the sum of the squares is close to one and continuous;
• FIG. 33 shows similarly the center left (CL) matrix element of the four channel decoder (and the Pro-Logic decoder), which is also the graph of the center right (CR) matrix element if left and right are interchanged, showing that the middle of the graph and the right and rear vertices have the value 1, the center vertex has the value 1.41, but in practice this element is scaled so the maximum value is one;
• FIG. 34 shows for comparison the center left matrix element in the decoder version 1.11, in which the middle value and the right and rear vertices have been reduced by 4.5dB, so that as cs increases, the center rises to the value of 1.41 in two slopes;
• FIG. 35 shows graphically as a solid curve the center attenuation needed for the LFL and LFR values according to the present invention if the energy of the center component of the input signal is to be preserved in the front three channels as steering increases toward the front, and also shows as a dotted curve the center values for a standard decoder;
• FIG. 36 shows graphically as a solid curve the value of GF needed for constant energy ratios with center attenuation GC, accordsing to the invention, and as a dashed curve the value of sin(cs) :,: corrl (the previous LFR element), while the dotted curve shows sin(cs), illustrating that GF is close to zero until cs reaches 30 degrees, and then increases sharply;
• LFR left front right
• FIG. 38 shows in perspective the center left (CL) matrix element with the added center boost function according to the invention, also showing the correction for panning along the boundary between left and center;
• FIG. 39 illustrates graphically the levels of the center output and the left output as a signal pans from center to left showing that with the correction the panning of the center, while not perfect, is reasonably close to the inverse of the left output (the values on the cs axis are inverted);
• FIG. 40 shows a block schematic of an active encoder according to the present invention.
• Preferred embodiments of the invention include a five channel and a seven channel decoder with maximum lateral separation, although reference will be made to general design principles that may be applied to decoders with other numbers of channels as well.
• the encoding will be assumed to follow the standard Dolby Surround matrix, and the decoder has four outputs such that the left output signal from the decoder comprises the left input times one; the center is the left input times 0.7 (strictly /O. ⁇ or 0.7071) plus the right input times 0.7; the right output signal is the right input signal times one; and the rear output is the sum of the left input times 0.7 and the right input times -0.7.
• FIG. 1 there is a simplified schematic of a passive Dolby surround matrix decoder 1 according to the prior art, in which these signal relationships are maintained.
• the A (LEFT) and B (RIGHT) audio signals are applied respectively to the input terminals 2, 4, and are buffered by unity gain buffer amplifiers 6 and 8 respectively. They are also combined in the above- specified ratios by signal combiners 10 and 12.
• the outputs of buffers 6, 8 appear at the LEFT (L) and RIGHT (R) output terminals 14, 16, respectively, and the outputs of signal combiners 10, 12, appear at the CENTER (C) and SURROUND (S) output terminals 18, 20.
• this matrix has constant gain in all directions, and all outputs are equal in amplitude when inputs are decorrelated.
• the passive matrix design it is possible to extend the passive matrix design to more than four channels. If we wish to have a left rear speaker, the appropriate signal can be made by using suitable matrix elements, but additional conditions are required to form a unique solution; the loudness of the decorrelated component of the signal should be equal in all outputs, and the separation should be high in opposite directions.
• the separation between two outputs is defined as the difference between the levels of a signal in one output and the signal in the other, expressed in decibels (dB).
• dB decibels
• the object of an active matrix is to increase separation between adjacent outputs when there is a directionally encoded signal at the decoder inputs.
• music we shall use the word "music" to denote any decorrelated signal of such complexity that both the directional control signals referred to previously and assumed to be derived from the stereophonic audio input signals are effectively zero.
• the output from the decoder for directional signals should have equal loudness regardless of the encoded direction. That is, the sum of the squares of the various outputs should be constant if a constant level directional component is moved through all directions. Most current art decoders do not achieve this criterion perfectly. There are loudness errors in all, but these errors are not significant in practice. This is the constant loudness criterion.
• the loudness of a music (i.e. decorrelated) component of an input signal should be constant in all output channels regardless of how the directional component of the input is moved, and regardless of the relative levels of the directional component and the music. This requirement means that the sum of the squares of the matrix elements for each output should be constant as the matrix elements change with direction. Decoders in the current art disobey this criterion in ways which are often noticeable. This may be called the constant power criterion.
• the signal intended to come from any direction in the front of the room, from left through center to right, should be boosted in level by 3dB relative to the level such a signal would have in a passive Dolby Surround matrix when there is little or no decorrelated component of the input signals (i.e. no music is present.) When music is the dominant input signal (no correlated components present,) the level is not boosted. Thus as the decoder makes the transition from a music only signal to a pure directionally encoded signal, the level of the directional signal in the front hemisphere should be raised.
• An active decoder matrix should have maximum lateral separation at all times, both during reproduction of decorrelated music signals and for music signals in the presence of a directionally encoded signal. For example if the music signal has violins only on the left and cellos only on the right, these locations should be maintained regardless of the strength or direction of a concurrently present directional signal. This requirement can only be relaxed when a strong directionally encoded signal is being removed from an output which should not reproduce it. Under these conditions, the music will drop in level unless the matrix elements are altered to add more energy to the affected channel from the direction opposite to the steered direction. This will reduce separation, but this separation reduction is difficult to hear in the presence of a strong directionally encoded signal.
• the encoder design in the above-referenced patent was used with some modification to make a number of commercially available decoders.
• the matrix design in the rear hemisphere for these decoders was developed heuristically, but generally meets the requirements stated above fairly well. There is, however, more “pumping” with music than would be optimal, and the leakage of steered signals between the left and right rear outputs is more than the desired level. In this context, “pumping” is audible variation of the music signal due to variation of the directional control signals responding to the direction of the directionally encoded signal. For both reasons, it was necessary to improve the decoder design, and this invention resulted from this design effort. It turns out that the requirements A through F above uniquely specify a matrix, which will be mathematically described below.
• the encoder assumed in the design of the decoder is a simple left-right pan pot.
• a standard sine-cosine curve is used, as described by equations ( 1) and (2) above. These may be restated in the form:
• any output signals intended for reproduction in the rear of the room should be identically zero.
• the output in both the left side and left rear outputs should be equal and smoothly rising, proportional to sin At.
• the output in the left side goes down 6dB and the output in the left rear goes up 2dB, keeping the total loudness, the sum of the squares of each output, constant.
• the left rear and right rear outputs have maximum separation for decorrelated music, since the matrix elements for the right input to the left rear output (and for the left input into the right rear output) are zero resulting in complete separation.
• the matrix elements used to achieve this signal cancellation are adjusted so that the music output is constant and has minimum correlation with the music signal in the left rear.
• the seven channel embodiment includes a time delay of about 15ms in the side channels, and in both versions the rear channels are delayed by about 25ms.
• a standard Dolby surround installation has all the surround loudspeakers wired in phase, and Dolby screening theaters are similarly equipped.
• the standard passive matrix described above with reference to FIG. 1, has a problem with the left rear and right rear outputs.
• a pan from left to surround results in a transition between L and L-R, and a pan from right to surround goes from R to R-L.
• the Fosgate 6-axis decoder described in U.S. Patent No. 5,307,415, among others, has this phase anomaly.
• the decoder of the present invention includes a phase shifter to flip the sign of the right rear output under full rear steering.
• the phase shift is made a function of the log ratio of center over surround, and is inactive when there is forward steering. Typical phase shifters for this purpose are described below with reference to FIGs. 5a and 5b.
• Real world encoders are not as simple as the pan pot mentioned above. However, by careful choice of the method of detecting the steering angle of the inputs, the problems with a standard four-channel encoder can be largely avoided. Thus even a standard film made with a four channel encoder will decode with a substantial amount of directional steering in the rear hemisphere.
• FIG. 2 which represents a standard encoder 21 according to the prior art, as shown in FIG. 1 of the prior Griesinger U. S. Patent No. 5,136,650, there are four input signals L, R, C and S (for left, right, center and surround, respectively,) which are applied to corresponding terminals 22, 24, 26 and 28 and signal combiners and phase shifting elements as shown.
• the left (L) signal 23 from terminal 22 and center (C) signal 25 from terminal 24 are applied to a signal combiner 30 in ratios 1 and 0.707 respectively; the right (R) signal 27 from terminal 26 and the center (C) signal 25 are similarly applied with the same ratios to signal combiner 32.
• the output 31 of signal combiner 30 is applied to a phase shifter 34, and the output 33 of signal combiner 32 is applied to a second identical phase shifter 38.
• the surround (S) signal 29 from terminal 28 is applied to a third phase shifter 36, which has a 90° phase lag relative to the phase shifters 34, 38.
• the output 35 of phase shifter 34 is applied to signal combiner 40, along with 0.707 times the output 37 of phase shifter 36.
• the output 39 of phase shifter 38 is combined with -0.707 times the output 37 of phase shifter 36 in the signal combiner 42.
• the outputs A and B of the encoder are the output signals 41 and 43 of the signal combiners 40 and 42 respectively. Mathematically, these encoder outputs can be described by the equations:
• the additional elements of the new encoder 48 are applied ahead of the standard encoder 21 of FIG. 2, described above.
• the left, center and right signals 51, 53 and 55 are applied to terminals 50, 52 and 54, respectively, of FIG. 3.
• an all-pass phase shifter, 56, 58 and 60 respectively, having a phase shift function (pif) (shown as ⁇ ) is inserted in the signal path.
• the left surround signal 63 is applied to input terminal 62 and then through an all-pass phase shifter 66 with phase shift function ⁇ -90°.
• the right surround signal 65 from input terminal 64 is applied to a ⁇ -90° phase shifter 68.
• the signal combiner 70 combines the left phase-shifter output signal 57 from phase shifter 56 with 0.83 times the left surround phase-shifted output signal 67 from phase shifter 66 to produce the output signal 71 labeled L, which is applied via terminal 76 to the left input terminal 22 of standard encoder 21.
• the signal combiner 72 combines the right phase-shifter output signal 61 from phase shifter 60 with -0.83 times the right surround phase-shifted output signal 69 from phase shifter 68 to produce the output signal 73 labeled R, which is applied via terminal 82 to the right input terminal 26 of standard encoder 21.
• the signal combiner 74 combines -0.53 times the left surround phase-shifter output signal 67 from phase shifter 66 with 0.53 times the right surround phase-shifted output signal 69 from phase shifter 68 to produce the output signal 75 labeled S, which is applied via terminal 80 to the surround input terminal 28 of standard encoder 21.
• the output signal 59 of the center phase shifter 58, labeled C, is applied via terminal 78 to the center input terminal 24 of standard encoder 21.
• the encoder of FIG. 3 has the property that a signal on any of the discrete inputs LS, L, C, R and RS will produce an encoded signal which will be reproduced correctly by the decoder of the present invention.
• a signal which is in phase in the two surround inputs LS, RS, will produce a fully rear steered input, and a signal which is out of phase in the two surround inputs will produce an unsteered signal, since the outputs A and B of the standard encoder will be in quadrature.
• the mathematical description of the encoder of FIG. 3 used in conjunction with the standard encoder of FIG. 2 may be given in the form:
• All current surround decoders which use active matrices control the matrix coefficients based on information supplied from the input signals. All current decoders, including that of the present invention, derive this information by finding the logarithms of the rectified and smoothed left and right input signals A and B, their sum A+B and their difference A-B. These four logarithms are then subtracted to get the log of the ratio of the left and right signals, 1/r, and the log of the ratio of the sum and difference signals, which will be identified as c/s, for center over surround.
• 1/r and c/s are assumed to be expressed in decibels, such that 1/r is positive if the left channel is louder than the right, and c/s is positive if the signal is steered forward, i.e. the sum signal is larger than the difference signal.
• the attenuation values in the five channel passive encoder above are chosen to produce the same value of 1/r when the LS input only is driven, it being understood that the simplified encoder is used to design the decoder when the angle t has been set to 22.5° (rear). In this case, 1/r is 2.41, or approximately 8dB.
• the two levels are compared in magnitude only, to determine whether the steering is front or back we need to know the sign of c/s, which is positive for forward steering and negative for rear steering.
• the input signals to the decoder are not derived from a pan pot but from an encoder as shown in FIG. 2, which utilizes quadrature phase shifters.
• the problem of specifying the matrix elements is divided into four sections, depending on what quadrant of the encoded space is being used, i.e. left front, left rear, right front or right rear.
• the left output is the matrix element LL times the left input plus the matrix element LR times the right input.
• FL(ts) which in our example decoder is assumed to be equal to cos(2ts).
• the center output should smoothly decrease as steering moves either left or right, and this decrease should be controlled by the magnitude of 1/r, not the magnitude of c/s. Strong steering in the left or right directions should cause the decrease. This will result in quite different values for the center left matrix element CL and the center right element CR, which will swap when the steering switches from right to left.
• the problem is that we want the left rear LRL matrix element to be 1 when there is no steering, and yet we want no directional output from this channel during either left or center steering. If we follow the method used above, we get matrix elements which give no output when the signal is steered to the left or center, but when there is no steering, the output will be the sum of the two input signals. This is a conventional solution, where there is poor separation when steering stops. We want full separation, which means LRL must be one and LRR must be zero with no steering.
• tl here is different from the angle defined previously for the center output.
• VGAs variable gain amplifiers
• GL, GC, GR and GS for left, center, right and surround respectively
• GSL and GSR are supplemental signals derived from these for the left and right surround VGA's.
• the coefficients here described use a linear combination of the G values to provide the left and right coefficients as a function of the two angles ts, derived from c/s, and tl, derived from 1/r, respectively.
• LS A ( 1 - GSL) - 0.5 (A + B) GC - 0.5 (A - B) GS - B x GL ...(44)
• RS B (1 - GSR) - 0.5 (A + B) GC + 0.5 (A - B) GS - A x GR ...(45)
• the center matrix elements are identical in rear steering as they depend only on angles derived from 1/r, and are not dependent on the sign of c/s.
• the side left and side right outputs should have full separation when steering is low or zero. However, the signal on the left side and rear outputs must be removed when there is strong left steering.
• right side and right rear outputs are inherently free of the left input when there is steering in the left rear quadrant, but we must remove signals steered center or rear, so terms must be included that are sensitive to c/s.
• Right side and right rear outputs are equal, except for different delays, and we have to solve:
• the decoder design meets all of the requirements set out at the start. Signals are removed from outputs where they do not belong, full separation is maintained when there is no steering, and the music has constant level in all outputs regardless of steering. Unfortunately, we cannot meet all of these requirements for the rear output in the rear quadrant.
• One of the assumptions must be broken, and the least problematic one to break is the assumption of constant music level as the steering goes to full rear.
• the standard film decoder does not boost the level to the rear speaker, and thus a standard film decoder does not increase the music level as a sound effect moves to the rear.
• the standard film decoder has no separation in the rear channels. We can get the rear separation we want only by allowing the music level to increase by 3dB during strong rear steering. This is in practice more than acceptable. Some increase in music level under these conditions is not audible — it may even be desirable.
• LSL cos(45 ° - tl) + RBOOSTdog c/s) - RSBOOST(te) ...(81)
• LSR -sin(45° - tl) ...(82)
• RSR cos(45° - tl) + RBOOSTdog c/s) - RSBOOST(te) ...(84) and for the rear outputs,
• RRL sin(45° - tl) ...(87)
• LFL LL matrix element
• tlr 90° - arctan(L ⁇ ) ...(89)
• trl 90° - arctan(r/l) ...(90)
• RFR cos ts + LFBOOST(t,rZ) ...(92)
• the decoder provides the left, center, right, left rear and right rear outputs, the left side and right side outputs being omitted. It is understood from the above mathematical description that the circuitry for the left rear and right rear outputs of the seven channel decoder can be obtained by similar circuitry to that for the left and right surround outputs shown, with an additional 10ms delay similar to the blocks 96 and 118 which implement 15ms delays.
• the input terminals 92 and 94 respectively receive the left and right stereophonic audio input signals labeled A and B, which may typically be outputs from the encoders of FIGs 2, 3, or 7, directly or after transmission/recording and reception/playback through typical audio reproduction media.
• the A signal at terminal 92 passes through a short (typically 15ms) delay before application to other circuit elements to be described below, so as to permit the signal processing which results in the 1/r and c/s signals to be completed in a similar time period, thereby causing the control signals to act on the delayed audio signals at precisely the right time for steering them to the appropriate loudspeakers.
• the A signal from terminal 92 is buffered by a unity gain buffer 98 and passed to a rectifier circuit 100 and a logarithmic amplifier 102.
• the B signal from terminal 94 is passed through a buffer 104, a rectifier 106 and a logarithmic amplifier 108.
• the outputs of the logarithmic amplifiers 102 and 108, labeled A" and B" respectively, are combined by subtractor 110 to produce the 1/r directional control signal, which is passed through switch 112 to the matrix circuitry described below.
• a time constant comprising resistor 114 and capacitor 116 is interposed in this path to slow down the output transitions of the 1/r signal.
• the B signal from terminal 94 is also passed through a 15ms delay for the reason stated above.
• the A and B signals from terminals 92 and 94 are combined in an analog adder 120, rectified by rectifier 122 and passed through logarithmic amplifier 124. Similarly, the A and B signals are subtracted in subtracter 126, then passed through rectifier 128 and logarithmic amplifier 130. The signals from the logarithmic amplifiers 124 and 130 are combined in subtracter 132 to produce the signal c/s, which is passed through switch 134. In the alternative position of switch 134, the signal passes through the time constant formed by resistor 136 and capacitor 138, which have identical values to the corresponding components 114 and 116. Thus far, the control voltage generation circuit has been described. As is typical of such circuits, the 1/r and c/s signals vary in proportion to the logarithms of the ratios between the amplitudes of left A and right B, and of center (sum) and surround (difference) of these signals.
• the matrix elements are represented by the circuit blocks 140 - 158, which are each labeled according to the coefficient they model, according to the preceding equations.
• the block 140 labeled LL performs the function described by equation (27), (54), (91) or (95) as appropriate. In each case, this function depends on the c/s output, which is shown as an input to this block with an arrow, to designate it as a controlling input rather than an audio signal input.
• the audio input is the delayed version of left input signal A after passing through the delay block 96, and it is multiplied by the coefficient LL in block 140 to produce the output signal from this block.
• the outputs of the several matrix elements are summed in summers 160 -
• the RS signal is passed through a variable phase shifter 170 before being applied to the output terminal 180.
• Phase shifter 170 is controlled by the c/s signal to provide a phase shift which changes from 0° to 180° as the signal c/s steers from front to rear.
• circuit elements 152 - 158, 166, 168 and 170 are duplicated, being fed from the same points as their corresponding elements shown in FIG 4, but with the coefficients LRL, LRR, RRL and RRR in blocks corresponding to 152 - 158 respectively, and with additional 10ms delays similar to blocks 96 and 118, which may be inserted either ahead of these blocks or after the corresponding summer elements to blocks 166 and 168.
• FIG. A Although an analog implementation is shown in FIG. A, it is equally possible, and may be physically much simpler, to implement the decoder functions entirely in the digital domain, using a digital signal processor (DSP) chip.
• DSP digital signal processor
• Such chips will be familiar to those skilled in the art, and the block schematic of FIG. 4 will be readily implemented as a program operating in such a DSP to perform the various signal delays, multiplications and additions, as well as to derive the signals 1/r and c/s and the angles tl and ts from these signals, to be used in the equations previously disclosed, so as to provide the full functionality of the decoder according to the present invention.
• FIG. 5a an analog version of the phase shifter 170 is shown.
• the input signal RS' is buffered by an operational amplifier 182 and then inverted by a second operational amplifier 184 with the input resistor 186 and equal feedback resistor 188 defining unity gain.
• the outputs of amplifiers 182 and 184 are respectively applied through variable resistor 190 and capacitor 192 to a third operational amplifier 196, which buffers the voltage at the junction of the variable resistor 190 and capacitor 192 to provide the output signal RS to terminal 180 of FIG. 4.
• This circuit is a conventional single pole phase shifter having an all-pass characteristic.
• variable resistor 190 is controlled by the c/s signal in such manner that the turnover frequency of the phase shifter is high when the signal is steered to the front, so that the rear output signals are out of phase (due to the matrix coefficients) but reduces as the signal steers to the rear, so that the rear output signals become in phase due to inversion of the right rear output RS.
• the phase shift is not the same at all frequencies, the psychoacoustic effect of this phase shifter is acceptable and reduces the phasiness of the rear signals substantially.
• FIG. 5b is shown a conventional variable digital delay element that may be used in implementing a digital embodiment of the delay block 170 of the circuit of FIG. 4.
• the gain value g is controlled by the value of control signal c/s so as to perform the same function as for the analog phase shifter of FIG. 5a.
• the signals applied to adder 200 are summed and delayed by delay block 202, the output of which is fed back through a multiplier 204 of gain g to one of the inputs of adder 200.
• the RS' signal is applied to the other input of adder 204 and also to multiplier 206, where it is multiplied by a coefficient -g.
• the output signal from delay block 202 is multiplied by (1 - g 2 ) in multiplier 208, and added to the signal from multiplier 206 in adder 210 to provide the RS signal at the output of adder 210.
• FIGs. 6a through 6e show graphically the variations of the various matrix coefficients of the decoder of FIG. 4 and its enhancements that are described by equations in the preceding section to the description of FIG. 4, for further clarification of the operation of this decoder.
• the curves A and B represent the variation of coefficients LL (LFL) and -LR (-LFR) respectively as the value of c/s ranges from OdB to about 33dB. These curves follow the sine - cosine law as derived in equations (27) and (28).
• the variation of RR (RFR) and RL (RFL) is similar in form for steering in the right front quadrant.
• the curves C and D respectively show the corresponding values of LFL and LFR for the decoder according to the previous Griesinger Patent No. 5,136,650 for comparison.
• the music component is 3dB too low, hence the new decoder curves A and B which meet at 0.71 provide constant music level, while the old curves do not.
• FIG. 6b are shown the curves E and F representing the center coefficients CL and CR under 1/r steering from center (OdB) to left (33dB).
• the left coefficient CL increases by 3dB while the right coefficient CR decreases to zero as the steering moves to the left. Similar considerations apply but in the opposite sense when the steering is to the right.
• LSL and LSR respectively as the ratio 1/r goes from OdB (no steering or center steering) to 33dB, representing full left steering.
• the LSL curve J reduces to zero, as it is removing left signal from the left surround channel, while the LSR signal increases so that the level of the music remains constant in the room.
• the matrix elements must total (in r.m.s. fashion) to 1 when the input has only a directional signal. This is achieved if they have values of cos 22.5 C or 0.92 and sin 22.5° or 0.38, as can be seen from the curves.
• 1/r can be zero dB either when the signal is steered fully rear, or when there is no steered component of the signal. In either case, the matrix relaxes to the full left-right separation that is desired.
• the curve L represents the RBOOST value tabulated above in TABLE 1 and used in equations (76) and (79), and subsequently.
• the value of LSL is too small when steering to full rear, so the value of RBOOST is added to it to keep the music level constant. Only LSL is boosted, so complete separation is maintained.
• the value of RBOOST depends only on c/s, as c/s varies from -8dB to -33dB (full rear) i.e. the x-axis of the graph is -c/s, with c/s in dB.
• curve M which represents the value of RSBOOST.
• this value is subtracted from the left side coefficient and half of it is added to the left rear component, when steering between left rear (-8dB) to full rear (-33dB).
• the axis is -(c/s in dB), and this curve goes from zero to 0.5, as expressed in equation (80) above.
• FIG. 7 there is shown an active encoder suitable for use in movie soundtrack encoding generally, and particularly with reference to the decoder embodiments presented above.
• the same five signals LS, L, C, R and RS are applied to the correspondingly numbered terminals 62, 50, 52. 54, 64 respectively as in the encoder of FIG. 3.
• a corresponding level detector and logarithmic amplifier to provide signals proportional to the logarithms of the amplitudes of each of these signals.
• These elements are numbered 212-230.
• the logarithmic signals are respectively labeled lsl, 11, cl, rl and rsl, corresponding to the inputs LS, L, C, R and RS. These signal levels are then compared in a comparator block (not shown), whose action is described below.
• Attenuators 254 and 256 attenuate the LS signal by factors of 0.53 and 0.83 respectively, and attenuators 258 and 260 attenuate the RS signal by factors of 0.83 and 0.53 respectively.
• Each of the five input signals passes through an all-pass phase shift network, the blocks labeled 232, 234, providing phase shift functions ⁇ and ⁇ -90° respectively for the attenuated LS signal from attenuators 254 and 256 respectively, blocks 236, 238, and 240 providing the phase shift function ⁇ to each of L, C and R signals respectively.
• a signal combiner 242 sums 0.38LS with -0.38RS to provide a center surround signal to phase shifter block 244, which has a phase shift function ⁇ .
• the phase shifter blocks 246 and 248 provide phase shift functions ⁇ >-90° and ⁇ respectively in the RS channel from attenuators 258 and 260 respectively.
• a similar matrix 252 sums the RS( ) signal with gain sin ⁇ ⁇ s , the RS( ⁇ -90°) signal with gain cos ⁇ ⁇ s , the R( ⁇ />) signal, the C( ⁇ ) signal with gain 0.707, and the S( ⁇ ) signal, to produce the right output B at terminal 46.
• the steering angles ⁇ LS and ⁇ ⁇ s are made dependent upon the log amplitude signals lsl, II, cl, rl and rsl in the following manner in this embodiment of the invention:
• ⁇ LS Whenever lsl is larger than any of the remaining signals, then ⁇ LS approaches 90°, otherwise, ⁇ S approaches 0. These values may be extremes of a smooth curve. Similarly, if rsl is larger than any of the other signals, ⁇ RS approaches 90 ° , otherwise ⁇ ⁇ s approaches 0.
• the particular advantage of this mode of operation is that when a signal is applied to the LS or RS input only, the output of the encoder is real, and produces an 1/r ratio in the decoder of 2.41: 1 (8dB), which is the same value produced by the simplified encoder and the passive encoder.
• FIG. 8 which shows a part of a decoder according to the invention having complex rather than real coefficients in the matrix
• the figure illustrates a method for generating a third control signal ls/rs (in addition to the signals 1/r and c/s generated by the decoder in FIG. 4), which is used for varying the additional phase shift network of FIG. 9 that is placed ahead of the decoder of FIG. 4 in order to effect the generation of complex coefficients in the matrix.
• a and B signals are now applied to terminals 300 and 302 respectively, instead of to terminals 92 and 94 of FIG. 4.
• An all-pass phase shift network 304 having the phase function ⁇ of frequency f, and a second all-pass phase shift network 306 having the phase function ⁇ (f)-90° receive the A signal from terminal 300.
• the phase shifted signal from 304 is attenuated by a factor -0.42 in attenuator 308 and the lagging quadrature phase shifted signal from 306 is attenuated by the factor 0.91 in attenuator 310.
• the outputs of attenuators 308 and 310 are summed in summer 312.
• the B signal at terminal 302 is passed through an all-pass phase shift network 314 so that the output of summer 312 is signal A shifted by 65° relative to signal B at the output of phase shifter 314.
• the output of summer 312 is passed through attenuator 316 with an attenuation factor 0.46, and to one input of a summer 318, where it is added to the phase-shifted signal B from shifter 314.
• the output of phase shifter 314 is attenuated by attenuator 320 with the same factor 0.46 and passed to summer 322 where it is added to the output of summer 312, the phase-shifted A signal.
• the particular choices of coefficients in attenuators 308, 310, 316 and 320 are made so that signals applied to the LS input only of the passive encoder will produce no output at the summer 308 and a signal applied to the RS input only will produce no output at the summer 322.
• the object thus is to design a circuit that will recognize as input of the decoder the case when the signal is only being applied to the left side or right side of the encoder. It does this by a cancellation technique, such that one or the other of the two signals goes to zero when the condition exists.
• the output of summer 318 is passed into level detection circuit 324 and log amplifier 326, while the output of summer 322 is passed through level detector 328 and logarithmic amplifier 330.
• the outputs of log amplifiers 326 and 330 are passed to subtracter 332 which produces an output proportional to their log ratio. This output may be selected by switch 334, or the output from the R-C time constant formed by resistor 336 and capacitor 338, which have values identical to the corresponding components shown in FIG. 4, may alternatively be selected by switch 334 and passed to terminal 340 as the steering signal ls/rs.
• the signal ls/rs will either be a maximum positive value when a signal is applied to the LS input of the passive encoder, or a maximum negative value when a signal is applied to the RS input.
• the purpose of the signal ls/rs is to control the input phases applied to the decoder of FIG. 4. For this reason, the network of FIG. 9 is interposed between the A and B signals applied to terminals 92 and 94 of FIG. 4.
• the circuit shown in FIG. 9 includes a phase shifter 342 of phase function ⁇ , which may be the same shifter as 304 in FIG. 8, followed by an attenuator 344 having the attenuation value cos ⁇ ⁇ s , while the phase shifter 346, which may be the same shifter as 306 in FIG. 8, of phase function ⁇ -90 is passed through attenuator 348 with attenuation factor sin ⁇ ⁇ .
• the outputs of attenuators 344 and 348 are summed by summer 350 to provide a modified A signal at terminal 352, which is to be directly connected to terminal 92 of FIG. 4.
• the B signal is applied to terminal 302 as in
• phase shifter 354 of phase function ⁇ and attenuator 356 of attenuation factor cos ⁇ S while in the other branch it passes through phase shifter 358 of phase function ⁇ -90 and attenuator 360 of attenuation factor sin ⁇ ljS .
• the signals from attenuators 356 and 360 are combined in subtracter 362 to provide a modified B signal at terminal 364, which is to be directly connected to the terminal 94 in FIG. 4.
• the result in the change of phase is to produce better separation between the LS and RS outputs of the decoder (as well as the LR and RR outputs in a 7-channel version) when only the LS or RS inputs of the passive encoder are being driven with signals.
• the relationship between the control signal ls/rs and the steering angle ⁇ LS is shown in the inset graph of FIG. 9. As ls/rs reaches 3dB, the angle ⁇ LS begins to change from 0° rising towards 65° at high values of ls/rs.
• An exactly complementary relationship applies to the other steering angle ⁇ RS which is controlled by the inverse of ls/rs, which we call rs/ls, so that when rs/ls exceeds 3dB, the value of ⁇ RS begins to increase from 0 moving towards an asymptote at -65° when rs/ls is at its maximum value.
• ⁇ LS and ⁇ RS vary, the matrix coefficients effectively become complex due to the phase changes at the inputs to the main part of the decoder shown in FIG. 4.
• FIG. 10 illustrates an alternative embodiment of an encoder that differs from that of FIG. 7 by simplifying the phase shift networks.
• the number of phase shift networks can by reduced by combining the real signals before sending them through the ⁇ phase shifter, thus resulting in only two ⁇ and two ⁇ -90° phase shift networks.
• the description of ⁇ and ⁇ ⁇ s is also simplified.
• ⁇ LS approaches 90° when lsl rsl is greater than 3dB, and otherwise is zero (just as in the decoder design).
• ⁇ ⁇ s approaches 90° when rsl/lsl is greater than 3dB, and otherwise is zero.
• FIG. 10 elements corresponding to those in the right half of FIG. 7, namely the attenuators 254-260 and the ⁇ -90 phase shifters 234 and 246 have been correspondingly numbered.
• the elements of FIG. 10 not so corresponding have also been numbered.
• the coefficients shown in signal combiner elements 242, 250 and 252 of FIG. 7 have been extracted from the signal combiners and applied separately to each of the relevant signals in attenuator elements 262-274, and that these signals thus modified are combined in simple summing devices 276-284, while the five ⁇ phase shifters 232, 236-240 and 248 have been replaced by two phase shifters 286-288.
• FIG. 10 the coefficients shown in signal combiner elements 242, 250 and 252 of FIG. 7 have been extracted from the signal combiners and applied separately to each of the relevant signals in attenuator elements 262-274, and that these signals thus modified are combined in simple summing devices 276-284, while the five ⁇ phase shifters 232, 236-240 and 248 have been replaced by two phase
• the signal path for the LS signal from terminal 62 of FIG. 7 passes as before through attenuator element 256 and ⁇ -90" phase shifter 234, then passing through the actively controlled attenuator 270 having attenuation factor cos ⁇ LH , this being the coefficient formerly shown in signal combiner 250 of FIG. 7.
• This signal is summed in summer 276 as one component of the signal output labeled A at terminal 44 of FIG. 7.
• the signal path for the RS signal at terminal 64 in FIG. 7 similarly passes through attenuator 258 and phase shifter 246, then through active attenuator 274 having attenuation coefficient cos ⁇ RS , formerly part of signal combiner 252 of FIG. 7, to summer 280 where it is one component of the signal labeled B at terminal 46 of FIG. 7.
• the signal path for the center signal C from terminal 52 of FIG. 7 passes first through attenuator 266 with attenuation coefficient 0.71, after which it is applied to summers 278 and 282.
• the L signal from terminal 50 of FIG. 7 is applied directly to summer 278.
• the R signal from terminal 54 of FIG. 7 is applied directly to summer 282.
• the LS signal is also applied through attenuator 254, and through active attenuator 268 with attenuation coefficient sin ⁇ L to the summer 278.
• the RS signal is also passed through attenuator 260 and active attenuator 272 with attenuation coefficient ⁇ ⁇ s to the summer 282.
• the LS signal passes through attenuator 262 of coefficient 0.38 and the RS signal passes through attenuator 264 of coefficient -0.38, both attenuated signals being summed in summer 284, before the result is applied to summer 278 with positive sign and summer 282 with negative sign.
• summer 278 The output of summer 278 is passed through ⁇ phase shifter 286 to summer 276, and the output of summer 282 is passed through ⁇ phase shifter 288 to summer 280, summers 276 and 280 respectively providing the signals A and B to terminals 44 and 46 of FIG. 7.
• FIG. 10 is also shown graphically the relationship between the angle ⁇ RS and the value of rs/ls (or -ls/rs) for signals steered in the right side quadrant. This angle affects the circuit elements 272 and 274, as indicated by the arrows. An exactly similar relationship exists between the steering angle ⁇ LS and the value of ls/rs, this angle affecting circuit elements 268 and 270.
• FIG. 11 an encoder is shown, that is very similar in construction to the encoder of FIG. 10. Those elements that are comparable in function are therefore numbered correspondingly. There are several new elements, the four gain control elements, variable attenuators 290-293, and two control signal generator elements 294, 295. The input and output terminals have been numbered in correspondence with FIG. 7.
• the purpose of the added gain control elements is to correct both the balance between strongly steered front signals and music, and the reduction of separation in response to simultaneous left side and right side signals.
• the Dolby Pro- Logic compatible type of decoder i.e. in this case one that meets criterion E, rather than criterion D, applies a boost of 3dB in the front channels. This boost is quite audible as a shift in the balance between dialog and music, for example.
• the three front signals L, C and R are passed through three variable attenuators 290-292 respectively having gain coefficients GL, GC and GR. These coefficients are controlled by steering control signals derived from the outputs of the encoder.
• the output signals A and B are fed into the inputs of a steering signal voltage generator 294 which comprises identical circuitry to that of the decoder of FIG. 4.
• the two steering voltages 1/r and c/s are thus derived, and will be identical to those generated in an active decoder.
• These two steering voltages affect the gain coefficients in the manner shown in FIGs. 12a and 12b.
• the signal 1/r and inverse r/1 control gains GL and GR respectively of elements 290 and 292, while gain GC of element 291 is controlled by c/s.
• a further improvement in the encoder of FIG. 11 is the addition of the gain coefficient GS of variable attenuator 293, which is controlled by the control voltage generation circuit 295.
• the gain coefficient GS acts upon the signal from summer 284, which is the difference signal between the left side and right side input signals (multiplied by 0.38)
• the purpose of this side difference signal is to provide the proper negative value of the c/s signal when there is a strongly steered left side or right side input to the encoder.
• this side difference signal reduces the separation between left side and right side inputs when both are present at the same time. This reduction in separation is particularly important in the case when the LS and RS inputs are nearly equal but uncorrelated, such as during music, applause, or surround effects like rain.
• the presence of correlation can be determined from the steering voltages derived from the left side and right side inputs to the encoder, using a control voltage generation circuit 295 similar to that in element 294, which thus produces the control signals ls/rs and cs/ss.
• the ls/rs steering voltage was also derived on the original version of the active encoder shown in FIG. 7, to control the values of ⁇ LS and ⁇ RS . While this feature is retained in the encoder of FIG. 11, additional circuitry determines the front-back components of the side signals.
• Both the ls/rs and cs/ss signals control the gain GS of attenuator element 293.
• the ls/rs signal also controls the steering angle tls in attenuators 270 and 272, and its inverse, rs/ls, controls the steering angle trs in attenuators 272 and 274.
• GS The value of GS is then determined by taking the larger of the absolute values of signals ls/rs and cs/ss, limiting this value to 7dB, dividing by 7, then subtracting the result from 1.
• the left front and right front outputs are reduced by an additional 3dB when there is rear steering on the same side.
• the front left signal is reduced by this amount as a signal pans from left to left side
• the right front signal is similarly reduced as a signal pans from right to right side.
• the variation in gain for the LL and LR matrix elements for left to left side steering is shown in FIGs 12b and 12c respectively. Similar curves apply to the right side steering.
• Another aspect of the decoder improvements is a special limiting correction that may be applied digitally to the 1/r and c/s directional control signals. This has the advantage of improving the speed and the accuracy of the steering.
• the 1/r and c/s signals are not independent, but follow a complementary path, shown in FIG. 13. If the logarithmic detectors act rapidly, this curve will be followed dynamically, but when a time constant is included, the value of the rising signal can increase rapidly, but the falling signal is usually changing at a slower speed. The result is that the falling signal is higher than it should be, reducing the dynamic separation. To correct this problem, the signal that is changing more rapidly is used to limit the other signal to follow the curve of FIG.
• FIG. 14 shows for reference the form of the left rear left (LRL) matrix element coefficient used in the matrix of FIG. 4, as implemented in the decoder according to the prior patent application No. 08/684,948.
• the value of this coefficient is plotted in three-dimensional format as the height of the ordinates with respect to the cs and Ir control signals, which are derived from the usual log- ratio detectors in the decoder of FIG. 4.
• the cs signal represents the ratio of center front to rear surround signal amplitudes and the Ir signal represents the ratio of left to right signal amplitudes.
• Each of these signals is encoded as an angle ranging from zero to 90 degrees.
• FIG. 17 shows the LFL matrix element as implemented in U.S. Patent No. 4,862,502, and in Dolby Pro-Logic, scaled so that the maximum value is 1.
• LFL 1 - 0.5*G(-cs) ...(106)
• LFR 0.5*G(-cs) ...( 107)
• the function G(x) is described in U.S. Patent No. 4,862,502, and specified in U. S. Patent No. 5,307,415. It varies from 0 to one as x varies from 0 to 45 degrees. In the previous patent application it is shown to be equal to l-tan(
• FIG. 19 is shown the square root of the sum of the squares of LFL and LFR from U.S. Patent No. 4,862,502, scaled so that the maximum value is one. Notice that the value is constant at 0.71 along the axis from unsteered to right. The unsteered to left rises 3dB to the value one, and the unsteered to center or to rear falls by 3dB to the value 0.5. The rear direction is identical to the center direction, but is not easily seen due to the perspective in this view.
• LFL cos(cs) ...(110)
• LFR -sin(cs) ...(111)
• LFL cos(-cs) ...(114)
• LFR sin(-cs) ...(115)
• FIG. 20 is shown the square root of the sum of the LFL and LFR matrix elements from the previous patent application, No. 08/684,948, scaled so the maximum value is 1. Note the constant value of 0.71 in the entire right half of the plane, and the gentle rise to one toward the left vertex.
• the decoder version 1.11 made several changes to these matrix elements. Keeping the basic functional dependence, an additional boost was added along the cs axis in the front, and a cut was added along the cs axis in the rear. The reason for the boost was to improve the performance with stereo music which was panned forward. The purpose of the cut in the rear was to increase the separation between the front channels and the rear channels when stereo music was panned to the rear.
• LFL (cos(cs) + 0.41 :;: G(/r)) :;: boostl(cs) .(116)
• LFR (-sin(cs)) ⁇ boostl(cs) ...(117)
• LFL (cos(cs))*boostl(cs) ...(118)
• LFR (-sin(cs))*boostl(cs) ...(119)
• LFL (cos(-cs) + 0.41*G(/r))/boost(cs) .(120)
• LFR (sin(cs)Vboost(cs) ...(121)
• LFL (cos(cs))/boost(cs) ...(122)
• LFR (sin(cs))/boost(cs) ...(123)
• Boost(cs) is given by corr(x) in the code below, in which comments are preceded by the % symbol:
• the boost along the boundary creates a panning error. Also the cut in the rear direction is not optimal. There are two areas where the performancecan be improved. The first is in the behavior of the steering along the boundaries between left and center, and between right and center. As a strong single signal pans from the left to the center, it can be seen in FIG. 21 that the value of the LFL matrix element increases to a maximum half-way between left and center. This increase in value is an unintended consequence of the deliberate increase in level for the left and right main outputs as a center signal is added to stereo music.
• LFL and LFR in the front right quadrant are similar, but without the +0.41*G term. These new definitions lead to the following matrix elements.
• the matrix elements used in the version 1.11 decoder — the ones above — result in the output in the front left channel being about -9dB when a signal is panned to the left rear position. This level difference is sufficient for good performance of the matrix, but it is not as good as it could be.
• LFL cos(t)*F(t) -/+ sin(t) :
• LFR - (sinG *F(t) +/- cos(t) ⁇ (sqrtd-F(t) ⁇ 2))) ...(129)
• FIG. 22 shows that the left front left matrix element has the correct amplitude along the left to center boundary, as well as along the center to right boundary.
• FIG. 23 shows the behavior of LFL and LFR along the rear boundary between left and full rear. The slight glitch is due to the absence of a point at 22.5 degrees.
• LFL cos(cs)/(cos(cs)+sin(cs)) - front_boundary_table(6 ) + 0.41*G(Zr) ...(136)
• the loudness in any given output of unsteered material presented to the inputs of the decoder should be constant, regardless of the direction of a steered signal wd ich is present at the same time.
• this requirement must be relaxed when there is strong steering in the direction of the output in question. That is, if we are looking at the left front output, the sum of the squares of the matrix elements must increase by 3dB when the steering goes full left.
• FIG. 26 shows the root mean squared sum of LFL and LFR, using the new design.
• FIG. 27 shows the square root of the sum of the squares of LFL and LFR including the correction to the rear level, viewed from the left rear.
• the unsteered (middle) to right axis has the value one
• the center vertex has the value 0.71
• the rear vertex has the value 0.5
• the left vertex has the value 1.41. Note the peak along the middle to center axis.
• the next concern addressed in the present invention is correcting the values of the rear matrix elements during front steering.
• LRL .71-.71*G(Zr)+.41 :i: .71*G(cs) ...(140)
• LRR -.71 ;1: G(Zr ) + .41*.71*G(cs) ...( 141)
• the gain of each output to the loudspeaker is adjusted so the sound power is the same at the listening position when a signal is presented to the decoder which has been encoded from the four major directions - left, center, right, and rear. In practice this means that the actual level of the matrix elements is scaled so the four outputs of the decoder are equal under conditions of full steering.
• the matrix elements from the 1989 patent are used, the same calibration procedure results in 3dB less sound power from the rear when the decoder inputs are uncoirelated.
• the elements for the rear outputs in the new design include a form of level boost when the outputs are fully steered - either to the left or right sides - or completely to the rear. Thus they follow the 1989 patent in terms of their surround level when they are unsteered.
• the solid curve shows the center matrix value as a function of cs + 1 in dB, assuming sound power ratios identical to stereo, and using Dolby matrix elements with 3dB less power in the rear than typically used.
• the dotted curve is the actual value of the center matrix elements in Pro-Logic. Notice that while the actual values give reasonable results for an unsteered signal and a fully steered signal, they are about 1.5dB too low in the middle.
• the solid curve shows the value of the center matrix elements assuming equal power ratios to stereo given the matrix elements and calibration actually used in Dolby Pro-Logic.
• the dotted curve shows the actual values of the center matrix elements in Pro-Logic. Notice that the actual values are more than 3dB too low for all steerings.
• the present invention also includes the creation of two independent rear outputs, as described below.
• the major problem with both the 1989 patent elements and the Dolby elements is that there is only a single rear output.
• the disclosure given in Griesinger's 1991 U.S. Patent No. 5,136,650 created two independent outputs at the sides, and the math in that patent was incorporated in the front left quadrant of the U.S. Patent Application No. 08/684,948 of July 1996.
• the goal of the elements in this quadrant was to eliminate the output of a signal steered from left to center, while maintaining some output from the left rear channel for unsteered material present at the same time.
• these matrix elements are very similar to those of the 1989 Griesinger U.S. Patent 4,862,502, but with the addition of a G(Zr) term in LRR and a GS(Zr) term in LRL.
• G(Zr) was included to add signals from the B input channel of the decoder to the left rear output, to provide some unsteered signal power as the steered signal was being removed.
• the formula for GS(Zr) turned out to be equal to G ⁇ 2(Zr), although a more complicated formula is given in the 1991 patent (5,136,650). The two formulae can be shown to be identical.
• the matrix elements are identical to the rear elements given in the 1989 patent (4,862,502), and were implemented in the version 1.11 decoder.
• LRL cos(cs) - GS(Zr) ...(154)
• LRR -sin(cs) - GR(Zr) ...(155)
• the peak in the sum of the squares along the boundary between left and center remains. In a practical design it is probably not very important to compensate for this error, but we can attempt to do so with the following strategy.
• xymin varies from zero to 22.5 degrees. If we multiply it by four, it will vary from zero to 90 degrees, and can be used below. In the front left quadrant
• LRL (cos(cs) - GS(Zr))/(l+.29 ⁇ sin(4 ⁇ xymin)) ...(158)
• LRR (-sin(cs) - GR(Zr))/(l+.29 ⁇ sin(4 xymin)) ...(159)
• FIG. 32 shows the square root of the sum of the squares of LRL and LRR using the new values for GR and GS. This factor shows up in FIG. 32 as a valley centered on zero steering. Note that except for the valley created by the "TV matrix" correction, the sum of the squares is close to one and continuous.
• the correction for TV matrix becomes tvcorr( ⁇ lr ⁇ ). Tvcorr(
• the steered component of the input should be removed from the left outputs - there should be no output from the rear left channel when the steering is toward the right or right rear.
• LRL cos(-cs) ⁇ tvcorr(
• cs ⁇ ) sri(-cs) ⁇ tvcorr(
• LRL cos(45-Zr) ⁇ sin(4 ⁇ (45-Zr))-sm(45-/r) : ' : cos(4 ⁇ (45-Zr))
• the two functions sra(x) and srac(x) are defined for 0 ⁇ Ir ⁇ 45.
• the version 1.11 decoder's LRR coefficient uses a better technique.
• LRR (-srac(lr) + sric(cs_bounded)) tvcorr(
• LRL ((sra(lr) + (sra(lr)-GS(lr)) ( 15-cs)/15) + sri(-cs)) ⁇ tvcorr(
• ); for cs 15 to 22.5
• LRL (sra(lr) + sri(-cs)) ⁇ tvcorr(
• LRL (sra(lr) + sri(cs) + rboost(cs)) ...(174)
• FIG. 33 shows the Center Left (CL) matrix element of the four channel and Dolby Pro-Logic decoder plotted in three dimensions. This is also the graph of the Center Right matrix element if we swap left and right. The middle of the graph, and the right and rear vertices have the value 1. The center vertex has the value 1.41. In practice this element is scaled so the maximum value is one.
• the version 1.11 implementation is based on the steering in the 1989 patent, but with a different scaling, and a different function of cs. We found that it was important to reduce the unsteered level of the center output, and a value 4.5dB less than the Pro-Logic level was chosen.
• the boost function (.41 ⁇ G(cs)) was changed to increase the value of the matrix elements back to the Pro-Logic value as cs increases toward center.
• the boost function in the version 1.11 decoder was chosen relatively arbitrarily.
• the new boost function of cs starts at zero as before, and rises with cs in such a way as CL and CR increase 4.5dB as cs goes from zero to 22.5 degrees.
• the increase is a constant number of dB for each dB of increase in cs.
• the function then changes slope, such that in the next 20 degrees the matrix elements rise another 3dB, and then hold constant.
• the newmatrix elements are equal to the neutral values of the old matrix elements.
• the new and the old matrix elements become equal.
• the output of the center channel is thus 4.5dB less than the old output when steering in neutral, but rises to the old value when the steering is fully to the center.
• FIG. 33 shows the Center Left matrix element in the Logic 7 decoder version 1.11. Note that, relative to the plot of FIG. 33, the middle value and the right and rear vertices have been reduced by 4.5dB. As cs increases the center rises to the value of 1.41 in two slopes. The solution for the center used in version 1.11 is not optimal.
• the center channel output must be derived from the A and B inputs to the decoder. While it is possible to remove either the A or the B input from the center channel output using matrix techniques, any time the steering is not biased either left or right, the center channel must reproduce the sum of the A and B inputs with some gain factor, either a boost or a cut. How loud should it be? The answer to this question depends on the behavior of the left and right main outputs.
• the matrix values presented above for LFL and LFR are designed to remove the center component of the input signals as the steering moves forward. We can show that if the input signal has been encoded forward with some kind of cross mixer, such as a stereo width control, the matrix elements given above (the 1989 elements of U.S. Patent No. 4,862,502, the July 1996 patent application elements, the version 1.11 decoder elements, and the new ones according to the present invention) all completely restore the original separation. The version 1.11 elements (with the level boost when cs is approximately 22.5) also restore the original separation.
• the solid curve shows the center attenuation needed for the new LFL and LFR if the energy of the center component of the input signal is to be preserved in the front three channels as steering increases toward the front.
• the dotted curve shows the center values for a standard decoder.
• the needed rise in the level of the center channel is quite steep — the rise is many dB of amplitude per dB of steering value. This steep change in amplitude is audible in practice.
• the relative balance of the center channel information in a popular recording is well preserved, if one is standing close to the center speaker the sudden changes in level can be annoying.
• the loudness of the center channel is extreme. We tested this curve and found the center balance to be excellent, but the front sound stage was dominated by the center loudspeaker, and left-right separation was minimal.
• center attenuation shown in figure 35 is derived assuming the matrix elements previously given for LFL and LFR. What if we used different elements? Specifically, do we need to be aggressive about removing the center component from the left and right front outputs?
• PLR (GP ⁇ 2+GF ⁇ 2) ⁇ (Lin ⁇ 2+Rin ⁇ 2) + (GP-GF) ⁇ 2 ⁇ Cin ⁇ 2 ...(192)
• PC GC ⁇ 2*(Lin ⁇ 2+Rin ⁇ 2) + 2 ⁇ GC ⁇ 2 ⁇ Cin ⁇ 2 ...(193)
• RATIO ((GP ⁇ 2 + GF ⁇ 2 + GC ⁇ 2) ⁇ (Lin ⁇ 2 + Rm ⁇ 2) + ((GP-GF) ⁇ 2
• Lin ⁇ 2 (Cin ⁇ 2/Lin ⁇ 2) ((gp(cs)-gf(cs)) ⁇ 2 + 2 ⁇ (gc(cs) ⁇ 2) + .5 ⁇ (cos(cs) - sin(cs)) ⁇ 2) / 2 ⁇ (gp(cs) ⁇ 2 + gc(cs) ⁇ 2 + gf(cs) ⁇ 2) + 1) ...(197)
• LFR left front right
• FIG. 38 shows the center left (CL) matrix element with the new center boost function GC(cs). Note the correction for panning along the boundary between left and center.
• FIG. 39 shows the levels of the center output and the left output as a signal pans from center to left. Note that with the correction the panning of the center, while not perfect, is reasonably close to the inverse of the left output. (The values on the cs axis are inverted).
• this encoder should be able to encode a 5.1 channel tape in a way that allows the encoded version to be decoded by a Logic 7 decoder according to the present invention with minimal inaccuracy.
• the encoded output should be stereo compatible —that is, it should sound as close as possible to a manual two-channel mix of the same material.
• One factor in this stereo compatibility should be that the output of the encoder, when played on a standard stereo system, should give identical perceived loudness for each sound source in an original five-channel mix.
• the apparent position of the sound source in stereo should also be as close as possible to the apparent position in the five channel original.
• the surround gain is gradually lowered 3dB to correspond to the European standard.
• a new architecture was developed to solve the problem with this tape. Although the encoding of this particular tape was only marginally improved, the new encoder is superior in its performance on a wide variety of difficult material.
• the original encoder was developed first as a passive encoder, and it performed reasonably well with a variety of input signals.
• the new encoder will also work in a passive mode, but is primarily intended to work as an active encoder (i.e. one in which the encoding depends on the types of signal presented to its inputs.)
• the active circuitry corrects several small errors inherent in the design. However, even without the active correction, the performance is better than that of the previously described encoder.
• the new encoder shown in block schematic form in FIG. 40, handles the left, center and right channels identically to the previous design, and identically to the Dolby encoder, providing that the center attenuation function fen in attenuator 302 is equal to 0.71 or -3dB.
• the left (L), center ⁇ and right (R) signals are presented to input terminals 50, 52, and 54 of the encoder circuitry, respectively.
• the left side (LS) and right side (RS) signals are presented to the input terminals 62 and 64 respectively.
• An additional signal LFE (for low frequency effects in a 5 + 1 mix) is applied to a new input terminal 370.
• the C and LFE signals pass through attenuator/gain elements 372, 374, respectively, where C is amplified by the factor fen and LFE by a factor of 2.0. These signals are each applied to both the summing circuits 278 and 282.
• the L signal is applied directly to summing circuit 278 and the R signal is similarly applied to summing circuit 282.
• the surround signals are also applied to these summing circuits, but only after some manipulation, which appears to be more complex than it really is.
• the functions fc() and fs() direct the surround channels either to a path (through phase shift elements 234 and 246) with a 90 degree phase shift relative to the front channels (which proceed through the phase shifters 286 and 288) or to a path with no relative phase shift.
• fc is one and fs is zero, so that the active path is through the 90 degree phase shifters.
• the LS signal passes unchanged through block 376 to attenuator 396, where it is multiplied by a 0.91 factor, then passes to the adder 406, where it is mixed with the cross-coupled RS signal from the attenuator 404 which has gain of -crx.
• the value of crx is typically 0.38. It controls the amount of negative cross feed for each surround channel.
• the signal then passes through a 90 degree phase shifter 234 and an adder 276, where it is mixed with the other signals from phase shifter 286 to this adder, and passes to the output terminal 44 as the "A" signal.
• the A and B outputs at terminals 44 and 46 respectively have an amplitude ratio of -.38/.91, which results in a steering angle of 22.5 degrees to the rear.
• the RS signal applied to terminal 64 similarly passes through attenuator 382 with unity gain to the inverting element 400, and then through an attenuator 402 with a gain of 0.91, as for the LS channel. This signal is then added to -crx times the unmodified RS signal in adder 408. As for the LS channel, the signal passes through a 90 degree phase shifter element 246 and thence to an adder element 280. The R, C and LFE signals after combination in summing circuit 282 pass through a phase shift element 288 into the adder 280, where they mix with the phase-shifted RS and cross-fed LS signals to provide the "B" output signal at terminal 46.
• the output level is unity, as the sum of the squares of 0.38 and 0.91 is one.
• the output of the encoder is simple when only one channel is driven, it becomes problematic when both surround inputs are driven at the same time. If we drive the LS and RS inputs with the same signal, a common practice in film, all the signals at the summing nodes are in phase, so the total level in the output is .38 + .91, or 1.29. This output is too strong by the factor of 1.29, or 2.2dB.
• Active circuitry (not shown, but similar to the active circuitry in the decoder) is included in the encoder to reduce the gain by the 2.2dB factor when this situation occurs, i.e. when the two surround channels are similar in amplitude and are in phase.
• the 3dB attenuation was carefully chosen through listening tests to produce a stereo compatible encoding. It was decided that the new encoder of the present invention should also incorporate this 3dB attenuation when classical music was being encoded, and that one could detect this condition through monitoring the relative levels of the front and surround channels in the encoder.
• a major function, therefore, of the function fc in the surround channels is to reduce the level of the surround channels in the output mix by 3dB when the surround channels are much softer than the front channels. Circuitry similar to that in the decoder is provided to compare the front and rear levels, and when the rear is less by 3dB, the value of fc is reduced to a maximum of 3dB. This maximum 3dB attenuation is reached when the rear channels are 8dB less strong than the front channels. This active circuit appears to work well. It makes the new encoder compatible with the European standard encoder for classical music. However, instruments which are intended to be strong in the rear channels are encoded with full level.
• the active circuits comprise elements for comparing the level and phase between the front and rear channels on each side, and for comparing the relative energy in the front and rear channels. These circuits are easily implemented in the form of log-ratio detectors and are well known to those skilled in the art. Dependent upon the outputs from these detectors, these active circuits
• Additional improvements to the encoder are likely to include a feature for the front channels such that when the two front channels are out of phase the encoder will not cause the decoder to place the sound in the rear, as at present, but will detect this condition and make the encoded output appear to be unsteered (i.e. a quadrature phase shift between the A and B channels will result.)

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• 1998
  • 1998-09-03 CN CNB988109131A patent/CN1214690C/zh not_active Expired - Lifetime
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