EP0971280A1 - Régulateur de tension et méthode pour la régulation de tension - Google Patents

Régulateur de tension et méthode pour la régulation de tension Download PDF

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Publication number
EP0971280A1
EP0971280A1 EP98401714A EP98401714A EP0971280A1 EP 0971280 A1 EP0971280 A1 EP 0971280A1 EP 98401714 A EP98401714 A EP 98401714A EP 98401714 A EP98401714 A EP 98401714A EP 0971280 A1 EP0971280 A1 EP 0971280A1
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EP
European Patent Office
Prior art keywords
power transistor
current
voltage
regulator
control terminal
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP98401714A
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German (de)
English (en)
Inventor
Ludovic Oddoart
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Freescale Semiconducteurs France SAS
Original Assignee
Motorola Semiconducteurs SA
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Motorola Semiconducteurs SA filed Critical Motorola Semiconducteurs SA
Priority to EP98401714A priority Critical patent/EP0971280A1/fr
Publication of EP0971280A1 publication Critical patent/EP0971280A1/fr
Withdrawn legal-status Critical Current

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    • G—PHYSICS
    • G05—CONTROLLING; REGULATING
    • G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
    • G05F1/00—Automatic systems in which deviations of an electric quantity from one or more predetermined values are detected at the output of the system and fed back to a device within the system to restore the detected quantity to its predetermined value or values, i.e. retroactive systems
    • G05F1/10—Regulating voltage or current
    • G05F1/46—Regulating voltage or current wherein the variable actually regulated by the final control device is dc
    • G05F1/56—Regulating voltage or current wherein the variable actually regulated by the final control device is dc using semiconductor devices in series with the load as final control devices
    • G05F1/565—Regulating voltage or current wherein the variable actually regulated by the final control device is dc using semiconductor devices in series with the load as final control devices sensing a condition of the system or its load in addition to means responsive to deviations in the output of the system, e.g. current, voltage, power factor
    • G—PHYSICS
    • G05—CONTROLLING; REGULATING
    • G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
    • G05F1/00—Automatic systems in which deviations of an electric quantity from one or more predetermined values are detected at the output of the system and fed back to a device within the system to restore the detected quantity to its predetermined value or values, i.e. retroactive systems
    • G05F1/10—Regulating voltage or current
    • G05F1/46—Regulating voltage or current wherein the variable actually regulated by the final control device is dc
    • G05F1/56—Regulating voltage or current wherein the variable actually regulated by the final control device is dc using semiconductor devices in series with the load as final control devices
    • G05F1/575—Regulating voltage or current wherein the variable actually regulated by the final control device is dc using semiconductor devices in series with the load as final control devices characterised by the feedback circuit

Definitions

  • the present invention relates to a voltage regulator and a method of regulating voltage, and in particular to a voltage regulator for a power amplifier of a portable communications device.
  • Voltage regulators for use with a power amplifier of a portable communications device such as a cellular telephone need to satisfy a number of important criteria. They must be able to generate the required output voltage quickly (i.e. they must have a short start-up time) without any significantly large overshoot. In addition, they must ideally be able to operate correctly when being supplied with a number of different supply voltages and when providing a number of different regulated output voltages.
  • the reason that external MOSFETs make the regulator particularly prone to overshoots is that the capacitance between the gate and source of the MOSFET (the gate-source capacitance, Cgs) varies with the voltage across the source and drain of the MOSFET (the drain-source voltage, Vds).
  • the amount of current which must be drawn away from the gate of the MOSFET in order to establish the correct voltage at the gate to generate the desired output voltage of the regulator varies with the gate-source capacitance Cgs and therefore also varies in dependence on the drain-source voltage, Vds. Since the regulator must be able to work with different supply voltages (e.g.
  • the regulator must be able to cater for the change in Cgs of the power MOSFET with different drain-source voltages (since the difference between the supply voltage and the output regulated voltage determines Vds). Furthermore, the relationship between Vds and Cgs is not even a straightforward linear relationship but a more complex inverse square relationship.
  • a voltage regulator for a power amplifier of a portable communications device comprising an operational amplifier and a power transistor connected together so as to perform a voltage regulation function with the output of the operational amplifier connected to a control terminal of the power transistor, wherein the power transistor has an associated capacitance such that the initial biasing of the power transistor requires a start-up current to be supplied to its control terminal to charge the associated capacitance, and wherein the voltage regulator additionally comprises start-up current generation means having a current node connected to the control terminal of the power transistor for supplying additional current to the control terminal of the power transistor during start-up of the regulator whereby the initial biasing of the power transistor is aided by the start-up current generation means.
  • a method of regulating voltage in a voltage regulator for a power amplifier of a portable communications device comprising an operational amplifier and a power transistor connected together so as to perform a voltage regulation function with the output of the operational amplifier connected to a control terminal of the power transistor, wherein the power transistor has an associated capacitance such that the initial biasing of the power transistor requires a start-up current to be supplied to its control terminal to charge the associated capacitance
  • the method comprising the steps of controlling the output voltage of the voltage regulator by supplying current to the control terminal of the power transistor, wherein the current is generated both by the op-amp and, during start-up of the regulator, by start-up current generation means having a current node connected to the control terminal of the power transistor for supplying additional current to the control terminal of the power transistor during start-up of the regulator whereby the initial biasing of the power transistor is aided by the start-up current generation means.
  • the start-up current generation means includes measurement means for measuring a property indicative of the associated capacitance of the power transistor, whereby the associated capacitance of the power transistor may be determined.
  • the start-up current generation means includes magnitude control means for varying the total amount of current supplied to the control terminal of the power transistor during start-up of the regulator in dependence upon the associated capacitance of the power transistor, whereby an appropriate current is supplied to the current node of the start-up current generation means for correctly biasing the power transistor during start-up of the regulator.
  • start-up current generation means preferably also includes current shaping means for shaping the current supplied to the control terminal of the power transistor such that the amount of current supplied by the start-up current generation means decays with time so as to minimise any overshoots.
  • the measurement means comprises means for measuring the voltage across the current terminals of the power transistor.
  • the power transistor is conveniently a P-channel Metal-Oxide Semiconductor Field Effect Transistor (P-MOSFET) having a gate (the control terminal) and a source and drain (the current terminals), there being a certain capacitance between the gate and the source (Cgs) which varies with the voltage applied across the drain and the source (Vds).
  • the measurement means of the start-up current generation means preferably comprises a P-MOSFET sense-transistor whose gate and source are connected to the drain and source of the power transistor respectively. In this way, a sense current which flows through the sense-transistor approximately mirrors Cgs of the power transistor, such that a suitable current for biasing the power transistor may be generated from the sense current.
  • the first is that the average power consumption of the voltage regulator may be greatly reduced since the power consumption of the operational amplifier may be greatly reduced.
  • the second advantage is that the amount of current required to correctly and quickly bias the power transistor with minimal overshoots is automatically generated without the requirement of an external RC circuit even where there are large variations in the supply voltage and/or the output regulated voltage.
  • a voltage regulator 1 for a power amplifier of a portable communications device is seen to comprise an operational amplifier (op-amp) 10, a power transistor 20 operating as the pass device of the voltage regulator, programmable feed-back means 30 and a de-coupling capacitor 40.
  • the op-amp 10 has a non-inverting input 11, an inverting input 12 and an output 13.
  • the power transistor 20, which in this case is a P-MOSFET, has a control terminal or gate 21, a first current terminal or source 22 and a second current terminal or drain 23.
  • a supply voltage input 2 to the regulator 1 is connected to the source 22, and a regulated voltage output 4 of the regulator 1 is connected to the drain 23 of the power transistor 20.
  • the regulator has a reference voltage, VBG, input 3 connected to the non-inverting input 11 of the op-amp 10.
  • the output 13 of the op-amp 10 is connected to the gate 21 of the power transistor 20.
  • the drain 23 of the power transistor 20 is connected to the inverting input 12 of the op-amp 10 via programmable feedback means 30; note that programmable feedback means 30 is simply a programmable voltage divider for outputting a divided down voltage from that input to it, the divisor value, B, being programmable.
  • programmable feedback means 30 is simply a programmable voltage divider for outputting a divided down voltage from that input to it, the divisor value, B, being programmable.
  • Such a programmable feedback means generally comprises a simple resistor ladder structure, one or more resistances of which may be bypassed or not by switching one or more bypass transistors on or off. Clearly by altering the ratio of resistances above and below the inverting input 12, the divisor B may be varied.
  • Decoupling capacitor 40 is connected between ground and the drain 23 of the power transistor 20.
  • the output 13 of the op-amp 10 adopts a high impedance state and the capacitance Cgs between the gate 21 and source 22 is not charged resulting in a substantially zero biasing voltage Vgs between the gate and source of the power transistor 20, which in turn results in the transistor adopting an off state.
  • decoupling capacitor 40 is also not charged and the voltage, Vreg, at the regulated voltage output 4 will be at ground regardless of the voltage, Vsup, applied to the supply voltage input 2 (note where, for example, Vsup is provided by one or more batteries, Vsup will be constantly applied to the Vsup input 2 of the regulator 1).
  • the op-amp 10 When the regulator 1 is turned on, operating current is supplied to the op-amp 10 which compares the voltages applied to its non-inverting and inverting inputs 11, 12, and attempts to produce at its output a voltage given by the difference between these input voltages multiplied by the gain of the amplifier, which in the present case is a large negative value. Since the voltage at the output 4 of the regulator 1 is initially substantially at ground, the voltage at input 12 will be lower than the positive reference voltage VBG applied to input 11 and as a consequence the voltage at output 13 will also attempt to go down (note that unless the input voltages are very almost equal, the output of the op-amp will be saturated, i.e. it will simply attempt to adopt the maximum or minimum voltage available to it from its power supply).
  • the effect of the output 13 going low is to commence charging of the associated capacitance Cgs of transistor 10.
  • the rate of charging is limited by the amount of current that the op-amp 20 is able to sink and this in turn depends upon the amount of operating current that is supplied to the op-amp.
  • the capacitance Cgs charges, the voltage at the gate decreases and therefore the power transistor 20 starts to turn on, enabling current flow through the transistor 20 and reducing the voltage drop thereacross.
  • the decoupling capacitor will start to charge up and the voltage at the regulated voltage output 4 of the regulator will increase.
  • the inverting and non-inverting inputs 12, 11 will be substantially equal and the regulator will be in a state of equilibrium in which the regulated voltage output 4 of the regulator is at the desired value.
  • the regulator 1 may be described as being in an on state, and the power transistor 10 as being correctly biased.
  • FIG. 2 there is shown a voltage regulator 100 for a power amplifier of a portable communications device, according to a preferred embodiment of the present invention.
  • the basic architecture of regulator 100 is the same as that of regulator 1 of Figure 1, except that regulator 100 includes start-up current generation means 200 which is described in greater detail below.
  • the other significant difference between regulators 1 and 100 is that op-amp 110 of regulator 100 is a much smaller op-amp than op-amp 10 of regulator 1 in terms of the amount of power which it consumes when it is on.
  • Op-amp 100 has a non-inverting input 111, an inverting input 112 and an output 113 connected in the same way as the corresponding elements of op-amp 10.
  • Like references have been used to describe corresponding elements in regulators 1 and 100 of Figures 1 and 2.
  • the start-up current generation means 200 comprises a sense transistor 210 which in this case is shown as a P-MOSFET, a 1:X current mirror 220, first and second voltage generating resistors 245, 255, a band gap follower 230, first and second comparators 240, 250 and first and second start-up transistors 270, 280.
  • Sense transistor 210 has a source 212 connected to the source 22 of the power transistor 20, a gate 211 connected to the drain 23 of the power transistor 20 and a drain 213 connected to master input(s) 221, 222 of the current mirror 220.
  • First voltage generating resistor 245 is connected between a slave output 223 of the current mirror 220 and the output 232 of the band gap follower circuit 230.
  • the input 231 of the band gap follower circuit 230 is connected to the reference voltage VBG.
  • the second voltage generating resistor 255 is connected between the output 232 of the band gap follower 230 and ground via a small current source 257.
  • First and second start-up reference voltage nodes 246, 256 are located downstream of the first and second voltage generating resistors 245, 255 (i.e. the resistors are between the band gap follower 230 and their respective start-up reference nodes).
  • First and second comparators 240, 250 each have a non-inverting input 241, 251 connected to the first and second start-up reference voltage nodes 246, 256 respectively and an inverting input 242, 252, both of which are connected to the inverting input 112 of op-amp 110 so as to receive the same input voltage, NEGIN.
  • Each output 243, 253 of the comparators is connected to the control terminal 271, 281 of the first and second start-up transistors 270, 280 respectively.
  • the current electrodes 272,273; 282,283 of the start-up transistors are connected between a current node 300 and ground respectively.
  • First and second start-up transistors 270, 280 have different physical characteristics (e.g.
  • Current node 300 is connected to the control electrode of power transistor 20 (between the control electrode of the power transistor 20 and the output 113 of op-amp 110.
  • op-amp 110 When it is desired to switch on regulator 100, operating power is supplied to op-amp 110, current mirror 220, band gap follower 230, comparators 240, 250 and current source 257 in order to switch these elements on. Thereafter, the basic regulator structure operates as normal to drive the output 113 of op-amp 110 low which attempts to bias the power transistor into the correctly biased conducting on-state.
  • op-amp 110 only consumes a small amount of power (and therefore effectively operates only as a low power comparator), it is not able to draw very much current away from the associated capacitance Cgs of the transistor. If it were not for the start-up current generation means 200, the operation of which is described below, this would lead to a very slow turn-on time of the regulator.
  • the start-up current generation means 200 also operates as follows.
  • This small current IE through resistor 255 is used to create a small voltage offset E in order to avoid a second regulation loop once the output voltage of the regulator is established.
  • the voltages VBGvar and VBG-E generated in this way are then compared by the first and second comparators respectively with NEGIN which is the voltage at the output of the feedback means 30.
  • VBG-E is only a little lower than VBG
  • the second comparator 250 will be on for almost all of the start-up time until NEGIN has very almost reached its final value of VBG.
  • the first comparator will remain on for a proportion of the start-up time which depends on how small Isense is.
  • the comparators 240,250 While the comparators 240,250 are on they switch on start-up transistors 270,280 respectively, which in turn causes first and second start-up currents I 1 and I 2 to flow from the current node 300.
  • the physical characteristics of the start-up transistors 270 and 280 are predetermined such that I 2 is a small value, while I 1 is a much larger value, the arrangement being such that in the worst case of the lowest anticipated value of Vds (corresponding to the largest value of Cgs) of the power transistor, the capacitance Cgs is correctly biased within an acceptably short period, in the knowledge that I 1 will be on for almost all of the start-up period.
  • the desired magnitude of I 2 is calculated on the basis of the lowest anticipated value of Cgs, and then the appropriate size for transistor 280 is determined accordingly.
  • the desired magnitude of I 1 is calculated on the basis of the largest anticipated value of Cgs, and then the appropriate size for transistor 270 is set accordingly.
  • FIGS 3 and 4 illustrate the start-up of regulator 100 under two extreme conditions of a large value of Vsup-Vreg and a low value of Vsup-Vreg respectively. It will be apparent from the above discussion and Figures 3 and 4 that the total current supplied to (or actually in this case drawn away from) the current node 300 in order to bias the power transistor 20 during start-up of the regulator depends upon the capacitance Cgs of the power transistor such that almost exactly the correct amount of current is drawn from the capacitance to ensure very quickly reaching the correct bias.
  • the amount of current drawn is initially large but reduces as the correct bias is approached; this ensures that very little overshoot occurs which is particularly important where the power amplifier supplied by this voltage regulator is formed using Gallium Arsenide which is preferred for power amplifiers used in portable communications devices.
  • the start-up current generation means 200 includes measurement means, in the form of sense transistor 210 for measuring a property indicative of the associated capacitance Cgs of the power transistor 20, whereby the associated capacitance of the power transistor may be determined and an appropriate current supplied to the current node 300 of the start-up current generation means 200 for correctly biasing the power transistor 20 during start-up of the regulator 100.
  • the start-up current generation means is preferably switched off again by suitable means (not shown).
  • suitable means not shown.
  • the start-up current generation means 200 is left on while the regulator is on the total power consumed by all of the elements 220, 230, 240, 245, 250, 255, 257 will be small compared with the power consumed by the op-amp of conventional regulator 1 of Figure 1. This is because the majority of the start-up current in regulator 100 is provided by start-up transistors 270, 280 which are turned off after start-up of the regulator. The power controlled by these transistors during start-up is much more than the total power consumed by all of the other elements of the regulator 100.

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  • Engineering & Computer Science (AREA)
  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
  • General Physics & Mathematics (AREA)
  • Radar, Positioning & Navigation (AREA)
  • Automation & Control Theory (AREA)
  • Continuous-Control Power Sources That Use Transistors (AREA)
EP98401714A 1998-07-07 1998-07-07 Régulateur de tension et méthode pour la régulation de tension Withdrawn EP0971280A1 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
EP98401714A EP0971280A1 (fr) 1998-07-07 1998-07-07 Régulateur de tension et méthode pour la régulation de tension

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EP98401714A EP0971280A1 (fr) 1998-07-07 1998-07-07 Régulateur de tension et méthode pour la régulation de tension

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Cited By (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
FR2807847A1 (fr) * 2000-04-12 2001-10-19 St Microelectronics Sa Regulateur lineaire a faible surtension en regime transitoire
EP1186983A2 (fr) * 2000-08-31 2002-03-13 STMicroelectronics S.r.l. Contrôleur de bande interdite à commutation
EP1231529A1 (fr) * 2001-02-09 2002-08-14 Atmel Nantes Sa Dispositif générateur d'une tension de référence précise
WO2005078932A1 (fr) * 2004-01-15 2005-08-25 Koninklijke Philips Electronics N.V. Circuit d'excursion haute
FR2872305A1 (fr) * 2004-06-24 2005-12-30 St Microelectronics Sa Procede de controle du fonctionnement d'un regulateur a faible chute de tension et circuit integre correspondant
CN102117089B (zh) * 2009-12-31 2013-04-17 财团法人工业技术研究院 低压降稳压器

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0731553A2 (fr) * 1991-09-30 1996-09-11 Siemens Aktiengesellschaft Suppresseur d'harmoniques actif
EP0749059A2 (fr) * 1995-06-14 1996-12-18 Philips Patentverwaltung GmbH Borne de contact de télécommunication avec régulateur de tension

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0731553A2 (fr) * 1991-09-30 1996-09-11 Siemens Aktiengesellschaft Suppresseur d'harmoniques actif
EP0749059A2 (fr) * 1995-06-14 1996-12-18 Philips Patentverwaltung GmbH Borne de contact de télécommunication avec régulateur de tension

Non-Patent Citations (1)

* Cited by examiner, † Cited by third party
Title
KMETZ G L: "SOFT-START REGULATOR STARTS AT OV", EDN ELECTRICAL DESIGN NEWS, vol. 41, no. 12, 6 June 1996 (1996-06-06), pages 104, 106, XP000622006 *

Cited By (13)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US6388433B2 (en) 2000-04-12 2002-05-14 Stmicroelectronics Linear regulator with low overshooting in transient state
EP1148405A1 (fr) * 2000-04-12 2001-10-24 STMicroelectronics Régulateur linéaire à faible surtension en régime transitoire
FR2807847A1 (fr) * 2000-04-12 2001-10-19 St Microelectronics Sa Regulateur lineaire a faible surtension en regime transitoire
EP1186983A3 (fr) * 2000-08-31 2003-11-12 STMicroelectronics S.r.l. Contrôleur de bande interdite à commutation
EP1186983A2 (fr) * 2000-08-31 2002-03-13 STMicroelectronics S.r.l. Contrôleur de bande interdite à commutation
EP1231529A1 (fr) * 2001-02-09 2002-08-14 Atmel Nantes Sa Dispositif générateur d'une tension de référence précise
FR2820904A1 (fr) * 2001-02-09 2002-08-16 Atmel Nantes Sa Dispositif generateur d'une tension de reference precise
US6650175B2 (en) 2001-02-09 2003-11-18 Atmel Nantes S.A. Device generating a precise reference voltage
WO2005078932A1 (fr) * 2004-01-15 2005-08-25 Koninklijke Philips Electronics N.V. Circuit d'excursion haute
JP2007518179A (ja) * 2004-01-15 2007-07-05 コーニンクレッカ フィリップス エレクトロニクス エヌ ヴィ プルアップ回路
FR2872305A1 (fr) * 2004-06-24 2005-12-30 St Microelectronics Sa Procede de controle du fonctionnement d'un regulateur a faible chute de tension et circuit integre correspondant
US7453249B2 (en) 2004-06-24 2008-11-18 Stmicroelectronics Sa Method for controlling the operation of a low-dropout voltage regulator and corresponding integrated circuit
CN102117089B (zh) * 2009-12-31 2013-04-17 财团法人工业技术研究院 低压降稳压器

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