EP0786921B1 - Digital demodulator - Google Patents

Digital demodulator Download PDF

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Publication number
EP0786921B1
EP0786921B1 EP96101105A EP96101105A EP0786921B1 EP 0786921 B1 EP0786921 B1 EP 0786921B1 EP 96101105 A EP96101105 A EP 96101105A EP 96101105 A EP96101105 A EP 96101105A EP 0786921 B1 EP0786921 B1 EP 0786921B1
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Prior art keywords
signal
phase
quadrature
phase signal
demodulator
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EP96101105A
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German (de)
French (fr)
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EP0786921A1 (en
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F.O. Dr. Witte
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TDK Micronas GmbH
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TDK Micronas GmbH
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Priority to DE59609450T priority Critical patent/DE59609450D1/en
Priority to EP96101105A priority patent/EP0786921B1/en
Priority to US08/792,924 priority patent/US5767739A/en
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04S—STEREOPHONIC SYSTEMS 
    • H04S1/00—Two-channel systems
    • H04S1/007—Two-channel systems in which the audio signals are in digital form

Definitions

  • the invention relates to a digital demodulator for a quadrature-modulated signal, which transmits a combination signal by means of amplitude and phase modulation.
  • Quadrature modulated signals are often used when in one Signals belonging to the transmission channel, but which are independent of one another are to be transferred.
  • One such application is the transmission of Stereo signals according to the C-QUAM standard, in which a sum signal over the Amplitude modulation and a difference signal as well as a pilot tone over the Phase modulation of the respective carrier is transmitted.
  • An example of one associated digital demodulator is published in DE 43 40 012 A1 described.
  • a quadrature signal source forms from the received one quadrature-modulated signal using a quadrature mixer, an in-phase signal and a Quadrature phase signal. Digitization can take place before or after Quadrature mixer done.
  • a coordinate converter By means of a coordinate converter, in particular according to The Cordic algorithm works from the digitized in-phase signal and the digitized quadrature phase signal, a magnitude signal and a phase signal are formed.
  • a control loop controlled by the phase signal controls the oscillator frequency of the Quadrature mixer exactly to the value of the carrier frequency, so that the in-phase signal and the quadrature phase signal are transformed into the baseband.
  • a remaining average phase deviation is corrected by the control loop also intervenes in the phase signal and adds or subtracts a correction signal there, that pulls the temporal average of the phase signal to the zero phase value.
  • Decoder which essentially contains a known stereo matrix, forms from the Amount signal and the phase signal, the left and right signal as well as the Pilot signal at 25 Hz.
  • the object of the invention is to provide an improved digital demodulator for such Quadrature modulated signals indicate the better of digital signal processing is adapted and places less demands on the quadrature signal source.
  • the main advantage of this arrangement is that the output signals of the Quadrature signal source, the digitized in-phase signal and the digitized Quadrature phase signal do not have to have the exact baseband position, but only must be in a relatively low frequency range.
  • the range of this deep Frequency range depends on the digitization frequency and should be as possible not be greater than a tenth of the digitization frequency.
  • This cheap Boundary conditions allow that a digital quadrature mixer in the simplest way can be realized by digital switchers because the quadrature modulated digital signal only to be multiplied by the values +1, -1 and 0.
  • the first control loop is advantageously controlled via the slope of the first Phase signal resulting from the difference between at least two in time neighboring samples.
  • control loops have one Integrator included. Accumulator loops are particularly suitable for this sufficient job capacity so that there is no overflow in normal operation.
  • control signal of the first and / or second control loop is of this type is designed that it as an additive or subtractive correction signal via a Adding circuit can be combined with the respective phase signal.
  • the two control signals can be added combine so that only a single adder for correction in the phase signal path is required.
  • the integrator for the first and second Control loop can be formed together by the two control signals in the adder Accumulator circuit are supplied. Its output then provides the common one Actuating signal.
  • the modification device corresponds to a predetermined signal characteristic, the inverse to the signal characteristic on the transmitter side.
  • the modification device can be a have nonlinear characteristics, e.g. is on the C-QUAM standard as a characteristic a tangent curve is prescribed for the receiver side.
  • the tangent course can be defined by a memory table or by a polynomial approximation as in the DE 43 40 012 already mentioned.
  • an input stage 1 receives quadrature modulated signal sq from an antenna, cable or other Facility.
  • a quadrature signal source 2 with connected oscillator 2.1 which as Mixing signal sx emits a digital signal with a predetermined frequency fx an in-phase signal I and a from the quadrature-modulated signal sq Quadrature phase signal Q, both signals I and Q being digitized.
  • the Digitization can be in the quadrature signal source 2 or already in the Entry level 1 take place.
  • C-QUAM Compact - Quadrature Amplitude Modulation
  • the amount of this pointer should always assume the value 1 + S, the value 1 representing the carrier with a constant size.
  • the size of the difference signal D affects only the phase position of the pointer M (t).
  • a pilot tone P with a frequency of 25 Hz is also applied to the difference signal D.
  • 5% modulation modulated the stereo detection and thus an automatic Stereo switching enabled.
  • the quadrature signal source 2 is followed by a coordinate converter 3 which from the in-phase signal I and the quadrature phase signal Q a magnitude signal b and a forms first phase signal p1.
  • the coordinate converter 3 carries out a conversion from Cartesian coordinates in polar coordinates. Is suitable for this implementation in particular the well-known Cordic algorithm, which uses a iterative approximation determined with arbitrary precision.
  • the quadrature mixing it is not necessary for the quadrature mixing to take place directly in the baseband. If the in-phase signal I and the quadrature phase signal Q are sampled at a frequency of 19 kHz, then it is sufficient for the demodulation according to the invention if the remaining rotational frequency ⁇ r of the complex pointer M (t) remains less than 2 kHz.
  • the difference between the mixing frequency fx and the carrier frequency f results in a residual frequency fr and thus a remaining rotational frequency ⁇ r of the complex pointer M (t).
  • the first phase signal p1 is not constant but increases or decreases constantly over time, cf. also Fig. 2.
  • This corresponds to a constant offset frequency ⁇ r which is brought to the value zero by means of a first control circuit 4, in that the mean slope mt of the first phase signal p1 is compensated by a first actuating signal c1 with an equally large negative slope.
  • the actuating signal c1 is added to the first phase signal p1 by means of a first adder 5 and thus forms a second phase signal p2, cf. also FIG. 3.
  • FIG. 3 In FIG.
  • the slope is formed by a difference former 6 from two successive sample values, which are then weighted and / or averaged by means of a first filter device 7.
  • the output of the first filter device 7 is integrated by means of an integrator 8, the output of which supplies the first actuating signal c1 to the first adder 5.
  • the difference generator 6 consists of a first retarder 6.1 and a subtractor 6.2.
  • the integrator 8 consists of an accumulator loop with a second adder 8.1 and a second delay 8.2.
  • the output signals of the two control loops 4, 9 are fed to the second adder 8.1 as inverted signals, so that the control direction in the first adder 5 is correct.
  • the compensation of the mean slope mp does not cause the second Phase signal p2 comes to lie exactly on the phase reference value on average over time.
  • the time average tm of the second phase signal p2 is easy in FIG. 3 rising straight line shown below the zero phase reference axis.
  • a second control loop 9 becomes the time average tm of the second phase signal p2 brought exactly to the zero phase reference axis. This is done by means of a second Filter device 10 and the integrator 8 achieved by the output signal of the first Adders 5 directly or via a modification device 11 to the input of the second Filter device 10 is supplied, at the output of a further input of the Integrators 8 lies.
  • the second control circuit 9 forms a second control signal c2, that by means of the first adder 5 additive / subtractive to the first Phase signal p1 and a first control signal c1 is added and thus forms a third phase signal p3 that is correct in terms of its slope and phase on average over time.
  • phase signal p2 supplies the input signal of the second Control loop.
  • the current deviations of the third phase signal p3 from the Zero phase reference position thus only correspond to the searched differential signal D and the pilot signal P.
  • the magnitude signals b and the third phase signal p3 the sought components L, R, P of the stereo combination signal educated.
  • the The third phase signal p3 is previously modified using the modification device 11, for example, by determining the associated tangent value. Because in the amount signal b the carrier amplitude is included, that is for the stereo matrix in the decoder 12 third phase signal p3 or the modified phase signal p3 'on the carrier amplitude normalized. This is done by means of a multiplier 13, the first input of which with the Magnitude signal b and its second input with the third phase signal p3 or p3 ' is fed.
  • the remaining rotation frequency ⁇ r of the complex pointer M (t) corresponds to a steady increase mp in the middle phase mp1, which is represented by a sawtooth-shaped, solid line.
  • the first phase signal p1 is preferably represented as a two's complement number whose lower or upper value limit corresponds to the phase angle - ⁇ or + ⁇ .
  • the steadily increasing phase mp1 thus jumps back from the phase value + ⁇ to the phase value - ⁇ .
  • the coupling of the respective phase value to the two's complement number representation has the great advantage that phase difference values are reproduced correctly, even if the phase has meanwhile overflowed.
  • the dashed area around the middle phase mp1 indicates the area in which the first phase signal p1 can be due to the modulation with the difference signal D and the pilot signal P.
  • phase 3 schematically shows the time profile of the second phase signal p2, which is obtained by a phase correction by means of the first control loop 4.
  • the middle phase mp2 has at most a very slight slope tm however, middle phase mp2 is not on the zero phase reference axis as required - at random, at best.
  • the second phase corrects the zero phase position Control circuit 9, which also suppresses the slight remaining slope tm.
  • the instantaneous phase of the second phase signal p2 is dashed phase range shown around the middle phase mp2.
  • the modulation vector M (t) rotating with the frequency ⁇ is shown in a complex pointer representation.
  • the modulation components 1 + S and D define the instantaneous amplitude and phase ⁇ of the pointer compared to a reference pointer rotating with a constant amplitude and a constant frequency.
  • the revolving reference pointer specifies the reference phase via the in-phase signal I.
  • the quadrature phase signal Q is perpendicular to this. From these two signals I, Q, the coordinate converter 3 determines the current length 1 + S and the current phase ⁇ of the pointer M (t).
  • the pointer representation is independent of the rotation frequency ⁇ . This representation applies both to the high-frequency quadrature signal sq and to the quadrature components I, Q, the associated reference pointer of which rotates at the low rotation frequency ⁇ r .
  • the implementation of the demodulator according to the invention can be done as a program in a processor, particularly in a monolithically integrated circuit, or as Circuit or in mixed form. It is irrelevant how the individual Functional units are realized in detail and whether the functional units are also different Serve purposes.

Description

Die Erfindung betrifft einen digitalen Demodulator für ein quadraturmoduliertes Signal, das mittels einer Amplituden- und Phasenmodulation ein Kombinationssignal überträgt.The invention relates to a digital demodulator for a quadrature-modulated signal, which transmits a combination signal by means of amplitude and phase modulation.

Quadraturmodulierte Signale werden gerne angewendet, wenn in einem Übertragungskanal zusammengehörige Signale, die aber unabhängig voneinander sein sollen, zu übertragen sind. Eine derartige Anwendung ist die Übertragung von Stereosignalen nach dem C-QUAM-Standard, bei der ein Summensignal über die Amplitudenmodulation und ein Differenzsignal sowie ein Pilotton über die Phasenmodulation des jeweiligen Trägers übertragen wird. Ein Beispiel für einen zugehörigen digitalen Demodulator ist in der Offenlegungschrift DE 43 40 012 A1 beschrieben. Eine Quadratursignalquelle bildet aus dem empfangenen quadraturmodulierten Signal mittels eines Quadraturmischers ein Inphasensignal und ein Quadraturphasensignal. Die Digitalisierung kann dabei vor oder nach dem Quadraturmischer erfolgen. Mittels eines Koordinatenumsetzers, der insbesondere nach dem Cordic-Algorithmus arbeitet, wird aus dem digitalisierten Inphasensignal und dem digitalisierten Quadraturphasensignal ein Betragssignal und ein Phasensignal gebildet. Ein vom Phasensignal gesteuerter Regelkreis steuert die Oszillatorfrequenz des Quadraturmischers exakt auf den Wert der Trägerfrequenz, so daß das Inphasensignal und das Quadraturphasensignal in das Basisband transformiert werden. Eine verbleibende mittlere Phasenabweichung wird dadurch korrigiert, daß der Regelkreis auch in das Phasensignal eingreift und dort ein Korrektursignal addiert oder subtrahiert, das den zeitlichen Mittelwert des Phasensignals auf den Nullphasenwert zieht. Ein Dekodierer, der im wesentlichen eine bekannte Stereomatrix enthält, bildet aus dem Betragssignal und dem Phasensignal das gesuchte Links- und Rechtssignal sowie das Pilotsignal bei 25 Hz.Quadrature modulated signals are often used when in one Signals belonging to the transmission channel, but which are independent of one another are to be transferred. One such application is the transmission of Stereo signals according to the C-QUAM standard, in which a sum signal over the Amplitude modulation and a difference signal as well as a pilot tone over the Phase modulation of the respective carrier is transmitted. An example of one associated digital demodulator is published in DE 43 40 012 A1 described. A quadrature signal source forms from the received one quadrature-modulated signal using a quadrature mixer, an in-phase signal and a Quadrature phase signal. Digitization can take place before or after Quadrature mixer done. By means of a coordinate converter, in particular according to The Cordic algorithm works from the digitized in-phase signal and the digitized quadrature phase signal, a magnitude signal and a phase signal are formed. A control loop controlled by the phase signal controls the oscillator frequency of the Quadrature mixer exactly to the value of the carrier frequency, so that the in-phase signal and the quadrature phase signal are transformed into the baseband. A remaining average phase deviation is corrected by the control loop also intervenes in the phase signal and adds or subtracts a correction signal there, that pulls the temporal average of the phase signal to the zero phase value. On Decoder, which essentially contains a known stereo matrix, forms from the Amount signal and the phase signal, the left and right signal as well as the Pilot signal at 25 Hz.

Aufgabe der Erfindung ist es, einen verbesserten digitalen Demodulator für derartige quadraturmodulierte Signale anzugeben, der besser an die digitale Signalverarbeitung angepaßt ist und geringere Anforderungen an die Quadratursignalquelle stellt.The object of the invention is to provide an improved digital demodulator for such Quadrature modulated signals indicate the better of digital signal processing is adapted and places less demands on the quadrature signal source.

Die Aufgabe wird durch die Erfindung entsprechend den Merkmalen des Anspruchs 1 wie folgt gelöst:

  • eine Quadratursignalquelle, die abhängig vom empfangenen quadraturmodulierten Signal ein digitalisiertes Inphasensignal und ein digitalisiertes Quadraturphasensignal in tiefer Frequenzlage liefert, wobei die Bandbreite des tiefen Frequenzbereiches nicht größer ist als ein Zehntel der Digitalisierungsfrequenz,
  • ein Koordinatenumsetzer, der aus dem digitalisierten Inphasensignal und dem digitalisierten Quadraturphasensignal ein Betragssignal und ein erstes Phasensignal bildet,
  • ein dem Koordinatenumsetzer nachgeschalteter erster Regelkreis, der die Steigung des ersten Phasensignals im zeitlichen Mittel auf den Wert Null oder einen Restwert regelt und damit ein zweites Phasensignal bildet,
  • ein dem Koordinatenumsetzer nachgeschalteter zweiter Regelkreis, der den zeitlichen Mittelwert des zweiten Phasensignals auf einen Phasenbezugswert, insbesondere eine Nullphasenlage, regelt und damit ein drittes Phasensignal bildet, und
  • ein Dekodierer, der aus dem Betragssignal und dem dritten Phasensignal mindestens eine digitalisierte Komponente des Kombinationssignals bildet.
The object is achieved by the invention according to the features of claim 1 as follows:
  • a quadrature signal source which, depending on the received quadrature-modulated signal, supplies a digitized in-phase signal and a digitized quadrature-phase signal in the low frequency range, the bandwidth of the low frequency range being no greater than a tenth of the digitization frequency,
  • a coordinate converter which forms a magnitude signal and a first phase signal from the digitized in-phase signal and the digitized quadrature phase signal,
  • a first control circuit downstream of the coordinate converter, which regulates the slope of the first phase signal to the value zero or a residual value on average over time and thus forms a second phase signal,
  • a second control circuit connected downstream of the coordinate converter, which regulates the time average of the second phase signal to a phase reference value, in particular a zero phase position, and thus forms a third phase signal, and
  • a decoder which forms at least one digitized component of the combination signal from the magnitude signal and the third phase signal.

Der wesentliche Vorteil dieser Anordnung besteht darin, daß die Ausgangssignale der Quadratursignalquelle, das digitalsierte Inphasensignal und das digitalisierte Quadraturphasensignal nicht die exakte Basisbandlage aufweisen müssen, sondern nur in einem relativ tiefen Frequenzbereich liegen müssen. Die Bandbreite dieses tiefen Frequenzbereiches hängt von der Digitalisierungsfrequenz ab und soll dabei möglichst nicht größer sein als ein Zehntel der Digitalisierungsfrequenz. Diese günstigen Randbedingungen erlauben, daß ein digitaler Quadraturmischer auf einfachste Weise durch digitale Umschalter zu realisieren ist, weil das quadraturmodulierte Digitalsignal lediglich mit den Werten +1, -1 und 0 zu multiplizieren ist. Bei einer exakten Transformation des quadraturmodulierten Digitalsignals in das Basisband wäre eine exakte Frequenzanpassung des digitalen Mischungssignals erforderlich, was nur über eine sehr aufwendige Sinus- und/oder Cosinus-Tabelle mit zwei aufwendigen digitalen Multiplizierern zu realisieren wäre. Eine analoge Ausführung des Quadraturmischers mit nachfolgender Digitalisierung des Inphasen- und Quadraturphasensignals ist natürlich ebenfalls möglich, wobei nach der Erfindung die Oszillatorfrequenz nicht nachgeregelt werden muß und daher unkritisch gegenüber der Frequenzlage und Driftänderungen ist. Die Erfindung vermeidet somit eine phasenstarre Nachführung des Quadraturmischers, aufwendige Sinus- und/oder Cosinus-Tabellen und aufwendige Multiplizierer bei der Quadraturmischung.The main advantage of this arrangement is that the output signals of the Quadrature signal source, the digitized in-phase signal and the digitized Quadrature phase signal do not have to have the exact baseband position, but only must be in a relatively low frequency range. The range of this deep Frequency range depends on the digitization frequency and should be as possible not be greater than a tenth of the digitization frequency. This cheap Boundary conditions allow that a digital quadrature mixer in the simplest way can be realized by digital switchers because the quadrature modulated digital signal only to be multiplied by the values +1, -1 and 0. With an exact Transforming the quadrature modulated digital signal into the baseband would be one exact frequency adjustment of the digital mix signal required, which is only about a very complex sine and / or cosine table with two complex digital Multipliers would be realized. An analog version of the quadrature mixer with subsequent digitization of the in-phase and quadrature-phase signal is natural likewise possible, the oscillator frequency not being readjusted according to the invention must be and is therefore not critical to the frequency situation and drift changes. The invention thus avoids a phase-rigid tracking of the quadrature mixer, complex sine and / or cosine tables and complex multipliers in the Quadrature mixing.

Eine vorteilhafte Ansteuerung des ersten Regelkreises erfolgt über die Steigung des ersten Phasensignals, die sich aus der Differenzbildung zwischen mindestens zwei zeitlich benachbarten Abtastwerten ergibt. Dies schließt selbstverständlich mit ein, daß weitere Abtastwerte zur Differenzbildung erfaßt werden können, wobei eine bessere Mittelwertbildung erreicht wird und Störgrößen besser unterdrückt werden können.The first control loop is advantageously controlled via the slope of the first Phase signal resulting from the difference between at least two in time neighboring samples. Of course, this includes that others Samples for difference formation can be acquired, with a better one Averaging is achieved and disturbances can be suppressed better.

Für die Genauigkeit der Regelung ist es ferner zweckmäßig, wenn die Regelkreise einen Integrator enthalten. Hierfür eignen sich insbesondere Akkumulatorschleifen mit genügender Stellenkapazität, damit im Normalbetrieb kein Überlauf stattfindet.For the accuracy of the control, it is also expedient if the control loops have one Integrator included. Accumulator loops are particularly suitable for this sufficient job capacity so that there is no overflow in normal operation.

Von Vorteil ist, wenn das Stellsignal des ersten und/oder zweiten Regelkreises derart ausgebildet ist, daß es als additives oder subtraktives Korrektursignal über eine Addierschaltung mit dem jeweiligen Phasensignal kombiniert werden kann. Bei geeigneter Ausführung der beiden Regelkreise lassen sich die beiden Stellsignale additiv kombinieren, so daß nur ein einziger Addierer zur Korrektur im Phasensignalpfad erforderlich ist. In gleicher Weise kann der Integrator für den ersten und zweiten Regelkreis gemeinsam ausgebildet sein, indem die beiden Stellsignale dem Addierer im Akkumulatorkreis zugeführt sind. Dessen Ausgang liefert dann das gemeinsame Stellsignal.It is advantageous if the control signal of the first and / or second control loop is of this type is designed that it as an additive or subtractive correction signal via a Adding circuit can be combined with the respective phase signal. at If the two control loops are designed appropriately, the two control signals can be added combine so that only a single adder for correction in the phase signal path is required. In the same way, the integrator for the first and second Control loop can be formed together by the two control signals in the adder Accumulator circuit are supplied. Its output then provides the common one Actuating signal.

Für die jeweilige Übertragungsnorm kann es erforderlich sein, daß das dritte Phasensignal vor dem Dekodierer mittels einer Modifizierungseinrichtung zu modifizieren ist. Die Modifizierungseinrichtung entspricht einer vorgegebenen Signalkennlinie, die invers zur Signalkennlinie auf der Senderseite ist. Die Modifizierungseinrichtung kann dabei eine nichtlineare Kennlinie aufweisen, z.B. ist bei dem C-QUAM-Standard als Kennlinie auf der Empfängerseite ein Tangensverlauf vorgeschrieben. Der Tangensverlauf kann dabei durch eine Speichertabelle definiert sein oder durch eine Polynomannäherung wie in der bereits genannten DE 43 40 012.It may be necessary for the respective transmission standard that the third Modify phase signal before the decoder by means of a modification device is. The modification device corresponds to a predetermined signal characteristic, the inverse to the signal characteristic on the transmitter side. The modification device can be a have nonlinear characteristics, e.g. is on the C-QUAM standard as a characteristic a tangent curve is prescribed for the receiver side. The tangent course can be defined by a memory table or by a polynomial approximation as in the DE 43 40 012 already mentioned.

Die Erfindung und vorteilhafte Ausgestaltungen werden nun anhand der Zeichnung mit mehreren Figuren näher erläutert:

  • Fig. 1 zeigt schematisch als Schaltung einen digitalen Demodulator nach der Erfindung,
  • Fig. 2 zeigt im Zeitdiagramm das zugehörige erste Phasensignal,
  • Fig. 3 zeigt im Zeitdiagramm das zugehörige zweite Phasensignal und
  • Fig. 4 zeigt einige Signale anhand einer komplexen Zeigerdarstellung.
    • The invention and advantageous embodiments are now explained in more detail with reference to the drawing with several figures:
  • 1 shows schematically as a circuit a digital demodulator according to the invention,
  • 2 shows the associated first phase signal in the time diagram,
  • 3 shows the associated second phase signal and
  • 4 shows some signals based on a complex pointer representation.

    • In der schematischen Darstellung von Fig. 1 empfängt eine Eingangsstufe 1 ein qudraturmoduliertes Signal sq von einer Antenne, einem Kabel oder einer sonstigen Einrichtung. Eine Quadratursignalquelle 2 mit angeschlossenem Oszillator 2.1, der als Mischungssignal sx ein digitales Signal mit einer vorgegebenen Frequenz fx abgibt, bildet aus dem quadraturmodulierten Signal sq ein Inphasensignal I und ein Quadraturphasensignal Q, wobei beide Signale I und Q digitalisiert sind. Die Digitalisierung kann dabei in der Quadratursignalquelle 2 oder bereits in der Eingangssstufe 1 erfolgen.In the schematic representation of FIG. 1, an input stage 1 receives quadrature modulated signal sq from an antenna, cable or other Facility. A quadrature signal source 2 with connected oscillator 2.1, which as Mixing signal sx emits a digital signal with a predetermined frequency fx an in-phase signal I and a from the quadrature-modulated signal sq Quadrature phase signal Q, both signals I and Q being digitized. The Digitization can be in the quadrature signal source 2 or already in the Entry level 1 take place.

    Zum Verständnis für das C-QUAM-Stereo-Übertragungsverfahren werden im folgenden, einige kurze Erläuterungen eingeschoben, vgl. hierzu Fig. 4. Die Abkürzung C-QUAM steht für "Compatibel - Quadratur Amplitude Modulation", einem AM-Stereoübertragungsverfahren, das von Motorola entwickelt wurde und zur Zeit insbesondere in USA und Australien verwendet wird. Für die Stereoübertragung wird aus der Links- und Rechtsinformation L bzw. R, wie bei nahezu allen Stereo-Standards, zuächst in ein Summen- und ein Differenzsignal S bzw. D gebildet: S = L + R und D = L - R. In order to understand the C-QUAM stereo transmission method, a few brief explanations are inserted below, cf. Fig. 4. The abbreviation C-QUAM stands for "Compatible - Quadrature Amplitude Modulation", an AM stereo transmission method that was developed by Motorola and is currently used particularly in the USA and Australia. For stereo transmission, the left and right information L or R, as in almost all stereo standards, first forms a sum and a difference signal S or D: S = L + R and D = L - R.

    Das modulierte Signal erhält man aus dem Realteil (=Re) und Imaginärteil eines komplexen Zeigers M(t), der sich entsprechend der Trägerfrequenz f mit der Rotationsfrequenz ω dreht. Der Betrag dieses Zeigers soll dabei immer den Wert 1+S annehmen, wobei der Wert 1 den Träger mit konstant angenommener Größe repräsentiert. Die Größe des Differenzsignals D wirkt sich ausschließlich auf die Phasenlage des Zeigers M(t) aus. Der Phasenwinkel ϕ des Modulationsvektors M(t) ergibt sich zu: ϕ = arctan (D/(1 + S)). The modulated signal is obtained from the real part (= Re) and imaginary part of a complex pointer M (t), which rotates with the rotation frequency ω according to the carrier frequency f. The amount of this pointer should always assume the value 1 + S, the value 1 representing the carrier with a constant size. The size of the difference signal D affects only the phase position of the pointer M (t). The phase angle ϕ of the modulation vector M (t) results in: ϕ = arctan (D / (1 + S)).

    Das auf die Trägeramplitude normierte C-QUAM-Signal kann somit durch folgenden Ausdruck beschrieben werden: M(t) = Re {(1 + S) Exp (jx (ωt + ϕ)}. The C-QUAM signal normalized to the carrier amplitude can thus be described by the following expression: M (t) = Re {(1 + S) Exp (j x (ωt + ϕ)}.

    Auf das Differenzsignal D wird ferner ein Pilotton P mit einer Frequenz von 25 Hz bei 5 % Aussteuerung moduliert, der eine Stereoerkennung und damit eine automatische Stereoumschaltung ermöglicht.A pilot tone P with a frequency of 25 Hz is also applied to the difference signal D. 5% modulation modulated, the stereo detection and thus an automatic Stereo switching enabled.

    In Fig. 1 schließt sich an die Quadratursignalquelle 2 ein Koordinatenumsetzer 3 an, der aus dem Inphasensignal I und dem Quadraturphasensignal Q ein Betragssginal b und ein erstes Phasensignal p1 bildet. Der Koordinatenumsetzer 3 führt eine Umsetzung von karthesischen Koordinaten in Polarkoordinaten aus. Für diese Umsetzung eignet sich insbesondere der bekannte Cordic-Algorithmus, der die gesuchten Werte über eine iterative Näherung mit beliebiger Genauigkeit bestimmt.In Fig. 1, the quadrature signal source 2 is followed by a coordinate converter 3 which from the in-phase signal I and the quadrature phase signal Q a magnitude signal b and a forms first phase signal p1. The coordinate converter 3 carries out a conversion from Cartesian coordinates in polar coordinates. Is suitable for this implementation in particular the well-known Cordic algorithm, which uses a iterative approximation determined with arbitrary precision.

    Wie eingangs erwähnt, ist es nicht erforderlich, daß die Quadraturmischung direkt in das Basisband erfolgt. Wenn das Inphasensignal I und das Quadraturphasensignal Q mit einer Frequenz von 19 kHz abgetastet werden, dann reicht es für die Demodulation nach der Erfindung aus, wenn die restliche Rotationsfrequenz ωr des komplexen Zeigers M(t) kleiner als 2 kHz bleibt.As mentioned at the beginning, it is not necessary for the quadrature mixing to take place directly in the baseband. If the in-phase signal I and the quadrature phase signal Q are sampled at a frequency of 19 kHz, then it is sufficient for the demodulation according to the invention if the remaining rotational frequency ω r of the complex pointer M (t) remains less than 2 kHz.

    Die Differenz aus der Mischungsfrequenz fx und der Trägerfrequenz f ergibt eine Restfrequenz fr und damit eine verbleibende Rotationsfrequenz ωr des komplexen Zeigers M(t). Sie bewirkt, daß das erste Phasensignal p1 nicht konstant ist sondern im zeitlichen Mittel konstant zu- oder abnimmt, vgl. auch Fig. 2. Dies entspricht einer konstanten Versatzfrequenz ωr die mittels eines ersten Regelkreises 4 auf den Wert Null gebracht wird, indem die mittlere Steigung mt des ersten Phasensignals p1 durch ein erstes Stellsignal c1 mit einer gleichgroßen negativen Steigung kompensiert wird. Das Stellsignal c1 wird mittels eines ersten Addierers 5 dem ersten Phasensignal p1 hinzugefügt und bildet damit ein zweites Phasensignal p2, vgl. auch Fig. 3. In Fig. 1 wird die Steigung durch einen Differenzbildner 6 aus zwei aufeinanderfolgenden Abtastwerten gebildet, die dann mittels einer ersten Filtereinrichtung 7 gewichtet und/oder gemittelt werden. Der Ausgang der ersten Filtereinrichtung 7 wird mittels eines Integrators 8 integriert, dessen Ausgang das erste Stellsignal c1 an den ersten Addierer 5 liefert. Der Differenzbildner 6 besteht aus einem ersten Verzögerer 6.1 und einem Subtrahierer 6.2. Der Integrator 8 besteht aus einer Akkumulatorschleife mit einem zweiten Addierer 8.1 und einem zweiten Verzögerer 8.2. Die Ausgangssignale der beiden Regelschleifen 4, 9 werden dem zweiten Addierer 8.1 als invertierte Signale zugeführt, damit die Regelrichtung beim ersten Addierer 5 stimmt.The difference between the mixing frequency fx and the carrier frequency f results in a residual frequency fr and thus a remaining rotational frequency ω r of the complex pointer M (t). It has the effect that the first phase signal p1 is not constant but increases or decreases constantly over time, cf. also Fig. 2. This corresponds to a constant offset frequency ω r which is brought to the value zero by means of a first control circuit 4, in that the mean slope mt of the first phase signal p1 is compensated by a first actuating signal c1 with an equally large negative slope. The actuating signal c1 is added to the first phase signal p1 by means of a first adder 5 and thus forms a second phase signal p2, cf. also FIG. 3. In FIG. 1, the slope is formed by a difference former 6 from two successive sample values, which are then weighted and / or averaged by means of a first filter device 7. The output of the first filter device 7 is integrated by means of an integrator 8, the output of which supplies the first actuating signal c1 to the first adder 5. The difference generator 6 consists of a first retarder 6.1 and a subtractor 6.2. The integrator 8 consists of an accumulator loop with a second adder 8.1 and a second delay 8.2. The output signals of the two control loops 4, 9 are fed to the second adder 8.1 as inverted signals, so that the control direction in the first adder 5 is correct.

    Die Kompensation der mittleren Steigung mp bewirkt aber noch nicht, daß das zweite Phasensignal p2 im zeitlichen Mittel exakt auf den Phasenbezugswert zu liegen kommt. Der zeitliche Mittelwert tm des zweiten Phasensignals p2 ist in Fig. 3 als leicht ansteigende Gerade unterhalb der Nullphasenbezugsachse dargestellt. Mittels eines zweiten Regelkreises 9 wird der zeitliche Mittelwert tm des zweiten Phasensignals p2 exakt auf die Nullphasenbezugsachse gebracht. Dies wird mittels einer zweiten Filtereinrichtung 10 und dem Integrator 8 erreicht, indem das Ausgangssignal des ersten Addierers 5 direkt oder über eine Modifizierungseinrichtung 11 dem Eingang der zweiten Filtereinrichtung 10 zugeführt wird, an deren Ausgang ein weiterer Eingang des Integrators 8 liegt. Als Ergebnis bildet der zweite Regelkreis 9 ein zweites Stellsignal c2, das mittels des ersten Addierers 5 additiv/subtraktiv dem ersten Phasensignal p1 und einem ersten Stellsignal c1 hinzugefügt wird und damit ein drittes Phasensignal p3 bildet, das bezüglich seiner Steigung und Phase im zeitlichen Mittel richtig liegt. Das zweite Phasensignal p2 liefert mit seinem Mittelwert mp2 das Eingangssignal des zweiten Regelkreises. Die momentanen Abweichungen des dritten Phasensignals p3 von der Nullphasenbezugslage entsprechen somit nur noch dem gesuchten Differenzsignal D und dem Pilotsignal P. Mittels eines Dekodierers 12 werden aus dem Betragssignal b und dem dritten Phasensignal p3 die gesuchten Komponenten L, R, P des Stereo-Kombinationssignals gebildet. Entsprechend dem Übertragungsstandard wird in der Regel zuvor das dritte Phasensignal p3 mittels der Modifizierungseinrichtung 11 modifiziert, indem beispielsweise der zugehörige Tangenswert bestimmt wird. Da im Betragssignal b die Trägeramplitude enthalten ist, wird für die Stereo-Matrix im Dekodierer 12 das dritte Phasensignal p3 bzw. das modifizierte Phasensignal p3' auf die Trägeramplitude normiert. Dies erfolgt mittels eines Multiplizierers 13, dessen erster Eingang mit dem Betragssignal b und dessen zweiter Eingang mit dem dritten Phasensignal p3 bzw. p3' gespeist ist.However, the compensation of the mean slope mp does not cause the second Phase signal p2 comes to lie exactly on the phase reference value on average over time. The time average tm of the second phase signal p2 is easy in FIG. 3 rising straight line shown below the zero phase reference axis. By means of a second control loop 9 becomes the time average tm of the second phase signal p2 brought exactly to the zero phase reference axis. This is done by means of a second Filter device 10 and the integrator 8 achieved by the output signal of the first Adders 5 directly or via a modification device 11 to the input of the second Filter device 10 is supplied, at the output of a further input of the Integrators 8 lies. As a result, the second control circuit 9 forms a second control signal c2, that by means of the first adder 5 additive / subtractive to the first Phase signal p1 and a first control signal c1 is added and thus forms a third phase signal p3 that is correct in terms of its slope and phase on average over time. The second With its mean value mp2, phase signal p2 supplies the input signal of the second Control loop. The current deviations of the third phase signal p3 from the Zero phase reference position thus only correspond to the searched differential signal D and the pilot signal P. By means of a decoder 12, the magnitude signals b and the third phase signal p3 the sought components L, R, P of the stereo combination signal educated. According to the transmission standard, the The third phase signal p3 is previously modified using the modification device 11, for example, by determining the associated tangent value. Because in the amount signal b the carrier amplitude is included, that is for the stereo matrix in the decoder 12 third phase signal p3 or the modified phase signal p3 'on the carrier amplitude normalized. This is done by means of a multiplier 13, the first input of which with the Magnitude signal b and its second input with the third phase signal p3 or p3 ' is fed.

    Es wird darauf hingewiesen, daß in Fig. 1 das zweite und dritte Phasensignal p2, p3 identisch sind, weil der Ausgang des ersten und zweiten Regelkreises 4, 9 durch den gemeinsamen Addierer 5 gebildet wird. Die Funktionsweise des Demodulators wird durch die getrennte Betrachtung von p2, p3 verständlicher.It is pointed out that in FIG. 1 the second and third phase signals p2, p3 are identical because the output of the first and second control loops 4, 9 through the common adder 5 is formed. The way the demodulator works is by considering p2, p3 separately.

    In Fig. 2 ist schematisch der zeitliche Verlauf des ersten Phasensignals p1 dargestellt. Der restlichen Rotationsfrequenz ωr des komplexen Zeigers M(t) entspricht eine stetige Zunahme mp der mittleren Phase mp1, die durch eine sägezahnförmige, durchgezogene Linie dargestellt ist. Das erste Phasensignal p1 wird vorzugsweise als Zweierkomplement-Zahlenwert dargestellt, dessen untere bzw. obere Wertegrenze dem Phasenwinkel -π bzw. +π entspricht. Die stetig zunehmende Phase mp1 springt somit gleichsam von dem Phasenwert +π auf den Phasenwert -π zurück. Die Ankopplung des jeweiligen Phasenwertes an die Zweierkomplement-Zahlendarstellung hat den großen Vorteil, daß Phasendifferenzwerte richtig wiedergegeben werden, auch wenn die Phase zwischenzeitlich übergelaufen ist. Der gestrichelte Bereich um die mittlere Phase mp1 gibt den Bereich an, in dem sich das erste Phasensignal p1 durch die Modulation mit dem Differenzsignal D und dem Pilotsignal P aufhalten kann.2 schematically shows the time profile of the first phase signal p1. The remaining rotation frequency ω r of the complex pointer M (t) corresponds to a steady increase mp in the middle phase mp1, which is represented by a sawtooth-shaped, solid line. The first phase signal p1 is preferably represented as a two's complement number whose lower or upper value limit corresponds to the phase angle -π or + π. The steadily increasing phase mp1 thus jumps back from the phase value + π to the phase value -π. The coupling of the respective phase value to the two's complement number representation has the great advantage that phase difference values are reproduced correctly, even if the phase has meanwhile overflowed. The dashed area around the middle phase mp1 indicates the area in which the first phase signal p1 can be due to the modulation with the difference signal D and the pilot signal P.

    In Fig. 3 ist schematisch der zeitliche Verlauf des zweiten Phasensignals p2 dargestellt, das durch eine Phasenkorrektur mittels des ersten Regelkreises 4 erhalten wird. Die mittlere Phase mp2 hat hierbei allenfalls noch eine ganz geringfügige Steigung tm. Die mittlere Phase mp2 liegt jedoch nicht wie erforderlich auf der Nullphasen-Bezugsachse - allenfalls zufällig. Die Korrektur der Nullphasenlage erfolgt durch den zweiten Regelkreis 9, der auch die geringfügige restliche Steigung tm unterdrückt. Die momentane Phase des zweiten Phasensignals p2 liegt dabei in dem gestrichelt dargestellten Phasenbereich um die mittlere Phase mp2.3 schematically shows the time profile of the second phase signal p2, which is obtained by a phase correction by means of the first control loop 4. The middle phase mp2 has at most a very slight slope tm however, middle phase mp2 is not on the zero phase reference axis as required - at random, at best. The second phase corrects the zero phase position Control circuit 9, which also suppresses the slight remaining slope tm. The The instantaneous phase of the second phase signal p2 is dashed phase range shown around the middle phase mp2.

    In Fig. 4 wird wie bereits erläuert in einer komplexen Zeigerdarstellung der mit der Frequenz ω rotierende Modulationsvektor M(t) dargestellt. Die Modulationskomponenten 1+S und D definieren dabei die momentane Amplitude und Phase ϕ des Zeigers gegenüber einem mit konstanter Amplitude und mit konstanter Frequenz umlaufenden Bezugszeiger. Beim hochfrequent übertragenen Quadratursignal sq ist dies der zugehörige Träger. Der umlaufende Bezugszeiger gibt über das Inphasensignal I die Bezugsphase vor. Senkrecht dazu steht das Quadraturphasensignal Q. Aus diesen beiden Signalen I,Q bestimmt der Koordinatenumsetzer 3 die momentane Länge 1+S und momentane Phase ϕ des Zeigers M(t). Die Zeigerdarstellung ist unabhängig von der Rotationsfrequenz ω. So gilt diese Darstellung sowohl für das hochfrequent übertragene Quadratursignal sq als auch für die Quadraturkomponenten I,Q, deren zugehöriger Bezugszeiger mit der niedrigen Rotationsfrequenz ωr umläuft.As already explained in FIG. 4, the modulation vector M (t) rotating with the frequency ω is shown in a complex pointer representation. The modulation components 1 + S and D define the instantaneous amplitude and phase ϕ of the pointer compared to a reference pointer rotating with a constant amplitude and a constant frequency. In the case of the high-frequency quadrature signal sq, this is the associated carrier. The revolving reference pointer specifies the reference phase via the in-phase signal I. The quadrature phase signal Q is perpendicular to this. From these two signals I, Q, the coordinate converter 3 determines the current length 1 + S and the current phase ϕ of the pointer M (t). The pointer representation is independent of the rotation frequency ω. This representation applies both to the high-frequency quadrature signal sq and to the quadrature components I, Q, the associated reference pointer of which rotates at the low rotation frequency ω r .

    Die Realisierung des Demodulators nach der Erfindung kann als Programmablauf in einem Prozessor, insbesondere in einer monolithisch integrierten Schaltung, oder als Schaltung oder in gemischter Form erfolgen. Es ist dabei unerheblich, wie die einzelnen Funtionseinheiten im Detail realisiert sind und ob die Funktionseinheiten auch anderen Zwecken dienen.The implementation of the demodulator according to the invention can be done as a program in a processor, particularly in a monolithically integrated circuit, or as Circuit or in mixed form. It is irrelevant how the individual Functional units are realized in detail and whether the functional units are also different Serve purposes.

    Claims (7)

    1. A digital demodulator for a quadrature-modulated signal (sq) which transmits a combination signal by amplitude and phase modulation, said digital demodulator comprising:
      a quadrature-signal source (2) which, in response to the received quadrature-modulated signal (sq), provides a digitized in-phase component (I) and a digitized quadrature component (Q) at a low frequency, the width of the low-frequency range being not greater than one tenth of the digitization frequency;
      a resolver (3) which converts the digitized in-phase component (I) and the digitized quadrature component (Q) into a magnitude signal (b) and a first phase signal (p1);
      a first feedback control loop (4) following the resolver (3), which, on a time average, maintains the slope of the first phase signal (p1) at the zero value or a residual value, thus forming a second phase signal (p2);
      a second feedback control loop (9) following the resolver (3), which maintains the time average of the second phase signal (p2) at a phase reference value, particularly at a zero phase value, thus forming a third phase signal (p3); and
      a decoder (12) which produces at least one digitized component (R, L, P) of the combination signal from the magnitude signal (b) and the third phase signal (p3).
    2. A demodulator as set forth in claim 1, characterized in that the slope (mp) of the first phase signal (p1) is formed from the difference between at least two temporally adjacent sample values.
    3. A demodulator as set forth in claim 1 or 2, characterized in that the first feedback control loop (4) and/or the second feedback control loop (9) comprise an integrator (8).
    4. A demodulator as set forth in any one of claims 1 to 3, characterized in that the first feedback control loop (4) and the second feedback control loop (9) form a first corrective signal (c1) and a second corrective signal (c2), respectively, with which the first phase signal (p1) and the second phase;signal (p2), respectively, are additively/subtractively changed in value.
    5. A demodulator as set forth in claim 3, characterized in that the integrator (8) is common to the first and second feedback control loops (4, 9).
    6. A demodulator as set forth in any one of claims 1 to 5, characterized in that the third phase signal (p3) is applied to the decoder (12) and/or to the second feedback control loop (9) through a modification device (11).
    7. A demodulator as set forth in claim 6, characterized in that the modification device (11) comprises a tangent-forming device.
    EP96101105A 1996-01-26 1996-01-26 Digital demodulator Expired - Lifetime EP0786921B1 (en)

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    DE59609450T DE59609450D1 (en) 1996-01-26 1996-01-26 Digital demodulator
    EP96101105A EP0786921B1 (en) 1996-01-26 1996-01-26 Digital demodulator
    US08/792,924 US5767739A (en) 1996-01-26 1997-01-21 Digital demodulator for quadrature amplitude and phase modulated signals

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    DE4434451A1 (en) * 1994-09-27 1996-03-28 Blaupunkt Werke Gmbh Amplitude demodulator
    KR100441196B1 (en) * 2002-01-14 2004-07-21 기가텔레콤 (주) Apparatus for continuous phase quadrature amplitude modulation and demodulation
    US7437299B2 (en) 2002-04-10 2008-10-14 Koninklijke Philips Electronics N.V. Coding of stereo signals
    DE102004020300B3 (en) * 2004-04-26 2005-09-22 Micronas Gmbh Pulsed signal method for determining a pulsed signal's scan-time point operates with a circuit structure for determining symbols from a digitized signal
    US10509295B2 (en) * 2017-03-15 2019-12-17 Elenion Technologies, Llc Bias control of optical modulators
    US10942377B2 (en) * 2018-10-08 2021-03-09 Cisco Technology, Inc. High swing AC-coupled Mach-Zehnder interferometer (MZI) driver

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    JPS6047513A (en) * 1983-08-26 1985-03-14 Nec Corp Frequency shift absorbing circuit
    DE3889326D1 (en) * 1988-05-27 1994-06-01 Itt Ind Gmbh Deutsche Correction circuit for a digital quadrature signal pair.
    US5249204A (en) * 1991-08-12 1993-09-28 Motorola, Inc. Circuit and method for phase error correction in a digital receiver
    DE4340012B4 (en) 1993-11-24 2004-04-22 Blaupunkt-Werke Gmbh demodulator
    US5497400A (en) * 1993-12-06 1996-03-05 Motorola, Inc. Decision feedback demodulator with phase and frequency estimation

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