EP0786921A1 - Digital demodulator - Google Patents

Abstract

From the digitised phase (I) and quadrature (Q) outputs of a quadrature signal source (2), a co-ordinate convertor (3) forms an amplitude signal (b) and a first phase signal (p1). Two control loops (4,9) zero the slope of the first phase signal and adjust its time-averaged value to the zero-phase position. The process produces a second and a third phase signal (p2,p3) of which the latter is used in conjunction with its tangent (p3'), and the amplitude signal in a decoder (12). f From these signals the left and right-hand channel signals (L,R), and the pilot signal (P), are obtained.

Description

The invention relates to a digital demodulator for a quadrature-modulated signal, which transmits a combination signal by means of amplitude and phase modulation.

Quadrature-modulated signals are often used when signals that belong together but are supposed to be independent of each other are to be transmitted in a transmission channel. One such application is the transmission of stereo signals according to the C-QUAM standard, in which a sum signal is transmitted via the amplitude modulation and a difference signal and a pilot tone via the phase modulation of the respective carrier. An example of an associated digital demodulator is described in DE 43 40 012 A1. A quadrature signal source forms an in-phase signal and a quadrature-phase signal from the received quadrature-modulated signal by means of a quadrature mixer. The digitization can take place before or after the quadrature mixer. A magnitude signal and a phase signal are formed from the digitized in-phase signal and the digitized quadrature phase signal by means of a coordinate converter, which operates in particular according to the Cordic algorithm. A control loop controlled by the phase signal controls the oscillator frequency of the quadrature mixer exactly to the value of the carrier frequency, so that the in-phase signal and the quadrature phase signal are transformed into the baseband. A remaining mean phase deviation is corrected in that the control loop also intervenes in the phase signal and there adds or subtracts a correction signal which pulls the temporal mean value of the phase signal to the zero phase value. A decoder, which essentially contains a known stereo matrix, forms the desired left and right signal and the pilot signal at 25 Hz from the magnitude signal and the phase signal.

The object of the invention is to provide an improved digital demodulator for such quadrature-modulated signals, which is better suited to digital signal processing is adapted and places less demands on the quadrature signal source.

• eine Quadratursignalquelle, die abhängig vom empfangenen quadraturmodulierten Signal ein digitalisiertes Inphasensignal und ein digitalisiertes Quadraturphasensignal in tiefer Frequenzlage liefert,
• ein Koordinatenumsetzer, der aus dem digitalisierten Inphasensignal und dem digitalisierten Quadraturphasensignal ein Betragssignal und ein erstes Phasensignal bildet,
• ein dem Koordinatenumsetzer nachgeschalteter erster Regelkreis, der die Steigung des ersten Phasensignals im zeitlichen Mittel auf den Wert Null oder einen Restwert regelt und damit ein zweites Phasensignal bildet,
• ein dem Koordinatenumsetzer nachgeschalteter zweiter Regelkreis, der den zeitlichen Mittelwert des zweiten Phasensignals auf einen Phasenbezugswert, insbesondere eine Nullphasenlage, regelt und damit ein drittes Phasensignal bildet, und
• ein Dekodierer, der aus dem Betragssignal und dem dritten Phasensignal mindestens eine digitalisierte Komponente des Kombinationssignals bildet.

The object is achieved by the invention according to the features of claim 1 as follows:

• a quadrature signal source which, depending on the received quadrature-modulated signal, supplies a digitized in-phase signal and a digitized quadrature-phase signal in the low frequency range,
• a coordinate converter which forms a magnitude signal and a first phase signal from the digitized in-phase signal and the digitized quadrature phase signal,
• a first control circuit downstream of the coordinate converter, which regulates the slope of the first phase signal to the value zero or a residual value on average over time and thus forms a second phase signal,
• a second control circuit connected downstream of the coordinate converter, which regulates the time average of the second phase signal to a phase reference value, in particular a zero phase position, and thus forms a third phase signal, and
• a decoder which forms at least one digitized component of the combination signal from the magnitude signal and the third phase signal.

The main advantage of this arrangement is that the output signals of the quadrature signal source, the digitized in-phase signal and the digitized quadrature phase signal do not have to have the exact baseband position, but only have to be in a relatively low frequency range. The bandwidth of this low frequency range depends on the digitization frequency and should preferably not be greater than one tenth of the digitization frequency. These favorable boundary conditions allow a digital quadrature mixer to be implemented in the simplest way by means of digital switches, because the quadrature-modulated digital signal only to be multiplied by the values +1, -1 and 0. With an exact transformation of the quadrature-modulated digital signal into the baseband, an exact frequency adjustment of the digital mixing signal would be necessary, which would only be possible using a very complex sine / cosine table with two complex digital multipliers. An analog design of the quadrature mixer with subsequent digitization of the in-phase and quadrature-phase signal is of course also possible, although according to the invention the oscillator frequency does not have to be readjusted and is therefore not critical to the frequency position and drift changes. The invention thus avoids a phase-rigid tracking of the quadrature mixer, complex sine / cosine tables and complex multipliers in the quadrature mixing.

The first control loop is advantageously controlled via the slope of the first phase signal, which results from the difference between at least two temporally adjacent samples. Of course, this also includes the fact that further samples can be acquired to form the difference, whereby a better averaging is achieved and disturbance variables can be better suppressed.

For the accuracy of the control, it is also expedient if the control loops contain an integrator. Accumulator loops with sufficient place capacity are particularly suitable for this, so that no overflow takes place in normal operation.

It is advantageous if the control signal of the first and / or second control circuit is designed such that it can be combined with the respective phase signal as an additive or subtractive correction signal via an adding circuit. With a suitable design of the two control loops, the two control signals can be combined additively, so that only a single adder is required for correction in the phase signal path. In the same way, the integrator for the first and second control loops can be designed jointly by feeding the two control signals to the adder in the accumulator loop. Its output then provides the common control signal.

It may be necessary for the respective transmission standard that the third Phase signal before the decoder is to be modified by means of an evaluation device. The evaluation device corresponds to a predetermined signal characteristic which is inverse to the signal characteristic on the transmitter side. The evaluation device can have a non-linear characteristic curve, for example in the case of the C-QUAM standard, a tangent curve is prescribed as the characteristic curve on the receiver side. The course of the tangent can be defined by a memory table or by a polynomial approximation as in DE 43 40 012 already mentioned.

• Fig. 1 zeigt schematisch als Schaltung einen digitalen Demodulator nach der Erfindung,
• Fig. 2 zeigt im Zeitdiagramm das zugehörige erste Phasensignal,
• Fig. 3 zeigt im Zeitdiagramm das zugehörige zweite Phasensignal und
• Fig. 4 zeigt einige Signale anhand einer komplexen Zeigerdarstellung.

The invention and advantageous embodiments are now explained in more detail with reference to the drawing with several figures:

• 1 shows schematically as a circuit a digital demodulator according to the invention,
• 2 shows the associated first phase signal in the time diagram,
• 3 shows the associated second phase signal and
• 4 shows some signals based on a complex pointer representation.

In the schematic representation of FIG. 1, an input stage 1 receives a quadrature modulated signal sq from an antenna, a cable or another device. A quadrature signal source 2 with connected oscillator 2.1, which emits a digital signal with a predetermined frequency fx as the mixing signal sx, forms an in-phase signal I and a quadrature phase signal Q from the quadrature-modulated signal sq, both signals I and Q being digitized. The digitization can take place in the quadrature signal source 2 or already in the input stage 1.

In order to understand the C-QUAM stereo transmission method, a few brief explanations are inserted below, cf. Fig. 4. The abbreviation C-QUAM stands for "Compatible - Quadrature Amplitude Modulation", an AM stereo transmission method that was developed by Motorola and is currently used particularly in the USA and Australia. For stereo broadcasting From the left and right information L or R, as with almost all stereo standards, first formed into a sum and a difference signal S or D: $$\text{S = L + R and D = L - R.}$$

The modulated signal is obtained from the real part (= Re) and imaginary part of a complex pointer M (t), which rotates with the rotation frequency ω according to the carrier frequency f. The amount of this pointer should always assume the value 1 + S, the value 1 representing the carrier with a constant size. The size of the difference signal D only affects the phase position of the pointer M (t). The phase angle ϕ of the modulation vector M (t) results in: $$\text{ϕ = arctan (D / (1 + S)).}$$

The C-QUAM signal normalized to the carrier amplitude can thus be described by the following expression: $${\text{M (t) = Re {(1 + S) Exp (j}}_{\text{x}}\text{(ωt + ϕ)}.}$$

A pilot tone P with a frequency of 25 Hz at 5% modulation is also modulated onto the differential signal D, which enables stereo detection and thus automatic stereo switching.

In Fig. 1, the quadrature signal source 2 is followed by a coordinate converter 3, which forms an absolute signal b and a first phase signal p1 from the in-phase signal I and the quadrature phase signal Q. The coordinate converter 3 carries out a conversion of Cartesian coordinates into polar coordinates. The well-known Cordic algorithm is particularly suitable for this implementation, which determines the searched values with an iterative approximation with arbitrary accuracy.

As mentioned at the beginning, it is not necessary for the quadrature mixing to take place directly in the baseband. If the in-phase signal I and the quadrature phase signal Q with a frequency of 19 kHz are sampled, then it is sufficient for the demodulation according to the invention if the remaining rotational frequency ω _{r of} the complex pointer M (t) remains less than 2 kHz.

The difference between the mixing frequency fx and the carrier frequency f results in a residual frequency fr and thus a remaining rotational frequency ω _{r of} the complex pointer M (t). It has the effect that the first phase signal p1 is not constant but increases or decreases constantly over time, cf. also Fig. 2. This corresponds to a constant offset frequency ω _r which is brought to the value zero by means of a first control circuit 4, in that the mean slope mt of the first phase signal p1 is compensated by a first actuating signal c1 with an equally large negative slope. The actuating signal c1 is added to the first phase signal p1 by means of a first adder 5 and thus forms a second phase signal p2, cf. also FIG. 3. In FIG. 1, the slope is formed by a difference former 6 from two successive sample values, which are then weighted and / or averaged by means of a first filter device 7. The output of the first filter device 7 is integrated by means of an integrator 8, the output of which supplies the first actuating signal c1 to the first adder 5. The difference generator 6 consists of a first retarder 6.1 and a subtractor 6.2. The integrator 8 consists of an accumulator loop with a second adder 8.1 and a second delay 8.2. The output signals of the two control loops 4, 9 are fed to the second adder 8.1 as inverted signals, so that the control direction in the first adder 5 is correct.

However, the compensation of the average slope mp does not yet cause the second phase signal p2 to be exactly at the phase reference value on average over time. The time average tm of the second phase signal p2 is shown in FIG. 3 as a slightly rising straight line below the zero phase reference axis. The time average tm of the second phase signal p2 is brought exactly to the zero phase reference axis by means of a second control loop 9. This is achieved by means of a second filter device 10 and the integrator 8, in that the output signal of the first adder 5 is fed directly or via an evaluation device 11 to the input of the second filter device 10, at the output of which there is a further input of the integrator 8. As a result, the second control circuit 9 forms a second control signal c2, which is additively / subtractively added to the first or second phase signal p1, p2 by means of the first adder 5 and thus forms a third phase signal p3 which is correct in terms of its slope and phase on average over time. With its mean value mp2, the second phase signal p2 supplies the input signal of the second control loop. The instantaneous deviations of the third phase signal p3 from the zero phase reference position thus only correspond to the sought difference signal D and the pilot signal P. By means of a decoder 12, the sought components L, R, P of the stereo combination signal are formed from the magnitude signal b and the third phase signal p3 . In accordance with the transmission standard, the third phase signal p3 is generally modified beforehand by means of the evaluation device 11, for example by determining the associated tangent value. Since the magnitude signal b contains the carrier amplitude, the third phase signal p3 or the modified phase signal p3 'is normalized to the carrier amplitude for the stereo matrix in the decoder 12. This is done by means of a multiplier 13, the first input of which is fed with the magnitude signal b and the second input of which is fed with the third phase signal p3 or p3 '.

It is pointed out that the second and third phase signals p2, p3 are identical in FIG. 1 because the output of the first and second control loops 4, 9 is formed by the common adder 5. The mode of operation of the demodulator is more understandable by considering p2, p3 separately.

2 schematically shows the time profile of the first phase signal p1. The remaining rotation frequency ω _{r of} the complex pointer M (t) corresponds to a steady increase mp in the middle phase mp1, which is represented by a sawtooth-shaped, solid line. The first phase signal p1 is preferably represented as a two's complement number whose lower or upper value limit corresponds to the phase angle -π or + π. The steadily increasing phase mp1 thus jumps back from the phase value + π to the phase value -π. The coupling of the respective phase value to the two's complement number representation has the great advantage that phase difference values are reproduced correctly, even if the phase has meanwhile overflowed. The dashed area around the middle phase mp1 indicates the area in which the first phase signal p1 is due to the modulation the difference signal D and the pilot signal P can stop.

FIG. 3 schematically shows the time profile of the second phase signal p2, which is obtained by a phase correction using the first control loop 4. The middle phase mp2 in this case still has a very slight slope tm. However, the middle phase mp2 does not lie on the zero-phase reference axis as required - at best by chance. The zero phase position is corrected by the second control circuit 9, which also suppresses the slight remaining slope tm. The instantaneous phase of the second phase signal p2 lies in the phase range around the middle phase mp2 shown in dashed lines.

As already explained in FIG. 4, the modulation vector M (t) rotating with the frequency ω is shown in a complex pointer representation. The modulation components 1 + S and D define the instantaneous amplitude and phase ϕ of the pointer compared to a reference pointer rotating with a constant amplitude and a constant frequency. In the case of the high-frequency quadrature signal sq, this is the associated carrier. The revolving reference pointer specifies the reference phase via the in-phase signal I. The quadrature phase signal Q is perpendicular to this. From these two signals I, Q, the coordinate converter 3 determines the current length 1 + S and the current phase ϕ of the pointer M (t). The pointer representation is independent of the rotation frequency ω. This representation applies both to the high-frequency transmitted quadrature signal sq and to the quadrature components I, Q, the associated reference pointer of which rotates at the low rotation frequency ω _r .

The demodulator according to the invention can be implemented as a program sequence in a processor, in particular in a monolithically integrated circuit, or as a circuit or in a mixed form. It is irrelevant how the individual functional units are implemented in detail and whether the functional units also serve other purposes.

Claims (7)

Digital demodulator for a quadrature modulated signal (sq), which transmits a combination signal by means of amplitude and phase modulation a quadrature signal source (2) which, depending on the received quadrature-modulated signal (sq), supplies a digitized in-phase signal (I) and a digitized quadrature-phase signal (q) in the low frequency range, a coordinate converter (3), which forms a magnitude signal (b) and a first phase signal (p1) from the digitized in-phase signal (I) and the digitized quadrature phase signal (Q), - a first control circuit (4) connected downstream of the coordinate converter (3), which regulates the slope (mp) of the first phase signal (p1) to the value zero or a residual value on average over time and thus forms a second phase signal (p2), - A second control circuit (9) connected downstream of the coordinate converter (3), which regulates the time average (tm) of the second phase signal (p2) to a phase reference value, in particular a zero phase position, and thus forms a third phase signal (p3), and - A decoder (12) which forms at least one digitized component (R, L, P) of the combination signal from the magnitude signal (b) and the third phase signal (p3). Demodulator according to claim 1, characterized in that the slope (mp) of the first phase signal (p1) is formed from the difference between at least two temporally adjacent samples. Demodulator according to claim 1 or 2, characterized in that the first and / or second control circuit (4, 9) contains an integrator (8). Demodulator according to one of claims 1 to 3, characterized in that the first and / or second control circuit (4, 9) forms a first and a second control signal (c1, c2) with which the first and / or second phase signal (p1 , p2) additive / subtractive in its value is changed. Demodulator according to claim 4, characterized in that the integrator (8) for the first and second control loops (4, 9) is present together. Demodulator according to one of Claims 1 to 5, characterized in that the third phase signal (p3) is fed to the decoder (12) and / or the second control loop (9) via an evaluation device (11). Demodulator according to Claim 6, characterized in that the evaluation device (11) contains a tangent generator.

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