EP0656737A1 - Prothèse auditive avec suppression du couplage acoustique - Google Patents

Prothèse auditive avec suppression du couplage acoustique Download PDF

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Publication number
EP0656737A1
EP0656737A1 EP94117510A EP94117510A EP0656737A1 EP 0656737 A1 EP0656737 A1 EP 0656737A1 EP 94117510 A EP94117510 A EP 94117510A EP 94117510 A EP94117510 A EP 94117510A EP 0656737 A1 EP0656737 A1 EP 0656737A1
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Prior art keywords
unit
input
filter
output
signal
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EP94117510A
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German (de)
English (en)
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EP0656737B1 (fr
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August Nazar Kälin
Pius Gerold Estermann
Bohumir Uvacek
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Sonova Holding AG
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Phonak AG
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R25/00—Deaf-aid sets, i.e. electro-acoustic or electro-mechanical hearing aids; Electric tinnitus maskers providing an auditory perception
    • H04R25/45—Prevention of acoustic reaction, i.e. acoustic oscillatory feedback
    • H04R25/453—Prevention of acoustic reaction, i.e. acoustic oscillatory feedback electronically
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04R—LOUDSPEAKERS, MICROPHONES, GRAMOPHONE PICK-UPS OR LIKE ACOUSTIC ELECTROMECHANICAL TRANSDUCERS; DEAF-AID SETS; PUBLIC ADDRESS SYSTEMS
    • H04R25/00—Deaf-aid sets, i.e. electro-acoustic or electro-mechanical hearing aids; Electric tinnitus maskers providing an auditory perception
    • H04R25/50—Customised settings for obtaining desired overall acoustical characteristics
    • H04R25/505—Customised settings for obtaining desired overall acoustical characteristics using digital signal processing

Definitions

  • the present invention relates to a hearing aid according to the preamble of claim 1 and an electrical model according to claim 29.
  • acoustic-electrical (ak / el) converter 1 shows an acoustic-electrical (ak / el) converter 1 with a downstream analog / digital (A / D) converter 3, a digital amplification filter section 5, which on the output side is connected to a digital / analog (D / A) converter 7, the latter acts on the electrical-acoustic (el / ak) converter 9.
  • Block 11 represents the acoustic-mechanical interference feedback with the generally time-variant transmission behavior h.
  • the feedback signal y (t) is superimposed on the useful signal v (t) and fed to the input of the ak / el converter 1, which on the output side, at times nT, supplies the discrete-time samples d (nT) required for digital processing.
  • a disadvantage of this procedure is that, given an assumed filter length of the compensator 15 of m steps, 2 m multiplications per sample value of the A / D converter 3 are necessary, which leads to an extremely complex system. This is particularly important with a view to the miniaturization required for hearing aids.
  • step length ⁇ of the LMS algorithm for maintaining the speech signal transmission is chosen to be as small as possible, which means that the adaptation of the compensator filter 15 to the interference feedback path 11 becomes correspondingly slow, which increases the possible increase in the gain on the path 5 , for reasons of stability.
  • a disadvantage of this procedure is the additional generator for the measurement signal and its necessary amplitude control to ensure a sufficient signal-to-noise ratio.
  • This procedure made it possible to increase the gain on the amplifier filter section by approx. 17dB with a compensator filter of the 32nd order.
  • the signal processing was carried out both on the amplification filter section and on the compensator in the frequency range, for which purpose the output signal of the A / D converter 3 was transformed into the frequency range by means of an overlapping orthogonal transformation (LOT) on the unit 17.
  • a corresponding inverse transformation (ILOT) at the unit 19 then again supplies the required signal u (nT) at the input of the el / ak converter 7.
  • time domain / frequency domain transformation is not carried out in front of the differential unit 13 f , as shown in FIG. 2, but rather that the difference is formed in the time domain, can surprisingly be obtained the required time invariance of the system.
  • suitably overlapping block division it is possible to implement the time domain / frequency domain transformations that are still used with significantly smaller block lengths, which in turn increases the compensation efficiency and therefore enables the gain on the gain filter section 5 f according to FIG. 2 to be increased drastically.
  • FIG. 3 shows a basic principle of the present invention or of the hearing aid device according to the invention on the basis of a signal flow / functional block diagram.
  • the reference symbols already used with reference to FIGS. 1 and 2 are used therein for the function blocks and signals already described there.
  • the difference signal r (nT) is converted at a LOT transformation unit 20 into the adaptation control signal E [k], which is fed to the adaptation input A f of the compensator filter 15 f . Because the time domain / frequency domain transformation takes place in the LOT transformation unit 20 in blocks of a predetermined number of samples from the difference signal r (nT), [k] denotes the number of the signal block appearing on the output side of the transformation unit 20.
  • the difference signal r (nT) is supplied to the amplification filter section 5 in the time domain and fed to the el / ak converter 9 via the D / A converter 7.
  • the D / A converter 7 is acted upon by the discrete-time output signal u (nT) of the amplification filter section 5.
  • This output signal u (nT) becomes a further orthogonal transformation unit 22 supplied and converted there from the time domain into the frequency domain.
  • the output signal of the transformation unit 22 is fed as an input signal to the input E f of the compensator filter 15 f .
  • the output signal ⁇ [k + 1] of said filter 15 f is transformed back into the time domain at a reverse transformation unit ILOT 24 and its output signal ⁇ (nT) is fed to the difference forming unit 13 as a discrete-time signal.
  • the amplification filter section 5 f is preceded by a transformation unit LOT 28 and the D / A converter 7 is a reverse transformation unit ILOT 26; the transformation unit 22 according to FIG. 3 is omitted.
  • FIG. 3 shows a first form of implementation which corresponds to the definition according to claim 2, namely in which a transformation unit LOT 20 or 22 is connected upstream of the signal input E f and the adaptation input A f of the compensator filter 15 f .
  • a preferred embodiment variant is that according to FIG. 4, which corresponds to the definition according to claim 3, according to which the adaptation input A f of the compensating filter 15 f and the input of the amplification filter section 5 f are preceded by a LOT transformation unit 20 or 28 and the input of the D / A converter 7 a corresponding ILOT reverse transformation unit 26.
  • the gain filter 5 f is also preceded by a LOT transformation unit 28, the input of the D / A converter 7 is an ILOT reverse transformation unit 26, and further the output of the compensator filter 15 f is followed by an ILOT reverse transformation unit 24.
  • These transformation or reverse transformation units 28, 24 and 26 operate in the mentioned preferred embodiment according to the "overlap-save” technique.
  • the LOT transformation unit 20 upstream of the adaptation input A f in particular according to FIG. 4, preferably works according to the "overlap-add" principle.
  • the time discrete differential signal r (nT) of a single LOT transform unit 30 is fed to here, from whose output signal both the adaption input A f supplied adaptation signal E [k] as well as that of the enhancement filter path 5 f supplied input signal R [k ] is formed.
  • the overlapping orthogonal transformations are preferably based on the DFT.
  • FIG. 6 shows a form of realization of the data transmission path between the time-discrete difference signal r (nT) on the output side of the difference forming unit 13 for the adaptation signal E [k] or the input signal R [k] to the amplification filter section 5 f according to FIG. 5.
  • an overlapping orthogonal transformation unit 30a based on the DFT follows the output of the difference formation unit 13 with the time-discrete difference signal r (nT).
  • the actual gain filter 40 follows first, which is followed by a delay unit 42 with corresponding intermediate storage.
  • the block signal U [k + 1] available on the output side is now fed on the one hand to the input E f of the compensator 15 f and on the other hand is subjected to an inverse DFT of the "overlap-save" type in the ILOT unit 26. Since the corresponding time signal u (nT) is delayed by a partial block length N, the numbering of U [k + 1] with the block number k + 1 is justified in retrospect.
  • FIG. 8 shows a preferred expansion variant of the compensator filter 15 f on the hearing aid according to the invention according to FIG. 5.
  • the block signals U [k + 1] to are thereby buffered with delay units of the type, as shown at 56 U [k + 1-L] provided and, based on this, with the aid of partial compensators, the first of which is referred to in FIG. 8 as unit 50, which produces the partial estimates ⁇ 1 [k + 1] to ⁇ L [k + 1], which in turn are provided in unit 52 for the overall estimate ⁇ [ k + 1] can be added.
  • the ILOT unit 24 in the preferred variant via an inverse DFT of the "overlap-save" type, then transforms back into the time domain.
  • the partial estimate ⁇ 1 [k + 1] arises at the output of the multiplication unit 64, on which the block signals U [k + 1] and the block weight ⁇ 1 [k + 1] act at the input.
  • the block weight H i [k + 1] represents the current estimate in the frequency range for the i-th subrange of length N of the discrete-time impulse response h of the acoustic-mechanical Noise feedback 11.
  • the estimate H i [k + 1] is updated in advance of the formation of ⁇ i, j [k + 1] with the aid of the old estimate H i [k].
  • the block signal acts for this purpose, again with reference to the partial compensator 1 U [k + 1-1] and the step size ⁇ [k + 1-1] to the multiplication unit 54, which on the output side is led to the multiplication unit 58 together with the block signal E [k].
  • the output of unit 58 is then in summation unit 60 according to the formula used to update H1 [k + 1].
  • j denotes the block location and i the partial compensator number.
  • the index (*) stands for conjugate complex.
  • any known method for guiding the step size ⁇ [k] can be used.
  • FIG. 9 shows a preferred variant today for generating the normalized step size ⁇ [k] according to FIG. 8, which is also used to stop the adaptation process.
  • this block signal before being fed to the multiplication unit 54 is used to calculate the current block signal ⁇ [k] by feeding the block signal U [k] to a power detection unit 70, which in turn has two Interpolation filter 72 resp. 74 acts.
  • these interpolation filters control the scaling unit 78, which finally supplies the scaling variable S [k] required for the normalization of the reference step size ⁇ 0 at the input of the multiplication unit 80.
  • the interpolation filters work according to the formula and are parameterized with ⁇ and c.
  • the index j denotes the block location.
  • the scaling variable S [k] is now used on the one hand via the output of the filter 72, in FIG. 9 as a block signal P U [k], to normalize the reference step size ⁇ 0, but on the other hand also via the output of the filter 74 in FIG. 9 referred to as block signal P U min [k], for freezing the adaptation process of individual frequency components when the power is insufficient.
  • the scaling variable S [k] is according to the formula formed, the j denoting the block location as usual.
  • FIG. 10 shows a further preferred variant which, with the use of partial compensators according to FIG. 8, significantly improves the speech quality, with otherwise the same parameters.
  • the estimate ⁇ i [k + 1] of the partial compensator i previously the multiplication with U [k + 2-i] in unit 64 of FIG. 8, via a projection unit 62.
  • the block weight ⁇ i [k + 1] is subjected to an inverse DFT (unit 82), then cleaned by zeroing the block locations with index N to 2N-1 (unit 84) and finally transformed back into the frequency range (unit 86).
  • the electrical-acoustic converter 9 is not linear in the sense that it no longer converts the input signal into the output signal linearly from certain input signal amplitudes.
  • the signal path via compensation filter 15 f should be modeled as exactly as possible over the signal path via function blocks 7, 9, 11, 1 and 3 and, according to the previous explanations, the nonlinearities mentioned on converter 9 should not can reproduce.
  • the maximum output level should also be adjustable in the hearing aid according to the individual needs of the user. The problem arises that the converter 9 is driven into its non-linear range, of course only if the individually set maximum output level can drive the converter in the mentioned range at all.
  • the gain filter 5 has a limiter unit which operates in the time domain and is preferably adjustable 90, which limits the output signal of the amplification filter 5 with respect to the amplitude so that the converter 9 is never driven into its non-linear range and which also allows the maximum output sound level at the converter 9 to be adjusted according to individual needs, in particular also lower, as is the case with is indicated by the double arrows.
  • this is achieved in that the amplification filter 5 f operating in the frequency range is followed by a unit 90 f , which limits the frequency components of the signal spectrum, taking into account their mutual phase position, in the frequency range in such a way that the output of the conversion unit 26 and the Digital / analog converter 7 produces a time-variable signal u (t) which never drives the converter 9 into the non-linear transmission range and which also allows the maximum individual modulation to be set.
  • FIG. 11 shows a further embodiment variant of the hearing aid according to the invention, which largely corresponds to that shown in FIG. 4, with the difference that the reverse transformation unit 26 according to FIG. 4, now 26a, is provided directly on the output side of the amplification filter unit 5 f and on the input side of the Compensation filter 15 f a LOT transformation unit 22a of the type already discussed is arranged.
  • FIG. 4 which, as explained above, as well as FIG. 5, represents a preferred embodiment variant of the inventive hearing aid, the provision of a limiter unit is only possible in the frequency range because such a unit is also in the signal path with the compensation filter 15 f must be effective.
  • the function block structure shown here enables the provision of a limiter unit 90 which operates in the time domain, which is much easier to implement than one which operates in the frequency domain.
  • FIG. 12 shows a preferred embodiment variant of the signal processing on the device according to FIG. 11 upstream of the compensation filter 15 f or downstream of the amplification filter 5 f .
  • the non-linearity of the electrical-acoustic transducer 9 is basically simulated, ie modeled, in the signal path with the compensation filter 15 f . This is implemented by a modeling unit 92, upstream of the transformation unit 22a according to FIG. 11 and therefore working in the time domain, and / or by a modeling unit 92 f , downstream of the transformation unit 22a and thus operating in the frequency domain.
  • This procedure ensures that, depending on the quality of the modeling unit 92, the limits of the unit 90 are set higher and the output signal can thus be increased by up to 6 dB compared to the embodiment variant in FIG. 11. If necessary, the limiter function of the unit 90 can also be stopped.
  • the modeling unit 92 can be implemented, for example, as a simplified Viennese model, as suggested in R. Isermann, "Identification of Dynamic Systems", Springer-Verlag, 2: 238, 1988.
  • the transformation into the time domain between the gain filter 5 f and the compensator filter 15 f also allows the addition of a nonlinear correction filter in the signal path with the gain filter 5 f in the same manner described above.
  • this is implemented by a modeling unit 94, connected downstream of the transformation unit 26a and thus operating in the time domain, and / or by a modeling unit 94 f , connected upstream of the transformation unit 26a and therefore operating in the frequency domain.
  • FIG. 13 shows the implementation of a loudspeaker model according to the invention in the time domain. It is used in particular in the hearing aid according to the invention, according to FIGS. 3 and 11 at the location of block 90 and according to FIG. 12 instead of blocks 92 or 90 and 94.
  • the pre-filter 100 with the transfer function F 1 ( ⁇ ), essentially with a low-pass characteristic.
  • the cutoff frequency ⁇ 1 in the Bode diagram of the filter characteristic, which is shown qualitatively in block 100, is approximately 0.8 kHz, the amplification
  • the asymptote slope S 1 is approximately 0 dB / DK.
  • the identification variables namely corner frequency ⁇ 1 and the asymptotic slopes S1 and S2, as well as the gain, for example at the corner frequency ⁇ 1, are identified by identifying the speaker or transducer 9 to be modeled.
  • a linear amplifier unit 102 is provided downstream of the prefilter 100, and the gain factor K is set to this.
  • a non-linear amplification unit 104 is provided downstream of the linear amplification unit 102.
  • the gain of the non-linear amplification unit 104 is one, so that the gain characteristic around the origin has a slope of one.
  • the nonlinear gain characteristic as is known from the loudspeaker or converter 9, exhibits saturation behavior.
  • the coefficients a, b, c, d and the gain K are in turn identified on the basis of the loudspeaker or converter 9 that is actually to be modeled.
  • a linear amplification unit 106 Downstream of the non-linear amplification unit 104, a linear amplification unit 106 is again provided, by means of which the amplification K of the linear amplification element 102 is compensated for - K ⁇ 1.
  • a filter unit 108 is provided downstream of it, essentially with a high-pass characteristic, which, as can be seen, essentially compensates for the frequency response of the pre-filter 100.
  • the loudspeaker modeling unit essentially consists of a linear amplifier part 102, 106, 100 and 108 and a non-linear amplification unit 104.
  • symptoms of saturation or limitation can also be based on another cause, namely on the drop in the battery voltage, which feeds the device according to the invention.
  • the aging of the battery that feeds the device causes, in particular at the D / A converter 7, a decrease in the signal amplification and a reduction in the modulation limit, i.e. the maximum analog modulation range becomes smaller with decreasing battery voltage.
  • the output impedance of the battery usually appears in series with the impedance of the electrical-acoustic converter 9.
  • the battery output impedance and thus the replacement image, maW which also includes the latter, follows the D / A converter 7, it changes the how was explained, to model non-linearities appearing on the output side of the converter 7.
  • the limiter unit 90 in the time domain or 90 f in the frequency domain by means of the instantaneous battery voltage and / or the instantaneous one To control battery impedance with regard to its limiting effect.
  • FIG. 14 This procedure is shown schematically in FIG. 14.
  • the current battery voltage U B and / or the current impedance is measured on a measuring unit 122 Z. ⁇ B measured, resulting in corresponding measurement signals e (U B ) or e ( Z. ⁇ B ).
  • These measurement signals control the limiter unit 90, analogously in the frequency range the limiter unit 90 f according to FIGS. 4, 5, 11 or 12, 14 and / or the model units 92, 92 f or 94, 94 f from FIGS. 12, 13, 14.
  • the measurement signals e are preferably used after digitization, for which purpose the measurement unit 122 is provided on the output side with an A / D converter (not shown).
  • the limiter limits and / or the model parameters are tracked in a controlling manner by the instantaneous battery output voltage or their instantaneous impedance.
  • model parameters on the model units 92 and 92 f , 94 and 94 f are modified in the function of the above-mentioned measured variables on the battery 120 or by means of values stored in tables that can be called up and activated by the current measured variables.
  • a gain loss on the D / A converter 7 is compensated for due to a decrease in the battery voltage: If the battery voltage decreases and thus the gain on the converter 7, the measurement signal e at block 7 the gain, compensatory, increased accordingly.
  • the battery voltage drop acts simultaneously as a signal limitation by a limiter and is best and preferably simulated by a battery output voltage-controlled limiter block 90 b in front of the loudspeaker model 92 or 92 f according to FIG. 14.
  • the blocks 90 can be omitted, as may 92 or 92 f blocks 94 and 94 f omitted when provision of the blocks, and there is a relatively low cost a battery voltage-independent, in this realization, stable Feedback suppression achieved according to the invention.
  • the function of the mentioned block 90 b can be taken over completely by providing the battery output voltage-controlled block 90 or 90 f according to FIGS. 4 and 5.
  • a non-linear model of the acoustic-electrical converter 1 possibly also taking into account the behavior of the A / D converter 3, between the output of the compensator filter 15 (FIG. 1) or 15 f (for example FIG. 11) and the subtraction input of the differential unit 13 switched, depending on the arrangement, operating in the frequency or time domain, as is entered at 91 or 91 f in FIG. 11.
  • a further improvement in the effect of the compensation filter section 15 f can be achieved by superimposing, if necessary, noise r in the time domain, as shown schematically in FIG. 15, on the output side of the gain filter 5 f .
  • the current signal spectrum on the output side of the amplification filter 5 f is examined on a spectrum detector 125, for example on how very individual spectral lines are superior in terms of performance, i.e. how much the spectrum profile is peaked, maW, generally, for example, the energy density distribution of the spectrum .
  • a spectrum detector 125 for example on how very individual spectral lines are superior in terms of performance, i.e. how much the spectrum profile is peaked, maW, generally, for example, the energy density distribution of the spectrum .
  • a predetermined limit profile such as a predetermined energy distribution from dominant spectral lines to other spectral lines
  • digital noise r is preferably coupled into the superimposition unit 129 via a noise generator 127.
  • a filter unit can, as shown at 133 in FIG. 16, preferably the noise generator 127, which controls the noise in such a way that it is sufficiently weak compared to the instantaneous useful signal transmitted at the converter 9, for example by 40 dB.
  • the noise can also optionally be coupled in in the frequency domain.
  • the noise generator 127 consists, for example, of a BPRN, in the frequency domain according to 127a in FIG. 17, for example, of a table with noise spectra or a noise algorithm.
  • the output signal of the amplification filter 5 f is examined on a spectrum shape detector unit 125a, and if the spectrum shape leaves a predetermined limit characteristic, the output signal of the noise generator 127, which is led via the linear filter 133, is represented by the signal u (as represented schematically by the activation unit 135). 15, preferably superimposed on the input side of the limiter unit 90.
  • the transmission behavior of the filter 133 is preferably controlled by the current spectrum.
  • FIG. 17 shows a preferred embodiment variant of the noise lock in the frequency range according to the dashed embodiment variant with block 131 of FIG. 15.
  • the spectrum on the output side of the amplification filter 5 f is examined on a spectrum shape detector unit 125 b , analogously to the unit 125 a of FIG. 16.
  • the output signal of a noise generator 127a in which, for example, noise spectra are stored in tables and can be called up, is transmitted via a shaping filter 137 the spectrum on the output side of the amplification filter 5 f is then superimposed, as shown schematically by the switch 135a, when the spectrum shape detector unit 125 b detects an instantaneous spectrum shape which makes the aforementioned noise switching necessary.
  • the noise in the frequency range is superimposed on an addition unit 129a.
  • the shaping filter 137 is in turn controlled by the current spectrum, for example on the output side of the gain filter 5 f .
  • the noise coupling with instantaneous spectrum-controlled amplitude and / or frequency distribution is also considered to be inventive.

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  • Acoustics & Sound (AREA)
  • Health & Medical Sciences (AREA)
  • Neurosurgery (AREA)
  • Otolaryngology (AREA)
  • Physics & Mathematics (AREA)
  • Engineering & Computer Science (AREA)
  • General Health & Medical Sciences (AREA)
  • Signal Processing (AREA)
  • Amplifiers (AREA)
  • Finger-Pressure Massage (AREA)
  • Adornments (AREA)
  • Stereophonic System (AREA)
  • Complex Calculations (AREA)
  • Filters That Use Time-Delay Elements (AREA)
EP94117510A 1993-11-10 1994-11-07 Prothèse auditive avec suppression du couplage acoustique Expired - Lifetime EP0656737B1 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
EP94117510A EP0656737B1 (fr) 1993-11-10 1994-11-07 Prothèse auditive avec suppression du couplage acoustique

Applications Claiming Priority (3)

Application Number Priority Date Filing Date Title
EP19930118186 EP0585976A3 (en) 1993-11-10 1993-11-10 Hearing aid with cancellation of acoustic feedback
EP93118186 1993-11-10
EP94117510A EP0656737B1 (fr) 1993-11-10 1994-11-07 Prothèse auditive avec suppression du couplage acoustique

Publications (2)

Publication Number Publication Date
EP0656737A1 true EP0656737A1 (fr) 1995-06-07
EP0656737B1 EP0656737B1 (fr) 1998-01-21

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EP19930118186 Withdrawn EP0585976A3 (en) 1993-11-10 1993-11-10 Hearing aid with cancellation of acoustic feedback
EP94117510A Expired - Lifetime EP0656737B1 (fr) 1993-11-10 1994-11-07 Prothèse auditive avec suppression du couplage acoustique

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EP19930118186 Withdrawn EP0585976A3 (en) 1993-11-10 1993-11-10 Hearing aid with cancellation of acoustic feedback

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US (1) US5661814A (fr)
EP (2) EP0585976A3 (fr)
AT (1) ATE162679T1 (fr)
DE (1) DE59405093D1 (fr)
DK (1) DK0656737T3 (fr)

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DE19922133A1 (de) * 1999-05-12 2000-11-30 Siemens Audiologische Technik Hörhilfsgerät mit Oszillationsdetektor sowie Verfahren zur Feststellung von Oszillationen in einem Hörhilfsgerät
WO2001095578A2 (fr) * 2001-10-05 2001-12-13 Phonak Ag Procede permettant de verifier la presence d'une composante de signal et dispositif servant a mettre en oeuvre ce procede
US6628794B1 (en) 1999-11-26 2003-09-30 Siemens Audiologische Technik Gmbh Method and apparatus for level limitation in a digital hearing aid
US6650124B2 (en) 2001-10-05 2003-11-18 Phonak Ag Method for checking an occurrence of a signal component and device to perform the method
FR2853804A1 (fr) * 2003-07-11 2004-10-15 France Telecom Procede de decodage d'un signal permettant de reconstituer une scene sonore et dispositif de decodage correspondant
EP1469702A2 (fr) * 2004-03-15 2004-10-20 Phonak Ag Suppression de la réaction acoustique
US7324651B2 (en) 2004-03-15 2008-01-29 Phonak Ag Feedback suppression

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EP0704118B1 (fr) * 1994-04-12 2003-06-04 Koninklijke Philips Electronics N.V. Systeme d'amplification de signaux a suppression d'echo amelioree
CN1106715C (zh) * 1996-02-27 2003-04-23 皇家菲利浦电子有限公司 一种信号编码和译码的方法及装置
US5909497A (en) * 1996-10-10 1999-06-01 Alexandrescu; Eugene Programmable hearing aid instrument and programming method thereof
DE59814316D1 (de) * 1998-01-14 2008-12-18 Bernafon Ag Schaltung und Verfahren zur adaptiven Unterdrückung einer akustischen Rückkopplung
JP3267556B2 (ja) * 1998-02-18 2002-03-18 沖電気工業株式会社 エコー除去装置および送話器
US6347148B1 (en) 1998-04-16 2002-02-12 Dspfactory Ltd. Method and apparatus for feedback reduction in acoustic systems, particularly in hearing aids
KR100363252B1 (ko) * 1999-04-30 2002-11-30 삼성전자 주식회사 다중대역 보청기를 위한 적응 피드백 제거장치 및 방법
ATE339865T1 (de) * 1999-07-19 2006-10-15 Oticon As Rückkopplungsunterdrückung unter verwendung von bandbreite-detektion
US6633202B2 (en) 2001-04-12 2003-10-14 Gennum Corporation Precision low jitter oscillator circuit
EP1251714B2 (fr) 2001-04-12 2015-06-03 Sound Design Technologies Ltd. Système digital de prothèse auditive
CA2382362C (fr) * 2001-04-18 2009-06-23 Gennum Corporation Communication intercanal dans un instrument auditif numerique multicanal
CA2382358C (fr) 2001-04-18 2007-01-09 Gennum Corporation Detecteur numerique quasi quadratique
US20020191800A1 (en) * 2001-04-19 2002-12-19 Armstrong Stephen W. In-situ transducer modeling in a digital hearing instrument
EP1284587B1 (fr) * 2001-08-15 2011-09-28 Sound Design Technologies Ltd. Appareil auditif reconfigurable à faible consommation d'énergie
DE10244184B3 (de) * 2002-09-23 2004-04-15 Siemens Audiologische Technik Gmbh Feedbackkompensation für Hörgeräte mit Systemabstandsschätzung
US7756276B2 (en) 2003-08-20 2010-07-13 Phonak Ag Audio amplification apparatus
AU2004201374B2 (en) 2004-04-01 2010-12-23 Phonak Ag Audio amplification apparatus
AU2003236382B2 (en) 2003-08-20 2011-02-24 Phonak Ag Feedback suppression in sound signal processing using frequency transposition
EP1716721A1 (fr) * 2004-02-11 2006-11-02 Koninklijke Philips Electronics N.V. Suppression de retour acoustique
US8553899B2 (en) 2006-03-13 2013-10-08 Starkey Laboratories, Inc. Output phase modulation entrainment containment for digital filters
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US8452034B2 (en) 2006-10-23 2013-05-28 Starkey Laboratories, Inc. Entrainment avoidance with a gradient adaptive lattice filter
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DE19922133A1 (de) * 1999-05-12 2000-11-30 Siemens Audiologische Technik Hörhilfsgerät mit Oszillationsdetektor sowie Verfahren zur Feststellung von Oszillationen in einem Hörhilfsgerät
DE19922133C2 (de) * 1999-05-12 2001-09-13 Siemens Audiologische Technik Hörhilfsgerät mit Oszillationsdetektor sowie Verfahren zur Feststellung von Oszillationen in einem Hörhilfsgerät
US6628794B1 (en) 1999-11-26 2003-09-30 Siemens Audiologische Technik Gmbh Method and apparatus for level limitation in a digital hearing aid
WO2001095578A2 (fr) * 2001-10-05 2001-12-13 Phonak Ag Procede permettant de verifier la presence d'une composante de signal et dispositif servant a mettre en oeuvre ce procede
WO2001095578A3 (fr) * 2001-10-05 2002-12-19 Phonak Ag Procede permettant de verifier la presence d'une composante de signal et dispositif servant a mettre en oeuvre ce procede
US6650124B2 (en) 2001-10-05 2003-11-18 Phonak Ag Method for checking an occurrence of a signal component and device to perform the method
FR2853804A1 (fr) * 2003-07-11 2004-10-15 France Telecom Procede de decodage d'un signal permettant de reconstituer une scene sonore et dispositif de decodage correspondant
EP1469702A2 (fr) * 2004-03-15 2004-10-20 Phonak Ag Suppression de la réaction acoustique
EP1469702A3 (fr) * 2004-03-15 2004-11-17 Phonak Ag Suppression de la réaction acoustique
US7324651B2 (en) 2004-03-15 2008-01-29 Phonak Ag Feedback suppression

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DK0656737T3 (da) 1998-09-14
US5661814A (en) 1997-08-26
EP0656737B1 (fr) 1998-01-21
EP0585976A3 (en) 1994-06-01
ATE162679T1 (de) 1998-02-15
DE59405093D1 (de) 1998-02-26
EP0585976A2 (fr) 1994-03-09

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