EP0656737A1 - Hearing aid with cancellation of acoustic feedback - Google Patents

Abstract

The acoustomechanical disturbing feedback in a hearing aid having an amplifying filter section is compensated by means of a compensator (15). In this case, the signals are joined in the frequency range at the compensator (15) and preferably also in the amplifying filter section (5). For this purpose, time range/frequency range transformation units (20, 28), or corresponding back-transformation units (26, 24) are connected downstream of a subtraction unit (13) to which the compensator signal is coupled. <IMAGE>

Description

The present invention relates to a hearing aid according to the preamble of claim 1 and an electrical model according to claim 29.

The problems that arise in particular due to the acoustic feedback between the electrical-acoustic transducer and the acoustic-electrical transducer of such hearing aids are known and are discussed in detail, for example, in EP-A-0 415 677, which in this regard is an integral part of the present description is explained.

An attempt was made to solve these problems in principle, as shown in FIG. 1.

1 shows an acoustic-electrical (ak / el) converter 1 with a downstream analog / digital (A / D) converter 3, a digital amplification filter section 5, which on the output side is connected to a digital / analog (D / A) converter 7, the latter acts on the electrical-acoustic (el / ak) converter 9.

Block 11 represents the acoustic-mechanical interference feedback with the generally time-variant transmission behavior h. The feedback signal y (t) is superimposed on the useful signal v (t) and fed to the input of the ak / el converter 1, which on the output side, at times nT, supplies the discrete-time samples d (nT) required for digital processing.

To suppress the interference-feedback signal y (t), for example in DK Bustamante et al., "Measurement and adaptive suppression of acoustic feedback in hearing aids", Proc. 1989 IEEE ICASSP, 3: 2017-2020, 1989, proposed a differential unit 13, via a compensator 15, to supply the estimate ŷ (nT) formed from the output signal of the gain filter section 5 by filtering with an m-stage FIR (finite impulse response) filter. With the help of the known LMS (least mean square) algorithm, the filter coefficients are changed iteratively until the output signal e (nT) no longer correlates with the estimate ŷ (nT). The signal e (nT) required for the adaptation is fed to the compensator 15 via the adaptation input A.

Assuming that the useful signal v (t) or v (nT) and the amplified signal u (t) or u (nT) are uncorrelated, which can be achieved by a suitable choice of the time delay DT in the digital gain filter of the path 5, it becomes this makes it possible to increase the amplification of the amplification filter 5 by 6 to 10 dB compared to hearing aid devices without a compensator 15.

A disadvantage of this procedure is that, given an assumed filter length of the compensator 15 of m steps, 2 m multiplications per sample value of the A / D converter 3 are necessary, which leads to an extremely complex system. This is particularly important with a view to the miniaturization required for hearing aids.

Furthermore, it is necessary that the step length μ of the LMS algorithm for maintaining the speech signal transmission is chosen to be as small as possible, which means that the adaptation of the compensator filter 15 to the interference feedback path 11 becomes correspondingly slow, which increases the possible increase in the gain on the path 5 , for reasons of stability.

In a further development of the procedure shown in FIG. 1, an attempt was then made to give the system a stationary measurement signal , as described for example from "Feedback Cancellation in Hearing Aids: Results from a Computer Simulation", JM Kates, IEEE Trans. on Signal Processing, Vol. 39, No. 3, March 1991, or EP-A-0 415 677 . A noise signal was fed into the system as a stationary measurement signal.

A disadvantage of this procedure is the additional generator for the measurement signal and its necessary amplitude control to ensure a sufficient signal-to-noise ratio.

This procedure made it possible to increase the gain on the amplifier filter section by approx. 17dB with a compensator filter of the 32nd order.

Due to the disadvantages resulting from the latter technique with measurement signal coupling, a procedure according to FIG. 2 was finally proposed, according to "Integrated Frequency-Domain Digital Hearing Aid With the Lapped Transform", S.M. Kuo and S. Voepel, Electronics Letters, Vol. 28, No. 23, November 1992.

Accordingly, the signal processing was carried out both on the amplification filter section and on the compensator in the frequency range, for which purpose the output signal of the A / D converter 3 was transformed into the frequency range by means of an overlapping orthogonal transformation (LOT) on the unit 17. A corresponding inverse transformation (ILOT) at the unit 19 then again supplies the required signal u (nT) at the input of the el / ak converter 7.

Because with a suitable time domain / frequency domain transformation, in particular with the discrete Fourier transform (DFT) and the discrete Hartley transform (DHT), the convolution at the compensator and amplification filters 15 _f and 5 _f, when transitioning into the frequency range into a multiplication, this procedure basically results in a reduction in the computational and hardware expenditure. In order to obtain a finite transformation length that can be realized, a subdivision of the discrete signal d (nT) on the input side of the transformation unit 17 into blocks of a given length is necessary. Unfortunately, the errors associated with this, compared with conventional folding, cannot be eliminated in the arrangement according to FIG. 2 even with an overlapping block division. They lead to a time-variant system, even if, together with the interference feedback h, the compensation filter 15 _{f is} time-invariant or frozen.

Therefore, a compromise had to be made by choosing long block lengths of, for example, 512 samples, which in turn leads to inefficient compensation via the compensating filter 15 _f . Accordingly, the achievable gain increase on amplifier filter section 5 _{f was} limited to less than 10 dB.

It is an object of the present invention to provide a hearing aid device of the type mentioned, in which, while maintaining the advantages of signal processing in the frequency domain, time invariance of the system with time-invariant interference feedback is guaranteed, in which the computational or hardware outlay is further minimized is to such a degree that the signal processing can be easily implemented under the extremely limited space available with hearing aids.

This is achieved, starting from the latter hearing aid device, in that it is designed according to the characterizing part of claim 1.

The fact that the time domain / frequency domain transformation is not carried out in front of the differential unit 13 _f , as shown in FIG. 2, but rather that the difference is formed in the time domain, can surprisingly be obtained the required time invariance of the system. In particular, when suitably overlapping block division is selected, it is possible to implement the time domain / frequency domain transformations that are still used with significantly smaller block lengths, which in turn increases the compensation efficiency and therefore enables the gain on the gain filter section 5 _f according to FIG. 2 to be increased drastically.

The invention, with its preferred embodiment variants specified in the further claims, is subsequently explained step by step using figures, for example, and finally presented using an implementation example.

Fig. 1

anhand eines Funktionsblockdiagrammes, vereinfacht, ein bekanntes Hörhilfegerät, bei welchem die Signalverarbeitung zeitdiskret erfolgt;

Fig. 2

in Darstellung analog zu Fig. 1, ein weiteres bekanntes Hörhilfegerät, bei welchem die Signalverarbeitung an Rückkopplungskompensator und Verstärkungsfilterstrecke gemäss Fig. 1 im Frequenzbereich durchgeführt wird;

Fig. 3

in Darstellung analog derjenigen der Fig. 1 und 2, eine erste Ausführungsvariante eines erfindungsgemässen Hörhilfegerätes;

Fig. 4

eine weitere bevorzugte Ausführungsvariante des Hörhilfegerätes nach Fig. 3, dargestellt analog zu den Fig. 1 bis 3;

Fig. 5

ausgehend von dem in Fig. 4 dargestellten Hörhilfegerät, eine weitere bevorzugte Ausführungsvariante des erfindungsgemässen Gerätes in Darstellung analog derjenigen der Fig. 1 bis 4;

Fig. 6

anhand eines vereinfachten Signalfluss-Funktionsblockdiagrammes eine bevorzugte Realisationsform der dem Adaptionseingang und der Verstärkungsfilterstrecke vorgelagerten Transformationseinheit gemäss Fig. 5;

Fig. 7

anhand eines vereinfachten Signalfluss-Funktionsblockdiagrammes eine bevorzugte Ausführungsvariante der Verstärkungsfilterstrecke am erfindungsgemässen Gerät gemäss Fig. 5;

Fig. 8

anhand eines vereinfachten Signalfluss-Funktionsblockdiagrammes eine bevorzugte Realisation des Kompensatorfilters am erfindungsgemässen Gerät gemäss Fig. 5;

Fig. 9

anhand eines vereinfachten Signalfluss-Funktionsblockdiagrammes die Bildung des Schrittgrössensignals in Funktion der erfassten Signalleistung, welches Schrittgrössensignal, wie in Fig. 9 bevorzugterweise gebildet, bei der Realisation des Kompensatorfilters nach Fig. 8 eingesetzt ist;

Fig. 10

eine bei der Realisation des Kompensatorfilters gemäss Fig. 8 bevorzugterweise eingesetzte Einheit in vereinfachter Signalfluss-Funktionsblockdarstellung;

Fig. 11

ausgehend von der Darstellung gemäss Fig. 4 eines erfindungsgemässen Hörhilfegerätes, eine heute besonders bevorzugte Ausführungsvariante anhand des bereits vorgestellten Funktionsblockdiagrammes;

Fig. 12

ausgehend von der besonders bevorzugten Ausführungsvariante gemäss Fig. 11, einen Teil einer Weiterentwicklung mit Modellierung des elektrisch-akustischen Wandlers im Zeit- und/oder Frequenzbereich;

Fig. 13

ein Funktionsblock/Signalflussdiagramm eines elektrischen, im Zeitbereich arbeitenden Lautsprechermodells, wie es vorzugsweise zur Berücksichtigung des Lautsprecher-Uebertragungsverhaltens am erfindungsgemässen Hörhilfegerät gemäss den Fig. 3, 11 oder 12 eingesetzt wird;

Fig. 14

ausgehend von der Darstellung nach Fig. 12, eine Weiterentwicklung des erfindungsgemässen Gerätes, bei welcher die Modellierung und/oder die Amplitudenlimitierung und/oder die Verstärkung in Funktion des IST-Zustandes einer Batterie geführt wird;

Fig. 15

ausgehend von einem Gerät nach Fig. 11, eine weitere Verbesserung durch gegebenenfalls selektiv gesteuerte Rauschsignalaufschaltung im Frequenz- oder Zeitbereich;

Fig. 16

eine bevorzugte Realisation der Rauschaufschaltung gemäss Fig. 15 im Zeitbereich;

Fig. 17

eine bevorzugte Realisationsform der Rauschaufschaltung nach Fig. 15 im Frequenzbereich.

Show:

Fig. 1

based on a functional block diagram, simplified, a known hearing aid in which the signal processing is time-discrete;

Fig. 2

in representation analogous to FIG. 1, another known hearing aid device in which the signal processing on the feedback compensator and amplification filter section according to FIG. 1 is carried out in the frequency range;

Fig. 3

in representation analogous to that of Figures 1 and 2, a first embodiment of a hearing aid according to the invention;

Fig. 4

a further preferred embodiment of the hearing aid device according to FIG. 3, shown analogously to FIGS. 1 to 3;

Fig. 5

starting from the hearing aid device shown in FIG. 4, a further preferred embodiment variant of the device according to the invention in a representation analogous to that of FIGS. 1 to 4;

Fig. 6

on the basis of a simplified signal flow function block diagram, a preferred form of realization of the transformation unit upstream of the adaptation input and the amplification filter section according to FIG. 5;

Fig. 7

based on a simplified signal flow function block diagram, a preferred embodiment variant of the amplification filter section on the device according to the invention according to FIG. 5;

Fig. 8

based on a simplified signal flow function block diagram, a preferred implementation of the compensator filter on the device according to the invention according to FIG. 5;

Fig. 9

on the basis of a simplified signal flow function block diagram, the formation of the step size signal as a function of the detected signal power, which step size signal, as preferably formed in FIG. 9, is used in the implementation of the compensator filter according to FIG. 8;

Fig. 10

a unit preferably used in the implementation of the compensator filter according to FIG simplified signal flow function block representation;

Fig. 11

starting from the representation according to FIG. 4 of a hearing aid according to the invention, a particularly preferred embodiment variant based on the functional block diagram already presented;

Fig. 12

starting from the particularly preferred embodiment variant according to FIG. 11, part of a further development with modeling of the electrical-acoustic transducer in the time and / or frequency range;

Fig. 13

3 shows a functional block / signal flow diagram of an electrical loudspeaker model operating in the time domain, as is preferably used to take into account the loudspeaker transmission behavior on the hearing aid according to the invention according to FIGS. 3, 11 or 12;

Fig. 14

starting from the representation according to FIG. 12, a further development of the device according to the invention, in which the modeling and / or the amplitude limitation and / or the amplification is carried out as a function of the current state of a battery;

Fig. 15

starting from a device according to FIG. 11, a further improvement by optionally selectively controlled noise signal application in the frequency or time domain;

Fig. 16

a preferred implementation of the noise lock according to FIG. 15 in the time domain;

Fig. 17

a preferred implementation of the noise lock according to FIG. 15 in the frequency domain.

3 shows a basic principle of the present invention or of the hearing aid device according to the invention on the basis of a signal flow / functional block diagram. The reference symbols already used with reference to FIGS. 1 and 2 are used therein for the function blocks and signals already described there.

3 and 4, the discrete-time difference signal r (nT) from the A / D-converted output signal d (t) of the ak / el converter 1 and the output signal of the compensator filter 15 _f educated. Only the difference signal r (nT) on the output side of the difference formation unit 13 is subjected to an overlapping orthogonal transformation LOT.

3, the difference signal r (nT) is converted at a LOT transformation unit 20 into the adaptation control signal E [k], which is fed to the adaptation input A _{f of} the compensator filter 15 _f . Because the time domain / frequency domain transformation takes place in the LOT transformation unit 20 in blocks of a predetermined number of samples from the difference signal r (nT), [k] denotes the number of the signal block appearing on the output side of the transformation unit 20.

3, the difference signal r (nT) is supplied to the amplification filter section 5 in the time domain and fed to the el / ak converter 9 via the D / A converter 7. On the input side, the D / A converter 7 is acted upon by the discrete-time output signal u (nT) of the amplification filter section 5. This output signal u (nT) becomes a further orthogonal transformation unit 22 supplied and converted there from the time domain into the frequency domain. The output signal of the transformation unit 22 is fed as an input signal to the input E _{f of} the compensator filter 15 _f . The output signal Ŷ [k + 1] of said filter 15 _f is transformed back into the time domain at a reverse transformation unit ILOT 24 and its output signal ŷ (nT) is fed to the difference forming unit 13 as a discrete-time signal.

In addition to the embodiment variant in FIG. 3, according to FIG. 4 not only the signal processing is carried out on the compensation filter 15 _f in the frequency range but also on the amplification filter section 5 _f . For this purpose, the amplification filter section 5 _{f is} preceded by a transformation unit LOT 28 and the D / A converter 7 is a reverse transformation unit ILOT 26; the transformation unit 22 according to FIG. 3 is omitted.

Basically, and according to the wording of claim 1, as explained with reference to FIGS. 3 and 4, in contrast to known procedures according to FIG. 2, the difference is formed at the difference forming unit 13 in the time domain, as a result of which the above-mentioned disadvantages with regard to the time variance of the procedure 2 are resolved.

This results in the possibility of working on the LOT transformation units 20, 22, 28 and, correspondingly, on the ILOT reverse transformation units 24, 26 with significantly smaller block lengths than is possible in the procedure according to FIG. 2, for example according to one preferred embodiment of the present invention with block lengths of blocks k of 128 samples.

3, as mentioned, shows a first form of implementation which corresponds to the definition according to claim 2, namely in which a transformation unit LOT 20 or 22 is connected upstream of the signal input E _f and the adaptation input A _{f of} the compensator filter 15 _f .

A preferred embodiment variant is that according to FIG. 4, which corresponds to the definition according to claim 3, according to which the adaptation input A _{f of} the compensating filter 15 _f and the input of the amplification filter section 5 _{f are preceded} by a LOT transformation unit 20 or 28 and the input of the D / A converter 7 a corresponding ILOT reverse transformation unit 26.

Two simple techniques are available for block formation and processing in overlapping orthogonal transformations, namely the "overlap-save" and "overlap-add" techniques. In this regard, reference can be made in full to the relevant literature, such as, for example, "Signal Processing with Lapped Transforms", Henrique S. Malvar, Artech House, Boston, 1992.

In a preferred embodiment of the present invention according to the wording of claim 5, as shown in FIG. 4, the gain filter 5 _{f is also} preceded by a LOT transformation unit 28, the input of the D / A converter 7 is an ILOT reverse transformation unit 26, and further the output of the compensator filter 15 _{f is followed by} an ILOT reverse transformation unit 24. These transformation or reverse transformation units 28, 24 and 26 operate in the mentioned preferred embodiment according to the "overlap-save" technique. In contrast, the LOT transformation unit 20 upstream of the adaptation input A _f , in particular according to FIG. 4, preferably works according to the "overlap-add" principle.

In particular, these preferred embodiment variants of the insert the block processing techniques lead to a further preferred form of realization of the hearing device according to the invention, as shown in FIG. 5.

In contrast to FIG. 4, the time discrete differential signal r (nT) of a single LOT transform unit 30 is fed to here, from whose output signal both the adaption input A _f supplied adaptation signal E [k] as well as that of the enhancement filter path 5 _f supplied input signal R [k ] is formed.

As mentioned, the overlapping orthogonal transformations are preferably based on the DFT.

FIG. 6 shows a form of realization of the data transmission path between the time-discrete difference signal r (nT) on the output side of the difference forming unit 13 for the adaptation signal E [k] or the input signal R [k] to the amplification filter section 5 _f according to FIG. 5.

Accordingly, an overlapping orthogonal transformation unit 30a based on the DFT follows the output of the difference formation unit 13 with the time-discrete difference signal r (nT). As shown with the indexing OA, it works according to the "overlap-add" principle. To this end, the error block e [k] is formed by dividing r (nT) into sub-blocks of length N, each of which, in the variant preferred here with N = 64, by adding zeros to a total block length, here from 2N = 128 values , be extended, ie

${\text{e [k] = (0 ... 0, r ((k + 1) NT), r ((k + 1) NT + T) ... r ((k + 2) NT-T))}}^{\text{T}}\text{.}$

gegeben, wobei j (von 0 bis 2N-1) die Nummer der Blockstelle bezeichnet.Its DFT, namely E [k], is, in the preferred variant according to FIG. 5, directly to the adaptation input A _{f of} the compensation filter 15 _f fed. The following blocks, that is to say the numbers k and k + 1, are made available via a delay unit 32 with corresponding intermediate storage. A partial overlay in the unit 34 then provides the block R [k] directly, but now the "overlap-save" type, which in the implementation previously described as the preferred variant, according to FIG. 5, is fed directly to the amplification filter section 5 _f . The overlay in unit 34 is given by the formula

${\text{R}}_{\text{j}}{\text{[k] = E}}_{\text{j}}{\text{[k] + (-1)}}^{\text{j}}{\text{E}}_{\text{j}}\text{[k-1],}$

given, where j (from 0 to 2N-1) denotes the number of the block location.

This procedure significantly reduces the necessary hardware and computing power.

According to FIG. 7, within the gain filter section 5 _f acted upon by R [k], the actual gain filter 40 follows first, which is followed by a delay unit 42 with corresponding intermediate storage. Here, the parameter d denotes the total delay of the system (from the output of the A / D converter 3 to the input of the D / A converter 7), normalized with the overlap parameter of the partial block length N. Due to the block processing, there is a minimum delay time of N samples , corresponding to a minimum d-value of 1. In the variant preferred here with a partial block length of N = 64 and a total block length of 2N = 128, using a single partial compensator (as will be explained in more detail below with reference to FIG. 8), d set to the value 2.

The block signal U [k + 1] available on the output side is now fed on the one hand to the input E _{f of} the compensator 15 _f and on the other hand is subjected to an inverse DFT of the "overlap-save" type in the ILOT unit 26. Since the corresponding time signal u (nT) is delayed by a partial block length N, the numbering of U [k + 1] with the block number k + 1 is justified in retrospect.

bereitgestellt und, davon ausgehend, mit Hilfe von Teilkompensatoren, deren erster in Fig. 8 als Einheit 50 bezeichnet ist, die Teilschätzungen Ŷ₁[k+1] bis Ŷ_L[k+1] erzeugt, die ihrerseits in Einheit 52 zur Gesamtschätzung Ŷ[k+1] addiert werden. Wie Fig. 5 zu entnehmen ist, erfolgt dann in der ILOT-Einheit 24, in der bevorzugten Variante über eine inverse DFT der "overlap-save"-Art, die Rücktransformation in den Zeitbereich.FIG. 8 shows a preferred expansion variant of the compensator filter 15 _f on the hearing aid according to the invention according to FIG. 5. The block signals U [k + 1] to are thereby buffered with delay units of the type, as shown at 56 $\text{U [k + 1-L]}$

provided and, based on this, with the aid of partial compensators, the first of which is referred to in FIG. 8 as unit 50, which produces the partial estimates Ŷ₁ [k + 1] to Ŷ _L [k + 1], which in turn are provided in unit 52 for the overall estimate Ŷ [ k + 1] can be added. As can be seen in FIG. 5, the ILOT unit 24, in the preferred variant via an inverse DFT of the "overlap-save" type, then transforms back into the time domain.

ausgeführt, wobei j die Blockstelle von 0 bis 2N-1 und i die Teilkompensatornummer von 1 bis L bezeichnen.With reference to the first partial compensator, the partial estimate Ŷ₁ [k + 1] arises at the output of the multiplication unit 64, on which the block signals U [k + 1] and the block weight Ĥ₁ [k + 1] act at the input. The multiplication is for each block position according to the formula

${\hat{\text{Y}}}_{\text{i, j}}{\text{[k + 1] = U}}_{\text{j}}{\text{[k + 2-i] H}}_{\text{i, j}}\text{[k + 1]}$

executed, where j denotes the block position from 0 to 2N-1 and i the partial compensator number from 1 to L.

und die Schrittweite $\text{µ[k+1-1]}$

auf die Multiplikationseinheit 54, welche ausgangsseitig zusammen mit dem Blocksignal E[k] auf die Multiplikationseinheit 58 geführt wird. Der Ausgang von Einheit 58 wird dann in der Summationseinheit 60 entsprechend der Formel

zur Aktualisierung von H₁[k+1] verwendet. Hierbei bezeichnet j wieder die Blockstelle und i die Teilkompensatornummer. Der Index (*) steht für konjugiert komplex.The block weight H _i [k + 1] represents the current estimate in the frequency range for the i-th subrange of length N of the discrete-time impulse response h of the acoustic-mechanical Noise feedback 11. The estimate H _i [k + 1] is updated in advance of the formation of Ŷ _{i, j} [k + 1] with the aid of the old estimate H _i [k]. The block signal acts for this purpose, again with reference to the partial compensator 1 $\text{U [k + 1-1]}$

and the step size $\text{µ [k + 1-1]}$

to the multiplication unit 54, which on the output side is led to the multiplication unit 58 together with the block signal E [k]. The output of unit 58 is then in summation unit 60 according to the formula

used to update H₁ [k + 1]. Here again j denotes the block location and i the partial compensator number. The index (*) stands for conjugate complex.

durch Wahl der Teilblocklänge N unabhängig von der tatsächlichen Impulsantwortlänge der Störrückkopplung 11 eingestellt werden kann. Damit ist ein "trade-off" zwischen Verzögerung D und der die Effizienz der Bearbeitung bestimmenden Teilblocklänge N möglich. Weiter lassen sich einzelne Teilbereiche der Impulsantwort h, beispielsweise entsprechend den akustischen Nah- und Fernbereichen, gezielt durch entsprechende Blockgewichte im Frequenzbereich beeinflussen.Working with partial expansion joints has the advantage of minimal delay $\text{D = N}$

can be set by selecting the partial block length N independently of the actual impulse response length of the interference feedback 11. A "trade-off" between delay D and the partial block length N which determines the efficiency of the machining is thus possible. Furthermore, individual sub-areas of the impulse response h can be influenced in a targeted manner by corresponding block weights in the frequency range, for example in accordance with the acoustic near and far ranges.

In principle, any known method for guiding the step size µ [k] can be used.

FIG. 9 shows a preferred variant today for generating the normalized step size μ [k] according to FIG. 8, which is also used to stop the adaptation process. For example, starting from 8, this block signal before being fed to the multiplication unit 54 is used to calculate the current block signal µ [k] by feeding the block signal U [k] to a power detection unit 70, which in turn has two Interpolation filter 72 resp. 74 acts. On the output side, these interpolation filters control the scaling unit 78, which finally supplies the scaling variable S [k] required for the normalization of the reference step size μ₀ at the input of the multiplication unit 80.

und sind mit γ und c parametrisiert. Der Index j bezeichnet, wie hier üblich, die Blockstelle. In der bevorzugten Realisierung wurde γ = 0.8 und c = 1 für das Filter 72 und γ = 0.995 und c = 0.2 für das Filter 74 gewählt.The interpolation filters work according to the formula

and are parameterized with γ and c. As is customary here, the index j denotes the block location. In the preferred implementation, γ = 0.8 and c = 1 were chosen for the filter 72 and γ = 0.995 and c = 0.2 for the filter 74.

If γ = 1 is selected for the interpolator 74, this interpolator is omitted, and there remains a block signal P _U ^min which is constant over time and which may be sufficient for various applications and which further reduces the hardware and computing outlay.

gebildet, wobei die j wie üblich die Blockstelle bezeichnen.The scaling variable S [k] is now used on the one hand via the output of the filter 72, in FIG. 9 as a block signal P _U [k], to normalize the reference step size μ₀, but on the other hand also via the output of the filter 74 in FIG. 9 referred to as block signal P _U ^min [k], for freezing the adaptation process of individual frequency components when the power is insufficient. The scaling variable S [k] is according to the formula

formed, the j denoting the block location as usual.

in Einheit 64 von Fig. 8, über eine Projektionseinheit 62 geführt. Hierzu wird beispielsweise das Blockgewicht Ĥ_i[k+1] einer inversen DFT unterworfen (Einheit 82), anschliessend durch Nullsetzen der Blockstellen mit Index N bis 2N-1 gereinigt (Einheit 84) und schlussendlich wieder in den Frequenzbereich zurücktransformiert (Einheit 86).FIG. 10 shows a further preferred variant which, with the use of partial compensators according to FIG. 8, significantly improves the speech quality, with otherwise the same parameters. For this purpose, the estimate Ĥ _i [k + 1] of the partial compensator i, previously the multiplication with $\text{U [k + 2-i]}$

in unit 64 of FIG. 8, via a projection unit 62. For this purpose, for example, the block weight Ĥ _i [k + 1] is subjected to an inverse DFT (unit 82), then cleaned by zeroing the block locations with index N to 2N-1 (unit 84) and finally transformed back into the frequency range (unit 86).

As is known, the electrical-acoustic converter 9 is not linear in the sense that it no longer converts the input signal into the output signal linearly from certain input signal amplitudes. In addition to the acoustic distortions caused by this, it should be taken into account that the signal path via compensation filter 15 _f should be modeled as exactly as possible over the signal path via function blocks 7, 9, 11, 1 and 3 and, according to the previous explanations, the nonlinearities mentioned on converter 9 should not can reproduce. In addition, the maximum output level should also be adjustable in the hearing aid according to the individual needs of the user. The problem arises that the converter 9 is driven into its non-linear range, of course only if the individually set maximum output level can drive the converter in the mentioned range at all.

For this reason, it is further proposed, as shown in dashed lines in FIG. 3, in this embodiment of the hearing aid device according to the invention that the gain filter 5 has a limiter unit which operates in the time domain and is preferably adjustable 90, which limits the output signal of the amplification filter 5 with respect to the amplitude so that the converter 9 is never driven into its non-linear range and which also allows the maximum output sound level at the converter 9 to be adjusted according to individual needs, in particular also lower, as is the case with is indicated by the double arrows.

In the embodiment variant according to FIG. 4, this is achieved in that the amplification filter 5 _f operating in the frequency range is followed by a unit 90 _f , which limits the frequency components of the signal spectrum, taking into account their mutual phase position, in the frequency range in such a way that the output of the conversion unit 26 and the Digital / analog converter 7 produces a time-variable signal u (t) which never drives the converter 9 into the non-linear transmission range and which also allows the maximum individual modulation to be set.

The same procedure is also implemented with the unit 90 _f in the embodiment variant according to FIG. 5.

FIG. 11 shows a further embodiment variant of the hearing aid according to the invention, which largely corresponds to that shown in FIG. 4, with the difference that the reverse transformation unit 26 according to FIG. 4, now 26a, is provided directly on the output side of the amplification filter unit 5 _f and on the input side of the Compensation filter 15 _f a LOT transformation unit 22a of the type already discussed is arranged. Although at first glance such an embodiment hardly seems to bring any advantages compared to that according to FIG. 4, it nevertheless opens up the possibility explained below.

As can be seen from FIG. 4, which, as explained above, as well as FIG. 5, represents a preferred embodiment variant of the inventive hearing aid, the provision of a limiter unit is only possible in the frequency range because such a unit is also in the signal path with the compensation filter 15 _f must be effective.

As shown in FIG. 11 at 90, the function block structure shown here enables the provision of a limiter unit 90 which operates in the time domain, which is much easier to implement than one which operates in the frequency domain.

This also allows simple expansion with units for compensating for non-linear effects, as described below.

In order to ensure a sufficiently accurate identification of the converter 9 by the compensator filter 15 _f , its modulation is initially limited in order to prevent it from being operated in the non-linear range. This naturally results in a corresponding reduction in the maximum possible signal amplification of the hearing aid according to the invention from transducer 1 to transducer 9.

FIG. 12 shows a preferred embodiment variant of the signal processing on the device according to FIG. 11 upstream of the compensation filter 15 _f or downstream of the amplification filter 5 _f . According to FIG. 12, the non-linearity of the electrical-acoustic transducer 9 is basically simulated, ie modeled, in the signal path with the compensation filter 15 _f . This is implemented by a modeling unit 92, upstream of the transformation unit 22a according to FIG. 11 and therefore working in the time domain, and / or by a modeling unit 92 _f , downstream of the transformation unit 22a and thus operating in the frequency domain.

This procedure ensures that, depending on the quality of the modeling unit 92, the limits of the unit 90 are set higher and the output signal can thus be increased by up to 6 dB compared to the embodiment variant in FIG. 11. If necessary, the limiter function of the unit 90 can also be stopped.

The modeling unit 92 can be implemented, for example, as a simplified Viennese model, as suggested in R. Isermann, "Identification of Dynamic Systems", Springer-Verlag, 2: 238, 1988.

The transformation into the time domain between the gain filter 5 _f and the compensator filter 15 _{f also} allows the addition of a nonlinear correction filter in the signal path with the gain filter 5 _f in the same manner described above. As can be seen from FIG. 12, this is implemented by a modeling unit 94, connected downstream of the transformation unit 26a and thus operating in the time domain, and / or by a modeling unit 94 _f , connected upstream of the transformation unit 26a and therefore operating in the frequency domain.

Of course, in the preferred implementation variants according to FIGS. 11 and 12, it is possible to replace the LOT units 20 and 28 by a single LOT transformation unit 30, as shown in FIGS. 5 and 6.

13 shows the implementation of a loudspeaker model according to the invention in the time domain. It is used in particular in the hearing aid according to the invention, according to FIGS. 3 and 11 at the location of block 90 and according to FIG. 12 instead of blocks 92 or 90 and 94.

It comprises a pre-filter 100 with the transfer function F 1 (ω), essentially with a low-pass characteristic. The cutoff frequency ω 1 in the Bode diagram of the filter characteristic, which is shown qualitatively in block 100, is approximately 0.8 kHz, the amplification | F 1 | at the cut-off frequency ω₁ is approx. 0dB. Likewise, the asymptote slope S 1 is approximately 0 dB / DK.

The identification variables, namely corner frequency ω₁ and the asymptotic slopes S₁ and S₂, as well as the gain, for example at the corner frequency ω₁, are identified by identifying the speaker or transducer 9 to be modeled.

ergibt sich zu:

$\text{y = x + ax² + bx³ + cx⁴ + dx⁵.}$

A linear amplifier unit 102 is provided downstream of the prefilter 100, and the gain factor K is set to this. A non-linear amplification unit 104 is provided downstream of the linear amplification unit 102. The characteristic of the non-linear gain function $\text{Y = Q (x)}$

results in:

$\text{y = x + ax² + bx³ + cx⁴ + dx⁵.}$

For small input signals, the gain of the non-linear amplification unit 104 is one, so that the gain characteristic around the origin has a slope of one. For a higher input signal deviation x, the nonlinear gain characteristic, as is known from the loudspeaker or converter 9, exhibits saturation behavior.

The coefficients a, b, c, d and the gain K are in turn identified on the basis of the loudspeaker or converter 9 that is actually to be modeled.

Downstream of the non-linear amplification unit 104, a linear amplification unit 106 is again provided, by means of which the amplification K of the linear amplification element 102 is compensated for - K⁻¹. A filter unit 108 is provided downstream of it, essentially with a high-pass characteristic, which, as can be seen, essentially compensates for the frequency response of the pre-filter 100.

The loudspeaker modeling unit, as shown in FIG. 13, essentially consists of a linear amplifier part 102, 106, 100 and 108 and a non-linear amplification unit 104.

In addition to the two causes already mentioned, namely deliberate limitation of the maximum output signal level of the converter 9 according to individual needs or modulation of the converter 9 in its converter-specific, non-linear saturation range, symptoms of saturation or limitation can also be based on another cause, namely on the drop in the battery voltage, which feeds the device according to the invention. The aging of the battery that feeds the device causes, in particular at the D / A converter 7, a decrease in the signal amplification and a reduction in the modulation limit, i.e. the maximum analog modulation range becomes smaller with decreasing battery voltage.

In addition, the output impedance of the battery usually appears in series with the impedance of the electrical-acoustic converter 9. Thus, towards the end of the battery life, the battery output impedance and thus the replacement image, maW, which also includes the latter, follows the D / A converter 7, it changes the how was explained, to model non-linearities appearing on the output side of the converter 7.

3, 4 or 5, the limiter unit 90 in the time domain or 90 _f in the frequency domain by means of the instantaneous battery voltage and / or the instantaneous one To control battery impedance with regard to its limiting effect.

_B gemessen, resultierend in entsprechenden Messsignalen e(U_B) bzw. e( $\overline{\text{Z}}$

_B). Diese Messsignale steuern die Limitereinheit 90, analog im Frequenzbereich die Limitereinheit 90_f gemäss den Fig. 4, 5, 11 bzw. 12, 14 und/oder die Modelleinheiten 92, 92_f bzw. 94, 94_f von Fig. 12, 13, 14. Selbstverständlich werden dabei bevorzugterweise die Messsignale e nach Digitalisierung eingesetzt, wozu die Messeinheit 122 ausgangsseitig mit einem A/D-Wandler (nicht dargestellt) versehen ist.Starting from the variant according to Figs. 11 and 12, this procedure is shown schematically in FIG. 14. At the output of the battery unit 120, which, as indicated schematically with "block powering", feeds the various active components in the blocks of the hearing aid according to the invention, the current battery voltage U _B and / or the current impedance is measured on a measuring unit 122 $\overline{\text{Z.}}$

_B measured, resulting in corresponding measurement signals e (U _B ) or e ( $\overline{\text{Z.}}$

_B ). These measurement signals control the limiter unit 90, analogously in the frequency range the limiter unit 90 _f according to FIGS. 4, 5, 11 or 12, 14 and / or the model units 92, 92 _f or 94, 94 _f from FIGS. 12, 13, 14. Of course, the measurement signals e are preferably used after digitization, for which purpose the measurement unit 122 is provided on the output side with an A / D converter (not shown).

As a result, in particular the limiter limits and / or the model parameters are tracked in a controlling manner by the instantaneous battery output voltage or their instantaneous impedance.

The model parameters on the model units 92 and 92 _f , 94 and 94 _f are modified in the function of the above-mentioned measured variables on the battery 120 or by means of values stored in tables that can be called up and activated by the current measured variables.

As further shown in FIG. 14, as a function of the measurement signals e mentioned, a gain loss on the D / A converter 7 is compensated for due to a decrease in the battery voltage: If the battery voltage decreases and thus the gain on the converter 7, the measurement signal e at block 7 the gain, compensatory, increased accordingly. The battery voltage drop acts simultaneously as a signal limitation by a limiter and is best and preferably simulated by a battery output voltage-controlled limiter block 90 _{b in} front of the loudspeaker model 92 or 92 _f according to FIG. 14.

With the provision of the Limiterblockes 90 according to FIG _b. 14, the blocks 90 can be omitted, as may 92 or 92 _f blocks 94 and 94 _f omitted when provision of the blocks, and there is a relatively low cost a battery voltage-independent, in this realization, stable Feedback suppression achieved according to the invention.

On the other hand, the function of the mentioned block 90 _{b can be taken over} completely by providing the battery output voltage-controlled block 90 or 90 _f according to FIGS. 4 and 5.

Taking into account the signal limitation caused by battery voltage drop by means of the controlled limiter blocks 90, 90 _f and 90 _b is of great importance in order to ensure the stability of the hearing aid with battery voltages which vary within wide limits.

In order to ensure the stability of the feedback suppression or compensation even in a very noisy environment where, for example according to FIG. 11, the acoustic-electrical converter 1 is overdriven and thus becomes non-linear, if necessary, a non-linear model of the acoustic-electrical converter 1, possibly also taking into account the behavior of the A / D converter 3, between the output of the compensator filter 15 (FIG. 1) or 15 _f (for example FIG. 11) and the subtraction input of the differential unit 13 switched, depending on the arrangement, operating in the frequency or time domain, as is entered at 91 or 91 _f in FIG. 11. For the model 91, 91 _f , the explanations apply analogously to those which were made with regard to the model 92, 92 _{f of} the electrical-acoustic transducer. Providing such a "microphone" model on a feedback-compensated hearing aid is considered to be inventive in itself, as is a "loudspeaker model" therefor, as shown, for example, in FIG. 13.

A further improvement in the effect of the compensation filter section 15 _f can be achieved by superimposing, if necessary, noise r in the time domain, as shown schematically in FIG. 15, on the output side of the gain filter 5 _f .

For this purpose, as shown in FIG. 15, the current signal spectrum on the output side of the amplification filter 5 _{f is} examined on a spectrum detector 125, for example on how very individual spectral lines are superior in terms of performance, i.e. how much the spectrum profile is peaked, maW, generally, for example, the energy density distribution of the spectrum . If the spectrum characteristic monitored at the unit 125 exceeds a predetermined limit profile, such as a predetermined energy distribution from dominant spectral lines to other spectral lines, then digital noise r is preferably coupled into the superimposition unit 129 via a noise generator 127. In order to reduce the audibility of this noise, a filter unit can, as shown at 133 in FIG. 16, preferably the noise generator 127, which controls the noise in such a way that it is sufficiently weak compared to the instantaneous useful signal transmitted at the converter 9, for example by 40 dB.

As further shown in dashed lines at 131 in FIG. 15, the noise can also optionally be coupled in in the frequency domain. If the noise is injected in the time domain, the noise generator 127 consists, for example, of a BPRN, in the frequency domain according to 127a in FIG. 17, for example, of a table with noise spectra or a noise algorithm.

16, starting from the representation of FIG. 15, shows a preferred form of implementation of the noise lock in the time domain. For this purpose, the output signal of the amplification filter 5 _{f is} examined on a spectrum shape detector unit 125a, and if the spectrum shape leaves a predetermined limit characteristic, the output signal of the noise generator 127, which is led via the linear filter 133, is represented by the signal u (as represented schematically by the activation unit 135). 15, preferably superimposed on the input side of the limiter unit 90. As shown with the control connection sc, the transmission behavior of the filter 133 is preferably controlled by the current spectrum.

FIG. 17 shows a preferred embodiment variant of the noise lock in the frequency range according to the dashed embodiment variant with block 131 of FIG. 15. The spectrum on the output side of the amplification filter 5 _f is examined on a spectrum shape detector unit 125 _b , analogously to the unit 125 a of FIG. 16. The output signal of a noise generator 127a, in which, for example, noise spectra are stored in tables and can be called up, is transmitted via a shaping filter 137 the spectrum on the output side of the amplification filter 5 _{f is} then superimposed, as shown schematically by the switch 135a, when the spectrum shape detector unit 125 _b detects an instantaneous spectrum shape which makes the aforementioned noise switching necessary. The noise in the frequency range is superimposed on an addition unit 129a.

The shaping filter 137 is in turn controlled by the current spectrum, for example on the output side of the gain filter 5 _f .

Basically, the noise coupling with instantaneous spectrum-controlled amplitude and / or frequency distribution is also considered to be inventive.

Claims (34)

Hearing aid with an acoustic / electrical (ak / el) converter with an output-side A / D converter and an electro-acoustic (el / ak) converter with an input-side D / A converter, a gain filter section between the A / D and D / A converters and an adaptive compensator filter (15 _f ), the signal input of which is operatively connected to the D / A converter input, the signal output of which is operatively connected to the one input of a difference formation unit (13), the second input of the difference formation unit (13) being connected to the output of the A / D Transducer (3) is operatively connected, its output acts on an adaptation input (A) of the compensator filter (15 _f ) and the input of the amplification filter section (5 _f ), the signal (E _f ) and the adaptation input (A _f ) of the Signals supplied to the compensator filter (15 _f ) on at least one transformation unit (20, 28) which carries out a fast orthogonal transformation are transformed from the time domain to the frequency domain, characterized in that that at least one transformation unit (22, 20; 20, 28) is arranged on the output side of the difference formation unit (13) and a reverse transformation unit (24) corresponding to the transformation unit acts between the output of the compensator filter (15 _f ) and the associated input of the difference formation unit (13). Device according to claim 1, characterized in that a respective transformation unit (22, 20) is arranged upstream of the signal input (E _f ) and the adaptation input (A _f ) of the compensator filter (15 _f ). Device according to claim 1, characterized in that the adaptation input (A _f ) of the compensator filter and the input of the gain filter (5 _f ) each have a transformation unit (20, 28) upstream and the input of the D / A converter (7) a reverse transformation unit (26). Apparatus according to claim 3, characterized in that a common transformation unit (30) is arranged upstream of the adaptation input (A _f ) of the compensator filter (15 _f ) and the input of the amplification filter section (5 _f ). Device according to one of claims 1 to 4, characterized in that a transformation unit (20, 28; 30; 30a, 32, 34) upstream of the input of the amplification filter (5 _f ), one downstream of the output of the compensator filter (15 _f ) and one (26) reverse transformation unit upstream of the D / A converter work in the "overlap-save" technique. Device according to one of claims 1 to 5, characterized in that a transformation unit (30; 30a) upstream of the adaptation input (A _f ) of the compensator filter (15 _f ) operates according to the "overlap-add" technique. Device according to one of claims 1 to 6, characterized in that the output of the difference-forming unit (13) is followed by a transformation unit (30a) which works according to the "overlap-add" technique, its output on the adaptation input (A _f ) of the Compensator filter (15 _f ) acts and is supplied to a block memory arrangement (32), in which the following signal blocks, which are formed according to the "overlap-add" technique, are successively stored, with assigned memory locations for assigned block locations being added with the correct sign on an addition unit (34) in such a way that the output block of the addition unit represents a block using "overlap-save" technology and the output of the addition unit (34) is fed to the input of the amplification filter section (5 _f ). Device according to one of claims 1 to 7, characterized in that the gain filter section (5 _f ) comprises a gain filter (40) and, downstream of it, a delay unit (42). Device according to one of claims 7 or 8, characterized in that the compensator filter (15 _f ) comprises: - The input (E _f ) of the compensator filter series-connected delay stages (56), - A number 1 ≦ i ≦ L of partial compensators (50), on which partial estimation signals

${\hat{\text{Y}}}_{\text{i}}\text{[k + 1] for 1 ≦ i ≦ L}$

are generated, where k denotes the block number, counted in the time domain / frequency domain transformation on the output side of the difference formation unit (13), - An addition unit (52), to which the partial estimation signals Ŷ _i [k + 1] of all 1 ≦ i ≦ L partial compensators (50) are added and the output of which forms the output of the compensator filter (15 _f ). Device according to claim 9, characterized in that each partial compensator (50) comprises: - One with the input (E _f ) of the compensator filter (15 _f ) via a number of delay stages (56) connected partial compensator input, the number of delay stages corresponding to the number of upstream partial compensators, each delay stage (56) having the input and the output of a Partial compensator (50) connects, a first multiplication stage (54) operatively connected to the output of the partial compensator, - downstream of the output of the first multiplication stage (54), an input of a second multiplication stage (58), the second input of which is operatively connected to the adaptation input (A _f ), - The output of the second multiplication stage (58) acts via an accumulation unit (60) on the one input of a third multiplication stage (64), the second input of which is operatively connected to the input of the partial compensator (50) and the output of which on the addition unit (52) works. Device according to one of claims 1 to 10, characterized in that the input of the amplification filter stage is preceded by a transformation unit, the output signal of which, in addition to the amplification filter path, acts on a power detection unit (70), the output signal of which when the energy of the signal at the output of the transformation unit unites exceeds the given threshold, controls the effectiveness of a signal at the adaptation input. Device according to Claim 11, characterized in that the output of a fourth multiplication unit (80) acts on the second input of the first multiplication stage (54), the input of which is supplied with a signal corresponding to a reference step size (µ₀), the second input of which is the output of a scaling unit (78), the latter of which the outputs of two interpolation filters (72, 74) are fed, both of which via the power detection unit (70) from the output signal are applied to the gain filter section. Apparatus according to claim 12, characterized in that instead of one of the interpolation filters (74) a time-constant signal is supplied to the scaling unit (78) (γ = 1). Device according to one of claims 12 or 13, characterized in that the output of the accumulation unit (60) and the input of the third multiplication stage (64) are interposed with a reverse transformation unit (82), a zeroing unit (84) and a transformation unit (86). Device according to one of claims 1 to 14, characterized in that an amplitude-limiting unit (90, 90 _f ) is connected upstream of the electrical-acoustic (el / ak) converter (9). Device according to one of Claims 3 to 14, characterized in that a transformation unit (22a) is connected upstream of the input of the compensation filter (15 _f ) and a reverse transformation unit (26a) and an amplitude limitation unit (90) are connected downstream of the output of the amplification filter section (5 _f ). Device according to one of claims 1 to 16, characterized in that the compensation filter (15 _f ) at least one unit working in the frequency and / or time domain, modeling the electro-acoustic transducer (9) and / or the acoustic-electric transducer (1) 91, 91 _f , 92, 92 _f ) is connected upstream and / or downstream. Device according to one of claims 3 or 4, characterized in that that the compensation filter (15 _f ) is preceded by a transformation unit (22a) and the amplification filter section (5 _f ) is followed by a reverse transformation unit (26a) and the transformation unit (22a) upstream of the compensation filter (15 _f ) is an electrical-acoustic converter (9) and / or the acoustic-electrical converter (1) upstream in the time domain modeling unit and / or the transformation unit (22a) a modeling the electrical-acoustic converter (9) and / or the acoustic-electrical converter (1) in the frequency domain Unit (92 _f ) is connected downstream. Device according to one of claims 1 to 18, characterized in that, upstream of the compensation filter (15 _f ) and downstream of the amplification filter (5 _f ), one each the electroacoustic transducer (9) and / or the acoustic-electric transducer (1) preferably modeling unit (92; 94) is provided in the time domain. Device according to one of claims 1 to 19, characterized in that it comprises at least one limiter unit (90, 90 _f , 90 _b ) operating in the time domain or frequency domain and is electrically powered by a battery, that further a measuring device for detecting the battery ACTUAL -Status (122) is provided, the output of which controls the limiter unit. Device according to one of claims 1 to 20, characterized in that the D / A converter has a gain control input, the device is battery-powered, a measuring device (122) for the current state of the supply battery (120) is provided, the output of which is on the Gain control input of the D / A converter is performed. Device according to one of claims 1 to 21, characterized in that that it comprises at least one unit (91, 91 _f , 92, 92 _f , 94, 94 _f ) that models the electrical-acoustic converter (9) and / or the acoustic-electrical converter (1), preferably in the time domain, is battery-powered and one Measuring device (122) for the current state of the battery (120), the output of which is led to parameter control inputs on the at least one modeling unit. Device according to one of Claims 1 to 22, characterized in that, preferably in the time domain, a noise signal (r) is fed (129, 135a) to the compensation filter (15 _f ) on the input side, preferably at least temporarily. Device according to claim 23, characterized in that the signal on the output side of the amplification filter is fed to a detection unit (125, 125 _a , 125 _b ), by means of which the current shape of its spectrum is examined to determine whether it fulfills a predetermined condition or not, and that Output signal of the detection unit controls the activation (135, 135 _a ) of the noise signal. Device according to claim 23, characterized in that the noise signal is supplied via a shaping filter which is controlled by the instantaneous spectrum of the output signal of the differential unit. Hearing aid at least according to the preamble of claim 1, optionally according to one of claims 1 to 25, characterized in that a noise generator (127a) is provided in the frequency range, the output signal of which is coupled in on the output side of the amplification filter section ( _5f ). Hearing aid according to claim 26, characterized in that the output of the noise generator (127a) is passed through a shape filter (137), which, as a control signal for its shaping behavior, is supplied with the current spectrum of a signal on the output side of the difference-forming unit (13). Hearing aid optionally according to one of claims 1 to 27, characterized in that, at least temporarily, a noise signal is superimposed on the signal transmitted electrically thereon, via a linear filter (133), the transmission behavior of which is controlled by the current spectrum of the electrically transmitted signal. Hearing aid with interference feedback compensation, possibly according to one of claims 1 to 28, characterized in that a unit (91, 91 _f ) modeling the transmission behavior of the acoustic-electrical transducer (1) of the device is provided in a compensation branch. The behavior of an electrical-acoustic transducer, in particular an electric transmission unit simulating a loudspeaker, preferably for or on a hearing aid according to one of claims 1 to 29, characterized in that it has a linear (100, 102, 106, 108) and a non-linear one (104) Transfer part includes. Transmission unit according to claim 30, characterized in that the linear transmission part comprises linear amplifiers and filters. Transmission unit according to Claim 31, characterized in that the linear transmission part comprises an input-side pre-filter, essentially with a low-pass characteristic, to which the non-linear transmission part (104) is connected, the latter being a compensation filter unit (108) is connected downstream with a frequency response essentially inverse to the frequency response of the pre-filter. Transmission unit according to claim 32, characterized in that the non-linear transmission unit (104) is preceded by a linear amplification element (102) and a linear amplification compensation element (106) compensating the amplification of the linear amplification element. Transmission unit according to claim 31, characterized in that the non-linear part has a transmission characteristic with saturation behavior.

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