EP0459571A1 - Langsam-Wellen-Mikrostreifenübertragungsleitung und Anordnung mit einer solchen Leitung - Google Patents

Langsam-Wellen-Mikrostreifenübertragungsleitung und Anordnung mit einer solchen Leitung Download PDF

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Publication number
EP0459571A1
EP0459571A1 EP91201234A EP91201234A EP0459571A1 EP 0459571 A1 EP0459571 A1 EP 0459571A1 EP 91201234 A EP91201234 A EP 91201234A EP 91201234 A EP91201234 A EP 91201234A EP 0459571 A1 EP0459571 A1 EP 0459571A1
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EP
European Patent Office
Prior art keywords
line
conductive layer
slow wave
called
lines
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Granted
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EP91201234A
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English (en)
French (fr)
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EP0459571B1 (de
Inventor
Patrick Société Civile S.P.I.D. Gamand
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Philips Research Suresnes
Koninklijke Philips NV
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Philips Research Suresnes
Philips Gloeilampenfabrieken NV
Koninklijke Philips Electronics NV
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Priority claimed from FR9006626A external-priority patent/FR2662858A1/fr
Priority claimed from FR9008598A external-priority patent/FR2664448A1/fr
Priority claimed from FR9102813A external-priority patent/FR2673766A1/fr
Application filed by Philips Research Suresnes, Philips Gloeilampenfabrieken NV, Koninklijke Philips Electronics NV filed Critical Philips Research Suresnes
Publication of EP0459571A1 publication Critical patent/EP0459571A1/de
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P3/00—Waveguides; Transmission lines of the waveguide type
    • H01P3/02—Waveguides; Transmission lines of the waveguide type with two longitudinal conductors
    • H01P3/08—Microstrips; Strip lines
    • H01P3/081—Microstriplines
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/18—Phase-shifters
    • H01P1/185—Phase-shifters using a diode or a gas filled discharge tube
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P5/00—Coupling devices of the waveguide type
    • H01P5/08—Coupling devices of the waveguide type for linking dissimilar lines or devices
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P9/00—Delay lines of the waveguide type

Definitions

  • the invention relates to a wave transmission line, in slow wave mode, of the so-called microstrip type, including a first so-called lower conductive layer acting as ground plane, a second so-called upper conductive layer in the form of a ribbon of transverse dimensions and longitudinal specific, and a third non-conductive material disposed between these two conductive layers.
  • the invention also relates to couplers formed from such lines.
  • the invention also relates to circuits including such a line.
  • the invention relates, among these circuits, to a transceiver device including an integrated circuit comprising a frequency duplexer for transmitting a first signal and receiving a second signal on a single pole.
  • the invention particularly finds its application in the production of integrable transmission lines, that is to say that can be included in integrated circuits, and more particularly in monolithic and microwave integrated circuits known under the name of MMIC's (of English: Monolithic Microwave Integrated Circuits).
  • the invention finds its application in the miniaturization of transmission lines and allows the increase in the integration density of integrated circuits including these lines, and / or the increase in the operating performance of these circuits. .
  • the invention finds its application in transmission and reception in the microwave domain by means of a single antenna, the transmitted signals being isolated from the signals transmitted by this single antenna by means of the integrated duplexer.
  • a microstrip type transmission line is described in the publication entitled: “Properties of Microstrip Line on Si-So2 System", by HIDEKI HASEGAWA, et alii, in “IEEE Transactions on Microwave Theory and Techniques, vol.MTT-19, N ° 11, November 1971, pp. 869-881 ".
  • a microstrip type line consists of a stacked structure formed of a metal layer acting as a ground plane, of a semiconductor layer of silicon (Si), of a dielectric layer of silica (SiO2 ) and a metallic ribbon of predetermined transverse dimension.
  • the third mode called “slow wave” appears when the operating frequency is low, of the order of 10 to 103 MHz, and when the resistivity of the semiconductor layer is also low, of the order of 10 ⁇ 4 to 102 ⁇ .cm.
  • this "slow wave” mode magnetic energy is distributed in the semiconductor layer, while electrical energy is stored in the dielectric layer. The sum of these energies is transmitted perpendicular to the layers, through the thin dielectric layer of silica (SiO2). The phase speed therefore decreases due to the transfer of energy to the semiconductor-dielectric interface (Si / SiO2).
  • phase constant is expressed in terms of normalized wavelengths: ⁇ g / ⁇ 0, a ratio which is equal to the speed of propagation in the line divided by the speed of light in a vacuum.
  • the upper limit frequency strongly depends on the resistivity of the semiconductor layer and becomes maximum when the resistivity reaches 10 ⁇ 1 ⁇ .cm, this frequency remaining less than GHz.
  • phase constant and the characteristic impedance of the line are also very dependent on the transverse dimension of the ribbon, and on the thickness of the semiconductor + dielectric layers separating the ground plane of the ribbon.
  • this document teaches that the operation in slow wave mode has high losses which could be reduced by constructing a multilayer structure between the ground plane and the ribbon, this multilayer structure being formed by the alternation of semiconductor layers and of layers. thin dielectrics to reduce skin effect losses. If such a multilayer structure were used to make a microstrip line operating in slow wave mode, then the dimension of the line could be reduced, which would make it possible to reduce the dimensions of integrated circuits with the line operating in the frequency domain of the order of GHz or lower.
  • a technical problem which currently arises is the monolithic integration of microwave circuits on semi-insulating substrate. Indeed, if a microwave circuit is not integrated monolithically, it is less efficient due to losses in the links between substrates, it operates at lower frequencies due to the parasitic capacitances which appear, it shows a higher consumption, and it is more expensive because it requires larger surfaces of semi-insulating substrates, and more manufacturing steps.
  • the known transmission lines necessary for producing the microwave circuits for example the microstrip lines operating in quasi-TEM mode, today occupy a large surface area on the substrates, making monolithic integration difficult, as soon as the circuit becomes complex.
  • MICs Microwave Integrated Circuits
  • the object of the present invention is therefore to propose a transmission line in slow wave mode of the MICRORUBAN type, in which the propagation structure is fully compatible with integrated circuits, for example with microwave integrated circuits and in particular with MMICs.
  • an object of the invention is to propose a transmission line in slow wave mode of the MICRORUBAN type whose characteristics are independent of the characteristics of the substrate.
  • An object of the invention is to provide such a line devoid of ground plane on the rear face of the substrate.
  • An object of the invention is to propose such a line whose losses are not higher than those of microstrip lines operating in TEM or quasi-TEM conventional mode.
  • An object of the invention is to provide such a line, the dimensions of which are several times smaller than those of lines operating in conventional TEM or quasi-TEM mode, for identical line characteristics.
  • An object of the invention is to propose such a line capable of being associated with microwave circuits.
  • An object of the invention is to propose such a line, the production method of which is in complete synergy with the production methods of all conventional integrated circuits whatever the semiconductor substrate chosen for this circuit, without increasing the number of steps required. processes, and using only layers or materials used in said processes.
  • the problems are solved by means of a circuit as described in the preamble of claim 1, characterized in that the transmission line has, longitudinally, a periodic structure, each period, of length l, being formed of a said bridge followed by a said pillar, in that each bridge consists of a section of the upper conductive tape, of length l1 ⁇ l, disposed on the surface of said first part of the third material, which is of dielectric nature , and that each pillar is a capacity.
  • the line according to the invention can then be included in an MMIC circuit with all the advantages already mentioned which result therefrom.
  • Another object of the invention is to provide a slow wave transmission line, the principle of which is based on such a periodic structure, the dimensions of which are further reduced and the performance of which is also improved, all by simply changing the design. in the step of drawing the masks of the integrated circuit.
  • the first conductive layer serving as ground plane has at least one recess respectively under each bridge.
  • This line has the property of having a higher deceleration than the previous line at equal frequency. This property makes it possible to produce, for the same application, even shorter lines, therefore even more easily integrated. When we know the problems linked to the integration of microwave lines, this result constitutes a first-rate industrial advantage, without any great additional technological difficulty.
  • Another object of the invention is to provide a coupler of the so-called Lange coupler type which is easily integrated, and in particular which is synergistically produced with current microwave integrated circuits, and whose performance is also improved compared to that we can expect known devices.
  • a Lange coupler is known to those skilled in the art by the publication "Integrated Stripline quadrature Hybrids", IEEE, MTT, Dec. 1969, pp.1150-1151.
  • This coupler is produced in microstrip technology, that is to say by means of microstrip conductors arranged on a first face of a substrate of given thickness, the second face of which receives the ground plane. Therefore, by this production method, this coupler is not fully compatible with current integrated circuit technologies.
  • This known coupler consists of an odd number, that is to say at least 3, of parallel transmission lines, connected 2 to 2 alternately to form an interdigitated structure.
  • the middle line is called the line main, and the coupler is completely symmetrical about the middle of the main line. In particular, its inputs and outputs are symmetrical.
  • the length L of the main line defines the operating frequency band of this coupler. This length L is of the order of a quarter of the wavelength ⁇ of the transported signal.
  • the operation of the Lange coupler is based on the following principle: an electromagnetic field coupling is formed between the parallel lines. This coupling is of the capacitive or inductive type depending on the relationships between the length L of the main line and the wavelength ⁇ of the signals which propagate in the coupler.
  • One of the aims of the invention is therefore to provide a Lange coupler whose design is compact and whose dimensions are minimized compared to those of known devices.
  • an integrated duplexer or active duplexer, is known from the publication entitled: “Distributed amplifiers as duplexer / low cross talk bidirectional element in S band” by OP LEISTEN, RJCOLLIER AND RNBATES in "Electronics Letters March 3, 1988 , Vol.24, N ° 5, p.264-265 ".
  • a transceiver device including an integrated circuit comprising a frequency duplexer for transmitting a first signal and receiving a second signal on a single pole, characterized in that the integrated frequency duplexer is a coupler directional of the aforementioned type, having two said first poles connected by electromagnetic coupling to two said second poles, in that one of said first poles constitutes an input for the first signal from a first amplifier, and the other says first an output for the second signal, which propagates towards the input of a second amplifier, and in that one of the said second poles constitutes an output for the first signal and an input for the second signal and the other said second poles is isolated.
  • FIGS. 1 and 2 to 6 This exemplary embodiment is illustrated by FIGS. 1 and 2 to 6.
  • Figure la shows a slow wave line seen from the above, of MICRORUBAN structure.
  • This line is produced on a substrate 10 which can be of any material whatsoever.
  • a substrate 10 which can be of any material whatsoever.
  • completely insulating, fully conducting, semi-insulating or semiconducting this unlimited choice of materials for producing the substrate makes it possible to apply the invention to all kinds of circuits, in all conceivable technologies, when the circuit comprises a transmission line.
  • Figure 2b shows a longitudinal section along the axis BB 'of the line of Figure 1a. This figure shows that, in example I, to make the contact of the parts 3 of the tape 12 with the dielectric layer 2, the tape 12 is collapsed at the level of the parts 3. On the contrary, in the suspended parts 4, the strip 12 is raised by a height e1 with respect to the upper surface of the dielectric layer 2.
  • the hanging parts 4 are the parts in which the propagation takes place.
  • the strip 12 is suspended above a dielectric 1, of relative permissiveness ⁇ r1 .
  • Figure 2a shows a cross section of the line along the axis AA 'of Figure 1a, at a bridge 4, and Figure 2c shows a cross section of the line along the axis CC' of Figure 1a , at the level of a pillar 13.
  • ⁇ 1, ⁇ 2 the propagation constants respectively in the BRIDGE part 4, and in the PILLAR part 13.
  • l1, l2 the BRIDGE, PILLAR lengths already defined as l1 + l2 l Z1, Z2 the characteristic impedances respectively in the BRIDGES 4 and PILLARS 13 parts.
  • FIG. 3 represents the equivalent diagram of a unit cell of the line, that is to say comprising a half BRIDGE, a PILLAR and a second half BRIDGE.
  • B is the susceptance of the discontinuity between BRIDGE 4 on dielectric 1 and PILLAR 13 MIM.
  • the line is in complete manufacturing synergy with an integrated circuit MMIC.
  • Table I brings together the preferred values of the parameters for implementing the line in this example I.
  • FIG. 1a also shows that the dielectric 2 has a length slightly greater than that of the ground plane 11 (which can be connected to ground by studs 21) to allow the realization of an input E by a stud 22a, and of an output O of the slow wave line by a pad 22b.
  • Figures 4, 5 and 6 give curves showing the performance of a line, obtained under the conditions where the elements of the line have the values given in table I.
  • Figure 4 shows the slow wave factor ⁇ 0 / ⁇ g as a function of frequency F in GHz. From this figure, we deduce that the relative effective permissiveness ⁇ reff is very high at low frequencies, frequencies for example less than 4 GHz, then remains almost constant between 4 and 20 GHz, with a value of the order of 20. This value must be compared with effective relative permissivity values known to a person skilled in the art for conventional MICRORUBANS lines, which are of the order of 6 to 8 when the line is made on alumina (Al2O3) or on a semiconductor.
  • FIG. 5 represents the real and imaginary parts, respectively Re (Z c ), and Im (Z c ), of the characteristic impedance Z c of this line.
  • the real part of the impedance Z c is extremely small. This line according to Example I will therefore find very interesting applications in the production of a low impedance line for an impedance transformer.
  • FIG. 6 shows on the one hand the losses ⁇ in the line, expressed in dB / cm, as a function of the frequency F in GHz, and on the other hand the losses ⁇ ′ in dB relative to the wavelength ⁇ g as a function of said frequency F. These losses per cm are slightly higher than those of a conventional MICRORUBAN line.
  • the slow wave line has a total length ⁇ reduced by approximately 2 times compared to a conventional MICRORUBAN line.
  • the performance of the slow wave line is not deteriorated compared to a conventional MICRORUBAN line, while on the contrary it has the advantage of being shorter, and therefore more easily integrated.
  • FIG. 1b seen from above and by FIG. 7 which is a section along the axis BB 'of FIG. 1b.
  • the dielectric layer 2 was continuous from one end to the other of the line.
  • the layer 2 is eliminated under the BRIDGES.
  • it is essential for producing the MIM structure of the pillars 13.
  • it was considered that, in example I, its influence under bridges 4 was negligible.
  • FIG. 1b This example is illustrated by FIG. 1b and by FIG. 8.
  • the slow wave line does not show any changes in the schematic representation seen from above and can therefore be illustrated by FIG. 1b.
  • Figure 8 is a section along the axis B-B 'of Figure 1b in this embodiment.
  • the dielectric 2 of the MIM structure of the pillars 13 has the same thickness as the dielectric 1 placed under the bridges 4.
  • the layer of dielectric 2 which could remain under the BRIDGES 4 in the example I should be excluded in this example III, as the possibility was shown in example II.
  • FIG. 1a This example can be illustrated by FIG. 1a, seen from above and by FIG. 9.
  • the slow wave line does not show any change in the schematic representation of FIG. 1a seen from above.
  • the ratios between the thicknesses e1 and e2 the ratios between the lengths l1 and l2 will be very different.
  • the main characteristic of this line is that the periodicity l shows a growth and in particular a geometric growth.
  • the growth factor can be included between 1 (1 being not included since we would then be in the case of the previous examples) and approximately 3.
  • Example I As regards the technology proper of such a line of periodicity l not constant, the person skilled in the art can preferably adopt that of Example I which is particularly easy to implement. But nothing prevents the creation of new variants by applying to this example V the teaching drawn from examples II to IV.
  • FIG. 1d This example is illustrated by FIG. 1d seen from above and by FIG. 11.
  • the conductive layer 11 itself has a periodic structure, of period l.
  • a diode 13 ′ polarized by a DC bias voltage V DD has been produced which can have different values.
  • the DIODE 13 ′ is more conveniently a field effect transistor with a Schottky gate, whose source S and the drain D short-circuited are brought to the DC bias voltage V DD and whose gate G is brought to ground M.
  • the substrate 10 is no longer arbitrary, as in the previous examples, but must include an active area 10a, of a semiconductor material, for example conductivity type N, the rest of the substrate 10b on either side of the active layer 10a being semi-insulating.
  • Regions 10a and 10b can be layers of material chosen from semiconductors such as: silicon (Si) or gallium arsenide (GaAs) for example.
  • the Schottky gate transistor 13 ′ is produced for example as follows: A semi-insulating layer 10b and regions 10a called active zones are produced by any means known to those skilled in the art of integrated circuits.
  • the active zones 10a are produced with a periodicity l chosen for the slow wave line.
  • the active areas 10a must have the dimensions necessary and sufficient to receive a Schottky gate field effect transistor. This technology is known to anyone skilled in the art of integrated circuits.
  • the conductive layer 11 is then produced. outside the active regions 10a, the conductive layer 11, the material of which is preferably chosen from metals capable of forming a Schottky grid, has the transverse dimension W1 determined as in the previous examples.
  • the metal layer 11 is on the other hand narrowed (see FIG. 1d). Longitudinally, along the axis BB 'of FIG. 1d, it has a dimension known as the gate width of the Schottky transistor and perpendicular to the axis BB', it has a small dimension of the order of ⁇ m known as the gate length of the transistor Schottky. Then ohmic contacts of a material 14 forming source pads S and drain D are disposed on either side of the gate G according to a conventional scheme of field effect transistor with Schottky gate.
  • the Schottky gate transistor 13 ′ is illustrated in FIG. 11 in section along the axis CC ′ in FIG. 1d.
  • the ribbon 12 is then produced, showing bridges 4 in the regions of the metal layer 12, where the latter has the dimension W1.
  • the ribbon 12 is divided into two parts 12a and 12b, the part 12a coming to establish the surface contact of the ohmic contact of source S, and the part 12b coming to establish the contact on the surface of the ohmic drain contact D for example .
  • the device is symmetrical with respect to the axis BB 'as well as with respect to the axis CC' of Figure 1d.
  • the parts 12a and 12b may consist of air bridges, or else a thin insulating dielectric layer such as the layer 2 described in the preceding examples may be provided at the same time under the bridges 4 and slightly projecting from the metal layer 11 in the Schottky grid regions, while leaving the ohmic contacts stripped on which the ribbon parts 12a and 12b come to rest and establish the electrical contact.
  • the sources S and drain D of the transistors 13 ′ are short-circuited and the Schottky gate G is grounded M via the metal layer 11.
  • connection line 15 to connect at least one ohmic contact S or D to an adjustable bias voltage V DD .
  • the strip 12, its parts 12a and 12b can be produced by any metal suitable for producing the second interconnection levels of the integrated circuits. Consequently, the connection line 15 which connects the ohmic contacts can be produced using the same technology.
  • phase ⁇ of the slow wave line is then electronically adjustable by adjusting the bias voltage V DD which varies the gate-source capacitance of the transistor 13 '.
  • FIG. 12 schematically seen from above.
  • FIG. 12 represents the connection of such a low impedance slow wave line and of reduced surface area, with a high impedance coplanar line.
  • coplanar line is meant a line made on the main face of the integrated circuit or MMIC, showing a central conductive tape of small transverse dimension disposed between two conductive tapes of larger transverse dimension.
  • the impedance of the coplanar line depends on the transverse dimension of the central conductive tape in which the distance which separates it from the two other tapes generally connected to a reference potential or mass is propagated.
  • the phase shift (generally expressed in wavelength, for example ⁇ / 4, ⁇ / 2) depends on the length of the line.
  • Coplanar lines can be used for both high impedance lines and low impedance lines. But, if the high impedance coplanar lines have dimensions compatible with integrated circuits, on the other hand, the low-impedance coplanar lines have dimensions, notably transverse, which occupy an enormous surface of the integrated circuit, which is entirely unfavorable for monolithic integration.
  • the low impedance slow wave line then makes it possible, by calculating its length and its characteristics appropriately, to form a line having for example the same phase shift as a coplanar line, ( ⁇ / 4, ⁇ / 2).
  • the part P1 delimited by broken lines is the low impedance slow wave line according to the invention, and the part P2 is a high impedance coplanar line as known to those skilled in the art.
  • a first metallization level will form the ground plane 11 of the slow wave line P1 separating into two ribbons to form the ground lines 11a and 11b of the coplanar line P2.
  • the slow wave line P1 will include, produced on the conductive layer 11, a dielectric layer 2, as already described, extending beyond the ground plane 11 of the slow wave line P1 in the regions necessary to avoid short circuits between the ground plane 11 and the line 12 produced subsequently.
  • the slow wave line P1 will include the ribbon 12, realizing as already described pillars 13 and BRIDGES 4, ribbon 12 which continues directly on the substrate 10 between the ground lines 11a and 11b to form the coplanar structure of the line P2.
  • the dielectric layer 2 it is generally necessary for the dielectric layer 2 to extend beyond the ground plane 11 of the slow wave line P1 on the side of the coplanar line P2 to avoid short circuits between the ground plane 11 and the line 12.
  • the dielectric layer 2 is also extended beyond the ground plane 11, and the strip 12 is provided with a output O as shown in Figures 1a, 1b, 1c.
  • the low impedance slow wave line had a conductive plane 11, connectable to ground, in contact with the upper main surface of the substrate.
  • contact with another ground plane made on the second face of the substrate, or rear face of the substrate can be made as is known to those skilled in the art under the name of "metallized hole”.
  • FIG. 13 we show an example of application of the impedance transformer described in example VII, to an integrated circuit.
  • the circuit includes a transistor, for example T1 field effect, having a gate G grille for receiving a signal F signal in a band of given frequencies, having a drain D1 connected to a DC bias voltage V D1 through a load R1, having an output O1 for said signal and having a source S1 for example connected to ground M.
  • a transistor for example T1 field effect, having a gate G grille for receiving a signal F signal in a band of given frequencies, having a drain D1 connected to a DC bias voltage V D1 through a load R1, having an output O1 for said signal and having a source S1 for example connected to ground M.
  • a circuit based on an impedance transformer P1 + P2 can be applied to the gate G1 of the transistor T1.
  • a high impedance line P2, for example ⁇ / 4 is connected by one end to the gate G1 and by its other end both to a low impedance line P1 slow waves according to the invention and to a DC bias voltage V G1 .
  • the low impedance line P1 is therefore connected at one end to both P2 and V G1 , and its other end is open in this application.
  • the slow wave line according to the invention has wide applications in integrated circuits of all kinds as well as in MMICs (microwave) because its operation can be, as we said, indifferent to the substrate, which it presents small dimensions compared to other lines having the same characteristics, and that it is compatible with all the integrated circuit technologies used to date.
  • FIGS. 14a, 14b, 14e, 2c This exemplary embodiment is illustrated by FIGS. 14a, 14b, 14e, 2c.
  • FIG. 14a shows a slow wave line seen from above, of MICRORUBAN structure, having first characteristics identical to those of the line of example II.
  • this line is produced on a substrate 10 which can be of any material whatsoever.
  • a substrate 10 which can be of any material whatsoever.
  • the structure also comprises, with respect to Example II, an essential element consisting of parts 5 in which the layer 11 of the ground plane, like the dielectric layer 2, are hollowed out under the suspended parts 4, so that the surface of the substrate 10 appears.
  • the recess 5 is unique under each suspended part 4, and the longitudinal dimension of the recess 5 is: l3 ⁇ l1
  • the value of l3 can approach that of l1 to within a few%, or be equal.
  • the hanging parts 4 are the parts in which the propagation takes place. In these parts, the strip 12 is suspended above a single dielectric 1, of relative permissiveness ⁇ r1 .
  • Figure 14e shows a cross section of the line along the axis AA 'of Figure 14a, at a bridge 4, and Figure 2c remains valid to show a cross section of the line along the axis CC' of FIG. 14a at the level of a pillar 13.
  • the value of the MIM capacities of the parts 13 is linked to l2, to e2 and ⁇ r2 .
  • the recesses 5 arranged in the bridge regions 4 play the role of inductors, making it possible to obtain an increase in the characteristic impedance of the line.
  • ⁇ 1, ⁇ 2 the propagation constants respectively in the BRIDGE part 4, and in the PILLAR part 13.
  • l1, l2 the BRIDGE, PILLAR lengths already defined as l1 + l2 l l3 the length of the recesses under the bridges equivalent to l1, Z1, Z2 the characteristic impedances respectively in the BRIDGES 4 and PILLARS 13 parts.
  • phase constant ⁇ is carried out in the same manner as described in Example I. It follows from these calculations that by choosing: l1, l2, l3 ⁇ r1 , ⁇ r2 e'1 and e2 W1 and W2 appropriately, the phase speed of the line is low. Hence the existence of the so-called slow wave mode already described in Example I.
  • This 3-layer structure was the result of constant teaching of the state of the art, and this teaching was an obstacle to an evolution making it possible to obtain an improvement compared to the slow wave structure mentioned above.
  • an increase in the characteristic line impedance has been achieved, by forming the recesses 5 in the ground plane M under the bridges, recesses 5 which increase the role inductive of the line constituting the bridge.
  • the recesses 5 indeed produce the desired favorable effect of additional deceleration, by acting both on the characteristic impedance of the line, on the thickness of dielectric e'1 under the bridges, on the value of the permissiveness ⁇ r1 since the only most favorable dielectric can be found under the bridges, and all this while benefiting from a technology which is easy to implement, the recesses 5 being produced during conventional stages of integrated circuit technology.
  • Table II below collates the preferred values of the parameters for implementing the line in this example X.
  • FIG. 14a shows that the other characteristics of the line of example X are very comparable to those of the line of examples I and II shown in FIGS. 1a and 1b.
  • FIG. 5e is also valid for representing the real and imaginary parts, respectively Re (Z c ) and Im (Z c ) of the characteristic impedance Z c of this line.
  • Figure 6 is also valid for showing the losses ⁇ in the line, expressed in dB / cm, as a function of the frequency F in GHz.
  • the curve ⁇ ′ in this figure 6 represents the losses in dB per wavelength.
  • the slow wave line has a total length ⁇ reduced compared to the line of Example I.
  • the reduction in lengths is inversely proportional to the deceleration factor R.
  • R was of the order of 2.5, while R was of the order of 4 in the line described in Example I.
  • R is of the order of 4.5.
  • the performance of the slow wave line according to the invention is not deteriorated, while it is notably shorter.
  • the present slow wave line structure produces losses evaluated at around 1dB.
  • FIG. 14c seen from above and by FIG. 14d which is a section along the axis BB 'of FIG. 14c.
  • a variant to this embodiment XI which proceeds from the same principle, is to provide for the capacities 13, capacities of different values, distributed alternately along the line. There is thus also obtained a period in the line period, and a consequent improvement in the line slowing factor.
  • one of the slow wave lines described above is applied to the production of a Lange coupler.
  • the coupler known from the publication IEEE, MTT, Dec. 1969, p.1150-1151 cited is constituted by at least 3 parallel lines connected 2 to 2 alternately to form an interdigitated structure.
  • the cited publication shows a 3 dB coupler with 5 transmission lines. An electromagnetic field coupling appears between the adjacent parallel lines.
  • FIG. 16a shows schematically this coupler.
  • FIG. 16b represents the same coupler seen from above, in a simplified manner, produced by means of layers specific to integrated circuits.
  • the coupler comprises two so-called input poles N1 and N2, and two so-called output poles N3 and N4.
  • the Lange coupler consists of 5 parallel microstrip lines including a so-called main line 110, electrically connected to lines 111 and 114, and two lines 112 and 113 electrically connected together, and forming a structure interdigitated by the fact that line 112 is arranged between lines 110 and 111 and line 113 between lines 110 and 114.
  • the coupler is symmetrical: that is to say that if N3 and N4 are inputs, then N1 and N2 are outputs .
  • Lines 110 and 111 are electrically connected directly to pole N1 by a simple conductor 101.
  • the lines 110 and 114 are electrically connected directly to pole N4 by a single conductor 104.
  • Line 112 and line113 are electrically connected to poles N2 and N3 respectively by single conductors 102 and 103.
  • poles N2 and N3 are electrically connected, by this assembly, crosswise with respect to the poles N1 and N4, as shown in Figure 16a and in Figure 16b.
  • the adjacent lines 110 and 112, the 110 and 113 are respectively parallel over a length L, while, in the interdigitated structure 110, 111, 112, the line 111 is parallel to the line 112 over a length equal to L / 2. It is the same in the interdigitated structure 110, 114, 113, where the line 114 is parallel to the line 113 over a length also L / 2.
  • the length L can be of the order of a quarter of the wavelength ⁇ of the signal transported according to the prior art.
  • Lines 111, 112, 110, 113, 114 of the Lange coupler can be produced using slow wave lines according to the invention.
  • the connections 115, 116, 117 and 118 are formed by means of a conductive layer arranged at a level different from the layers 11 and 12, with openings on the layer 12 at the appropriate places to form the electrical connection with layer 12 according to a technique known as VIA well known to those skilled in the art, and with portions of insulating layers in the parts where on the contrary the electrical connection is not desired with layers 11 or 12.
  • the other simple connections can be formed by means of parts of the conductive layer 12.
  • FIG. 16c represents an enlarged part of the coupler of FIG. 16b, in which it appears that the lines used by way of nonlimiting example to produce the coupler of example XII, are those described in example X.
  • FIG. 16d represents an enlarged part of the coupler of FIG. 16b, in which it appears that the recesses 5 of the parallel lines, for example 112, 110, 113, 114 can be grouped together, to form a single recess 5, the bridges 4 being respectively opposite for all the lines, and the pillars 13 also.
  • This device has a technological advantage over the previous one, due to its simplicity of construction; in fact the mask relating to the recesses 5 is less critical to position.
  • This coupler then accepts the same operating principle as the known coupler. By making the lines necessary for the formation of such a Lange coupler, by means of the slow wave lines according to the invention, we also obtain the advantages that this device is very efficient and much more compact, compatible with circuit projects. integrated with high density, and of a low cost for the applications general public, in the field of television or the automobile for example.
  • FIG. 17 shows on the curve M, the adaptation of the coupler in dB as a function of the frequency F, and on the curve K the coupling in dB, as a function of the frequency F.
  • a device conventional transceiver includes an input Q1 for a first signal V1, at the frequency F1, propagating through an amplifier ⁇ 1 then through a duplexer 50, to an antenna A, then to the outside environment .
  • This signal is applied to the N1 pole of the duplexer 50 and exits to the N3 pole of this duplexer 50.
  • This device further comprises an output Q2 for a second signal V2 at the frequency F2.
  • This signal is first picked up by the same antenna A then it propagates through the duplexer 50, in which it enters the pole N3 and through an amplifier ⁇ 2 towards the output Q2.
  • this coupler there are circulated in this coupler two signals V1 and V2 at two different frequencies F1 and F2.
  • the Lange coupler is broadband, greater than 1 octave, the difference between the frequencies F1 and F2 is not a drawback if it is less than this passband, for example less than 1 octave.
  • the length L of the main line will be chosen as a function of the wavelength ⁇ of the weakest signal, generally V2.
  • the increase in the number of fingers makes it possible to increase the coupling factor and to decrease the losses in the coupler.
  • the losses are 3 dB; with 6 fingers (or 7 lines), the losses are 2 dB, etc ...
  • the first signal V1 at the frequency F1 is applied to the pole N1 of the Lange coupler as shown in FIG. 16b, and exits through the pole N3, to be emitted then by an antenna A to the outside environment.
  • the second signal V2, at the frequency F2, picked up by the antenna is applied to the Lange coupler on the same pole N3 (so as to solve the problem of using a single antenna), and leaves the coupler by the pole N2.
  • the fourth pole N4 of the Lange coupler is connected to ground through an impedance Z C.
  • the conductor 101 (or pole N1) is an input
  • the conductor 102 (or pole N2) is an output
  • the conductor 104 (or pole N4) is isolated
  • the conductor 103 (or pole N3) is both an input and an output.
  • the conductor 103 is for example only an input and the conductors 101 and 102 are then only phase-shifted outputs, the conductor 104 being insulated.
  • the coupler is connected as shown in FIG. 19 on the one hand to the antenna A and on the other hand to the amplifiers ⁇ 1 and ⁇ 2.
  • the signal V1 at the frequency F1 to be transmitted is processed by an amplifier ⁇ 1 with high gain and high insulation, and the signal V2 of frequency F2 received is processed by a low noise ⁇ 2 amplifier.
  • the operation of the transceiver device is as follows: The signal V1 to be transmitted at the frequency F1 first enters the node Q1 of the transmitter-receiver device, then is processed by the amplifier ⁇ 1.
  • the signal V1 to be transmitted at the frequency F1 also passes directly by conduction into the characteristic impedance Z c connected to the output pole N4;
  • the signal to be transmitted V1 at the frequency F1 then propagates from the pole N3 of the coupler to the outside environment by means of the antenna A.
  • the latter receives the second signal V2 at another frequency F2, of amplitude generally much lower than the first signal V1 of frequency F1.
  • This second signal V2 passes by conduction, directly from the input-output pole N3 to the output pole N2. Then the second signal V2 is processed as already said by the low noise amplifier ⁇ 2 and leaves the device at the node Q2.
  • duplexer 50 a coupler with branches, as described for example in the publication "Millimeter wave engineering and applications" by P. BHARTIA and IJ BAHL at John Wiley and Sons, New-York (A Wiley-Interscience Publication) p.355, or even in the publication de Microwave Journal, July 1988 p.119 and pp.122-123, entitled “Microstrip Power Dividers at mm-wave frequencies” by Mazen Hamadallah (p.115).
  • a branch coupler comprises two sections of line 201 and 202 of length L and of impedance Z c ⁇ 2, connected at each of their ends by two sections of line 203 and 204 of impedance Z c and of length L.
  • each of impedance Z c In series with the first line sections 201 and 202, there are line sections to form the poles N1 and N2 on the one hand, and N3 and N4 on the other hand, each of impedance Z c .
  • L ⁇ / 4 where ⁇ would be the wavelength of the only input signal applied to a pole, for example N3.
  • the N4 pole would be isolated.
  • a direct signal would be collected on the N1 pole and a coupled signal on the N2 pole.
  • a type of slow wave lines is chosen chosen from those described above, and on the other hand, as shown in FIG. 19, two input signals, one V1, are applied as before. on pole N1, the other V2 on pole N3 (via the single antenna A).
  • the pole N4 is the isolated pole, the pole N2 is the output pole for the signal V2 and the pole N3 is the output pole for the signal V1.
  • amplifiers ⁇ 1 and ⁇ 2 are added to the coupler to optimize the results.
  • Example XIII The technology used is the same as in Example XIII, and the results are identical except that which concerns the bandwidth which is less wide.
  • the branch coupler can be provided with several branches parallel to the branches 201 and 202.
  • the surface occupied by the device according to Example XIV is also slightly greater than that occupied by the device according to Example XIII, but this device is nevertheless perfectly integrable.
  • a generator 58 of signal V1 to the frequency F1 said local oscillator OL whose signal is applied to the amplifier ⁇ 1 possibly formed by two medium power amplifiers ⁇ '1, ⁇ ''1, then the signal is applied to the pole N1 of the coupler 50.
  • the pole N3 is applied to the antenna A
  • the pole N4 is connected to the ground via the impedance Z c , for example 50 ⁇
  • the pole N2 is connected to the input of the amplifier ⁇ 2 possibly formed by two low noise amplifiers ⁇ '2, ⁇ ''2.
  • the circuit according to the invention is both integratable and perfectly capable of working at such high frequencies. It therefore fully meets these conditions, however severe they may be.

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  • Semiconductor Integrated Circuits (AREA)
  • Programmable Controllers (AREA)
EP91201234A 1990-05-29 1991-05-24 Langsam-Wellen-Mikrostreifenübertragungsleitung und Anordnung mit einer solchen Leitung Expired - Lifetime EP0459571B1 (de)

Applications Claiming Priority (6)

Application Number Priority Date Filing Date Title
FR9006626 1990-05-29
FR9006626A FR2662858A1 (fr) 1990-05-29 1990-05-29 Ligne de transmission en mode a ondes lentes, du type microruban et circuit incluant une telle ligne.
FR9008598A FR2664448A1 (fr) 1990-07-06 1990-07-06 Dispositif emetteur-recepteur incluant un circuit integre comprenant un circulateur/duplexeur de frequences.
FR9008598 1990-07-06
FR9102813A FR2673766A1 (fr) 1991-03-08 1991-03-08 Lignes de transmission en mode a ondes lentes, du type microruban, et coupleur forme de telles lignes.
FR9102813 1991-03-08

Publications (2)

Publication Number Publication Date
EP0459571A1 true EP0459571A1 (de) 1991-12-04
EP0459571B1 EP0459571B1 (de) 1995-09-20

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US (1) US5369381A (de)
EP (1) EP0459571B1 (de)
JP (1) JPH05500896A (de)
DE (1) DE69113116T2 (de)
WO (1) WO1991019329A1 (de)

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EP0795921A3 (de) * 1996-03-12 1997-09-24 Siemens Aktiengesellschaft Funksprechgerät
CN102099957A (zh) * 2008-07-15 2011-06-15 松下电器产业株式会社 慢波传输线路

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US7398054B2 (en) 2003-08-29 2008-07-08 Zih Corp. Spatially selective UHF near field microstrip coupler device and RFID systems using device
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US9935680B2 (en) 2012-07-30 2018-04-03 Photonic Systems, Inc. Same-aperture any-frequency simultaneous transmit and receive communication system
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EP0637042A2 (de) * 1993-07-27 1995-02-01 Texas Instruments Incorporated Vorrichtung, die ein Eingangssignal beeinflusst
EP0637042A3 (de) * 1993-07-27 1998-05-27 Texas Instruments Incorporated Vorrichtung, die ein Eingangssignal beeinflusst
EP0795921A3 (de) * 1996-03-12 1997-09-24 Siemens Aktiengesellschaft Funksprechgerät
CN102099957A (zh) * 2008-07-15 2011-06-15 松下电器产业株式会社 慢波传输线路

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US5369381A (en) 1994-11-29
DE69113116D1 (de) 1995-10-26
DE69113116T2 (de) 1996-04-18
EP0459571B1 (de) 1995-09-20
WO1991019329A1 (fr) 1991-12-12
JPH05500896A (ja) 1993-02-18

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