DE4339191A1 - Energy supply control to an electric discharge machine - Google Patents

Abstract

The method for machining a workpiece by supplying electric energy in the form of current pulses into machining gap between an electrode and the workpiece in a dielectric medium is characterised by the following steps: (a) operation of switching devices to control a current setting signal corresponding to a pulse to be supplied to the machining gap; (b) superposition of a current component to compensate the drone current produced by the switching operation, with this component combined with the pulse being supplied to produce a resultant current; and (c) the resultant current is fed into the machining gap.The appts. includes two switching systems corresponding to the first and second (superimposed) currents delivered into the machining gap, as well as current detection, signal processing/issuing and other auxiliary devices.

Description

The present invention relates to a method and a Device for controlling the energy supply with one electrical discharge working machine (hereinafter referred to as EDM machine), which is a machining energy a working gap between an electrode and a Feed workpiece, which is provided in a dielectric is.

An EDM machine delivers a pulse with one constant current to a working gap to a workpiece melt and remove molten material from it, and around the workpiece due to the energy discharge to edit. Generally, the following are described four conventional Power supply circuitry used to To deliver constant current pulse.

A known circuit arrangement for a first energy supply device is shown in FIG. 54. This arrangement is described, for example, in Japanese Laid-Open Patent Publication No. SHO-62-27928, entitled "Pulse Generator Used in an Electric Eroding Machine Tool".

In Fig. 54, reference numeral 1 denotes an electrode, 2 a workpiece, 3 a control circuit for a switching device 4 , 4 a switching device, 5 a power supply for supplying a machining current, 6 a diode which causes a residual current to flow, 7 a current detection resistor, 8 a and 8 b leakage inductances of the wiring, 9 a comparator, 10 an envelope signal generator, and 18 a servo device for executing the servo control of the electrode 1 .

The operation of this circuit will now be described. Before a discharge is started, the switching device 4 is in the conductive state and a machining voltage is applied to the working gap between the electrode 1 and the workpiece 2 by the power supply 5 . After the start of the discharge, a pulse command 16 , which corresponds to a machining current waveform to be applied to the working gap, is issued to the envelope signal generator 10 by a controller (not shown in FIG. 54). The pulse command 16 is emitted by the envelope signal generator 10 as envelope signals 13 , 14 . Fig. 55 shows the shapes of the envelope signals 13, 14. In the comparator 9 , the current flowing in the working gap is detected by the current detection resistor 7 in order to obtain a current machining current value 15 , and thereby to compare the envelope signals 13 , 14 with the current machining current value 15 , and to output a control signal 12 to the control circuit 3 . The control circuit 3 turns the switching element 4 on / off under the control of the control signal 12 to regulate the machining current so that it is within a predetermined range of values. If the current processing current value 15 exceeds the envelope signal 13 , the switching device 4 is switched off. In contrast to this, the switching device 4 is switched on when the current machining current value 15 drops below the envelope signal 14 . In the method described above, the machining current is controlled or regulated.

In this method, the rate of rise of the waveform of the machining current through the current detection resistor 7 and the size of the inductors 8 a, 8 b of a machining power supply apparatus defined so that therefore, the resistance and the inductances are used as loads for performing the shift control.

A second conventional circuit arrangement for a power supply device is shown in Fig. 58, which is described, for example, in Japanese Laid-Open Patent Application SHO-157-33949 as "pulse generation circuit controlled for generation by intermittent electric discharge". This power supply device has been improved in the rising and falling speeds of the machining current compared to the first power supply device so as to ensure accelerated operation. In Fig. 58 an auxiliary power supply 28, a first switching device 4, a current detector 24, a reactor 22 and a diode 23 constitute a first auxiliary circuit. A power supply 5 , the auxiliary power supply 28 , the first switching device 4 , the current detector 24 , the reactor 22 , an electrode 1 , a workpiece 2 and a second switching device 20 form a main circuit.

The operation of this circuit will now be described. In the first auxiliary circuit, the switching device 4 is operated by a control circuit 27 under control of the detector signal of the current detector 24 . The control circuit 27 carries out the switching control or regulation of the switching device 4 in such a way that the current flowing in the current detector 24 is kept constant. In this case, the reactor 22 inserted into the circuit keeps the current flowing in the first auxiliary circuit constant.

This second energy supply device is provided with a second switching device 20 , which is used exclusively for switching the discharge pulse on / off. When the discharge pulse is turned off, a current flows within a predetermined range in a stable state in the first auxiliary circuit, and as soon as the discharge is initiated, the machining current is supplied from the first auxiliary circuit. This enables the current to rise extremely quickly. The current during the discharge flows in the main circuit, which consists of the power supply 5 , the auxiliary power supply 28 , the first switching device 4 , the current detector 24 , the reactor 22 , the electrode 1 , the workpiece 2 and the second switching device 20 . When the discharge has stopped, the current which has flowed in the reactor 22 of the main circuit flows to the second diode 23 in the first auxiliary circuit so as to quickly interrupt the current of the working gap.

A first diode 25 is provided for the purpose of increasing the efficiency of the power supply by forming a second auxiliary circuit and causing the current flowing in the reactor 22 to return to the power supply 5 when both the first switching device 4 and the second switching device 20 can be switched off. The second auxiliary circuit is formed by the first diode 25 , the current detector 24 , the reactor 22 , the second diode 23 and the main power supply 5 . Fig. 59 shows a machining current waveform generated by the second power supply device.

Furthermore, there is a third conventional circuit arrangement for a power supply device shown in Fig. 60 and described, for example, in Japanese Patent Laid-Open No. HEI-2-34732 entitled "Control Method for EDM Machining Power Supply" . In Fig. 60, reference numerals 30 a to 30 e denote driver devices which cause switching devices 32 a to 32 e to conduct, and form a logic circuit 35 . Reference numerals 33 a to 33 e denote limiting resistors which control a machining current and which each have different values. A detector 36 for detecting the start of a discharge is arranged between an electrode 1 and a workpiece 2 . This detector 36 transmits a discharge detection signal 37 to the logic circuit 35 . The logic circuit 35 selects the switching devices 32 a to 32 e, which are to be driven under the control of the output signal of an oscillator 34 and the discharge detection signal 37 .

The operation of this circuit will now be described. In the circuit, a power supply 5 is provided to supply a current, and a parallel connection of circuits, each comprising series connections of the switching devices 32 a to 32 e and the current limiting resistors 33 a to 33 e, is connected in series with the power supply 5 . The resistance values of the current limiting resistors 33 a to 33 e, which differ from one another, are selected so that they are a power of 2, that is to say a factor of 1, 2, 4, and so on. If a square wave signal with a constant current value and a duration t _{P is} to be supplied, as shown in FIG. 61, some of the switching devices 32 are switched on by their associated driver circuit 30 to cause a current to flow through the associated current limiting resistors 33 . When the discharge is started, a machining current is applied to the working gap via selected resistors 33 . A differential voltage between the output voltage of the main power supply 5 and the discharge voltage generated at the working gap between the electrode 1 and the workpiece 2 is applied to each current limiting resistor so as to determine the current flowing in the current limiting resistor. Since the discharge voltage is generally constant, the machining current is determined solely by the selection of the current limiting resistors.

Furthermore, as shown in Fig. 62, the slew rate of a current waveform can be controlled. By switching the switching devices 32 on / off continuously after the discharge current has risen to a point indicated by reference numeral 48 in Fig. 62, the current can be increased further, but only in such a way that its slope is further reduced . Such deliberate control of the discharge current waveform is often performed to enable finer control or regulation of the machining process.

Furthermore, there is a circuit arrangement for a fourth conventional power supply device, which is shown in FIG. 63, which is described, for example, in US Pat. No. 4,306,135. In this figure, reference numeral 49 indicates a fixed current-limiting resistor, 50 a semiconductor amplifier, for example a FET, 5 a switching device for turning on / off of the semiconductor amplifier 50 to produce a discharge pulse on / off, 52 a digital signal which determines the current waveform of the discharge pulse, 53 a digital / analog converter which converts the digital signal into an analog signal, 54 an amplifier for driving the amplifier 50 , and 55 a limiting resistor for the amplifier 54 .

The operation of this circuit is described below. For the on / off clock of the discharge pulse, a signal is output by the oscillator 21 in order to drive the switching device 51 . The current supplied to the working gap between the electrode 1 and the workpiece 2 after the discharge has occurred is determined by the resistance values of the fixed resistor 49 and the semiconductor amplifier 50 . For example, when an FET is used as the semiconductor amplifier 50 , it can be operated as a variable resistor.

The characteristic of the FET is as shown in Fig. 64. If VGS is optionally set, ID is kept constant if VGS changes slightly. This is because the FET ensures that the machining current is regulated so that it is kept constant, regardless of a slight change in the power supply voltage 5 . For this reason, the current is stable during discharge, and so-called pulse interruption is unlikely to occur, in which discharge stops after half the pulse, thereby providing extremely stable processing.

Furthermore, changing a signal for the gate G of the FET 50 within a single pulse provides an optional signal form and ensures a constant current characteristic for a command value G, which ensures particularly stable processing.

The conventional EDM machining energy supply with the structure described above has the following specified disadvantages.

Namely, since the "pulse generator used in the EDM machine tool" described in Japanese Patent Laid-Open Publication No. SHO-62-27928 tries to keep the machining current value in the switching control substantially within the specified range the machining current waveform 47 , as shown in Fig. 55, hum. This hum is generally several amps wide. Examples of machining current pulses generated under different conditions are shown in FIGS. 56 and 57. Fig. 56 shows a high machining current setting value, that is, a current setting for a so-called rough or pre-machining. With a current waveform 47 of this example b is the ripple-width (which is approximately equal to the distance between command values 13 and 14) is small relative to a peak value 13 of the machining current command value, and therefore, causes no particular machining error. However, if the command value of the current peak value is decreased, as shown in Fig. 57, the lower limit value 14 of the command value is no longer significant, and the originally rectangular desired waveform becomes triangular, as indicated by reference number 47 c. As a result, the pulse cannot be sustained for a desired period of time and becomes intermittent. This signal form is not able to provide a desired processing result.

Due to the hum in the current waveform, the Current waveform that are regulated by the switching regulation not for regulating a microcurrent waveform suitable, for example in the finishing or fine machining, and cannot achieve the desired processing.

Furthermore, the "pulse generating circuit which is controlled to generate intermittent electric discharges" as described in Japanese Laid-Open Utility Model Publication No. SHO-57-33949 is designed to some extent eliminate the difficulties in the procedure described in the Japanese Publication of Laid-Open Patent No. SHO-62-27928. That is , the reactor (choke coil) 22 inserted into the circuit of FIG. 58 more simply maintains the current constant, and the ripple width of the current waveform may be considerably less than that of the circuit of FIG. 54. In general, the insertion leaves of a reactor keep the current constant more easily, but has the disadvantage that no (sufficient) rise and fall speeds can be provided. However, in the circuit of Fig. 58, the first auxiliary circuit is used to ensure the peak current value beforehand, and after the start of the discharge pulse, the second, separate switching device 20 is used to cause the discharge current to conduct or not conduct, thereby the rates of rise and fall are improved. However, the auxiliary power supply 28 is required for this purpose.

This auxiliary power supply 28 is probably provided with a considerably larger output capacity than the power supply 5 , which serves as the main power supply, since it can have a lower output voltage, but its output current must be substantially equal to the working current. Another disadvantage of this procedure is the difficulty in making the circuit available at low cost.

Furthermore, the method described in Japanese Patent Application Laid-Open No. SHO-57-33949 has a disadvantage in that, unlike the method disclosed in Japanese Patent Application Laid-Open No. No. SHO-62-27928, it is difficult to provide an optional discharge pulse waveform at the same time as controlling its rise and fall speeds so that only the square wave signal shown in Fig. 59 can be provided.

Furthermore, the procedure described in "Tax Procedure for an EDM machining power supply "according to the Japanese publication of a published patent with No. HEI-2-34732 is described so that it is the Overcomes difficulties with the procedures exist in the Japanese publication of a published patent no. SHO-62-27928 and in the Japanese publication of a disclosed Utility model with the number SHO-57-33949 are described.

The procedure described in Japanese Patent Laid-Open No. HEI-2-34732 uses the constant voltage power supply and the resistor inserted therein to regulate the machining current value without regulating or controlling the discharge machining current by switching regulation. Therefore, the provided discharge current waveform virtually no current Brum on, as shown in Fig. 61 shown, and the on / off of the resistors 33 a to 33 d in the circuit at high speed allow it, the direction indicated by reference numeral 48 rising speed and selectively adjust the current waveform as shown in FIG. 62.

However, the disadvantage of this procedure is that the current is not controlled or regulated directly, but is regulated as a function of the resistance value which limits the current, as a result of which the discharge current value changes in accordance with the output voltage of the energy supply 5 . In other words, if a predetermined current value has been set, the same processing state cannot be made available if the voltage of the power supply changes.

Furthermore, it is known that the discharge gap between the electrode 1 and the workpiece 2 physically acts as a constant voltage load of approximately 25 V. For this reason, the differential voltage between the output voltage of the power supply 5 and the voltage drop of 25 V of the discharge gap is mainly applied to the current limiting resistors 33 and is consumed as thermal energy. Namely, as a power supply for the EDM machine, this approach cannot avoid a reduction in the power supply efficiency compared to the procedures disclosed in Japanese Patent Laid-Open No. SHO-62-27928 and Japanese Patent Application Laid-Open No. SHO-57 -33949 are described. This prevents downsizing of the power supply device and also makes it difficult to provide the same functions inexpensively. As explained above, the device in the information document 3 has the disadvantages that the machining current is not simply kept constant, that the energy supply efficiency is poor, and that this poor energy supply efficiency leads to considerable dimensions and high costs of the device.

In addition, according to U.S. Patent No. 4,306,135 some disadvantages of the procedure that are in the Japanese publication of a published patent with No. HEI-2-34732.

Namely, the method described in U.S. Patent No. 4,306,135 uses a semiconductor amplifier instead of the plurality of current limiting resistors used in the procedure described in Japanese Patent Laid-Open No. HEI-2-34732 are. In this prior art, since the FET is used as the semiconductor amplifier , the constant current characteristic shown in Fig. 64 is provided. Therefore, a constant current with respect to the change in the output voltage of the power supply 5 can be maintained and regulated, and moreover, the constant current regulation or control can be exercised during the persistence of the discharge pulse, which enables extremely stable processing. In the sense that the pulse current can be kept constant, more stable processing can be achieved compared to the switching power supplies in the procedures disclosed in Japanese Patent Laid-Open No. SHO-62-27928 and Japanese Patent Laid-Open Utility model with the number SHO-57-33949 are described. In addition, since the resistance value at the time of the start of discharge is extremely small compared to the Japanese publication of a laid-open patent No. HEI-2-34732, the discharge current can be increased more quickly.

However, the differential voltage between the output voltage of the power supply 5 and the machining gap voltage is applied to the semiconductor amplifier 50 as a whole. Therefore, the thermal energy to be consumed by the semiconductor amplifier 50 is large. Compared to conventional electrical parts, the semiconductor is easily influenced, in particular by heat, and is important in terms of the design of the heat dissipation. However, the method used in U.S. Patent No. 4,306,135 generates significant heat because it not only uses the semiconductor as the switching device, but also uses it as a variable resistor in an active area, so this method does not cause the flow of a high current, and it is very difficult to design a circuit which can achieve a rough machining which requires a current peak of several tens of amperes or more.

The method described in U.S. Patent No. 4,306,135 has a disadvantage in that a high Electricity sufficient for rough or preprocessing, cannot be controlled or regulated.

The present invention is therefore based on the object to overcome these difficulties in that a Power supply to an EDM machine (one with electrical discharge working machine) which is a little hum at one Machining current pulse, allows a Microcurrent is easily generated during finishing, and it also allows a Power supply device has small dimensions and low cost due to its extremely high Energy supply efficiency.

The procedure for controlling the energy supply for one EDM machine, according to the present invention, switches the switchgear in an optional cycle under the Control of the current command value signal on / off, accordingly the waveform of the current pulse which is sent to the working gap should be created, whereby an optional form of a Current pulse is applied to the working gap, and the Current component to compensate for the hum component, the generated by switching at the time the current is supplied is superimposed on the optional form of the current and the Working gap fed.  

The second processing circuit according to the invention superimposes the current, which is the difference between the Current command value signal and the current from the first Machining circuit corresponds, so-called Current component to compensate for the hum component, the generated by switching the first processing circuit the current from the first processing circuit, and delivers the resulting current to the working gap.

The first DC power source according to the invention supplies the Working gap with a current based on the signal, which by subtracting the predetermined value from the Current command value signal is obtained, and the second DC source superimposes the current according to the Difference between the current command value signal and the current from the first direct current source the current from the first DC power source, and delivers the resulting current the working gap.

The first DC power source according to the invention supplies the Working gap with the current based on the Current command value signal, and the second current source superimposes the current according to the positive difference between the current value from the first DC source and the current command value signal the current from the first DC source, and sends the resulting current the working gap, and beyond that, the third one Current source the current according to the negative difference between the current value from the first DC source and the current command value signal of the circuit from the first DC source, and sends the resulting current the working gap.

The energy supply control or regulatory process according to the Invention provides the output current level and Output current hum the constant current supply section  sets the addition result of the set Output current level and output current hum than that Constant current supply output current command signal fixed, compares the output current command signal with the output current of the constant current supply section and controls the regulator of the Constant current supply section according to the result this comparison.

The power supply device according to the invention comprises the ripple current setting device, which a Has modulation device to the set Signal frequency of the ripple current setting value according to the set value of the output current level adjuster to modulate to the set signal frequency of the Ripple current setting value decrease when the set Output current level adjuster level is high, and around the set signal frequency of the Ripple current setting value increase when the set Output current level adjuster level is low.

The power supply device according to the invention provides the output current level and the output current hum of the Constant current supply section, that sets Addition result of the set output current level and of the output current hum as the output current command signal of the constant current supply section, compares that Output current command signal with the output current of the Constant current supply section, and controls that Switching device of the constant current supply section according to the result of this comparison. The Gate device turns off noise from the Switching the first switching device on / off in the Constant current supply section originates.  

The power supply device according to the invention provides the output current levels of the constant current supply sections one, the first output current hum and the second Output current hum that is 180 ° from the first Output current hum is out of phase, that puts Result of addition of the output current level and the first Output current hum as the first output current command signal of the first constant current supply section determines that Addition result of the set output current level and of the second output current hum as the second Output current command signal of the second Constant current supply section, compares the first Output current command signal with the output current of the first Constant current supply section and controls the switching device of the first constant current supply section according to the result of this comparison, and compares the second output current command signal with the output current of the second constant current supply section and controls or controls the switching device of the second Constant current supply section according to the result of this comparison.

The power supply device according to the invention provides the output current level and the output current hum of the Constant current supply section loads that Result of addition of the output current level and the Output current hum (ripple) than that Constant current supply output current command signal section, compares the output current command signal with the output current of the constant current supply section and outputs the signal that the first switching device of the Constant power supply section turns off accordingly the result of this comparison, and still gives the Clock device the signal that the first Switching device of the constant current supply section turns on when a predetermined period of time has elapsed,  after the comparison device has output the signal, which is the first switching device of the Constant power supply section turns off.

The power supply control or regulation process according to the Invention causes the current to decrease the Output current suppressed when switched off of the first switching device occurs when the current to Addition is fed to keep the current so that it does not is abruptly reduced, which reduces hum become.

The series connection of the second DC power supply, where the voltage is equal to the electrical Discharge voltage (eroding voltage) is or slight less, the third switching device and the diode in the Initiate power supply device according to the invention the current which is the reduction of the output current suppresses the off state of the first Switchgear occurs when the current is fed to the addition becomes to keep the current at such a value that it is not abruptly reduced, causing hum be reduced.

In the power supply device according to the invention there is an increase or decrease in the output current between the current command value and the overcurrent command value, if a current is too high due to a short circuit or the like flows over the electrode and the workpiece, thereby preventing the flow of a short circuit current.

The power supply control method according to the invention holds the current at such a level that a sudden increase is avoided if the output current is at a predetermined Current level is controlled or regulated, whereby Hum can be avoided.  

The series connection of the third DC power supply, where the tension is the working gap with a tension can supply the higher than the electrical Discharge voltage is, and lower than the voltage that is supplied by the first DC power supply the fourth switching device and the diode in the Power supply device according to the invention keep the Current at such a value that an abrupt increase is avoided if the output current is on a predetermined current level is controlled or regulated, whereby Hum can be avoided.

The power supply device according to the invention uses the first switching device and the fourth switching device to control or regulate the output current, and, after the output current has reached the specified value, the fourth switchgear uses the output current control or regulate, whereby a current control or control is carried out with low hum.

The series connection of the fourth DC power supply, where the tension on the slope of the leading edge of the Current in the power supply device according to the Invention is adapted, reduces hum when the Output current is increased, so the desired leading edge to provide the electricity.

The power supply device according to the invention works in such a way that it switches off the hum of the current, and between the time when the electrical discharge occurs until the time the current reaches the Command value reached.

The series connection of the fifth DC power supply, where the tension is the working gap with a tension can supply, which is higher than the voltage, which of the  first DC power supply is supplied, and des sixth switching device in the power supply device according to the invention works so that the reliable electrical discharge is generated.

The power supply device according to the invention switches the sixth switching device under the control of one High voltage pulse signal when the electrical Discharge was delayed, so that reliable production electrical discharge is ensured.

The power supply device according to one of the various embodiments of the invention Low voltage power supplies as theirs Use DC power supplies, such as the first DC power supply, the fourth DC power supply, and so on, and enables such an effective use of energy supplies.

The invention is illustrated below with reference to drawings illustrated embodiments explained in more detail what other advantages and features emerge. It shows:

Fig. 1 is a diagram for explaining a first embodiment of the present invention;

Figure 2 (a) to (d) current waveform charts in the first embodiment.

Fig. 3 is a current waveform diagram in the first embodiment;

Figure 4 (a) to (e) operational timing charts for explaining the operation of the circuit in the first embodiment.

Fig. 5 is a current waveform diagram in the first embodiment;

Fig. 6 is a circuit diagram for explaining a second embodiment of the present invention;

Fig. 7 is a circuit diagram for explaining a third embodiment of the present invention;

Fig. 8 is a rectangular current waveform diagram in the third embodiment;

Figure 9 is a triangular current waveform diagram in the third embodiment.

FIG. 10 is a circuit diagram for explaining a fourth embodiment of the present invention;

Figure 11 is a square wave current waveform diagram in the fourth embodiment.

12 is a triangular current waveform diagram in the fourth embodiment.

Fig. 13 is a triangular signal current waveform diagram in the fourth embodiment;

FIG. 14 is a diagram for explaining a fifth embodiment of the present invention;

FIG. 15 is a diagram for explaining a sixth embodiment of the present invention;

FIG. 16 is a diagram for explaining a seventh embodiment of the present invention;

FIG. 17 is a current waveform diagram in the fifth embodiment;

FIG. 18 is a current waveform diagram in the sixth embodiment;

FIG. 19 is a current waveform diagram in the seventh embodiment;

FIG. 20 is a diagram for explaining an eighth embodiment of the present invention;

Fig. 21 (a) to (e) are waveform diagrams and timing charts used for describing the operation of the eighth embodiment;

Fig. 22 (a) to (c) are waveform diagrams used for describing the modification of the eighth embodiment;

Figure 23 is a diagram for explaining a ninth embodiment of the present invention.

Fig. 24 is a graph of the Vf characteristic for describing the operation in the ninth embodiment;

Figure 25 (a) to (d) are waveform diagrams and timing charts used for explaining the operation in the ninth embodiment.

Figure 26 is a diagram for explaining a tenth embodiment of the present invention.

Fig. 27 (a) to (g) are waveform diagrams and timing diagrams of the tenth embodiment are used for describing the operation;

FIG. 28 is a diagram for explaining an eleventh embodiment of the present invention;

Figure 29 (a) to (d) are waveform diagrams used for describing the operation in the eleventh embodiment.

Figure 30 (a) to (c) are waveform diagrams used for describing the operation in the eleventh embodiment.

FIG. 31 is a diagram for explaining a twelfth embodiment of the present invention;

Figure 32 (a) to (c) is a waveform diagram and a timing chart used to describe the operation in the twelfth embodiment.

Fig. 33 is a main circuit diagram for explaining a thirteenth embodiment of the present invention;

Figure 34 is a diagram for explaining a control or regulating circuit in the thirteenth embodiment of the present invention.

FIG. 35 is a diagram for explaining an example of a clock circuit in the thirteenth embodiment of the present invention;

Fig. 36 is a circuit diagram for explaining an alternative example of the clock circuit in the thirteenth embodiment of the present invention;

FIG. 37 is timing charts used for describing the operation of the timer circuit in the thirteenth embodiment of the present invention;

Fig. 38 waveform diagrams and timing diagrams which are used to describe a main operation in the thirteenth embodiment of the present invention;

Fig. 39 waveform diagrams and timing charts used for explaining the operation at the time of a short circuit of a working gap in the thirteenth embodiment of the present invention;

Fig. 40 is a circuit diagram for explaining an alternative example of the control or regulating circuit in the thirteenth embodiment of the present invention;

Fig. 41 are waveform diagrams used to describe the actual operation in the thirteenth embodiment of the present invention;

Fig. 42 are waveform diagrams used to describe the actual operation in the thirteenth embodiment of the present invention;

43 is a main circuit diagram for explaining a fourteenth embodiment of the present invention;

44 is a diagram for explaining a control or regulating circuit in the fourteenth embodiment of the present invention.

Fig. 45 waveform diagrams and timing diagrams which are used to describe a main operation that occurs in the fourteenth embodiment of the present invention;

46 is a main circuit diagram for explaining a fifteenth embodiment of the present invention;

47 is a diagram for explaining a control or regulating circuit in the fifteenth embodiment of the present invention.

Fig. 48 waveform diagrams and timing diagrams which are used to describe a main operation in the fifteenth embodiment of the present invention;

49 is a main circuit diagram for explaining a sixteenth embodiment of the present invention;

Fig. 50 is a diagram for explaining a control or regulating circuit in the sixteenth embodiment of the present invention;

FIG. 51 is waveform diagrams and timing diagrams which are used to describe a main operation according to the sixteenth embodiment of the present invention;

Fig. 52 is a main circuit diagram for explaining a seventeenth embodiment of the present invention;

FIG. 53 is a main circuit diagram for explaining an eighteenth embodiment of the present invention;

54 is a diagram for explaining a first device according to the prior art.

Fig. 55 is a current waveform diagram generated in the circuit of Fig. 54;

Fig. 56 is a current waveform diagram used to describe the disadvantages of the first prior art device;

Fig. 57 is a current waveform diagram used to describe the drawbacks in the first prior art device;

FIG. 58 is a circuit diagram for explaining a second device according to the prior art;

Fig. 59 is a current waveform diagram generated by the circuit in Fig. 58;

Figure 60 is a diagram for explaining a third device according to the prior art.

Fig. 61 is a current waveform diagram generated by the circuit in Fig. 60;

Fig. 62 is a current waveform diagram generated by the circuit in Fig. 60;

FIG. 63 is a circuit diagram for explaining a fourth device according to the prior art; and

FIG. 64 is a current characteristic diagram of a semiconductor amplifier used in the circuit of FIG. 63.

A first embodiment of the present invention will be described below based on FIGS. 1 to 5. Fig. 1 is a circuit diagram of a power supply for an electric discharge machine (discharge machine) according to the first embodiment of the present invention. In Fig. 1, reference numeral 1 denotes an electrode, 2 a workpiece 4, a first switching device, 5 a power supply, 6 is a diode, which causes a leakage current to flow, 7 a current detector, 100 a to 100 c second switching device, 101 a to 101 c current limiting resistors whose resistance values are in the ratio of powers of 2, i.e. 1: 2: 4, 102 a current command value signal setting unit which is used to peak, duration, stop time, shape and the like of a current waveform to a current command value signal generator 103 , 103 a current command value signal generator that generates a current command value signal set by the current command value signal setting unit 102 , and 104 denotes the output thereof.

105 denotes a signal converter which performs addition / subtraction of a predetermined value to / from the output signal 104 or multiplies the output signal by a predetermined rate, 106 denotes its output signal, 107 denotes a first signal adder / subtractor which detects a difference between the output signal 106 and the current value of a detector signal 112 , which is determined by the current detector 7 , 108 denotes its output signal, 109 denotes a logic circuit which outputs a logic signal for switching on / off the first switching device 4 under the control of the output signal 108 , 110 denotes the latter Output signal, and 111 denotes an amplifier for driving the first switching device 4 under the control of the output signal 110 .

112 denotes a detector signal which indicates the instantaneous value of a current provided by the current detector 7 , 113 denotes a second signal adder / subtractor which processes a difference between a command value from the current command value signal generator 103 and the instantaneous value of the detector signal 112 from the current detector 7 , 114 denotes an analog / digital converter which converts the output signal of the second signal adder / subtractor 113 into a digital signal, 115 a to 115 c denotes amplifiers which amplify each bit of the digital signal of the analog / digital converter 114 to the switching devices 100 a to drive 100 c, 116 denotes a working gap voltage signal for detecting the start of a discharge, and 117 denotes a resistor.

In the present embodiment, the power supply 5 , the first switching device 4 , the current detector 7 , the resistor 117 and the working gap are connected in series so that they form a first processing circuit. Furthermore, form a plurality of series circuits of the current-limiting resistors 101 a to 101 c and the second switching devices 100a to 100 c, a second processing circuit which is connected parallel to first processing circuit, to supply the working gap with a current which is superimposed on the current of the first processing circuit . In addition, the logic circuit 109 and the amplifier 110 form a first control or regulating circuit, and the analog / digital converter 114 and the amplifier 115 a to 115 c form a second control or regulating circuit.

The operation of this circuit will be described below with reference to Figs. 2 (a) to (d) to Fig. 5. First, before starting machining, a user working with the EDM machine sets the shape, duration, stop time, and the like of a current waveform from the setting unit 102 for the current command value signal generator 103 . These values can be specified by a program from an NC control device or the like. The current command value signal generator 103 generates signals that are necessary for the power supply device in actual processing, for example the clock of the application of a voltage to the working gap, under the conditions of the setting unit 102 . In other words, the current command value signal generator 103 generates a current command value signal that specifies the operation so that, for example, the machining current waveform is output when a discharge is started after the voltage is applied to the working gap and the voltage is reapplied when a predetermined one The time period of a stop time has passed after the machining current waveform has stopped.

The signal 104 , which is output from the current command value signal generator 103 , is converted into the output signal 106 by the signal converter 105 . This output signal 106 is converted so that its level corresponds to that of the output signal 104 or has a slightly lower signal level. A difference between this output signal 106 and the signal 112 , which is detected by the current detector 107 (the current flowing in the first processing circuit) is then treated by the first signal adder 107 and serves as the output signal 108 .

This output signal 108 is further sent to the logic circuit 109 . This logic circuit 109 is designed such that it outputs the operating clock of the first switching device 4 under the control of the output signal 108 , that is to say as the output signal 110 , and the logic circuit 109 supplies an output signal "1" for switching on the first switching device 4 when that of the Current detector 7 detected current value is smaller than the command value given by signal 106 and provides an output signal "0" for switching off the first switching device 4 when the current value detected by current detector 7 is greater than the command value given by signal 106 . The amplifier 111 drives the first switching device 4 under the control of the output signal 110 .

Fig. 2 (a) illustrates the foregoing description of the format of an operating current waveform. A form enclosed by 104 a to 104 c represents the command value of the current waveform corresponding to the output signal 104 in FIG. 1. If it is now assumed that the signal converter 105 does not perform any signal conversion, the output signal 106 matches the output signal 104 . If the first switching device 4 is conductive, a discharge is started, the discharge current begins to flow, and the current supplied by the first switching device 4 begins to increase, as indicated by 121 in FIG. 2 (a).

A predetermined upper threshold 118 a exists in the logic circuit 109 , the output of which switches from "1" to "0" in order to switch off the first switching device 4 (cf. clock 119 in FIG. 2 (a)) when the differential signal 108 is on "0" is switched, and its output switches from "0" to "1" in order to switch on the first switching device 4 (this is the clock 120 in FIG. 2 (a)) when the differential signal 108 reaches a lower threshold value 118 b . The current, which is caused to flow by the first switching device 4 , therefore goes up and down between 118 a and 118 b in FIG. 2 (a). A difference between 118 a and 118 b defines a threshold range. When the duration of the pulse ends, the current drops towards zero, as indicated by 122 . As described above, the current waveform provided by the first switching device 4 has a ripple as indicated by 121 , 119 , 120 , 122 and so on in Fig. 2 (a). This is a current hum.

In order to compensate for this current hum component so that the signal almost again corresponds to the original output signal 104 , that is, the current waveform command value, the present embodiment is provided with the second processing circuit in addition to the first processing circuit which includes the first switching device 4 and so on Series connections of the second switching devices 100 a to 100 c and the resistors 101 a to 101 c exist in parallel to the first processing circuit.

A difference between the output signal 104 , which serves as the current command value, and the detector signal 112 , which serves as the current current value, as a result of the current hum or a rise delay of the current from the first switching device 4 , is determined by the second adder / subtractor 113 and on passed the analog / digital converter 114 . Here, the difference is converted into a digital signal by the analog / digital converter 114 . In the present embodiment, the resistors 101 a to 101 c are designed so that they are in a ratio of 1: 2: 4.

Therefore, the current values flowing in the resistors 101 a to 101 c behave as 4: 2: 1. Since the resistance values of the resistors 101 a to 101 c are chosen so that they represent powers of 2, as explained above, eight different current values can be used be adjusted 114 in accordance with the combinations of the three outputs of the analog / digital converter, that is, as the logic signals which are sent to the second switching device 100 a to 100 c. By changing this combination, the current, which corresponds to the difference between the current command value and the current current value, is supplied by the resistors 101 a to 101 c.

FIGS. 4 (a) to (e) show the operation clocks of the preceding embodiment, wherein Fig. 4 (a) shows the voltage waveform of the machining gap, Fig. 4 (b) shows the current waveform of the machining gap, and a hatched region 129 by the Current is compensated, which is supplied by the second processing circuit. Fig. 4 (c) shows the operating cycle of the first switching device 4 , and Fig. 4 (d) shows the operating cycles of the second switching devices 100 a to 100 c. Fig. 4 (e) shows the on / off table of the second switching devices 100 a to 100 c, which are switched on / off, under the control of the signals which are output by the logic circuits 115 a to 115 c, with "1" indicating that the switching device is switched on and "0" indicating that the switching device is switched off.

First, in a state in which the voltage is applied to the working gap before the start of the discharge, the first switching device 4 and the second switching devices 100 a to 100 c are all switched on.

When the start of discharge is determined at 124 in Fig. 4 (a), the current waveform begins to rise as shown in Fig. 4 (b). At the same time, the current decreases due to the resistors 101 a to 101 c. Therefore, the second switching devices 100 a to 100 c are switched on / off (in a suitable manner so that the current component is compensated for in accordance with the difference between the current command value and the current current value as a result of the delay in the rise in the current of the first switching device 4 ), for example in Fig. 4 (e) shown to reduce the current supplied by the resistors 101 a to 101 c. When the current value has almost reached the command value, the current from the first switching device 4 exceeds the upper threshold value 118 a, whereby the first switching device 4 is switched off (clock 131 in FIG. 4 (c)). Then the current from the first switching device 4 begins to decrease in the direction of the lower threshold value 18 b.

The hum generated here is determined by the second adder / subtractor 113 in Fig. 1, and the second switching devices 110 a to 100 c are selectively switched on by the analog / digital converter 114 , which causes the resistors 101 a to 101 c to the current to deliver, which compensates the current hum. Although Fig. 4 (d) shows an example that only the switching device 100 is a turned on / off, in order to compensate the ripple, and the other resistors are of course selected when the ripple is considerably larger. If a second switching device and a resistor, that is to say a resistor, is a ratio of the resistance value corresponding to 0.5 with respect to 101 a to 101 c, in order to carry out a smaller current compensation, then the current compensation is carried out by a plurality of switching devices and the hum continues decreased.

Current waveforms for small current setting values are shown in Figs. 2 (b) to 2 (d). If the current setting value is small in FIG. 2 (b), the operation is identical to that for a large current value as in FIG. 2 (a). Fig. 2 (c) shows that the current setting value is even smaller and is below a switching threshold range. The switching threshold values are designated by 132 a and 132 b. When the lower through 132 b nominated threshold was set so that it has been gradually reduced from the current peak value by a predetermined value, it may be lower than "0", as indicated by the dashed line in Fig. 2 (c). In this case, the lower threshold, which is originally 132 b, can be changed to 132 c for controlling the regulation. Because the threshold range is small at this time; the switching frequency of the first switching device 4 is higher compared to the switching frequency in FIGS. 2 (a) and (b).

As the switching frequency increases, switching losses in the case of a semiconductor amplifier increase. Therefore, more reliable circuit operation can be performed by making the adjustment as shown in Fig. 2 (d) instead of arbitrarily decreasing the threshold range to increase the switching frequency. Therefore, at the small current setting below the switching threshold range, the first switching device 4 is kept switched off, and the current is only supplied by the second processing circuit, which is formed by the series circuits of the second switching devices 100 a to 100 c and the resistors 101 a to 101 c. At this time, since the current setting value is small, the reduction in power efficiency resulting from the fact that the power supply switching control is not used is not a big problem.

There is another difficulty in setting the switching threshold. Fig. 3 shows that an upper switching threshold 333 b is less than a set current peak value 134 a. The present invention has been described based on the assumption that both are set to the same value. Namely, the coefficient of the signal converter 105 in FIG. 1 is "1" and no conversion is performed.

If the upper threshold for switching matches the peak value of the current command value, there will be a time delay until the current begins to decrease since the switching is actually performed after the current exceeds the upper threshold. In short, a switching current flows above the upper threshold. While the reduction in the switching current below the current command value at 134 b is compensated for by the second processing circuit, which is formed by the series connections of the second switching devices 100 a to 100 c and the resistors 101 a to 101 c, the excess switching current cannot be compensated. Therefore, there is a method in which the upper threshold is selected to be lower than the peak current of the current command value, taking into account the overshoot of the current value due to the switching delay. In Fig. 3, the current value is at 134 a is higher than the upper threshold 133 b. However, if the upper threshold 133 b is set slightly smaller than the peak value 104 b of the current command, the value of the total current does not exceed the current command value.

While a square wave signal was used as the current command in the above description, the current waveform used in electrical discharge machining is not limited to a square wave signal b 99999 00070 552 001000280000000200012000285919988800040 0002004339191 00004 99880. It is known that the use of a selectable slew rate type triangular current waveform as shown in Fig. 5 very well suppresses electrode wear. The present embodiment allows easy generation of such a current waveform with such a selectable shape. Of course, since the power supply efficiency is high, it is easy to reduce the size and cost of the power supply, and in addition, an exact current waveform can be provided as the desired waveform. In Fig. 5, the switching threshold values 135 a and 135 b were set so that they match a desired waveform 135 a. Namely, the upper threshold value 135 a is adapted to the signal shape, and the lower threshold value 135 b that is set lies at a value which is smaller than the upper threshold value 135 a by a predetermined value. In this way, a waveform that is different from the square wave signal can be easily provided.

As described above, the Power supply device for the electrical EDM machine according to the first embodiment switched energy supply with high Power supply efficiency to easily get the one you want Reduction in size and cost of To achieve energy supply. The addition of the Resistor circuit for compensating the current hum, the occurs when the power supply is switched, switches the Current hum and enables stable processing, even with a small current.

A second embodiment of the present invention will be described below based on FIG. 6. In this embodiment, the second processing circuit, which is formed by the plurality of series circuits of the second switching devices 100 a to 100 c and the resistors 101 a to 101 c and is used for current compensation in the first embodiment, is replaced by a second processing circuit which consists of a A FET 136 and a resistor 137 are connected in series. As described in connection with FIGS. 63 and 64, which show the prior art, the FET 136 can be operated as an adjustable resistor by changing the type of control. Whereas the second switching devices 100 a to 100 c were only used for the purpose of switching in the first embodiment, an analog control or regulation therefore allows the three second switching devices 100 a to 100 c and the three resistors 101 a to 101 c, provided in the first embodiment are replaced with an FET 136 and a limiting resistor 137 , as shown in FIG. 6. The other structure is the same as that in the first embodiment.

Since the FET 136 is operated under the control of an analog voltage signal, the analog / digital converter 114 required in the first embodiment is not required. After the output signal of the second subtractor 113 has been amplified by the amplifier 139 , the gate of the FET 136 can be driven directly. As a result, the number of components can be reduced and current compensation can be carried out at the same time without the analog signal being converted into a digital signal. Therefore, a so-called current compensation shortage due to a quantization error as shown in Fig. 4 in the first embodiment does not occur. An insufficient current component can be compensated extremely precisely.

In addition to the switching control section constituting the first switching device 4 (the first processing circuit), the constant current characteristic of the FET 136 also allows the compensation circuit (the second processing circuit) to provide a constant current. Therefore, the current supplied to the working gap can be kept constant, whereby the machining is stabilized and an extremely excellent machining characteristic is ensured.

In addition, the constant current characteristic of the FET 136 can increase the current response speed compared to the method in which the resistors are used in the first embodiment, and thereby provides a machining power supply device which has excellent characteristics in both accuracy and hum. with respect to the waveform command value.

When the FET 136 is used, the driving clock of the switching device 4 and the relevant upper and lower threshold values are the same as in the first embodiment.

A use of the switched power supply that like explained above a high Has energy supply efficiency leads to the Power supply device for the electrical EDM machine according to the second embodiment made it easy can achieve the dimensions and cost of Energy supply are reduced. In addition, the Addition of the semiconductor gain circuit which the Current hum compensated for when switched Power supply occurs, reducing the number the parts so that only a few parts are required, to turn off a hum, and to deliver one stable current, even in the case of a small current. In addition, the rate of current rise is extreme high. Therefore, a stable one can be used for any current value Achieve editing.

A third embodiment of the present invention will be described below with reference to FIGS. 7 to 9. In this embodiment, a first processing circuit consists of a collector circuit for electrical energy, which is formed by series connection of the energy supply 5 and an auxiliary power supply 28 , for supplying processing energy, and the first switching device 4 , the current detector 7 , the reactor 22 and the first diode 23 for intermittently supplying and collecting electrical energy from the power supply 5 , a third switching device 20 which is connected in such a way that it supplies the working gap with an output current from the electrical energy collector circuit in order to apply the output current to the processing gap in pulsed form, and one second diode 6 connected to return a residual current to the electrical energy collector circuit generated in the working gap when the third switching device 20 is turned off.

As in the first embodiment, the middle series connections of the current limiting resistors 101 a to 101 c and the second switching devices 100 a to 100 c form a second processing circuit. This second processing circuit is connected in parallel to the first processing circuit in order to supply the working gap with a current which is superimposed on the current from the first processing circuit.

Regardless of whether or not a current flows in the working gap, a current always flows in the electrical energy collector circuit, and the current is turned on / off by the third switching device 20 . Therefore, the use of the reactor 22 reduces the times associated with turning the power on and off, that is, the rise time and the fall time of the power in the working gap. These rising and falling speeds may exceed those of the switched power supply according to the first and second embodiments which do not use the reactor 22 , which enables further stabilized processing.

In Fig. 7, 29 denotes a capacitor, and 141 an amplifier for detecting the instantaneous value of the current flowing in the reactor 22 . 142 denotes a third adder / subtractor that processes a difference between the current value of the electrical energy collector circuit and the current command value 104 , and 143 denotes a logic circuit that outputs a clock signal for turning on / off the third switching device 20 in response to the output sign of the output signal 104 . The rest of the arrangement is the same as in the first embodiment and therefore will not be described again here. In addition, the switching devices 4 , 20 and 100 a to 100 c, which are represented by switching symbols, are more generally designated in FIG. 4 and are completely identical to the switching devices previously shown in FIG. 6.

Fig. 8 shows the timing relationship between the third switching device 20 and the current waveform when the present embodiment is used for forming a square wave signal. In the present embodiment, upper and lower threshold values 145 a, 145 b for switching are also set during the stop time in order to keep a circulating current 146 within a predetermined range. The advantage is that if a next pulse rises after the stop time, the current waveform can rise to the command value at extremely high speed.

Fig. 9 is a timing chart of a signal different from a square wave signal, such as a kind of a triangle signal. In order to deliberately set the current rise so that it is slow with this signal form, the circulating current 146 is selected to be higher than the current command value in the original rise stage 147 of the current signal form. Therefore, in the initial stage, the switching device is switched off as indicated at 144 , so that the current is not supplied from the circulation circuit to the working gap. The desired waveform can be provided by supplying the current from the resistance circuit that is required for the initial stage of the current waveform. Something like this could not be achieved at all with the conventional, switched circuit and with conventional circulation circuit systems.

A use of the switched power supply, and also the provision of the circuit for circulating the excess current leads to energy supply, as described above to the fact that the power supply device for the electric discharge machine according to the third embodiment has an extremely high energy supply efficiency, and therefore simply the dimensions and cost of the Energy supply can be reduced. The addition the resistance circuit, which compensates for the current hum, that occurs with the switched energy supply, allows you to turn off a hum, and the Carrying out stable machining, even with one small stream. The addition of the circulation circuit poses moreover, make sure the machining current pulse width can be controlled precisely, resulting in a stable and very leads to reproducible processing.

A fourth embodiment of the present invention will be described below with reference to FIGS. 10 to 13. In Fig. 10, a first processing circuit consists of a collector circuit for electrical energy, the series connection of the power supply 5 and the auxiliary power supply 28 for supplying processing energy, and the first switching device 4 , the current detector 7 , the reactor 22 , the third switching device 20 and the first diode 23 for intermittent delivery and for collecting electrical energy from the power supply 5 , and the second diode 6 is formed, which is connected so that it returns to the electrical energy collector circuit a residual current which is generated in the working gap when that third switching device 20 is turned off. The third switching device 20 is connected so that it supplies the working gap with an output current from the collector circuit for electrical energy, and supplies the output current to the working gap in a pulsed form.

As in the second embodiment, a series circuit of an FET 136 and a resistor 137 forms a second processing circuit. This second processing circuit is connected in parallel to the first processing circuit in order to supply the working gap with a current which is superimposed on the current from the first processing circuit.

In the embodiment of FIG. 10, the collector circuit or storage circuit for electrical energy does not have the auxiliary power supply 28 that was required for the third embodiment shown in FIG. 7. However, since the current increase of the FET circuit can take place at a sufficiently high speed, a circulation or cycle need not be carried out continuously. Instead, with the aid of the high-speed operation performed by the FET, the omission of the circulating auxiliary power supply 28 allows a further reduction in the size and cost of the power supply.

Fig. 11 shows the operation of the circuit according to the present embodiment. Although the command waveform is rectangular, the circulating current value gradually decreases as indicated by 150 because the auxiliary power supply 28 for supplying power is absent during the stop time. However, the rest of the recycle stream, as well as the high speed operation of the FET, provide sufficiently high rise and fall speeds of the stream. In Fig. 11, 148 a denotes an upper threshold, and 148 b a lower threshold.

Fig. 12 shows the operation performed when a kind of a triangular signal is used as a current command. While, as indicated by 153 , the circulating current decreases during the stop time, sufficient current is provided because the next pulse's rising current value is small. In Fig. 12, 151a denotes an upper threshold, and 151b a lower threshold.

Fig. 13 shows that the circulating current is higher than the rising current value of the pulse. In this case, the FET circuit as long as supplies the power to a circulating current value 156 exceeds a current command value 155th While the current supplied by the FET is high, the time is so short that the device temperature does not rise so much that a problem could arise with the operation of the semiconductor. As shown in FIG. 6 with respect to the second embodiment, the FET shares the amount of heat with the resistance directly introduced, and therefore, in terms of temperature design, there is no difficulty in preventing the temperature of the FET device from rising in that regard. In this case, the thermal energy consumed by the FET is reduced to a minimum throughout the circuit and the circuitry is simplified to allow rapid response and delivery of current in response to a command for a selectable waveform. The circuit, which is itself designed to form a constant current circuit, therefore ensures extremely stable processing even when a micro current is supplied. In Fig. 13, 154 a denotes an upper threshold, and 154 b a lower threshold.

The use of switched power supply, and that Providing the circuit to circulate the excess current lead to energy supply as described above to the fact that the power supply device for the Electric eroding machine according to the fourth embodiment has an extremely high energy supply efficiency, and thereby simply the dimensions and cost of the Energy supply can be reduced. The addition the semiconductor amplifier circuit, which the current hum  compensated for with a switched energy supply occurs, causes an electrical hum to be switched off, and stable even in the case of a small current Editing can be done. It also allows the semiconductor gain circuit that the The rate of current rise is sufficiently high without an auxiliary power supply is used in the circulation circuit is reduced so that the number of parts in the circuit is, and so a formation of the circuit at low Cost is achieved. The addition of the circulation circuit also provides exact control or regulation of the Machining current pulse width safely, creating a stable and very reproducible processing can be achieved.

A third embodiment of the present invention will be described below with reference to FIGS. 14 and 17. In Fig. 14, the processing circuit by a first current source 157 and a second current source 158 is formed. The first power source 157 has a higher power supply efficiency than the second power source 158 , and the second power source 158 has a faster response time than the first power source 157 . In order to best utilize the properties of the two current sources and to compensate for their disadvantages, the currents supplied to the working gap are superimposed on one another in the present embodiment.

In Fig. 14, reference numeral 102 denotes a current waveform command value, 161 denotes a signal representing it, 162 denotes an adjustment circuit, 174 denotes a subtraction circuit which subtracts the value of δ1 from the command value signal 161 and sets the subtraction result as a new command value signal 163 , 190 denotes a first signal adder / subtractor which processes a difference between the command value signal 163 and a signal 164 which represents a current detected by the current detector 160 which is supplied from the first current source 157 to the working gap, 165 represents its output signal, 166 denotes a first control circuit that outputs a signal 167 under the control of the signal 165 to control the first current source 157 , 171 denotes a second signal adder / subtractor that processes a difference between the command value signal 161 and the signal 164 , 168 whose A output signal, 169 denotes a second control circuit that outputs a signal 170 under the control of the signal 168 to control the second power source 158 , and 161 represents a switching device through which the first power source 157 performs turn-on / turn-off control of the current from the second power source 158 is supplied, with a predetermined clock.

Referring to Figs. 14 and 17 of operation of the present embodiment is described below. The signal 161 representing the current waveform command value is used as a reference, and this reference value is divided into two values which are sent to the respective current source 157 , 158 as the command values. At this time, the new command value 163 , which is determined by causing the subtracting circuit 174 to subtract the value of δ1 set by the setting circuit 162 from the reference signal 161 , is sent to the first power source 157 as the command value cleverly. Furthermore, a difference between the reference current waveform command signal 161 and the current value supplied from the first current source 157 , that is, a component insufficient for the reference current waveform, is sent to the second current source 158 as the command value. As a result, the insufficient current value for the working gap in the first current source 157 is supplied by the second current source 158 which has a high response speed, whereby a stable current is supplied to the working gap so as to achieve stable processing.

The result of subtracting the value of δ1 from the reference command value by the subtracting circuit 174 was set as the command value for the first current source 157 to some extent generate a current shortage. Unless insufficient current has to be compensated by the second current source 158 and the current value of the first current source 157 is higher than the command value of the reference current waveform, the second current source 158, which has a high response rate, cannot compensate, so that the current response to the current current Working gap is slow, which leads to unstable machining.

A sixth embodiment of the present invention will be described below with reference to FIGS. 15 and 18. In Fig. 15, the processing circuit consists of the first current source 157 and the second current source 158 , as in the fifth embodiment. The first power source 157 has a higher power supply efficiency than the second power source 158 , and the second power source 158 has a higher response speed than the first power source 157 . In order to best use the features of the two current sources and to compensate for their disadvantages, their currents supplied to the working gap are superimposed on one another in the present embodiment. In Fig. 15, 172 denotes an adjustment circuit, and 173 denotes a multiplier circuit which multiplies the command value signal 161 by a value "k" which is larger than "0" and not larger than "1" and the result of the multiplication as a new one Command value signal 163 sets. The rest of the arrangement is identical to that of the fifth embodiment and will not be described again here.

Referring to Figs. 15 and 18, the operation of the present embodiment will be described. The original current waveform command value 102 and the signal 161 representing it are used as a reference, and this reference value is divided into two parts, which are supplied as the command values to the respective current source 157 , 158 . At this time, the result obtained by causing the multiplier 173 to multiply the reference signal 161 by the constant ratio "k" which is larger than "0" and not larger than "1" is caused as the command value sent to the first power source 157 . Furthermore, a difference between the reference current waveform command signal 161 and the current value supplied from the first current source 157 , that is, a component insufficient for the reference current waveform , is supplied as the command value to the second current source 158 . As a result, the insufficient current value for the working gap in the first current source 157 is supplied by the second current source 158 which has a high response speed, thereby supplying a stable current to the working gap so as to achieve stable processing.

The result of multiplying the reference command value by the value "k", which is greater than "0" and not greater than "1" by means of the multiplier circuit 173 , was set as the command value for the first current source 157 to be , to some extent, one To generate electricity shortages. If the second current source 158 does not have to compensate for insufficient current and the current value of the first current source 157 is above the command value of the reference current waveform , no compensation can be carried out by the second current source 158 , which has a high reaction speed, so that the Current reaction to the working gap is slow, which leads to unstable processing. If the reference current waveform 102 is such that it changes the current value at its center, the multiplication of the constants causes the reference value to be divided according to the command current value then prevailing, whereby the excellent power supply efficiency of the first current source 157 can be maximized.

A combination of the energy supply that a current control or Control system, which has a high Has energy supply efficiency, and the Power supply that is a power control or regulation system which has a high reaction rate, such as explained above, causes the Power supply device for the electrical EDM machine according to the fifth and sixth Embodiments of a power supply available provides a high energy supply efficiency has, and a high reaction rate, and none Electricity hum generated, and the like, creating a stable Processing through a compact, inexpensive Energy supply is achieved.

A seventh embodiment of the present invention will be described below with reference to FIGS. 16 and 19. In Fig. 16, the processing circuit by the first current source 157, the second current source 158 and a third current source 175 is formed. Here, the third current source 175 is connected in such a way that it supplies a current in a direction opposite to the working gap in one of the first current source 157 and the second current source 158 . The first power source 157 has a higher power supply efficiency than the second power source 158 and the third power source 175 , and the second power source 158 and the third power source 175 have a higher response speed than the first power source 157 .

In order to make optimum use of the characteristics of the two types of power sources and to take advantage of their disadvantages, their currents supplied to the working gap are superimposed on one another and, moreover, negatively superimposed to reduce an excess component in the present embodiment. In Fig. 16, 177 denotes a first signal adder / subtractor that performs processing and outputting a difference between the command value signal 161 and the signal 164 which represents a current detected by the current detector 160 and from the first current source 157 the working gap is supplied, 166 denotes a first control circuit which outputs the signal 167 under the control of the output signal of the first signal adder / subtractor 166 to control the first current source 157 , 178 denotes a second signal adder / subtractor, the one Difference between command value signal 161 and signal 164 processed, 179 denotes its output signal, and 180 denotes a second control circuit which outputs signals 170 and 176 under the control of signal 179 to control the second and third current sources 158 , 175, respectively.

The operation of the present embodiment will now be explained with reference to FIG. 19. The reference current waveform command value 102 also serves as a current command value for the first current source 157 . In the case of a switched power supply according to FIG. 16, the current waveform that is supplied to the working gap by the first current source 157 has a signal shape designated by 167 a, 167 b in FIG. 15.

It differs from the reference current waveform 102 . Therefore, an insufficient component 170 is provided to the second current source 158 as the command value, and an excess component 176 is provided to the third current source 175 as the command value. Since the third current source 175 has an opposite power supply direction compared to the first and second current sources 157 , 158 , the difference between the command value and the current current value generated for some reason in the first current source 157 can be determined by the second and third current sources 158 , 175 can be compensated. The compensation provided by these high speed response power supplies improves the rise and fall speeds of the current supplied to the working gap and results in extremely constant power supply formation, thereby stabilizing processing and achieving fast processing .

A combination of energy supply with a current control or Control system, which has a high Has energy supply efficiency, and the two Power supplies that have a power control or regulation system which, as explained above, is high Reaction speed has the result that the Power supply device for the electrical EDM machine according to the seventh embodiment Power supply that provides a high Energy supply efficiency and high Response speed, no current hum and generated the like, and achieved a power supply that has a simple control or regulating circuit system, which ensures stable machining with a compact, inexpensive energy supply is achieved.

An eighth embodiment of the present invention will be described below with reference to FIGS. 20 to 22. In Fig. 20, a constant current supply section 200 , which is formed by a first switching device 201 , a first diode 204 and a reactor 203 , is connected to a power supply E0 to supply a DC voltage and gives a current to a turn-on / turn-off output current section 210 . The constant current supply section 200 consists of a voltage drop interrupter consisting of the first switching device 201 , the first diode 204 and the reactor 203 , and a second diode 202 is connected between its input and its output. Furthermore, it is provided with a current detector 205 which detects the current of the reactor 203 . The output current switch-on / switch-off section 210 is formed by a series connection of a second switching device 211 , a third diode 212 and a voltage source 213 and by a fourth diode 214 . The output of the output current switch-on / switch-off section 210 supplies machining energy between the electrode 1 and the workpiece 2 , which are provided in the dielectric, in order to carry out electrical erosion machining, that is to say machining with an electrical discharge.

Furthermore, this device has a comparator 232 , which adds a signal 209 of a ripple current setting section 205 to a signal 208 of an output current command section 230 , in order to compare a resultant addition signal 216 with the signal of the current detector 205 , which cuts the current of the reactor 203 in the constant current supply 200 recorded . Furthermore, this device has a gate driver circuit 206 which controls the first switching device 201 to control the output current of the constant current supply section 200 to a predetermined current value, and further has a gate driver circuit 215 which switches the second switching device 211 on / off to turn on / off the signal of a discharge command section 240 so as to control the output current on / off section 210 .

Fig. 21 (a) shows the signal of the discharge command portion 240. A pulse signal 260 turns on the switching device 211 of the output current on / off section 210 to apply an unloaded voltage 261 between the electrode and the workpiece 2 , as shown in Fig. 21 (b). Then, when a discharge occurs between the electrode 1 and the workpiece 2, the no-load voltage changes to become a discharge voltage as indicated by reference numeral 262 .

When the discharge occurs, a current flows from the power supply E0 to the electrode 1 and the workpiece 2 through the first switching device 201 , the reactor 203 , the second switching device 211 and the fourth diode 214 . The signal 208 in Fig. 21 (c) denotes the signal from the output current level setting section 230 which is output in synchronism with the start of the discharge. The signal of the ripple current setting device 250 , which is output by the ripple current setting device 250 , is also denoted by 209 in FIG. 21 (d), likewise synchronized with the start of the discharge. Furthermore, with 216 in Fig. 21 (e) denotes a signal which the Brummstromeinstellschaltung is obtained by adding 250 of the signal 208 of the Ausgangstrompegeleinstellabschnitts 230 and the signal 209 (hereinafter referred to as the addition signal).

When the discharge is started, the output current of the constant current supply section 200 increases with the time constant of the inductance of the reactor 203 in the circuit, and the detection value of the output current is as indicated by a signal 207 . The addition signal 216 and the current detector value 207 are constantly compared. When the detector signal 207 drops below the addition signal 216 , the comparator 232 outputs a signal which keeps the first switching device 201 switched on. If the detector signal 207 rises above the addition signal 216 , the comparator 232 outputs a signal which keeps the first switching device 201 switched off.

The details of the constant current control or regulation are described below. Fig. 22 shows (a) increases, a waveform 263 in Fig. 21 (e). If the current value is less than the addition signal 216 , the waveform continues to increase, as indicated by 264 , and the current detector value 207 increases in accordance with the time constant of the inductance. The addition signal 216 continues to change starting from the start of the discharge. If it crosses the waveform of the addition signal 216 during its rise, as indicated at 265 , the current detector value 207 drops below the value of the addition signal 216 and is switched off. Therefore, switching is forced to repeat in the vicinity of current detector value 207 and ultimately to be kept within a current hum setting peak value 266 . The triangular signal used as the vibration signal for the current hum setpoint 209 in the description of the present embodiment may be a square wave signal 267 or a sine wave 268 as in Fig. 22 (b) or 22 (c) to produce the same effect.

A ninth embodiment will be described below with reference to FIGS. 23 to 26. A current hum setting device 210 in FIG. 23 has a Vf converter that converts the signal of the current command section 230 to a frequency (hereinafter referred to as Vf conversion), and is configured to turn on the signal 208 of the current setting section 230 supplies the Vf converter, and a signal which results from the addition of the Vf-converted signal 209 and the current setting value 208 to the comparator 232 , and is further designed so that the frequency of the transmission signal 209 in response to the signal 208 of the current setting section 230 is changed. The rest of the arrangement is identical to that of the eighth embodiment and will not be described again here.

Fig. 24 is an example of the characteristic of Vf shows conversion. The signal 208 of the current setting section 230 and the frequency of the transmission signal 209 are almost inversely proportional to each other. The frequency of the current is preset to be higher when the level of signal 208 is lower, and the hum is set to a maximum when the level of signal 208 is at a maximum. Since there is a limit to the frequency response of the first switching device 201 when the level of the signal 208 decreases to some extent, a maximum frequency value fmax is set to prevent the frequency from increasing when the level of the signal 208 reaches or falls below a predetermined value this drops. Accordingly, a minimum frequency value fmin is set to prevent the frequency from reaching a predetermined value or rising beyond it.

Fig. 25 (a) shows the addition of stream 210 and the current detecting signal 207 at a time when the current peak value is high. The addition signal 210 has a low frequency due to the high peak value, and therefore a switching cycle 271 of the first switching device 201 is long, as shown in Fig. 25 (b), which increases the on-time and off-time accordingly, and increases a hum ΔI1. Fig. 26 (c) shows the addition flow 216 and the current detecting signal 207 at a time when the current peak value is low. If the current peak value is low, the frequency of the addition signal 216 is high, the switching cycle 271 of the first switching device 201 is short, as shown in FIG. 25 (d), and therefore the switch-on time and the switch-off time decrease, which increases a hum ΔI2. By modulating the frequency of the addition signal in accordance with the current peak value as described above, the hum can be reduced, which ultimately provides a uniform processing accuracy.

A tenth embodiment will now be described with reference to FIGS. 26 and 27. In FIG. 26, the current hum setting device 270 has a device for outputting a synchronization signal 273 . This synchronization signal 273 and an output signal 272 of a comparator 232 are applied to a gate in order to drive the first switching device 201 by means of an output 281 of the gate. The gate consists of a first NAND circuit 276 which receives the synchronization signal 273 , which is output by the current hum setting device 270 , and the output signal 272 of the comparator 272 , a second NAND circuit 277 , which the synchronization signal 273 and the output signal 272 via inverters 285 , and from an RS flip-flop 278 , which receives an output signal 279 of the first NAND circuit 276 at its reset terminal (reset) and receives an output signal 280 of the second NAND circuit 277 at its set terminal (set). The other arrangement is identical to that of the ninth embodiment and will not be described again here.

The Fig. 27 (a) to (g) show a timing chart in the tenth embodiment, and with reference to this timing chart, the operation will be described below. Fig. 27 (a) shows a square wave synchronization signal 273 from the ripple current adjuster 270 , Fig. 27 (b) shows a signal 209 from the ripple current adjuster 270 , Fig. 27 (c) shows the detector signal 207 and the addition signal 216 of the current waveform, and Fig. 27 (d) shows the output signal 272 of the comparator 232nd When the current detector value 207 exceeds the addition signal 216 , the output of the comparator is switched low to switch off the first switching device 201 . However, noise 274 occurs at the time of normal switching, as shown in Fig. 27 (c).

Therefore, as shown in Fig. 27 (d), the comparator 232 continuously compares the current detector value 207 and the addition signal 216 , and compares the noise 274 and the addition signal 216 , and the output signal 272 of the comparator 232 occurs due to the influence of the noise 274 Power-on / power-off repeat section 275 which results in malfunction. For this reason, as shown in FIGS. 27 (e) and (f), the output signal 272 of the comparator 232 and the square wave synchronization signal 273 of the current hum setting device 270 of the first NAND circuit 276 , the inverter 285 and the second NAND circuit 277 input, and the outputs 279, 280 of the first and second NAND circuit 276 is input 277 to the flip-flop 278, to invert the flip-flop 278 only once with respect to the high and low value of the square-wave signal, whereby a switching signal 280 may be provided according to Fig. 27 (g), to disable an unstable operation in the portion of noise 274th Therefore, an exact switching operation can be performed when noise 274 is present at the input of the comparator 232 .

An eleventh embodiment will be described below with reference to FIGS. 28 to 30. The arrangement of a first circuit 290 and a second circuit 291 is identical to the above-described circuit according to the eighth embodiment, and will not be described again here. In Fig. 28, the first circuit 290 and the second circuit 291 are connected in parallel to the working gap, and are provided with a detector device 205 which portion of the output current of the first constant current supply 200 determines, as well as a detector device 305 which the output current from a second constant current supply section 300 notes. The output signal of the output current level setting device 230 , which forms the command for the output currents of the first and second constant current supply sections, is added to the signal of the first ripple current setting device 250 (hereinafter referred to as the first addition signal 351 ), which forms the command command for the ripple current of the output current of the first constant current supply section 200 , and is compared by the first comparator 232 with the output signal of the detector device 205 (hereinafter referred to as the first detector signal 352 ), which detects the output current of the first constant current supply section 200 .

The output signal of the detector device 305 , which detects the output current of the second constant current supply section 300 , represents a second detector signal 253. A first ripple current setting output signal 250 , which represents the command instruction for the ripple current of the output current of the first constant current supply section 200 , is inverted by an inverter 354 . The setting value of the inverting device 354 is a value that is 180 ° out of phase. The setting value 208 of the first / second output current level setting device is added to the inversion signal and generates a second addition signal 355 . The addition signal 355 and the detector value 353 of the second detector device are compared by a second comparator 332 .

The operation will now be described with reference to a timing chart of Figs. 29 (a) to (d). Fig. 29 (a) shows a switch (ON) for the second switching device 215 of the first constant current circuit 200 and a second switching device 315 of the second constant current circuit 300, which the discharge command section is turned 240 under the control of a command 360th Fig. 29 (b) shows the output signal 208 which is output from the output current level setting means 230 in synchronism with the start of the discharge. At this time, the output current level signal 208 is set to a value that is about half of a desired output value. Fig. 29 (c) 352 shows the detector signal of the first constant-current supply section and the first addition signal 351st The first output signal 352 and the first addition signal 351 are compared by the first comparator 232 . If the detector signal 352 of the first constant current supply section 200 is lower than the first addition signal 351 , the gate driver circuit 206 switches on the first switching device 201 . In contrast, when the first output signal 352 is higher than the first addition signal 351 , the first switching device 201 is turned off by the gate driver circuit 206 .

Fig. 29 (d) shows the second output signal 353 and the second addition signal 355th The second output signal 353 and the second addition signal 355 are compared by the second comparator 332 . If the second output detector signal 353 is lower than the second addition signal 355 , the first switching device 301 of the second constant current supply device 300 is switched on by a gate driver circuit 306 . In contrast, when the second output signal 353 is higher than the second addition signal 355 , the first switching device 301 is turned off by the gate driver circuit 306 .

Fig. 30 (a) shows an output current 362 of the first circuit, and Fig. 30 (b) shows an output current 363 of the second circuit. In order to cause a flow of the output currents of the circuits, the outputs of the constant current supply sections are applied to the gap formed between the electrode 1 and the workpiece 2 by their respective second switching devices 211 , 311 corresponding to the desired pulse width which is provided by an output 360 of the Discharge command section 250 is set. Since the hum of the output signal 352 of the constant current supply section 200 in the first circuit 290 is determined by the addition signal 351 , the output current 362 of the first circuit 290 has a hum width ΔI1. Since the output signal 353 of the constant current supply section in the second circuit 291 is controlled by the second addition signal 355 which is 180 ° out of phase with the addition signal 351 which controls the hum of the constant current supply section 200 in the first circuit 290 , the hum of the output current 363 of the second circuit 291 flowing in the working gap is 180 ° out of phase with the hum of the output current 362 of the first circuit.

At this time, the hum ΔI2 of the output current 363 has almost the same width as the hum ΔI1 of the first output current. Therefore, as shown in FIG. 30 (c), a working gap current 361 flowing in the working gap represents a current resulting from the addition of the output currents 362 and 363 , and serves to offset the hum part of the output currents 362 and 363 in relation to each other, and therefore a hum ΔI3 becomes comparatively extremely small. This circuit system provides a low hum current and provides a discharge current that has a shape that is almost identical to the current level command setting.

A twelfth embodiment will be described below with reference to Figs. 31 and 32 (a) to (c). In Fig. 31, 400 denotes a clock that outputs a signal 401 shown in Fig. 32 (b). The rest of the arrangement is substantially identical to that of the eighth embodiment, except that the ripple current adjuster 250 is not provided, and therefore will not be described again here.

Based on timing charts in Figs. 32 (a) to (c), the operation of the present embodiment will be described below. In Fig. 32 (a), 208 denotes the signal of the output current level setting section 230 , which is output in synchronism with the start of the discharge. 401 in FIG. 32 (b) denotes the output of the clock 400 .

When the discharge begins, the output current of the constant current supply section 200 increases in accordance with the time constant of the inductance of the reactor (choke coil) 203 in the circuit. When the detector signal 207 rises above the signal 208 , the comparator 232 outputs a signal which switches on the first switching device 201 . If a predetermined period of time has then expired, the clock generator 400 outputs a signal which switches on the first switching device 201 . Then, the detector signal 207 rises above the signal 208 , and the comparator 232 outputs the signal that turns off the first switching device 201 . This results in a current detector value indicated by signal 207 in Fig. 32 (c).

If the inductance value of the reactor (choke coil) 203 and the on / off time of the clock generator 400 are selected appropriately, this circuit system also provides a low hum current and delivers a discharge current, the shape of which is practically identical to the current level command setting.

A thirteenth embodiment of the present invention will be described below with reference to FIGS. 33 to 42. Fig. 33 is a main circuit diagram showing the thirteenth embodiment, where E1 denotes a first DC power supply, and 201 denotes a first switching device which is on / off by the gate drive circuit 206. The reactor 203 is placed between the first switching device 201 and the second switching device 211 , and the electrode 1 and the workpiece 2 are connected to the second switching device 211 and the first DC power supply E1. The first diode 202 is connected between the connection point of the first switching device 201 and the reactor 203 and the first DC power supply E1, and the second diode 204 is connected between the connection point of the first energy supply E1 and the first switching device 201 and the connection point of the reactor 203 and the second Switching device 211 switched, in a direction in which the current flows to the first DC power supply E1.

A series connection of the third diode 212 and the DC power supply 213 is connected between the side of the electrode 1 of the second switching device 211 and the negative voltage side of the first DC power supply E1. The current detector 205 is connected so that it detects the current flowing in the reactor 203 . A series circuit of the second DC power supply E2, which has a voltage that can supply the working gap with a voltage that is substantially less than or equal to an electrical discharge voltage, a third switching device 501 and a diode 502 are connected in parallel to the first diode 202 . This third switching device 501 is turned on / off by a gate driver circuit 503 .

It is noted that in this embodiment, the first switching device 201 , the first diode 202 and the reactor 203 form the constant current supply section 200 , and the output current turn-on / turn-off section 210 composed of the second switching device 211 and a series connection of the third diode 212 and the DC power supply 213 exists.

FIG. 34 shows a control circuit of the gate driver circuits 503 , 206 and 215 shown in FIG. 33, wherein a first comparator 504 compares a current command value S1 with a current detector value I1 of the current detector 205 and outputs a signal to the input terminal of a clock generator circuit 512 . A second comparator 505 compares an overcurrent command value 507 , which is provided by connecting a direct voltage 506 in series with the current command value S1, with the current detector value I1 of the current detector 205 , and outputs a signal to the reset terminal R of a first flip-flop 508 out. The output of the first comparator 504 is inverted by an inverter 509 , and the result of the inversion is connected to the set terminal S of the first flip-flop 508 .

Meanwhile, an electric discharge signal H1, which is output from an NC device, turns the second switching device 211 on and off under the control of the gate driver circuit 215 . The AND state of the electrical discharge signal H1, the output of the clock circuit 512 , and the output of the first flip-flop 508 is determined by an AND circuit 511 in order to switch the first switching device 201 on / off under the control of the gate driver circuit 206 . Furthermore, the AND state of the electrical discharge signal H1 and the output of the first flip-flop 508 is determined by an AND circuit 510 in order to switch the third switching device 501 on / off under the control of the gate driver circuit 503 .

Figs. 35 and 36 show specific examples of the clock circuit 512 in Fig. 34. In the example of the clock circuit 512 of Fig. 35, the output of the first comparator 504 is connected to the input terminal IN. A MOSFET 512 A closes / opens a capacitor 512 C in a time constant circuit which is formed by a resistor 512 B and the capacitor 512 C. If this circuit are shown in Fig. 37 in which an input signal (a) and an output signal (b), the output of the first comparator is switched 504 at a point 600 in the input signal (a) to the high level, the MOSFET switches 512 A to close capacitor 512C , thereby switching output signal (b) to a low level.

When the output (b) of the first comparator 504 is switched to a low level at a point 601 in the input (a), the MOSFET 512A turns off to open the capacitor 512C , causing the output (d) to pass the threshold of a buffer exceeds 512 D, and is switched to a high level at a point 602 , after a period 603 of certain length, by the time constant circuit consisting of the resistor 512 B and the capacitor 512 C. The output OUT is switched to a low level when the output of the first comparator 504 is switched to a high level, and the output OUT is switched to a high level by a certain time after the output of the first comparator 504 is switched to a low level. The clock circuit 512 operates in the manner described above. In the example of the clock circuit 512 of Fig. 36, which and a flip-flop 512 F is formed by a logic circuit comprising a monostable multivibrator 512 E, is the process of switching of the output OUT to the high level by the monostable at the multi-vibrator 512 E set time identical to the process of Fig. 37.

Timing diagrams and waveform diagrams of FIG. 38 to illustrate the operation of this thirteenth embodiment. In Fig. 38, (a) shows the electric discharge signal H1, (b) shows an output voltage waveform, (c) shows the waveform of the current command value S1 output from a controller (not shown) of the electric discharge machine, (d) shows an output current waveform , (e) shows the on / off state of the first switching device 201 , (f) shows the on / off state of the third switching device 501 , (g) shows the current through state of the diode 202 , (h) shows the current through state of the diode 502 , (i) shows the output state of the first comparator 504 , (k) shows the output state of the clock circuit 512 , (l) shows the output state of the first flip-flop 508 , and (m) shows the output state of the second comparator 505 .

When the electrical discharge signal H1 is turned on at a point 700 in the waveform (a), the gate driver circuit 215 turns on the second switching device 211 . Since the first switching device 201 in FIG. 33 is turned on at this time, as shown in (e), the voltage of the first DC power supply E1 is applied between the electrode 1 and the workpiece 2 , as shown at (b). The gap between the electrode 1 and the workpiece 2 is filled with a dielectric liquid, such as oil or water, and is extremely precisely controlled by a servomechanism, an NC device, and the like (not shown). When a dielectric breakdown occurs in this extremely small gap, an electrical discharge is generated between the electrode 1 and the workpiece 2 . This is indicated by 701 in waveform (a), and the output voltage in (b) serves as an electrical discharge voltage 702 . This electrical discharge voltage is practically constant between approximately 25 and 30 V. As soon as the electrical discharge occurs, the current begins to flow between the electrode 1 and the workpiece 2 . The current denoted by 703 in (d) flows through the first DC power supply E1, the first switching device 201 , the reactor (choke coil) 203 and the second switching device 211 , and increases rapidly because the voltage of approximately 80 V of the first DC power supply E1 is higher is as the electrical discharge voltage 702 .

When the output current, i.e. the current of the reactor 203 , has reached the current command value S1, at a point 704 , as shown in (i), the output of the first comparator 504 is switched to a high level. Therefore, according to (k), the output of the clock 512 is switched to a low level and the first switching device 201 is switched off, as shown in (e). When the output of the first comparator 504 is switched to a high level, the output of the clock circuit 512 is only switched to a low level for a predetermined period of time, which is denoted by 705 in (k), and is then switched to a high level, whereby the first switching device 201 is switched on again.

During this preset time of the clock generator circuit 512 , that is to say during the period 705 when the first switching device 201 is switched off, the current from the second DC power supply E2 is applied between the electrode 1 and the workpiece 2 , via the third switching device 501 , which has already been switched on Diode 502 and reactor 203 . Since the second DC power supply E2 is set to such a voltage that it is as large as or slightly larger than the electrical discharge voltage 702 , the output current decreases slightly as indicated at 707 (d). This is due to the small change in current due to the reduced terminal voltage of the reactor 203 . This is repeated to cause the output current to follow the current command value S1 as shown in (d).

When the current command value S1 reaches a certain value, as indicated at 703 (c), the output current slowly decreases as indicated by 709 (d), but then increases rapidly, as shown at 710 (d). Since the time of the drop at 709 is the set time of the clock generator 512 as at 707 , the switching frequency of the first switching device 201 does not drop below the time period 709 when 710 approaches zero. As the current denoted by 709 slowly decreases, the hum of the output current becomes small. By connecting the second DC power supply E2 while the first switching device 201 is switched off for a certain period of time, a power supply device for an electric discharge machine can be provided which provides an output current waveform with low hum components.

Fig. 39 shows waveform charts and timing charts used to describe the operation in which a short circuit between the electrode 1 and the workpiece 2 occurs during the generation of an electrical discharge. If a short circuit occurs at a point 711 when the discharge electric current flows as in Fig. 38, the output voltage (b) drops below the voltage of the second DC power supply E2, whereby the output current (d) increases as indicated by 712 . When this current, which also represents the current of the reactor 203 , increases and reaches the overcurrent detection value 507 , which is denoted by 713 (d), the output of the second comparator 505 is switched to a high level, which also corresponds to the current detector value I1 of the current detector 205 the overcurrent command value 407 , which results from the addition of the voltage of a DC power supply 505 to the current command value S1.

Therefore, the output Q of the first flip-flop 508 is switched to a low level, and the third switching device 501 is switched off by the AND circuit 510 . It is pointed out that the first switching device 201 is already switched off at this point in time. Therefore, the output current is provided by the second diode 202 , the reactor 203 and the second switching device 211 and decreases as indicated at 714 . If the output current continues to decrease, down to the current command value S1, the output of the first comparator 504 is turned low at (i) at a point 715 , whereby the signal inverted by the inverter 509 sets the first flip-flop 508 so that the output Q is switched to a high level. This turns on the third switching device 501 so as to increase the output current. Therefore, when a short circuit occurs between the electrode 1 and the workpiece 2 , the output current increases and decreases between the current command value S1 and the overcurrent command value 507 , and does not increase rapidly like a (common) short circuit current, thereby preventing the electrode 1 and the workpiece 2 are damaged by a high current.

If the short circuit condition is overcome for any reason, the output voltage will return to the discharge electrical current at a point 716 , as shown at 717 (b). Therefore, the output current (d) decreases sharply, and when it has decreased to the current command value S1 as shown at 718 , the output of the first comparator 504 is switched to a low level and the first flip-flop 508 is set by the inverter 509 , whereupon its output Q is switched high, the third switching device 501 is switched on, and the output current slowly decreases, as shown at 719 . The clock circuit 512 outputs a low level signal during the set time after the output of the first comparator 504 is switched to a low level and returns to its normal operation.

Fig. 40 shows a modification of the control or regulating circuit for the FIG. 34 shown gate driver circuits 206, 215, 503, and illustrates a method for adding the DC voltage 506 of the circuit in Fig. 34 to the current command value S1 with the aid of an adder 513, to determine overcurrent command value 507 . With this modification, operation identical to that of the circuit of Fig. 34 can be performed.

Fig. 41 illustrates waveform diagrams represents, in which were actual output currents measured when the peak value of the current command value in accordance with S1 was 35 amperes in the apparatus of the thirteenth embodiment. (a) shows the changes in the current command value 51 , the peak value of which is 35 amperes. (b) shows a waveform at a time when the voltage of the first DC power supply E1 is 80 V and the voltage of the second DC power supply E2 is 0 volts, (c) shows a waveform at a time when the voltage of the first DC power supply E1 is 80 V is and the voltage of the second DC power supply E2 is 15 V, and (d) shows a waveform at a time when the voltage of the first DC power supply E1 is 80 V and the voltage of the second DC power supply E2 is 30 V. These diagrams show that although the voltage of the second DC power supply E2 was zero, that is to say in the conventional device in which the second DC power supply E2 was not present, hum components 604 of almost 16 amperes occurred, but the hum is extremely small, such as at (c) where the hum 605 is 7 amps when the voltage of the second DC power supply E2 is 15 V and as in (d) where the hum 606 is 2 amps when the voltage of the second DC power supply E2 Is 30 V.

Furthermore. 42 shows waveform diagrams, in which actual output currents were measured when the peak value of the current command value S1 is 10 amperes in the apparatus according to this thirteenth embodiment. (a) shows the changes in the current command value S1, the peak value of which is 10 amperes. (b) shows a waveform at a time when the voltage of the first DC power supply E1 is 80 V and the voltage of the second DC power supply E2 is 0 V, (c) shows a waveform at a time when the voltage of the first DC power supply E1 Is 80 V, and the voltage of the second DC power supply E2 is 15 V, and (d) shows a waveform at a time when the voltage of the first DC power supply E1 is 80 V and the voltage of the second DC power supply E2 is 30 V. These diagrams show that while hum component 604 of nearly 10 amperes occurred when the voltage of the second DC power supply E2 was 0 V, the hum component was considerably smaller, as shown in (c), where hum 605 was 5 ampere when the voltage of the second DC power supply E2 is 15 V, or as in (e) where the ripple portion 606 is almost zero when the voltage of the second DC power supply E2 is 30 V.

Particularly in (b), where the output current value on a leading edge 607 of the current is zero, electrical discharge machining performed with such a waveform results in stopping the generated electric discharge, which does not allow the current waveform of the command value (a) to be output. If the second DC power supply E2 has a voltage of 30 V, a current waveform similar to the command value can be provided, as indicated at 608 , and the hum components are almost zero, as indicated at 606 . Although 30 V was selected here for the second DC power supply E2, the third switching device 501 , the diode 502 , the DC resistance value of the reactor (choke coil) 203 and the switch-on voltage of the second switching device 211 are present, and therefore there is actually a voltage that has a value which is obtained by subtracting the turn-on voltage from the 30 V of the second DC power supply E2, the DC voltage source which causes the current to flow to the voltage across the electrode 1 and the workpiece 2 . It is pointed out that the desired goal can be achieved if the voltage applied to the working gap by the second DC power supply E2 is approximately 1 to 2 V higher than the electrical discharge voltage.

Referring to Figure, the. 43 to 45, a fourteenth embodiment of the present invention will be described below. Fig. 43 is a main circuit diagram showing the fourteenth embodiment, wherein E1 denotes a first DC power supply, and 201, a first switching device which is turned off by the gate driver circuit 206 on / off. The reactor 203 is connected between the first switching device 201 and the second switching device 211 , and the electrode 1 and the workpiece 2 are connected to the second switching device 211 and the first DC power supply E1. The first diode 202 is connected between the connection point of the first switching device 201 and the reactor 203 and the first DC power supply E1, and the second diode 204 is connected between the connection point of the first DC power supply E1 and the first switching device 201 and the connection point of the reactor 203 and the second Switching device 211 switched in the direction in which the current flows to the first DC power supply E1.

A series connection of the third diode 212 and the DC power supply 213 is connected between the electrode side of the second switching device 211 and the negative voltage side of the first DC power supply E1. The current detector 205 is connected to detect the current flowing in the reactor 203 . A series connection of the second DC power supply E2, which has a voltage that can supply the working gap with a voltage that is substantially equal to or lower than the electrical discharge voltage, of the third switching device 501 and the diode 502 is connected in parallel to the first diode 202 . This third switching device 501 is turned on / off by the gate driver circuit 503 . A series connection of a third DC power supply E3, which has a voltage that can supply the working gap with a voltage that is higher than the electrical discharge voltage and lower than a voltage that is supplied by the first DC power supply, with a fourth switching device 514 and a diode 515 is connected in parallel to the first diode 202 . This fourth switching device 514 is turned on / off by a gate driver circuit 516 .

Fig. 44 shows a control circuit of the gate driver circuits 503 , 516 , 206 and 215 shown in Fig. 43, wherein the first comparator 504 compares the current command value S1 with the current detector value I1 of the current detector 205 , and a signal outputs the input terminal of the clock circuit 512 . The second comparator 505 compares the overcurrent command value 507 , which is provided by connecting the DC voltage 506 in series with the current command value S1, to the current detector value I1 of the current detector 205 , and outputs a signal to the reset terminal R of the first flip-flop 508 . The output signal of the first comparator 501 is inverted by the inverter 509 , and the result of the inversion is applied to the set terminal S of the first flip-flop 508 .

The electrical discharge signal H1 switches the second switching device 211 on and off under the control of the gate driver circuit 215 . The AND state of a current rise signal H2, the output signal of the clock circuit 512 , and the electrical discharge signal H1 is determined by an AND circuit 519 to turn on / off the first switching device 201 under the control of the gate driver circuit 206 . The AND state of the output of an OR circuit 517 , which provides the OR state of the current rise signal H2 and the output of the clock circuit 512 , the electrical discharge signal H1 and the output of the first flip-flop 508 , is determined by an AND circuit 518 determined to turn on / off the fourth switching device 514 under the control of the gate driver circuit 516 . Furthermore, the AND state of the electrical discharge signal H1 and the output of the first flip-flop 508 is determined by the AND circuit 510 in order to switch the third switching device 501 on / off under the control of the gate driver circuit 503 .

The timing charts and waveform charts shown in Fig. 45 show the operation of this fourteenth embodiment. In Fig. 45 (a) shows the electrical discharge signal H1, (b) shows the output voltage waveform, (c) shows the waveform of the current command value S1, the controller (not shown) of one of the electrical discharge machine is issued (d) shows the output current waveform, (e) shows the on / off state of the first switching device 201 , (f) shows the on / off state of the third switching device 501 , (g) shows the on state of the diode 202 , (h) shows the Current on state of diode 502 , (i) shows the output state of the first comparator 504 , (j) shows the current rise signal H2, (k) shows the output state of the clock circuit 512 , (n) shows the on / off state of the fourth switching device 514 , and (o) shows the on state of the diode 515 .

When the electrical discharge signal H1 is turned on at a point 700 in Fig. 45 (a), the gate driver circuit 215 turns on the second switching device 211 . At point 700 , the current rise signal H2 is also switched on. Since the first switching device 201 in Fig. 43 is turned on at this time, as shown at (e), the voltage of the first DC power is supply E1 between the electrode 1 and the workpiece 2 is applied as a no-load voltage, as in (b). The gap between the electrode 1 and the workpiece 2 is filled with the dielectric fluid, for example oil or water, and controlled extremely precisely by a servomechanism, an NC device and the like (not shown). If a dielectric breakdown occurs in this extremely small gap, an electrical discharge is generated between the electrode 1 and the workpiece 2 . This is indicated by 701 in FIG. 45 (a), and the output voltage in (b) serves as the electrical discharge voltage 702 . This electrical discharge voltage is approximately constant between approximately 25 and 30 V. As soon as the electrical discharge occurs, the current begins to flow between the electrode 1 and the workpiece 2 . The current drawn by 703 (d) flows through the first DC power supply E1, the first switching device 201 , the reactor (choke coil) 203 , and the second switching device 211 , and rises rapidly as the voltage of the first DC power supply E1 is approximately 80 V higher is as the electrical discharge voltage 702 .

When the output current, i.e. the current of the reactor 203 , has reached the current command value S1, the output of the first comparator 504 is switched to a high level at a point 704 , as shown in (i). Therefore, the output of the clock circuit 512 is switched to low level according to (k), and as shown in (e), the first switching device 201 is switched off. When the output of the first comparator 504 is switched to a low level, the output of the clock circuit 512 is switched to a low level only for a predetermined period of time, which is denoted by 705 in (k), and then switched to a high level, thereby the first switching device 201 is switched on again.

During this predetermined period of time of the clock circuit 512 , i.e. during the period 705 when the first switching device 201 is switched off, the fourth switching device 514 shown in FIG. 45 (n) is already switched on, as a result of which the current from the third DC power supply E1 is in place is delivered between the electrode 1 and the workpiece 2 by the fourth switching device 514 , the diode 515 , the reactor 203 and the second switching device 210 , which is already switched on. Since the voltage of the third DC power supply E3 is set to be gradually higher than the electrical discharge voltage 702 , the output current increases slightly, as indicated in (d) at 707 . This occurs due to the slight increase in current because the terminal voltage of the reactor (choke coil) 203 is slightly high. This is repeated to cause the output current to follow the current command value S1 as shown in (d).

If the current rise signal H2 in (j) is switched to a low level at a point 720 (c) of FIG. 45, where the current command value S1, which is rising, reaches a certain value, the output current slowly decreases as shown at 709 because that third switching device 501 is on, but the output current also rises slowly, as shown at 721 , since the third DC power supply E3 is connected when the fourth switching device 514 is on. At this time, since the fall time denoted by 709 is the set time of the clock circuit 512 as shown at 707 and the current rises slowly, the switching frequency is about twice the period of this signal 709 , the ripple portions of the output current are low, and is Switching frequency low. Therefore, a power supply device for an electric discharge machine is provided in this manner, which provides an output current waveform with a low hum when the inductance value of the reactor (choke coil) 203 is low. Specifically, when the current rise signal H2 is turned on to increase the current command value, the current is supplied from the third DC power supply E3 to cause the output current to increase further when the first switching device 201 is turned off, thereby providing a power supply device for an electric discharge machine which provides a low hum output current waveform.

A fifteenth embodiment of the present invention will be described below with reference to FIGS. 46 to 48. Fig. 46 is a main circuit diagram of the fifteenth embodiment in which E1 a first DC power supply designated, and 201, a first switching device which is turned off by the gate driver circuit 206 on / off. The reactor 203 is connected between the first switching device 201 and the second switching device 211 , and the electrode 1 and the workpiece 2 are connected to the second switching device 211 and the first DC power supply. The first diode 202 is connected between the connection point of the first switching device 201 and the reactor 203 and the first DC power supply E1, and the second diode 204 is connected between the connection point of the first DC power supply E1 and the first switching device 201 and the connection point of the reactor 203 and the second Switching device 211 switched in the direction in which the current flows to the first DC power supply E1.

A series connection of the third diode 212 and the DC power supply 213 is connected between the side of the electrode 1 of the second switching device 211 and the negative voltage side of the first DC power supply E1. The current detector 205 is connected so that it detects the current flowing in the reactor 203 . A series connection of the second DC power supply E2, which has a voltage that can supply the working gap with a voltage that is substantially equal to or lower than the electrical discharge voltage, of the third switching device 501 and the diode 502 is connected in parallel to the first diode 202 . This third switching device 501 is turned on / off by the gate driver circuit 503 . A series circuit of a variable fourth DC power supply E4, a fifth switching device 521 and a diode 522 is connected in parallel to the first diode 202 . This fifth switching device 521 is switched on / off by a gate driver circuit 520 .

Fig. 47 shows a control circuit of the gate driver circuits 503 , 520 , 206 and 215 shown in Fig. 46, wherein the first comparator compares the current command value S1 and the current detector value I1 of the current detector 205 , and a signal to the input terminal of the clock circuit 512 issues. The second comparator 505 compares the overcurrent command value 507 , which is provided by connecting the DC voltage 506 in series with the current command value S1, with the current detector value I1 of the current detector 205 , and outputs a signal to the reset terminal R of the first flip-flop 508 . The output signal of the first comparator 504 is inverted by the inverter 509 , and the inverted result is connected to the set terminal S of the first flip-flop 508 .

The electric discharge signal H1 switches the second switching device 211 on / off under the control of the gate driver circuit 215 . The AND state of an unloaded voltage signal H3 and the electrical discharge signal H1 is determined by an AND circuit 524 in order to switch the first switching device 201 on / off under the control of the gate driver circuit 206 . The AND state of the output of the clock circuit 512 , the electrical discharge signal H1 and the output of the first flip-flop 508 is determined by an AND circuit 523 in order to switch the fifth switching device 521 on / off under the control of the gate driver circuit 520 . Furthermore, the AND state of the electrical discharge signal H1 and the output of the first flip-flop 508 is determined by the AND circuit 510 in order to switch the third switching device 501 on / off under the control of the gate driver circuit 503 .

Time charts and waveform charts shown in Fig. 48 show the operation of this fifteenth embodiment. In Fig. 48 (a) shows the electrical discharge signal H1, (b) the output voltage waveform, (c) the waveform of the current command value S1, the controller (not shown) of one of the electrical discharge machine is output, (d) the output current Waveform, (e) the on / off state of the first switching device 201 , (f) the on / off state of the third switching device 501 , (g) the current conduction state of the diode 202 , (h) the current conduction state of the diode 502 , ( i) the output state of the first comparator 504 , (j) the unloaded voltage signal H3, (k) the output state of the clock circuit 512 , (n) the on / off state of the fifth switching device 521 , and (o) the current continuity state of the diode 522 .

When the electrical discharge signal H1 is turned on at a point 700 in FIG. 48 (a), the gate driver circuit 215 turns on the second switching device 211 . The unloaded voltage signal H3 is also switched on at point 700 . Since the first switching device 201 shown in FIG. 46 is switched on at this time, as shown in (e), the voltage of the first DC power supply E1 between the electrode 1 and the workpiece 2 is applied as an unloaded voltage, as in (b) shown. The current command value S1 in the present embodiment may also have a waveform that only provides a command for the peak value of the electric discharge current, as shown in (c).

The gap between the electrode 1 and the workpiece 2 is filled with the dielectric fluid, for example oil or water, and is controlled extremely precisely by a servomechanism, an NC device, and the like (not shown). If a dielectric breakdown occurs at this extremely narrow gap, an electrical discharge is generated between the electrode 1 and the workpiece 2 . This is shown at 701 in FIG. 48, and the output voltage in (b) serves as the electrical discharge voltage 702 . This electrical discharge voltage is approximately constant between 25 and 30 V. As soon as the electrical discharge occurs, the current begins to flow between the electrode 1 and the workpiece 2 and the unloaded voltage signal H3 is switched to a low level, as shown in (j). Therefore, as shown in (e), the first switching device 201 is turned off.

Since the output current indicated by 722 (d) flows through the fourth DC power supply E4, the fifth switching device 521 , the reactor 203 and the second switching device 211 , and the fourth DC power supply E4 is at the preset voltage (approximately 25 to 100 V), increases the output current on the leading edge of the output current as shown at 722 (d), which is determined by a difference voltage between the voltage of the fourth DC power supply E4 and the voltage across the electrode 1 and the workpiece 2 , and by the inductance value of the reactor (the Inductor) 203 . At 723 (d), where it is indicated that the voltage of the fourth DC power supply E4 has been increased, the change in the voltage of the fourth DC power supply E4 provides the leading edge of the output current with a desired slope. Since this leading edge of the output current has no hum, even an output current at a low level is not set to zero, so as to ensure stable electrical discharge machining.

Then, when the output current, that is, the current of the reactor 203 , has reached the current command value S1, the output of the first comparator 504 is switched to a high level at a point 724 , as shown in (i). Therefore, the output of the clock circuit 512 shown in (k) is switched to a low level, and the fifth switching device 521 is turned off as shown in (n). When the output of the first comparator 504 is switched to a low level, the output of the clock circuit 512 is only switched to a low level for a predetermined period of time, which is denoted by 705 (k).

During this predetermined period of time of the clock circuit 512 , i.e. during the period 705 when the fifth switching device 521 is switched off, the third switching device 501 shown in (n) is already switched on, as a result of which the current from the second DC power supply E2 to the location between the electrode 1 and the workpiece 2 is supplied by the third switching device 501 , the diode 502 and the reactor 203 . Since the voltage of the second DC power supply E2 is selected to be slightly lower than the electrical discharge voltage 702 , the output current gradually decreases as shown at 727 in (d). Then, at a point 726, the output of the clock circuit 512 is switched to a high level, whereby the fifth switching device 521 is switched on again to increase the current. This is repeated to cause the output current to follow the current command value S1 as shown in (d).

In this fifteenth embodiment, the output current increases with a certain slope, as shown at 722 and 723 , during the current rise so as not to generate hum, and the fifth switching device 521 remains on during this period so that no switching is required. After reaching the command value, the output current slowly decreases, as indicated at 727 , the decrease time denoted by 727 being the set time of the clock circuit 512 , denoted by 705 , the switching frequency being approximately the period of 705 , the hum components of the output current being low , and the switching frequency is low. This makes it possible to provide an energy supply device for an electric eroding machine which provides an output current waveform with a low hum when the inductance value of the reactor (the dro 13179 00070 552 001000280000000200012000285911306800040 0002004339191 00004 13060sselspule) 203 is low.

A sixteenth embodiment of the present invention will be described below with reference to FIGS. 49 to 51. Fig. 49 is a main circuit diagram of the sixteenth embodiment, in which a series connection of a fifth DC power supply E5, a sixth switching device 526 and a resistor 529 is connected in parallel to the working gap formed by the electrode 1 and the workpiece in the thirteenth embodiment. The second switching device 211 is provided with a diode 527 in order to prevent current flow in the wrong direction. The fifth DC power supply E5 should have a voltage which can supply the working gap with a voltage which is higher than that which the first DC power supply supplies. It is noted that 525 denotes a gate drive circuit of the switching device 526 .

Fig. 50 shows a control circuit of the gate driver circuits 503 , 206 , 215 and 525 of Fig. 49, wherein the AND state of a high voltage pulse signal A4 and the electrical discharge signal H1 is determined by an AND circuit 528 by the sixth Switching device 526 on / off under the control of the gate driver circuit 525 in the control circuit shown in FIG. 34 according to the thirteenth embodiment. It is noted that the other arrangement is identical to that of the control circuit shown in FIG. 34 according to the thirteenth embodiment, and therefore the description is not repeated here.

The timing charts and waveform diagrams shown in Fig. 51 show the operation of this sixteenth embodiment. In Fig. 51, (a) shows the electrical discharge signal H1, (b) the output voltage, (d) the output current, (e) the on / off state of the first switching device 201 , (f) the high voltage pulse signal, and (g) the On / off state of the sixth switching device 526 . When the electrical discharge signal H1 is turned on at a point 700 in FIG. 51 (a), the gate driver circuit 215 turns on the switching device 211 . Is turned on because at this time, the switching device 201 in FIG. 49, as shown in (e), the voltage of the first DC power supply E1 between the electrode 1 and the workpiece 2 is applied as an unloaded voltage, as indicated at 728 in (b).

The gap between the electrode 1 and the workpiece 2 is filled with the dielectric fluid, for example oil or water, and is controlled extremely precisely by a servomechanism, an NC device, and the like (not shown). If a dielectric breakdown occurs in this extremely small gap, an electrical discharge is generated between the electrode 1 and the workpiece 2 . However, sometimes the electric discharge does not easily occur by itself, which leads to an unstable state of machining by electric discharge.

To prevent this, if the electrical discharge does not occur within the time period 729 after the electrical discharge signal H1 is turned on, the high voltage pulse signal H4 is provided as shown in (f) at a point 730 to turn on the sixth switching device 526 (g). and thereby output the voltage of the fifth DC power supply E5. This voltage is shown in (b) at 731 . The actual voltage of the fifth DC power supply E5 is at least 150 to 300 V.

After the electrical discharge is generated, the high voltage pulse H4 is switched to a low level at a point 701 in order to switch off the sixth switching device 526 , whereby the voltage of the first DC power supply E1 is switched between the electrode 1 and the workpiece by the already switched on first switching device 201 and the output current increases. At the moment this electrical discharge occurs, a current flows in the resistor 529 , but the resistor 529 consumes practically no power because the sixth switching device 526 switches off immediately. Since the voltage of the DC power supply E5 of 150 to 300 V is higher than the voltage of the first DC power supply E1 of 80 V, the electrical discharge occurs reliably, which stabilizes the machining by an electrical discharge. It is noted that the other operations are the same as in the thirteenth embodiment and therefore will not be described again here.

Although the present sixteenth embodiment has been described using the device in which the series connection of the fifth DC power supply E5, the sixth switching device 526 and the resistor 529 are connected in parallel to the working gap which is formed by the electrode 1 and the workpiece 2 at the thirteenth embodiment, but the series connection of the fifth DC power supply E5, the sixth switching device 526 and the resistor 529 can also be connected in parallel to the working gap formed by the electrode 1 and the workpiece 2 in the fourteenth or fifteenth embodiment, for the same effects to evoke. Of course, in this case, the fifth DC power supply E5 is set to a voltage higher than that of the other DC power supplies E1, E2, E3 and E4.

A seventeenth embodiment of the present invention will be described below with reference to FIG. 52. Fig. 52 is a main circuit diagram of the seventeenth embodiment in which the second DC power supply E2, a sixth DC power supply E6, a seventh DC power supply E7, and an eighth DC power supply E8 are connected in series, one end of the series connection of the third switching device 501 and the diode with the Connection point of the second DC power supply E2 and the sixth DC power supply E6 is connected, and the other end of which is connected to the connection point of the reactor (the choke coil) 203 and the diode 202 . Furthermore, one end of the series connection of the fourth switching device 514 and the diode 515 is connected to the connection point of the sixth DC power supply E6 and the seventh DC power supply E7, and their other end is connected to the connection point of the reactor 203 and the diode 202 . Furthermore, one end of the first switching device 201 is connected to the connection point of the seventh DC power supply E7 and the eighth DC power supply E8, and its other end is connected to the connection point of the reactor 203 and the diode 202 . In addition, one end of the series connection of the sixth switching device 526 and the resistor 529 is connected to one end of the eighth DC power supply E8, and the other end thereof is connected to the electrode 1 (or the workpiece 2 ). Note that in Fig. 52, reference numeral 527 denotes a diode.

Because in the thirteenth, fourteenth and sixteenth Embodiment the relationship of the tension of the first, second, third and fifth direct current power supply E1, E2, E3, E5 to the electrical discharge voltage are such that E5 <E1 <E3 electrical discharge voltage E2 applies, the Voltage of the sixth DC power supply E6 as "E3-E2" are set to be the seventh DC power supply E7 as "E1-E6-E2", and the the eighth DC power supply E8 as "E5-E7-E6-E2 = E5-E1". In detail, the DC voltages are E2, E6, E7 and E8 20 to 30 V, 5 to 15 V, 40 to 50 V, or 70 to 230 V, so that the DC power used supplies have low voltages, and the Power supplies with high efficiency can be used can.

In the present embodiment, when the operation of the thirteenth embodiment is to be performed, the fourth switching device 514 and the sixth switching device 526 can be turned off, and the first switching device 201 and the third switching device 501 can be controlled on / off as in the thirteenth embodiment. If it is desired to operate as in the fourteenth embodiment, the sixth switching device 526 can be turned off and the first switching device 201 , the third switching device 501 and the fourth switching device 514 can be controlled on / off as in the fourteenth embodiment. On the other hand, when it is desired to carry out the operation of the sixteenth embodiment, the fourth switching device 514 can be turned off, and the first switching device 201 , the third switching device 501 and the sixth switching device 526 can be controlled on / off as in the sixteenth embodiment. Since the details of these operating procedures can be easily understood on the basis of the explanations already given of the operating procedures, this is not repeated here.

Referring to Fig. 53, an eighteenth embodiment of the present invention will be described below. Fig. 53 is a main circuit diagram of the eighteenth embodiment, and this embodiment is a combination of the fourteenth and sixteenth embodiments. Namely, it is the series connection of the third DC power supply E3, the fourth switching device 514 and the diode 515 in the fifteenth embodiment in parallel with the first diode 202 of the sixteenth embodiment.

The operation of this embodiment follows simply from the already described the operational sequence, and will therefore not repeated here.

The switchgear at the thirteenth to eighteenth Transistors used by any embodiment The device can be realized electrically can be turned on / off, and can by such  Switching devices such as MOSFETS, IGBTs and SITs are replaced, with the same effects.

The comparators, clock circuits, flip-flops, command values, AND circuits and inverters that are analog based as the Control circuits at the eighth to seventeenth Embodiment are provided by DPSs (digital Signal processors), microprocessors and the like Digital base to be replaced to have the same effects achieve.

It is apparent that the one described above present invention a power supply available provides a high energy supply efficiency has a desired electrical discharge current generated, has a high reaction speed, and extremely low current hum available. Especially the present invention provides a compact, low-cost energy supply provided, which ensures stable processing.

The entire disclosure of each foreign patent application, their foreign priority in the present application was so claimed by reference to the including this application as if it were completely included in it.

The present invention was based on at least one preferred embodiment to some extent specific details are described, however, will be discussed noted that the description of the preferred Embodiment took place only as an example, and that numerous changes in the details and arrangement of the Let components perform without the essence and scope deviate from the invention, which results from the entirety of present registration documents.

Claims (32)

1. Method for machining a workpiece using an energy supply and switches for supplying pulsed electrical energy to a working gap between an electrode and a workpiece in a dielectric, for machining the workpiece, characterized by the following steps:
Switching the switching devices on / off in a selectable cycle under the control of a current command value signal which corresponds to a selectable signal form of a current pulse which is to be supplied to the working gap;
Superimposing a current component to compensate for a current hum generated by switching at the time the current is supplied, the current component being superimposed on the selectable waveform of a current pulse to produce a resulting current; and
Supply of the resulting current to the working gap. 2. Process for machining a workpiece Claim 1, characterized in that the Current component by selectively switching several parallel resistors is provided. 3. A method for machining a workpiece according to claim 1, characterized in that the current component by selective actuation of an analog switching device in Row with at least one resistor available is provided.   4. A method for machining a workpiece according to claim 1, characterized in that further electrical Energy collected and intermittent the working gap is fed. 5. Energy supply device for an electrical eroding machine, for supplying electrical energy with a predetermined pulse shape to a working gap between an electrode and a workpiece, characterized by:
a first processing circuit having a power supply for generating a first current and supplying processing energy to the working gap, a first switching device and a first resistor, the power supply, the first switching device and the first resistor being connected in series;
current detector means for detecting the first current flowing in the first processing circuit;
a second processing circuit which has at least one series circuit which has at least one second circuit device and is connected in parallel with the first switching device and the first resistor in the first processing circuit in order to supply the working gap with a second current which corresponds to the first current from the first processing circuit is overlaid;
means for setting a current command value signal according to the waveform of a current pulse to be supplied to the working gap;
a first signal addition / subtraction device for processing and outputting a difference between at least a part of the current command value signal and a part of the output signal from the current detector device;
a first control device for outputting a signal to the first switching device in the first processing circuit in accordance with the output signal of the first signal addition / subtraction device;
second signal addition / subtraction means for processing and outputting a difference between the current command value signal and the output signal of the current detector means; and
a second control device for outputting a switching signal to one or more of the at least one second circuit devices in the second processing circuit in accordance with the output signal of the second signal addition / subtraction device. 6. Power supply device for an electrical EDM machine according to claim 5, characterized in that that the at least one second circuit device a series connection of a second switching device and has a second resistor. 7. Power supply device for an electrical EDM machine according to claim 5, characterized in that that the at least one second circuit device has a semiconductor amplifier. 8. Power supply device for an electrical eroding machine according to claim 5, characterized in that the first processing circuit comprises:
an electrical energy collector circuit comprising the first switching device, an inductor and a first diode connected in series to collect and intermittently supply electrical energy from the power supply;
a third switching device which is connected to supply the work gap with an output current from the collector circuit for electrical energy in order to supply the output current to the work gap in pulses; and
your second diode, connected to return a residual current to the electrical energy collector circuit that is generated in the working gap when the third switching device is turned off. 9. Power supply device for an electrical EDM machine according to claim 8, characterized in that that the second processing circuit at least one Has semiconductor amplifier that is parallel to Collector circuit for electrical energy in the first Machining circuit is switched to the working gap to supply with the second current. 10. Energy supply device for an electrical eroding machine for supplying pulsed electrical energy to a working gap between an electrode and a workpiece, which are provided in a dielectric, characterized by:
a first current source for supplying a first current to the working gap, the current being defined by a pulse shape and response speed;
current detector means for detecting the first current supplied from the first current source to the working gap;
a second current source connected in parallel with the first current source and operable to supply the working gap with a second current superimposed on the first current to form a resulting current, the second current source having a higher output current response speed has as the first power source;
means for setting a current command value signal according to the waveform of the current pulse to be supplied to the working gap;
arithmetic means for arithmetically modifying the current command value signal;
a first control device for providing the output signal of the arithmetic device as a current command value to the first current source; and
a second control device for outputting a difference between the current command value signal and an output signal of the current detector device as the current command for the second current source. 11. Power supply device for an electrical Eroding machine according to claim 10, characterized characterized in that the arithmetic means a Means for subtracting a predetermined value of the current command value.   12. Power supply device for an electrical Eroding machine according to claim 10, characterized characterized in that the arithmetic means a Means for multiplying a predetermined one Value that is greater than 0 and not greater than 1 with has the current command value. 13. Energy supply device for an electrical eroding machine for supplying a pulsed electrical current to a working gap between an electrode and a workpiece, characterized by:
a first current source for supplying a current pulse, defined by an output reaction rate, to the working gap;
current detector means for detecting a current supplied to the working gap from the first current source;
a second power source connected in parallel with the first power source and configured to supply the working gap with a current superimposed on the current from the first power source and configured to have a higher output current response rate than that first power source;
a third current source connected in parallel with the first current source and capable of supplying current in a direction opposite to the current supply direction of the second current source and having a higher output current response speed than the first current source;
means for setting a current command value signal according to the waveform of the current pulse to be supplied to the working gap;
a first controller for outputting the current command value signal to the first current source;
a second control device for outputting a first polarity difference between the current command value signal and a current signal, which is detected by the current detector device, as a current command for the second current source; and
a third control device for outputting a second polarity difference between the current command value signal and a current signal, which is detected by the current detector, as a current command for the third current source. 14. A method of controlling a power supply for an electric discharge machine having a constant current supply section provided with at least a first switching device and an output current turn-on / turn-off section having a second switching device for machining energy a working gap between an electrode and a Add workpiece that are provided in a dielectric, characterized by the following steps:
Setting the output current level and the output current hum of the constant current supply section;
Setting the addition result of the set output current level and the output current hum as the output current command signal of the constant current supply section, and comparing the output current command signal with the output current of the constant current supply section; and
Controlling the switching device of the constant current supply section in accordance with the result of the comparison step. 15. Energy supply device for an electrical eroding machine, characterized by:
a constant current supply section having at least a first switching device and an output current on / off switching section provided with a second switching device for supplying machining energy to a working gap between an electrode and a workpiece provided in a dielectric;
detector means for detecting the output current of the constant current supply section and for outputting a detection value;
output current level setting means for setting the value of the output current level of the constant current supply section;
ripple current setting means for setting the value of the output current ripple of the constant current supply section;
comparing means for comparing a set value obtained by adding the set value of the ripple current setting means to the set value of the output current level setting means with the detection value of the detector means; and
a device for switching on / off control of the first switching device in the constant current supply section according to the comparison of the comparison device. 16. Power supply device for an electrical Eroding machine according to claim 15, characterized characterized in that the ripple current setting device has a modulation device to the set signal frequency of the hum current setting value according to the setting value of the output current level to modulate the adjustment device. 17. Energy supply device for an electrical eroding machine, characterized by:
a constant current supply section having at least a first switching device and an output current on / off switching section provided with a second switching device for supplying machining energy to a working gap between an electrode and a workpiece provided in a dielectric;
detector means for detecting the output current of the constant current supply section;
output current level setting means for commanding the output current level of the constant current supply section;
a ripple current setting means for outputting a ripple current set value setting signal to set the ripple of the output current of the constant current supply section;
means for outputting a synchronization signal which is synchronized with the ripple current setting value signal of the ripple current setting device;
comparing means for comparing a set value obtained by adding the set value of the ripple current setting means to the set value of the output current level setting means with the detector value of the detector means;
a gate device for receiving the output signal of the comparison device and the synchronization signal in order to switch off noise which is generated when the first switching device is switched on / off; and
a device for switching on / off control of the first switching device in accordance with the output signal of the gate device. 18. Energy supply device for an electrical eroding machine, characterized by:
Constant current supply sections which have at least first switching devices and output current switch-on / off sections which are provided with second switching devices for supplying machining energy to a working gap between an electrode and a workpiece which are located in a dielectric;
a first constant current supply section;
a second constant current supply section;
first detector means for detecting the output current of the first constant current supply section;
a second detector means for detecting the output current of the second constant current supply section;
output current level setting means for setting the output current levels of the first and second constant current supply portions;
first ripple current setting means for setting the output current ripple of the first constant current supply section;
a second ripple current setting device for setting a set value which is 180 ° out of phase with the set value of the first ripple current setting device;
first comparing means for comparing a setting value obtained by adding the setting value of the first ripple current setting device to the setting value for the first and second constant current supply portions for the output current level setting device with the detector value of the first detector device;
second comparing means for comparing a setting value obtained by adding the setting value of the second ripple current setting means to the setting value for the first and second constant current supply portions for the output current level setting means with the detector value of the second detector means; and
a device for switching on / off control of the first switching devices in the first and second constant current supply sections in accordance with the comparison results of the first and second comparison devices. 19. Energy supply device for an electrical eroding machine, characterized by:
a constant current supply section having at least a first switching device and an output current on / off section provided with a second switching device for supplying machining energy to a working gap between an electrode and a workpiece which are in a dielectric;
detector means for detecting the output current of the constant current supply section;
output current level setting means for commanding the output current level of the constant current supply section;
comparison means for comparing the set value of the output current level setting means with the detector value of the detector means to output a signal that turns off the first switching device in the constant current supply section in accordance with the result of the comparison;
clock means for receiving the comparison output of the comparison means and outputting a signal that turns on the first switching device in the constant current supply section when a predetermined period of time passes after the comparison device outputs a signal that switches off the first switching device in the constant current supply section; and
a device for switching on / off control of the first switching device according to the output signal of the clock device. 20. A method for controlling a power supply for an electric discharge machine, which has a constant current supply section with a first direct current power supply, a first switching device, a diode and a choke coil, for supplying machining energy to a working gap between an electrode and a workpiece which is in a dielectric Fluid, characterized by the following steps:
Turning on / off the first switching device at desired intervals to supply the working gap with the current corresponding to a current command value signal from the first DC power supply; and
Adding a current to suppress the decrease in the output current that occurs in the switched-off period of the first switching device. 21. Power supply device for an electrical EDM machine, characterized by a constant current supply section that a first Has DC power supply, a first Switchgear connected to one pole of the first DC power supply is connected, a Choke coil in series with the first switching device is connected, a first diode, one end of which the other pole of the first DC power supply  is connected, and its other end with the Connection point of the first switching device and the Inductor is connected to an output current Power on / off section with a second Switching device is provided to a machining energy Working gap between one electrode and one Feed workpiece into a dielectric fluid are arranged, a series connection of a second DC power supply, which is a voltage has the working gap with a voltage can supply that is essentially equal to one discharge voltage or lower third switching device and a second diode connected in parallel to the first diode of the constant current supply section are switched. 22. Power supply device for an electrical eroding machine according to claim 21, characterized by:
a first comparison device for comparing a current command value with a current detector value of the inductor and for generating an inverted output signal;
a second comparison device for comparing an overcurrent command value with the current detector value of the inductor;
a clock device whose input is connected to the output of the first comparison device; and
a first state memory device which has a reset input which is connected to the output of the second comparison device and a set input which is connected to the inverted output signal of the first comparison device;
wherein the first switching device is controlled by the product of the output signal of the clock device, the output signal of the first state storage device and an electrical discharge signal, the third switching device is controlled by the product of the output signal of the first state storage device and the electrical discharge signal, and the second switching device is controlled by the electrical discharge signal is controlled; and
wherein the first switching device is turned off for a period of time set by the timing means when the current detector value of the inductor exceeds the current command value, and the third switching device is also turned off when the current detector value of the inductor exceeds the overcurrent command value. 23. A method of controlling or regulating a power supply for an electric discharge machine, which has a constant current supply section having a first direct current power supply, a first switching device, a diode and a choke coil, for supplying machining energy to a working gap between an electrode and a workpiece, the are arranged in a dielectric fluid, characterized by the following steps:
Supplying the working gap from a second DC power supply with a voltage which is higher than an electrical discharge voltage and lower than a voltage supplied by the first DC power supply, and switching off the first switching device when an output current is at a predetermined current level; and
Switching on / off a switching device that differs from the first switching device at desired intervals in order to control the current from the second DC power supply. 24. Power supply device for an electrical EDM machine, characterized by a constant current supply section that a first Has DC power supply, a first Switchgear connected to one pole of the first DC power supply is connected, a Choke coil in series with the first switching device is connected, and a diode, one end of which is connected to the other pole of the first DC power supply is connected, and the other end to the Connection point of the first switching device and the Inductor is connected, and one Output current turn-on / turn-off section using a second switching device is provided to Machining energy a working gap between one Feed electrode and a workpiece in one dielectric fluid are arranged, one Series connection of a second DC power supply which is a voltage which has the working gap with a voltage can supply that is essentially equal to one discharge voltage or lower, a third switching device and a diode in parallel Diode of the constant current supply section switched and a series connection of a third DC power supply which is a voltage having to supply the working gap with a Voltage sufficient that is higher than the electrical Discharge voltage is, and lower than a voltage, that of the first DC power supply is delivered, a fourth switching device and one  Diode parallel to the diode of the Constant current supply section is switched. 25. Power supply device for an electric eroding machine according to claim 24, characterized in that a first comparison device is further provided to compare a current command value with a current detector value of the inductor and to generate an inverted output signal, a second comparison device for comparing an overcurrent command value with the current detector value Choke coil, a clock device, the input terminal of which is connected to the output of the first comparison device, and a first state memory device, which has a reset input terminal which is connected to the output of the second comparison device, and a set input terminal which is connected to the inverted output of the first comparison device is connected;
wherein the fourth switching device is controlled by the output signal of the clock device and a current increase signal, the output signal of the first state storage device and an electrical discharge signal;
the first switching device is controlled by the product of the output signal of the clock device, the current rise signal and the electrical discharge signal;
the third switching device is controlled by the product of the output signal of the first state storage device and the electrical discharge signal;
the second switching device is controlled by the electrical discharge signal; and
furthermore, the fourth switching device is switched off for a period of time which is set in the clock device when the current detector value of the inductor exceeds the current command value, and the third and fourth switching devices are also switched off when the current detector value of the inductor exceeds the overcurrent command value. 26. Power supply device for an electrical eroding machine, characterized by a constant current supply section which has a first direct current energy supply, a first switching device which is connected to a pole of the first direct current energy supply, a choke coil which is connected in series with the first switching device, and a diode, one end of which is connected to the other pole of the first DC power supply, and the other end of which is connected to the connection point of the first switching device and the choke coil, and has an output current turn-on / turn-off section provided with a second switching device to supply machining power to a working gap between an electrode and a workpiece, which are arranged in a dielectric fluid, wherein the device further comprises:
a series circuit of a second DC power supply which has a voltage which can supply to the working gap a voltage which is substantially equal to or lower than an electrical discharge voltage,
a third switching device and a diode connected in parallel to the diode of the constant current supply section; and
a series circuit of a fourth DC power supply, which can change a voltage, a fourth switching device and a diode, in parallel with the diode of the constant current supply section. 27. Energy supply device for an electrical eroding machine according to claim 26, characterized in that the following are further provided:
a first comparison device for comparing a current command value with a current detector value of the inductor;
a second comparison device for comparing an overcurrent command value with the current detector value of the inductor;
clock means having an input terminal connected to the output of the first comparator; and
a first state memory device which has a reset input terminal which is connected to the output of the second comparison device and a set input terminal which is connected to the inverted output of the first comparison device;
wherein the fourth switching device is controlled by the product of the output signal of the clock device, the output signal of the first state storage device and an electrical discharge signal, the first switching device is controlled by the product of an unloaded voltage signal and the electrical discharge signal, the third switching device by the product of the output signal of the first State storage device and the electrical discharge signal is controlled, the second switching device is controlled by the electrical discharge signal, and further the fourth switching device is turned off for a period of time set by the timing device when the current detector value of the inductor exceeds the current command value, and the third and fourth Switching device can also be switched off if the current detector value of the inductor exceeds the overcurrent command value. 28. Energy supply device for an electrical eroding machine, characterized by:
a constant current supply section provided with a first DC power supply, a first switching device which is connected to a pole of the first DC power supply, a choke coil which is connected in series with the first switching device, and a diode whose one end is connected to the other pole of the first DC power supply is connected, and the other end is connected to the connection point of the first switching device and the reactor, and one
Output current on / off section provided with a second switching device for supplying machining energy to a working gap between an electrode and a workpiece arranged in a dielectric fluid, the device further comprising:
a series circuit from a second DC power supply having a voltage that can supply the working gap with a voltage that is substantially equal to or lower than an electrical discharge voltage;
a third switching device and a diode, connected in parallel to the diode of the constant current supply section; and
a series circuit of a third DC power supply, which has a voltage that can supply the working gap with a voltage that is higher than a voltage supplied by the first DC power supply, a fourth switching device and a resistor, which are connected in parallel to the working gap . 29. Power supply device for an electrical eroding machine according to claim 28, characterized in that the following are further provided:
a first comparison device for comparing a current command value with a current detector value of the inductor;
a second comparison device for comparing an overcurrent command value with the current detector value of the inductor;
a clock device whose input terminal is connected to the output of the first comparison device; and
a first state memory device which has a reset input terminal which is connected to the output of the second comparison device, and a set input terminal which is connected to the inverted signal of the output of the first comparison device;
wherein the first switching device is controlled by the product of the output signal of the clock device, the output signal of the first state storage device and an electrical discharge signal, the third switching device is controlled by the product of the output signal of the first state storage device and the electrical discharge signal, the second switching device is controlled by the electrical discharge signal is switched off, and furthermore the first switching device is switched off for a period of time which is set by the clock device if the current detector value of the inductor exceeds the current command value, and the third switching device is also switched off if the current detector value of the inductor exceeds the overcurrent command value, and that fourth switching device is turned on by the product of a high voltage pulse signal and the electrical discharge signal. 30. Power supply device for an electrical eroding machine, characterized by a constant current supply section which has a first direct current energy supply, a first switching device which is connected to a pole of the first direct current energy supply, a choke coil which is connected in series with the first switching device, and a diode, one end of which is connected to the other pole of the first DC power supply, and the other end of which is connected to the connection point of the first switching device and the choke coil, and an output current turn-on / turn-off section provided with a second switching device to supply machining power to a working gap between an electrode and a workpiece, which are arranged in a dielectric fluid, wherein
the first DC power supply includes a plurality of DC power supplies that have predetermined voltages and are connected in series with each other, one of the plurality of DC power supplies being a DC power supply that has a voltage that can supply the working gap with a voltage that is substantially equal to or less than an electrical discharge voltage is;
and one end of a series connection of a third switching device and a diode is connected to the connection point of the second DC power supply and a third DC power supply, and its other end is connected to the connection point of the first switching device and the inductor. 31. Power supply device for an electrical Eroding machine according to claim 30, characterized characterized that one of the several DC power supplies as a fourth DC power supply is used, the one Has tension that the working gap with a Can supply voltage that is higher than the electrical Discharge voltage and is lower than a voltage those composed of the first DC power supply supplied with the second DC power supply with one end of a series connection of a third  Switchgear and a diode with the connection point the third DC power supply and one fifth DC power supply, unlike the second DC power supply, is connected, and their other end to the connection point of the first Switching device and the choke coil is connected. 32. Power supply device for an electrical Eroding machine according to claim 30, characterized characterized that a sixth DC power supply to the first DC power supply is connected that a End of series connection of a fifth switching device and a resistor with an end of the sixth DC power supply is connected, and her other end either to the electrode or to the Workpiece is connected.