DE2361715C3 - Synchronous rectifier and amplifier - Google Patents

Description

V _

20th

and a negative feedback capacitor (Q, see Fig. 6) may exist, with exact mutual Compensation of the amplifier input currents must be dispensed with.

The invention relates to an arrangement for synchronous alignment with the correct sign and at the same time with adjustable amplification of an alternating voltage, preferably in combination with a previous one Separation of this alternating voltage from a superimposed direct voltage.

This task exists in many measuring devices, e.g. B. J5 for synchronized AC voltage amplifiers (so-called> ■ 'ock' ::% <amplifiers) or for the rectification of the modulated signal in DC voltage chopper amplifiers. In the latter case, the alternating voltage to be rectified is usually small and approximately w rectangular.

In addition to self-evident, generally desired properties such as simple and reliable construction, etc., the main requirement of such synchronous rectifiers is that the AC input voltage in a; one half of the period is transmitted proportionally, regardless of its amplitude, without a sign reversal, in the other half of the period with the same proportionality factor but with reversed sign -Switching period half to the other result in increased residual alternating voltage at the output, which can subsequently be sifted out as much as required, usually not without disadvantages - for example for the cut-off frequency of the device. The requirement for a proportionality that is as independent as possible of the input amplitude down to amplitude zero *> o means that the rectifier itself should not generate any voltage at its output when the input signal is zero. Above all, the changes in such an interference voltage over time or temperature are harmful. * '

There are numerous solutions to this problem, mostly quite complex and / ixjer imperfect, z. B. Ring modulators and two-way transistor chopper circuits with a downstream amplifier. In the German Disclosure Document 21 50 888 a rectifier is already proposed, which is free of residual voltage Switch, e.g. B, a field effect transistor, and a negative feedback operational amplifier However, this circuit has the following disadvantages:

1. With this circuit it is impermissible to divert the input AC voltage from the superimposed one DC voltage, which occurs almost always in electronic AC voltage amplifiers, by a coupling capacitor - as the simplest possibility - in the input of this rectifier circuit to separate. Namely, the rectifier circuit would remove this capacitor when it occurs an input AC voltage according to the present time constant more or less charge quickly by an amount proportional to the alternating voltage amplitude. The one at the entrance the DC voltage generated by the rectifier circuit is chopped up and overtaken z. B. in the case a rectangular input alternating voltage a superimposed alternating voltage of the same at the output in addition to the desired direct voltage Amplitude, i.e. one resulting between zero in one half of the period and that double the value of the alternating voltage amplitude in the other half of the period pulsating voltage (Half-wave rectification). The obvious way out, behind the coupling condenser for discharge Switching a resistor to ground of the direct current would also have been difficult Disadvantages: In principle, this measure cannot eliminate, but only alleviate the cause disturbing charging effect. In order for this improvement to be sufficient in practice, the Leak resistance very small compared to the not arbitrary with regard to the amplifier interference currents large operational amplifier wiring resistances be. This means that a very large (expensive) coupling capacitor is required, and so is the output of the AC voltage preamplifier becomes very strong due to the low-ohmic leakage resistance burdened

2. The ratio of the two connected to the inverting input of the op amp Resistances must have a certain value close to 1 - with an ideal switch exactly 1 - so that a full-wave rectification is achieved with the same gain factor for both halves of the period will. This amplification factor is then 1, i.e. the circuit does not amplify the Most applications require amplification of the rectified voltage must by a further downstream amplifiers can be obtained. First, ι increases the effort, secondly, the incorrect voltage (offset voltage) of the post-amplifier and its drift add up not weakened to the corresponding values of the rectifier operational amplifier.

3. A compensation of the operational amplifier input currents (through the same source resistances at both operational amplifier inputs) of the rectifier operational amplifier in both halves of the period can not. A disturbing residual AC voltage at the output of the rectifier circuit even if there is no or small, precisely square-wave AC input voltage is the consequence.

The object of the present invention is to be achieved while avoiding those just mentioned Disadvantages I to 3 with the lowest possible technical effort a phase-controlled rectification and Build amplification circuit.

The basic idea of the present invention is to provide an operational amplifier with a Network of resistors and one or more electronic or mechanical switches so too switch that the operational amplifier not only alternately inverting with the same gain factor (in one half of the period) and non-inverting (in the other half of the period) and thus the Input signal rectified ", but also amplified at the same time if necessary (gain factor V by dimensioning selectable between 1 and high, limited by the operational amplifier properties Values), the ohmic load of the circuit input in both period halves keeps the same (which is a capacitive coupling of the input allowed) and the most complete possible compensation of both of them Input currents (same ohmic load on both inputs of the operational amplifier) to the to avoid three aforementioned disadvantages of the known circuit. Switching between the inverting and the non-inverting operation of the operational amplifier with negative feedback in both working phases can be done in various ways and through mechanical or preferably electronic switches. 7. B. field-effect transistors controlled with square-wave voltages.

A simple and advantageous implementation possibility is shown in FIG. 1. The input voltage u ' _e is applied to the rectifier circuit via the capacitor C to separate the direct voltage components. This is controlled by a switch with the forward resistance r, for example in the form of a field effect transistor, the gate of which is connected to a suitably selected square-wave voltage of the desired phase position relative to the input signal u. If the switch is open (resistance can be regarded as infinitely large, even with

the operational amplifier OP amplifies in an inverting manner, ie reversing the sign, with the factor V = ■ - * ».

Both inputs of the operational amplifier OPI are connected to ground, the inverting one is virtual. The resistors R. and R- are almost currentless and have no significant influence on the circuit. Only the correct compensation of the OP input currents takes place with the help of these resistors. If the switch is closed, the non-inverting OP input is almost at the input voltage u * via the voltage divider from around R ^ with r <R-. The negative feedback of the now non-inverting OP has the same voltage , so that now the resistance R _e except for the small voltage drop at r than almost

This operation of an operational amplifier is in the form of a non-amplifying pure rectifier (with a different Wiring) already from the aforementioned German Offenlegungsschrift No. 21 50 888 known that it is also with the

1 ^ Hf

alternating inverting and non-inverting operational amplifier operation is, however, not specified there and evidently not recognized in such an abstract way.

electroless, only the differential input resistance of the OP decreasing Shut between the two OP inputs played a minor role. A longer calculation shows, first, that the ohmic load on the capacitor C is exactly the same in both halves of the period with an ideal OP , that is, that this capacitor does not experience any direct current charging due to the change in the amplitude of the alternating voltage signal u ₍

R. ~ R ₁ .

is chosen, regardless of the other resistors R. R _g and especially from r.

· -. Secondly, this calculation shows that, even if a finite switch on-state resistance r is taken into account in the case of an ideal operational amplifier, the requirement for selectable, for both period halves of the same gain V of different signs

: ■ 'is exactly fulfilled. when dimensioning:

and

D.

I -

2r

Third, it shows. that with this dimensioning, together with R * = R _e , the third requirement for correct input current compensation of the OP is automatically in a good, practically sufficient approximation. that is, according to the same ohmic source resistances at its two inputs, is fulfilled in both halves of the period. The relative deviation of the two source resistances from their mean value is only about ^, i.e. z. B. 1% at r = 100 Ω and R _r =

what in view of the usual at least as high deviations of the two OP input currents from each other means no significant deterioration.

in Kaenn / lo

after reinforcement is needed, the resistance /? _ as Special case chosen to be infinitely large, d. H. be omitted. The above formulas result from the corresponding Transformation for the negative feedback resistance:

R.

and for reinforcement:
1

V =

Ir R.

i.e. a value closer to 1 (slight signal weakening during rectification).

Another, on the whole, less favorable execution

bO possibility shows F i g. 2. This is similar to the circuit in the above-mentioned laid- open specification 21 50 888, but differs in function from this by the selectable gain> 1 and by the same ohmic load on the input (of the coupling capacitor Q in both halves of the period. In contrast to the circuit according to F i g. 1, however, does not mean that the ohmic source resistances of the two OP inputs to the input current

compensation possible for both halves of the period. Therefore, depending on the quality of the OP, the lowest possible wiring resistances must be selected, provided that the ratio of these resistances to the exemplary and environment-dependent switch resistances η and ο (resistance of one field effect transistor when switched on) allows this.

The mode of operation of this circuit according to FIG. 2 is that of FIG. I explained similarly, can therefore be explained here in abbreviated form. The inverting working phase is when the switch 1 is switched on with the on -state resistance n. Both OP inputs then have almost ground potential except for the fraction n / (Ri + n) of the input voltage u, s from the voltage divider consisting of R ₁ and r will. R is again almost no current and is of minor importance in this phase. The gain V is due to the voltage drop at Γ: here somewhat smaller than Κ, ν / ί ,. (Gen. iue formula see below). In the non-inverting phase switch 1 is open, so that the input voltage (/, · apart from the small voltage drop of the OP input current at / ?, at the non-inverting OP input. The gain is

The resistors R, and / ?, are almost currentless and wm of subordinate importance in this work phase. The switch 2, which is switched synchronously but out of phase with switch I, with the on-resistance r- is closed in this phase and, by connecting the correctly dimensioned bleeder resistor R, between the ground and the output side of the coupling capacitor Γ ensures that it is connected to the in both working phases (halves of the period) same ohmic load is connected. So not inadmissibly charged DC-like dependency ·? of the amplitude of the AC input voltage u, -Without Rq , the coupling capacitor C would be almost currentless in this non-inverting phase, while

Zero) in the inverting phase the capacitor current is approximately equal to the quotient of u .. and the parallel connection of R, - and R,. in order to then decay to zero in the subsequent inverting phases until the capacitor C is charged on the output side to the arithmetic mean value of the alternating voltage component of u ', -. taken over the half of the period belonging to the inverting phase. This DC voltage would be converted into a harmful output AC voltage by the rectifier circuit, which works as a chopper. The two requirements for the same gain V and the same ohmic load on the coupling capacitor C in both half-periods are assumed, ideally OP. exactly fulfilled with the following dimensions:

R. = (I - Il R

R. =

R ₁ , =

R ₁

R _r - Zr ₁

R _r - R _r

R ₁₁ - R-

Especially when using the rectifier circuit according to FIG. I or 2 as a demodulator in DC voltage chopper amplifiers, a mixed proportional-integrating negative feedback of the rectifier operational amplifier OP can be a significant advantage, because this makes the false voltages and drifts of subsequent DC voltage amplifier modules practically completely harmless - without chopper residues and similar higher-frequency input interference voltages are unnecessarily amplified - and because the loop gain of the chopper amplifier for low frequencies and especially for DC voltage can be brought to extremely high values without additional effort, in favor of linearity and, with conventional electrometer negative feedback, also to the detriment of the input resistance of the chopper amplifier. In order to achieve this mixed proportional-integrating effect, it is sufficient to use the negative feedback resistance A \. (see Fig. I and 2) a capacitor C \ - to be connected in series, so R _f by the arrangement according to F i g. Replace 3 / u. The proportional part of the gain remains the same in both half-periods, the constant of integration differs somewhat (in the inverting phase the higher current = JJ 'charges the negative feedback con-

dcnsator C _{f on} faster than the smaller current = _R 'in

the non-inverting phase.) This effect is harmless and results in a constant square wave signal u, - no additional output square wave voltage. instead, instead of a linear increase in u, a kinked, but ready-to-jump output voltage, made up of pieces with different slopes, that is, a certain function that cannot be differentiated everywhere. Fig. 4 shows an example. In the case of the circuit according to FIG. I must also accept that the OP current compensation is less accurate when a negative feedback capacitor Cf is interposed, because the direct current source resistance of the inverting input increases from approximately RR? almost to Hpn Wpri ß nhnp rlaR sirh Her Oiiellharz of the non-inverting input changes. If a greater limitation of the DC voltage amplification is desired than that brought about by the finite open amplification of the rectifier operational amplifier OP , a (relatively high-resistance) resistor R _n according to FIG. 5 parallel to Q or, according to FIG. 6, parallel to the series connection of Q and a resistor R \ - . (In this case, R _f R _r = R _f must be dimensioned.) If there is no external DC negative feedback. with integrating sound, such a parallel resistor R _: or another direct voltage negative feedback path must be provided.

The advantages that can be achieved with the invention are in particular the possibility of using simple means to achieve selectable amplification and / or integration and, at the same time, synchronous rectification with simple capacitive separation of the input AC voltage from a DC voltage superimposed on it, and in the case of an embodiment according to FIG. 1 largely compensates for errors caused by input currents of the operational amplifier used, regardless of the model; to be able to.

For this purpose 2 sheets of drawings

Claims (1)

R_ = - 1 - It R. ok Patent claims: 1, synchronous rectifier using a switch, preferably a field effect transistor and a negative feedback operational amplifier, which is operated alternately (preferably during one half of the switching period) inverting and (in the other half of the period) non-inverting with the same gain factor, characterized in that this Gain factor - even beyond 1 - can be set between slightly below 1 and high values by dimensioning, so that the synchronous rectifier can work as an amplifier at the same time, so that the ohmic load on the circuit input can be set to the same size in both switching phases by suitable dimensioning (enables capacitive Coupling) and that the two operational amplifier input currents can be largely compensated for by almost the same ohmic load on the two operational amplifier inputs, achieved by connecting the inverting input of the Operat ion amplifier (OP, see Fig. 1) via a first, technologically selectable resistor (Re) with the circuit input, via a second resistor (R _g ) with the output, whereby the gain factor K = - = - is determined and ^w via a third resistance stand (R-) with the SigB'I reference zero point ("earth") and by connecting the non-inverting input of the operational amplifier (OP) via a fourth resistor (R ₊ ), which with js consideration of the above-mentioned equality of the ohmic Loads as exactly as possible that should be chosen equal to the first resistance (Re) , with »E. Depending on and via a switch controlled synchronously with the measurement signal (e.g. field effect transistor, the smallest possible residual resistance r when switched on) with the circuit input, the third resistance being exactly or approximately according to the relationship 40 or when using the appropriate division R + = R _t ( see above) according to t \ _ - IT "" is measured. 2. Synchronous rectifier according to claim 1, characterized in that, as a special case, the third resistor (R =) is omitted (/ ?, - «, connection interrupted), so that the value for the second resistor should be aimed for (approximation: R ₀ = R _p ) and gain the value V = assumes just below 1 (slight weakening of the signal). 3. Synchronous rectifier using, inter alia, a switch, preferably a field effect transistor, and a negative feedback operational amplifier, which is operated alternately (preferably during half of the switching period) inverting and (in the other half of the period) non-inverting with the same gain factor, characterized in that, that this gain factor V - even beyond 1 - can be set between 1 and high values by dimensioning, so that the synchronous rectifier can be operated as an amplifier at the same time and that the ohmic load on the circuit input is the same in both switching phases through suitable dimensioning can be set (enables capacitive coupling), achieved by connecting the inverting input of the operational amplifier {OP, see FIG. 2) via a first resistor (Rc) to the circuit input, via a second resistor (R _g ) to the output and via a third resistor (R-) to the signal reference zero point (»earth«), furthermore by connecting the non- inverting input of the operational amplifier (OP) via a fourth resistor (R _r ) to the circuit input and via a first switch controlled synchronously with the measurement signal (e.g. field effect transistor, the smallest possible residual resistance η when switched on) to »earth« and by connecting the Circuit input via a fifth resistor (Rq) in series with a second switch (e.g. field effect transistor, residual resistance r ₂ if possible), which is controlled synchronously in antiphase (always one switch off, the other on) with "earth", whereby R and R, 'm technologically reasonable limits can be freely chosen, but the relationships + r, R. R ₁ + R_ • κ, - should be adhered to as precisely as possible. 4. synchronous rectifier according to claim I to 3, characterized in that instead of the pure proportional gain a mixed proportional-integrating transfer function is introduced while increasing the DC voltage gain almost to the value of the gain of the non-negative feedback operational amplifier (OP) by the ohmic The negative feedback resistor (R ₁ ,) is replaced by the series connection of a resistor of the same size (R _f ) with a negative feedback capacitor (Cf) (see Fig. 3), whereby exact mutual compensation of the amplifier input currents must be dispensed with. 2361 7 \ 5 5, synchronous rectifier according to claim 1 to 3, characterized in that the proportional gain is replaced by a mixed proportional-integrating transfer function, increasing the DC voltage gain to an adjustable higher value, but maintaining DC negative feedback from the output to the inverting input of the operational amplifier (OP), achieved by replacing the ohmic negative feedback resistor (R _g ) with ι ο a circuit that either consists of the series connection of a resistor of the same size (R _e ) with the parallel connection of a negative feedback capacitor (Cg) and a parallel resistor (R _p ) ( s, Fig. 5) or from the parallel connection of a parallel resistor (R _p ) with the series connection of a negative feedback resistor