DE1937412B2 - BALLISTICS CALCULATOR FOR CALCULATING FIRE CONTROL SIGNALS - Google Patents

Description

The invention relates to a ballistic calculator for the calculation of fire control signals depending on the target distance and non-standard conditions, which are characteristic of the deviation of the environmental conditions and the ammunition properties from standard values, with a distance multiplier to modify the distance data as a function of the non-standard conditions and to generate a normalized one Range signal, a time-of-flight function generator and an elevation angle function generator for multiplying the normalized distance signal with non-linear functions of the normalized distance and a time-of-flight multiplier and an elevation angle multiplier to modify the output signals of the time-of-flight or the elevation angle function generator as a function of non-standard conditions and to generate a time-of-flight signal and a ballistic Elevation angle signal.

It is known, for example, from US Pat. No. 3,373,263 for generating fire control signals To determine target distance, target direction and target speed and a ballistics calculator to supply the data for the azimuth angle, the elevation angle and the distance of the target. In addition, further data can be entered into such a computer partly automatically and partly by hand that are important for solving the ballistic equations. So the ballistics calculator can solve the ballistic equations on the basis of this data, he needs a multitude of Function generators. Although it is also known through a modification of the entered into the computer Target range for different types of guns to get appropriate output signals, but acts this is only a rough approximation that is not acceptable if it is the one from the ballistics calculator The data provided lead to a very high probability of being hit with the first shot should.

Compared to computers of this type, which require a large number of function generators, the ballistics calculator has of the type mentioned above, which is the subject of an earlier patent application P 19 29 300.6-15 forms, the advantage that it has a significantly simplified structure because it is because of the standardization of the Distance signal as a function of non-standard conditions with only two permanently programmed function generators gets by without the resulting deviations from the exact solution of the ballistic equations would be so large that they significantly affect the accuracy of the hit would.

The invention is based on the object of the ballistic computer according to the earlier patent application to develop that the hit probability, by taking into account a variety of influencing factors is further increased without the computer complicated circuit arrangements and in particular required a large number of additional function generators.

This object is achieved according to the invention in that at least one of the multiplication devices an ammunition type multiplier and one connected in series with the ammunition type multiplier Correction circuit includes, of which the type of ammunition multiplier with the signal supplied to it a transfer function characteristic of the type of ammunition multiplied and the correction circuit a multiplier that fed to it Signal multiplied by a function of the deviations of the environmental conditions from the standard values, and a summing circuit for combining the output signals of the multiplier with it supplied input signal includes.

The ballistic calculator according to the invention makes use of a special type of approximate solution for the ballistic equations which deviates from the approximations customary up to now. It goes without saying that the ballistic equations can be solved for a set of fixed values for the variables exactly according to the quantities of interest, namely the attachment and lead angles. However, since all variables can change within a certain range and there is a different solution for each set of variables, a fire control computer would be extremely large if an exact solution of the equations were to be provided for even approximately all possible combination values. It is therefore more expedient to start from a solution of the ballistic equation for certain standard values and to take into account deviations of the variables from these standard values by means of approximations. A practical approximation is a Taylor series expansion restricted to the linear terms. Such a linear approximation gives the Taylor series F (u ₀ , V ₀ , w ₀ ) + F '(11, v ₀ , W ₀ ) (uu ₀ ) + for a function of several variables F (u, v, w) F '{v, u ₀ , W ₀ ) (v - V ₀ ) + F' (w, w ₀ , V ₀ ) (w - W ₉ ). A

to calculate such a series of well-trained computers would have to be able to use the function F (M ₀ , v ₀ , w ₀ ) for standard conditions and the partial derivatives F ' (m, v ₀ , w ₀ ), F' ( v, u ₀ , W ₀ ) and F ' (w, M ₀ , ü ₀ ) separately, each of the partial derivatives with the corresponding change in their variables, i.e. with (m - M ₀ ) or (v - V ₀ ) or (w - w ₀ ) and then add the four terms obtained in this way. Accordingly, it was necessary to create function generators for each of the partial derivatives, to provide multipliers to multiply these partial derivatives with the changes in the variables and finally to add the products of the partial derivatives and the changes in the variables to the function. A further disadvantage of this method is that a change in one variable has no influence on the effect which is caused by the change in another variable.

A Taylor series expansion was used in the construction of the ballistics computer according to the invention assumed, but it was assumed that the equations to be solved are in the

Environment of the standard values around an exponential function of the general form

acts. The first derivative of such an expon - broken and furthermore (u - M ₀ ) denoted by Au,

tial function is

r in) = - f (u).

so you get

(«) = F {« o) + f '(uo) Ju.

If now again f (u) is developed as a Taylor series. If the above given value for f '(u) is now

unwinds and takes off the row after the first link, so the result

(«) = F (" o) + -J (U ₀ ) Au = / (U ₀ ) (l + k ^). The approximation is then obtained for the function of several variables F (u, v, w) given above

(m, v, w) = F (M ₀ , V ₀ , W ₀ ) (l + k -

\ U

Au. Av Aw \

'1 h m ).

ν w J

With this approximation, non-linear functions no longer occur and the above can be used Also write the equation as follows

F (u, v, W) = F (U ₀ , V ₀ , W ₀ ) (l + kf) (l + / f) (l +

This form of approximation is particularly well suited for a calculation because the signal characterizing the function f (u ₀ , v ₀ , w ₀ ) can be passed through several multipliers one after the other, which each perform the multiplication with the factors formed by the expressions in brackets. This results in a considerable simplification of the structure of the ballistics computer. In addition, the approximation is also improved because the multiplication of the expressions in brackets results in mixed products, i.e. members of a higher order, which guarantee a better approximation than would only be possible with the linear members. The correction circuit of the ballistic computer according to the invention provides for a multiplication with terms of the form 1 + k (Au / u).

The invention is described below with reference to the exemplary embodiment shown in the drawing described and explained in more detail. Show it

IA and IB are block diagrams of a ballistics computer according to the invention in connection with a resolver connection,

F i g. 2 the circuit diagram of an adjustable ballistic term multiplier, as it is in the computer according to FIGS. 1A and 1B are used,

F i g. 3 shows the schematic circuit diagram of a function generator as it is in the arithmetic circuit according to FIG F i g. 1A is used,

F i g. 4 is a diagram of a signal that is generated by the function generator according to FIG. 3 is generated,

F i g. 5 shows the circuit diagram of a sequence and clamping circuit as it is in the arithmetic circuit according to FIG. 1B Is used, and

F i g. 6 shows the circuit diagram of an analog switch as it appears in the arithmetic circuit according to FIG. 1B use finds.

In the case of the FIG. 1A and 1B illustrated embodiment of a ballistic computer, distance information R sin ω t and information about deviations from standard conditions are entered into a computer, which generates an attachment angle signal ε, a time-of-flight signal t _f , a ballistic drift signal η ₀ and a wind deflection angle η _κ according to ballistic equations which are then used to generate fire control signals such as elevation E and deflection D.

In addition to the distance, the ballistic properties of the projectile and the ambient conditions must also be known in order to be able to use the ballistic Equations to generate fire control signals. For example, it is necessary to see the effects of ballistic conditions such as mass, initial speed, shape, size and spin the projectile as well as environmental conditions such as the air density, the air temperature, the air pressure, the cross wind, the propellant charge temperature, the angular speed of a pivoting movement etc. to know.

Since some of the ballistic properties change with different projectiles or with different ammunition, the resulting signals, such as the attachment angle signal ε, the time-of-flight signal t _f , the ballistic drift signal η ₀ and the wind deflection angle η _{κ will} also change for each projectile. It was found that the non-linear equations for ballistic flight, from which the angle of attack ε ₀ and the flight time t _f0 for standard conditions are derived, for a large number of projectiles from a first adjustable multiplier 954, the range signal R with an individual ballistic term multiplied or processed for each individual projectile, which contains an individual normalization transfer function I / R " ₀ for generating a normalized distance signal R / R" _o , second function-generating circuits 956 and 958 in two channels, the normalized distance signal R / R " after Process a single specific transfer function, which applies to the ballistics of a plurality of projectile types, to generate the non-linear function signals f _e (R / R _n ) and f _t (RfR _n ), which relate to the selected projectile type, and third variable multipliers 960 and 962 for multiplying or processing the non-linear Fu nction signals with individual second ballistic terms or normalization

canceling transfer functions ε _π0 and t " ₀ , which are assigned to the selected storey, can be realized. It can be stated that the ballistic terms or transfer functions R _n , ε _η0 and t _n0 can be considered constant for a set of standard conditions.

For example, the approximation equation for the attachment angle s ₀ under standard conditions, which is the angle at which a projectile device, such as a cannon, must be raised above the line of sight to the target,

«B =

mg

2RKJm

2K ₀ V ₁ I.

oo

( 2RKo / m V

U)

The equation for the flight time t _{f0 of} the projectile under standard conditions is

tfO =

K ₀ V ₀ ,

(e

RKJm

(2)

In these equations is

R = distance,

K = pd ² K _D ,

K _D = drag coefficient,

p = air density,

d = bullet diameter,

m = bullet mass,

F ₀ = initial velocity of the projectile,

g = acceleration due to gravity.

An additional index 0 indicates that there is

is the value under standard conditions.

The ballistic drift r \ _d is according to the equation

30th

= - K _d

(3 a)

proportional to the attachment angle ε. In this equation, K _{D is} a term that depends on the moment of inertia and the rotational speed of the projectile as well as the coefficients of lift and momentum, which quantities can be determined for each ammunition. The wind deflection angle η _κ is given by the equation

η _κ = V _w K _w t _f (lB)

(3 b)

determined, in which V _{w is} the crosswind speed, K _{w is} a coefficient dependent on the ammunition and (1 - B) is a factor that has yet to be explained.

As already stated, signals which are a function of ballistic equations can be generated by circuitry which processes the range information R in accordance with the transmission conditions for ballistic terms and normalized functions.

For example, the attachment angle ε ₀ for standard conditions is generated by a circuit arrangement that generates the distance signal R according to the following approximation equation

with the transfer functions

«" 0 =

mg

mg

2K ₀ V ₀ I 2P ₀ (PV ₀ IK

and m ~ Ko ~

K-m-

d ² ■ P ₀ (Tl ' ² - K _t )

(5)

(6)

processed in which

T = air temperature,

P = air pressure,

K _n = ballistic constant of the projectile,

K, = temperature coefficient.

A signal relating to the time of flight t _f0 under standard conditions is generated by a circuit arrangement which processes the range signal R in accordance with the following approximation equation

20th i / o =

. (RfR J

with the transfer function

K ₀ V ₀

"-DO

From these equations it can be seen that when the circuit arrangements are _implemented , the transfer functions R n0, ε _η0 and i " _{0 are} ballistic terms which are assigned to the ballistic conditions and the environmental conditions for a specific projectile or ammunition and for each fixed set of Standard conditions can be treated as constant, whereas / _f {R / R _n0 ) and f _t (R / R „o) are functions that are independent of the type of bullet and environmental conditions or, in other words, common to all types of bullets. The advantage is that only one such non-linear function generator, which applies to a large number of projectile types, needs to be set up for each ballistic signal and that signals for deviations from standard conditions can be fed to the computer outside the function generator, as will be explained below will.

As can be seen in more detail from the block diagram according to FIG. 1A, the range signal R is first normalized by multiplying by a normalization term for standard conditions or an individual transfer function i / R _n0 , which is assigned to a selected or several types of ammunition. The multiplication takes place with the aid of a ballistic constant multiplier circuit 954, which is also referred to below as a ballistic term multiplier 954 and can be set for any type of ammunition.

As will be explained in more detail later, the normalization term l / R _{n0 is corrected} for changes i / AR _n , which result from changes AP in the air pressure and changes Δ T in the air temperature, by adding the term with a partial multiplication factor (1 - B) is multiplied in which

B = ARJR _n

is. As a result, the equations for the partial with respect to deviations from standard

309 528/214

conditions corrected values of the attachment angle and the flight time t _fl

(10)

^ν π0

2 V _c

00

_e 2 (R / R _M ) (l-ARJR ") _ j

- 1

(H)

then results

«Ι = ^ε « ο

e ^1R l ^R ° - \

2RfR _n

- 1

(12)

In an analogous way one obtains

t -

Oo

(R / R _M ) (l-ARJR _n )

_n ) _ Λ

As indicated above in equation 6, the normalization term l / R n0 is a function of the air temperature T and the air pressure P, so that the factor B by the part of the circuit _{arrangement according to FIG.} 1A that includes summing amplifier 940. The summing amplifier 940 receives a signal A Tj T ₀ for the measured air temperature at one input and multiplies it by a temperature constant K _T. The summing amplifier 940 also receives a signal Δ P / P ₀ for the measured air pressure at a second input. These signals are summed and it then becomes the output signal B of amplifier 940

B =

R "

_4 £ _ _K ~ "4,

AT

40

ΔΤ

T ₀ ΔΡ

Po

= TT ₀ ,

= Standard air temperature and

= PP ₀ ,

= Standard air pressure.

IO The partially corrected attachment angle signal B ₁ is accordingly further corrected for deviations from standard conditions, and the fully corrected attachment angle signal ε is then given by the following equation:

(15)

15th

20th = S ₁ (I + B) (I-2H)

The partially corrected time-of-flight signal is also used for deviations from standard conditions further corrected and becomes too

(17)

The factor H can be obtained by the circuit of FIG. IA are generated, in which a signal ATJT received _g0 for the measured propellant charge temperature of an adjustable Ballistiktermmultiplikator 942, with its own Ballistikterm K ₈ multiplied for each selected ammunition and then an input of a summing amplifier 948 is supplied will. A signal EFC for the effective tube wear is received by an adjustable ballistic term multiplier 944, multiplied by a separate ballistic term K _e for each selected ammunition and passed through a function generator 946. The function generator 946 can approximate the function by two or more rectilinear sections, as will be described in more detail below. This function generator generates an output signal / (KJLFC) which is fed to another input of the summing amplifier 948. The resulting output of the summing amplifier 948 is the factor H according to equation

H =

Δ V ₀

45 = f (KßFQ - K ₁

ΔΤ _η

(19) (20)

As shown later on the basis of FIG. 1A and 1B will be explained in more detail, the partially corrected attachment angle and time-of-flight signals E ₁ and t _{n are} completely corrected for the deviations from standard conditions by further using the partial multiplication factors (1 + B) and (1 - nH) are multiplied, of which one in the manner indicated above by changes in the normalization term 1 / R " _O due to variations AT and AP of the air temperature or the air pressure and the other of changes Δ V _{0 of} the initial speed due to a change in the effective pipe wear EFC and variations Δ T ₉ in the propellant charge temperature depends, so that

H =

AV ₀ V ₀

The factor η has the value 1 or 2, depending on whether there is a linear or quadratic function of V ₀ .

65 _{According to the equations discussed above, the range signal} R / R n0 which is available at the output of the ballistic term multiplier 954 and is normalized to standard conditions is passed through a secondary multiplier 966 , in which it is multiplied by the factor B , so that an output signal is generated which is a inverting input of an operational amplifier 968 is supplied. The output of ballistic term multiplier 954 is also fed directly to a non-inverting input of operational amplifier 968 to provide the corrected range signal

R / R _n0 (lB) = R / R _n

to create.

The secondary multiplier forms part of a main-secondary multiplier such as that from U.S. Patent 3,493,737 is known.

The corrected distance signal RfR _n is then fed to function generators 956 and 958 , which

are arranged parallel to one another and generate signals characteristic of the ballistic functions f _t . (R / R _n ) and f _{t (R / R n).} These function signals are then fed to ballistic term multipliers 960 or 962, which multiply these function signals by the transfer functions e " ₀ or t n0 for the selected _{type of projectile.} As a result, the output signal £ " ₀ f _e (R / R _n ) of the ballistic constant multiplier 960 relates to the partially corrected attachment angle S ₁ for the specific projectile at the specific distance R according to the ballistic equation (4) given above. The output signal t " _o f _t (R / R") of the ballistic constant multiplier 962 is also characteristic of the partially corrected flight time t _{fl of} the selected projectile at the specific range R according to equation (7) above.

As shown in FIG. 1A, a main multiplier 970, belonging to the third of the series-arranged devices for multiplication by the linear approximation of a partial derivative, receives from the summing amplifier 948 the signal f (K _e EFC) - K ₁₁ IT _g / T _g and forms from this the term H, which is characteristic of changes in the muzzle velocity Δ V ₀ / V ₀ and which is used together with the term B , which is dependent on temperature and pressure, to generate the partially corrected attachment angle and time-of-flight signals S ₁ and t _fl with the aid of the in F i G. 1B to completely correct the circuit arrangement illustrated. The partially corrected top angle signal S ₁ is passed through a first secondary multiplier 972, in which it is multiplied by the temperature and pressure-dependent element B in order to generate an output signal S ₁ B which is fed to an input of an operational amplifier 974. The partially corrected top angle signal S ₁ is also fed directly to another input of the operational amplifier 974, so that its output signal assumes the value S ₁ (I + B). This signal is then passed through a second secondary multiplier 976 in which it is multiplied by the muzzle velocity dependent term H to produce a signal S ₁ (I + B) H which is applied to an input of an operational amplifier 978 to be used inverted and multiplied by a factor of 2. The signal S ₁ (1 + B) is also fed directly to another input of the operational amplifier 978, so that this operational amplifier generates a top angle signal ε , which is corrected for conditions deviating from the standard and has the following form according to equation 15:

ε = S ₁ (I + B) (I-2H).

Operational amplifier 986 is supplied. The further corrected time-of-flight signal J ^ ₁ (I + B) is also fed directly to another input of the operational amplifier 986, which generates a time-of-flight signal t _f completely corrected for deviations from standard conditions, which according to equation 17 corresponds to the expression

is equal to. This riding-in-flight signal t _s is fed to a main multiplier 988 for use as a time-of-flight multiplication signal Z, as will be described in greater detail later.

The computer also generates a parallax correction signal α to compensate for the distance between the line of sight and the pipe axis. For this purpose, the distance signal R is fed to a function generator 990 and a rectifier 992 (FIG. 1A). As will be explained in more detail later, the output of the rectifier is fed to a breakpoint selector 994 which determines the slope gain and offset of the function generator 990 to produce a parallax correction signal α for the distance jR that obeys the following equation:

D _p

α =

(21)
(22)

= D _p (l / Rl / R _c ) = D _p -β.

In these equations is

D _p = the distance between the line of sight

and the pipe axis,
R = the measured distance and
R _c = the cutting distance at which the line of sight intersects the pipe axis.

β = (1 / R - 1 / R _1. ) is called the parallax coefficient.

The following applies to the parallax correction in the elevation

(23) (24)

Here, the distance D _pe between the line of sight and the barrel axis in elevation is a constant for every gun arrangement, for example that of a tank.

For a parallax correction in the azimuth (deflection) applies

RR _c

The partially corrected time-of-flight signal t _fl is passed analogously through a first secondary multiplier 980, in which it is multiplied by element B supplied by the main multiplier 964, so that an output signal t _fl B is produced which is fed to an input of an operational amplifier 982. The partially corrected time-of-flight signal t _n is also fed directly to another input of the operational amplifier 982, in which the further corrected time-of-flight _{signal Jy 1} (I + B) is generated. This further corrected time-of-flight signal is passed through a second secondary multiplier 984, in which it is multiplied by the term H , so that a signal t _n (\ + B) H is generated which is an inverting input of a

55 _a \ =

(25)
(26)

Here, too, D _{pd is} a constant for any gun arrangement, for example that of a tank.

The ballistics calculator will now be described in more detail below. The range signal R is fed to the ballistic term multiplier 954 ^ u- shown in FIG. 2 is shown in detail in order to generate the normalized range signal R / R n0 for a selected one of several _{projectile types.} The 954 ballistic term multiplier is an operational

amplifier, which has a number of η parallel input resistance branches which are used for the amplification. Setting serve and each of which contains one of the surface field effect transistors 996 to 996 η, which is connected in series with one of the resistors 998 to 998 η. The resistor branches are connected to an input of an amplifier 1000 . The index η denotes the circuit elements in the η-th resistor branch and is equal to a corresponding number of the floors. During operation, only one of the surface field effect transistors 996 to 996 η is switched on by a positive voltage + V, which is fed to the gate electrode via one of the ν resistors 1002 to 1002 η, while all other of the transistors 996 to 996 π are switched on by a negative voltage - V are blocked, which is fed to their gate electrodes via the corresponding resistors 1002 to 1002 ″ .

It is now assumed that the selected projectile or the selected ammunition is assigned a linear transfer function 1 / R " _O, which is determined in the ballistic term multiplier 954 by the sum of the series resistances in the circuit branch between the cathode and the anode of the switched-on field effect transistor 996 and the Resistor 998 between transistor 996 and the input of operational amplifier 1000 is input. In operation, an ammunition selector switch 1004 to 1004 η is set so that a voltage + V is fed to the gate electrode of the field effect transistor 996 via the resistor 1002 in order to switch this transistor on, while all other transistors, such as the transistor 996 n, are connected to theirs via the switch Gate electrodes receive a voltage - V and are therefore blocked.

The operational amplifier Looo is compensated for an amplification factor of 1, and its output is connected via a feedback resistor 1006 and one of its inputs so that the gain of Ballistiktermmultiplikators 954 the ratio of the value of the feedback resistor 1006 to the sum of the resistances between the cathode and Anode of the switched-on field effect transistor 996 and the resistor 998 is proportional and can be expressed by the term 1 / R " _O. The received range signal R is multiplied by the ballistic term l / R _n0 so that the output of the ballistic term multiplier 954 is a normalized range signal R / R _n0 under standard conditions for the selected ammunition. Any gain resulting from the resistance circuits containing the switched-off field effect transistors can be disregarded, since the resistance between the cathode and the anode is very high in relation to the other circuit resistances. The offset of the operational amplifier 1000 can be set with the help of the center tap of a potentiometer 1008 , at which the tapped voltage is essentially 0 V and from which it is fed to an input of the operational amplifier 1000 via a resistor network. For other storeys the transfer function i / R " _{0 has} a different value, because the values of the resistors 998 to 998 η are chosen so that they result in the respective transfer function for different storeys. Accordingly, the selected, standardized distance signal R / R " _{o is} specifically assigned to the selected projectile. The output signal R / R _n0 is then fed to the ballistic function generators 956 and 958.

Before the normalized distance signal R / R " _{o is} fed to the function generators, it is corrected for temperature and air pressure conditions deviating from the standard in a main- sub-multiplier 964, 966 with time division. The multiplier with time division can again be an electronic multiplier as described in US Pat. No. 3,493,737 and also in the publication "Electronic Analog and Hybrid Computers", McGraw Hill 1964, pp. 268-281 is. With such a multiplier, the output signal is the mean value of a pulse train whose duty cycle is equal to the ratio of two variables and whose amplitude is controlled by another variable. Such a main-sub-multiplier with time division consists of the main multiplier 964 and the sub-multiplier 966. A main multiplier can be used to operate a larger number of sub-multipliers, as shown in FIG. IA is shown.

F i g. 3 illustrates the function generator 956 which generates an output function f _t (R / R _n ) using a number of curve segments. The function generator comprises a break point selector 1044 and a switched resistor network 1046 which contains a number of resistor branches which are selectively summed in dependence on output signals provided by the break point selector 1044. The selectively summed resistances are connected to the input of an operational amplifier 1048 to vary its gain approximately in accordance with the exponential transfer function common to a variety of ammunition types, such as that shown in FIG. 4 shown function f _F (R / R _n ).

The function generator according to FIG. 3 can be used with either a direct voltage or an alternating voltage as the analog input voltage, namely the normalized distance signal RfR _n or (R / R _n ) sin ω t. With an analog AC input voltage, a four-way switch 1050 is in the AC position, so that both the distance input signal (R / R _n ) sin ω t supplied to input terminal 1052 and the reference AC voltage supplied to terminal 1054 , which have the same frequency and phase as the distance input signal has, can be converted with the aid of precision rectifiers 1056 and 1058 into output DC voltages which are proportional to the rms value of the corresponding AC voltages. The precision rectifier 1056 generates a positive DC output voltage that is proportional to the normalized range signal R / R _n , while the precision rectifier 1058 generates a negative DC output voltage that is proportional to the reference AC voltage. The normalized distance signal R / R " and the reference alternating voltage are fed directly to the switched resistor network 1046 because they can be used in it without having previously been converted into direct voltages.

With an analog input DC voltage and a constant DC reference voltage, the switch 1050 is in the DC position, as shown in FIG. 3 is illustrated. The four-way switch is in the AC position mentioned above

1050 from the in FIG. 3 switched position illustrated. When the switch 1050 is in the DC position, the normalized DC distance signal RfR _{n is} fed directly from the input terminal 1060 to the break point selector 1044 and the switched resistor network 1046. The constant DC reference voltage is fed from the input terminal 1062 directly to the switched resistor network 1046 and also to the break point selector 1044 via an inverter 1064, because a negative (-) DC reference voltage is required for the correct operation of the break point selector 1044.

The circuit arrangement according to FIG. 3 is described below for the use of the normalized DC voltage distance signal RfR _n and a reference DC voltage. From the previous and the following explanations it becomes clear how the function generator would work with a normalized AC input signal [RfR _n ) sin ω t and a reference signal that has the same function as the reference DC voltage discussed below.

The breakpoint selector 1044 includes a number of operational amplifiers 1066, 1068 and 1070. Each operational amplifier has an inverting input (2), a non-inverting input (3) and an output (6). Resistor 1072 connects the non-inverting input to ground to minimize the bias current error inherent in operational amplifiers. Comparison circuits, each consisting of resistors 1074 and 1076, 1078 and 1080 as well as 1082 and 1084, are connected in parallel between the input terminal 1060 and the output of the inverter 1064 via the switch 1050 and are used to receive and compare the normalized distance signal RfR _n and the constant DC reference voltage. The connection points of the resistors of each comparison circuit are each connected to one of the inverting inputs of the operational amplifiers 1066, 1068 and 1070. Each of the operational amplifiers 1066, 1068 and 1070 produces a negative output signal when a positive voltage is applied to its inverting input. In the case of a distant target, the distance and, as a result, the amplitude of the normalized range signal RfR _{n is} greater than in the case of a close-up target. The size of the resistors 1074, 1076, 1078, 1080, 1082 and 1084 is chosen so that with a continuous increase in the distance R first the operational amplifier 1066, then, with a further increase in the distance R, the next operational amplifier 1068 and finally, if the distance R increases even further, the operational amplifier 1070 is turned on. The switch-on points for the operational amplifiers are reached when the normalized distance signal RfR _{n reaches} a voltage V _A , a voltage V _B and a voltage V _c , which are represented by the points a, b and c in FIG. 4 are illustrated. When the voltage level V _{A is} exceeded by the normalized distance signal RfR _n , the output voltage of the operational amplifier 1066 becomes low or negative. The output signal of the operational amplifier 1066 is fed through a resistor 1084 to the base of a pnp transistor 1086 in order to turn on the transistor 1086 and to cause the collector voltage to change from a negative value to a positive value due to the voltage drop at the collector resistor 1088. The signal picked up at the collector of transistor 1086 is referred to below as switching or breakpoint selection signal A and at this time has a positive potential.

If, as a result of a further increase in the target range R, the normalized range signal RfR _n exceeds the voltage level V _B , the output signal of the operational amplifier 1068 becomes low or negative. The output of operational amplifier 1068 is fed through a resistor 1090 to the base of a pnp transistor 1092 to turn on transistor 1092 and cause the voltage across collector resistor 1094 to change from a negative to a positive value. The signal picked up at the collector of transistor 1092 is referred to below as switching or breakpoint selection signal B and has a positive potential at this point in time.

If the target distance increases further, the amplitude of the normalized distance signal RfR _{n also increases} further. When the normalized distance signal RfR _n exceeds the voltage level V _c , which is illustrated by the point c in FIG. 4, the output signal of the operational amplifier 1070 becomes low or negative. The output signal of the operational amplifier 1070 is fed through a resistor 1096 to the base of a pnp transistor 1098 in order to turn this transistor on. When transistor 1098 is conductive, the voltage across collector resistor 1100 changes from a negative to a positive value. The signal tapped at the collector of transistor 1098 is referred to below as switching or breakpoint selection signal C and is positive at this point in time.

These switching or breakpoint selection signals a, b and c are fed to the resistor network 1046 in order to change the gain of the operational amplifier 1048 selectively.

The resistor network 1046 contains a first set of parallel resistor branches 1102 to which the normalized distance signal RfR _{n is} fed and which respond to the breakpoint selection signals a, b and c for setting the slope gain, ie the slope factor a _{t of} the element apc of the function / (RfR _n ), which is generated at the output of the operational amplifier 1048. The x-factor of the element α, χ is the analog input voltage RfR _n , which is fed to a resistor 1104 and the cathodes of field effect transistors 1126, 1136 and 1150 via the switch 1050. This analog input voltage RfR _n or the jR / l? In connection with the resistance that is offered to the inverting input of the operational amplifier 1048 by the first set of parallel resistor branches 1102, the _n factor has, through the previously described operation of the circuit arrangement, the generation of the element a, oc of the function f _€ (RfR _n ), which is formed at the output of the operational amplifier 1048, the result. The resistor network 1046 also contains a second set of parallel resistor branches 1112, which are connected in common via the switch 1050 to the input terminal 1062 in order to receive the reference voltage and which respond to the reference voltage and the breakpoint selection signals a, b and c to set the set the intended gain gain so that the coordinates of the function f _f (RfR _n ) in points α, b and c

309 528/214

4, which represent the voltages V _A , V _B and V _c . These coordinate intersections determine the ordinate intersections or voltages b ₂ , b ₃ and b _A. The intersection voltage foj occurs before the break point selection voltage V _A , because it is part of the equation a, oc + ft, - of the first curve segment. The generation of the term bi of the non-linear function f _e (R / R _n ), which represents the intersection voltages ^ ₁ , b ₂ , b ₃ and fo ₄ , at the output of the operational amplifier 1048 is achieved by the action of the resistor to the inverting input of the operational amplifier 1048 is offered by the second set of parallel resistor branches in cooperation with the reference voltage.

If the amplitude of the input voltage analogous to a distance is smaller than the break point voltage V _A , only the rise resistor 1104 and an intersection resistor 1114 are connected to the input of the operational amplifier 1048, because there are no positive break point selection signals a, b or c that correspond to one of the Field effect transistors contained in both the first set of parallel resistor branches 1102 and the second set of parallel resistor branches 1112 could turn on. As a result, the output of the operational amplifier 1048 is given by the equation

fAR / Rn) = O ₁ X + b,

of the first curve segment.

The operational amplifier 1048 includes an amplifier 1116 and has a feedback resistor 1118 which connects its output (6) to its inverting input (2). In addition, the non-inverting input (3) is connected to a reference potential through a resistor 1120 in order to minimize the bias current error inherent in such operational amplifiers. The combination of the rise resistor 1104 and the intersection resistor 1114 has the effect of a resistor that determines the gain factor, so that the gain of the operational amplifier corresponds to the ratio of the resistance value of the feedback resistor 1118 to the sum of the parallel resistance values of the rise resistor 1104, to which the input signal RfR _n is applied, and the Intersection of resistor 1114, at which the reference voltage is applied, is proportional.

Under these conditions, the function generator 956 generates a first one, shown in FIG. 4 shown curve element until the increasing analog input voltage is indicated by the point α in F i g. The voltage F _{4 shown in} FIG. 4 exceeds. If the amplitude of the normalized distance signal RjR _n exceeds the first break point or switching voltage V _A , the positive switching or break point selection signal α is generated by the break point selector 1044 and via resistors 1122 and 1124 the gate electrodes of the field effect transistors 1126 and 1128 of the first and the second set of parallel resistor branches 1102 and 1112 supplied to turn on these field effect transistors. The sum of the parallel-connected resistance values of the resistor 1122 and the resistance branch, including the resistance between the cathode and the anode of the switched-on field effect transistor 1126 and the resistor 1130 effectively reduces the rise resistance of the operational amplifier 1048 ", whereby the rise of the second curve segment between the voltages V _A and V _B , which are shown as points α and b in Fig. 4. In addition, the sum of the parallel resistance values of the resistor 1114 and the resistance branch which provides the resistance between the cathode and the anode of the activated field effect transistor 1128 and the resistor 1132, to the reference voltage in order to bias the gain of the operational amplifier 1048 so that a projection of the second curve segment intersects the ordinate of the diagram according to FIG. 4 at a certain point b ₂ , not shown.

In the same way, the curve segment between the inflection point voltages V _B and V _c , which are indicated by the points b and c in FIG. 4, generated when the intersection selection signal B changes from a negative potential to a positive potential and is supplied to the gate electrode of a field effect transistor 1136 through a resistor 1134 and to the gate electrode of a field effect transistor 1140 through a resistor 1138. The positive switch or breakpoint selection signal B turns on the field effect transistors 1136 and 1140, so that their respective cathode-anode resistors and the resistors 1142 and 1144 are connected in parallel to the resistor branches described above in order to further reduce the rise resistance of the operational amplifier 1048 and thereby increasing the rise of the third curve segment between the voltage levels V _B and V ₀ , which are shown in FIG. 4 are illustrated by points b and c. Furthermore, in cooperation with the reference voltage, the gain of the operational amplifier 1048 is further biased so that a projection of the third curve segment intersects the ordinate of the diagram according to FIG. 4 at a predetermined point b _{3 (} not shown). The fourth curve segment, which follows the break point voltage V ₀ , which starts from point c in FIG. 4, is generated when the switch or breakpoint selection signal c changes from a negative to a positive voltage and is fed through resistors 1146 and 1148 to the gate electrodes of field effect transistors 1150 and 1152 to turn these field effect transistors on. The resistor branches, which contain the resistance between the cathode and the anode of the field effect transistor 1150 or 1152 and the resistors 1154 or 1156, are each connected in parallel to the resistors described above in order to continue to increase the gain of the operational amplifier 1048 and to continue to increase the operational amplifier biased so that the fourth curve segment, when projected, the ordinate of the diagram of FIG. 4 intersects at a further point & 4, not shown.

It should be emphasized that it is possible to check the accuracy of the curve approximation with the aid of this function generator to increase by adding an additional comparison circuit for each additionally desired curve segment, an additional operational amplifier and transistor circuit for the breakpoint selector 1044 and an additional resistor branch in the first and second set of parallel resistor branches 1102 and 1112, each with a field effect transistor and a resistor can be provided. Which he-

Creation of additional curve segments and additional kinks would approximate the desired continuous curve, which reproduces the required function, better. The resulting output signal f _F (R / R ") at the output terminal of operational amplifier 1048 approximates the function described by the equations above.

The resulting output signal £ ■ (R / R ") at the output terminal of the operational amplifier 1048 is received by the adjustable ballistic _term multiplier 960 (FIG. 1 A), in which it is multiplied by the linear transfer function ε π0, which removes the normalization. In construction, the adjustable ballistic term multiplier 960 is the same as the adjustable ballistic term multiplier 954 of FIG . 2, apart from the fact that the values of the resistors 998 to 998 η and of the feedback resistor 1006 are selected according to the transfer function ε _η0 for each ammunition. The output signal of the adjustable ballistic term multiplier 960 is proportional to the partially corrected attachment angle E ₁ according to equations 10 to 12.

The ballistic function generator 958 for generating the non-linear ballistic function f _t (R / R “) and the adjustable ballistic term multiplier 962 for multiplying the function with the transfer function i _n0 according to FIG. 1A are designed in the same way as the ballistic function generator 956 of FIG . 3 or the adjustable ballistic term multiplier 954 according to FIG. 2, except that the values of the resistances contained therein are chosen so that the resulting ballistic function and the multiplication term approximate the continuous curve and the expression given by equation 13 well.

As above with reference to FIG. 1B, the partially corrected attachment angle signal S ₁ and the partially corrected time-of-flight signal t _{fl are} each passed through a pair of secondary multipliers in order to determine them further with regard to deviations from standard conditions according to the linear approximations (1 + B) and (1 -2H) and (1 -H) to correct partial derivatives. In more detail, the partially corrected crown angle signal E _{1 is passed} through slave multiplier 972, operational amplifier 974, slave multiplier 976, and operational amplifier 978 to produce the crown angle output signal

+ B) (I -2H)

(15)

for conditions deviating from the standard.

The partially corrected time-of-flight signal t _n is passed through the secondary multiplier 980, the operational amplifier 982, the secondary multiplier 984 and the operational amplifier 986 to produce the time-of-flight signal corrected for deviations from standard conditions

t, = t _n (\ + B) (\ - H) (17) _fo

to accomplish.

The four secondary multipliers 972, 976, 980 and 984 again belong to multipliers as they are known from US Pat. No. 3,493,737. These secondary multipliers work together with the main multipliers 964 and 970 already mentioned. -

The parallax coefficient β is generated by the function generator 990, the rectifier 992 and the breakpoint selector 994 , which are designed in the same way as the circuit arrangement according to FIG. 3. However, since the parallax coefficient β = 1 / R-1 / R _{C has} an inverse dependence on the distance, it is generated in dependence on the distance in such a way that the function generator reduces the number of parallel resistance branches as the distance increases , that is exactly the opposite of that on hand F i g. 3 described mode of operation.

As shown in FIG. 1B, the elevation control signal E and the deflection control signal D for controlling the fire control system in terms of height or deflection are generated on the basis of the attachment angle signal ε and the time-of-flight signal t _f by using these signals for the ballistic drift η _ά , the wind deflection angle η _κ , the tower angular velocity%, parallax ß, jumping movements Dj, Ej and inclination D _d , E _d can be corrected.

The attachment angle signal ε is determined by a ballistic term multiplier 1160 of the type shown in FIG. 2 by multiplying it by a selected ballistic drift constant K _d for each selected ammunition. This ballistic term multiplier 1160 can contain an instantaneous multiplier of the type mentioned above in order to generate a drift angle signal r \ _d according to equation 3a. This drift angle signal η _ά is then fed to a summing input of an operational amplifier 1162 , whereupon this circuit uses this ballistic drift angle signal to correct the elevation control signal E and the deflection control signal D , as will be explained in more detail shortly.

The crosswind signal V _w received by the anemometer is passed through a variable ballistic term multiplier 1164 of the type previously described by multiplying it by a crosswind coefficient K _w for each selected ammunition to produce a signal V _W K _W. This signal is then passed through a secondary multiplier 1166 in which it is multiplied by the factor B to produce an output signal V "K _W B which is applied to the inverting input of a summing operational amplifier 1168 . The signal V _W K _W is also fed directly to a non-inverting input of the operational amplifier 1168 so that an output signal V _w K _w (I-B) is generated to correct for deviations from standard conditions. This signal is passed through a secondary multiplier 1170 by multiplying it by the time-of-flight signal Z (Z = t _f ) . The wind deflection angle signal η _ν is fed from the output of the secondary multiplier 1170 to a second input of the summing operational amplifier 1162.

The azimuth angular velocity Co _{1 of} the turret is fed through a follow-up and clamp circuit 1172 in which it is stored as a DC signal when a control signal RTLL indicating a lead lock is received. As can be seen from FIG. 5, an arrangement typical of such a follower and clamp circuit comprises a switch 1182 to which the direct current signal W _{1 is} fed. The input of an amplifier 1184 is connected to ground via a suitable resistor 1186 and to switch 1182 via a resistor 1188 . Furthermore, the output of the amplifier 1184 is via a series circuit consisting of a feedback resistor 1190 and a switch 1192

and also connected to the input via a capacitor 1194. The switches 1182 and 1192 are responsive to the signal from a switch driver 1198 which in turn is responsive to the control signal RTLL. The switches 1182 and 1192 are closed for filter operation and open for a hold operation when the RTLL signal is terminated. In all modes of operation, switch 1192 is closed in the absence of an RTLL signal. When the sample and hold circuit enable of signal RTLL is used, switch driver 1198 causes switches 1182 and 1192 to open so that the angular velocity is stored _{as DC signal W 1.}

This DC signal is then passed through a secondary multiplier 1174 of the type described above in which it is multiplied by the time-of-flight multiplication signal Z to form _{a kinematic lead angle signal Cu 1} t _f. If there is a Einkreiselsystem, this kinematic lead angle signal is coytf through an analog switch 1176 a as a single kinematic gyro lead angle signal% the other input of the operational amplifier 1162 is supplied to the correction of the deflection signal D and the elevation signal E.

Depending on the type of system, there may be a second vortex velocity signal co _e , which runs through a sequence and clamping circuit 1262 and a secondary multiplier 1264 in the same way as the angular velocity signal, so that a second kinematic gyro lead angle signal m _e t _{f is} formed which then fed to a second analog switch 1176 b.

The in F i g. 6 illustrated analog switches 1176 a and 1176 b receive the one kinematic gyro lead angle signal% = ω ₁ ί _/ at an input terminal 1204 and optionally the optional second kinematic gyro lead angle signal m _e t _f at a second input terminal 1206. The control signal RTLL is at a Input terminal 1212 received. This control signal RTLL is normally at a low level when the computer is not in the CALCULATE mode, so that the input signals at terminals 1208 and 1210, which are at ground potential, reach output terminals 1214 and 1216.

When the control signal RTLL, which is fed to the terminal 1212, is low, the diode circuit 1218 is acted upon in the forward direction, so that one end of the base current resistor 1220 is at ground potential. This ground potential signal is passed through a threshold circuit 1222, which contains two diodes connected in series, to turn off an npn transistor 1224. Furthermore, a base-emitter resistor 1226 is connected between the base and the emitter of the transistor 1224. A bias network comprising resistors 1228 and 1230 connected between the collector and a +12 V terminal and a resistor 1232 is connected between the +12 V terminal and a + 50 V terminal at one end and the base of a transistor 1234 in order to keep the base voltage to 12 V ± Δ V, 1224. for example, when the transistor 1224 is depending on the state of the transistor in its normally locked state, the base of the transistor 1234 applied voltage greater than +12 V, and the transistor 1234, which is part of a differential amplifier, is blocked. When the transistor 1234 is blocked, the other transistor 1236 of the differential amplifier is conductive, which is due to the voltage at the emitter, which results from the current flow through the emitter resistor 1238, and the potential formed at the collector resistor 1240.

In this state of the differential amplifier, the surface field effect transistors 1242 and 1244 are switched on by a signal which is fed to their gate electrodes by the collector voltage which is present at the collector resistor 1240. When the surface-effect transistors 1242 and 1244 are switched on, the signal received at the input terminals 1208 and 1210 is conducted with ground potential via the cathode-anode paths of the field-effect transistors 1242 and 1244 to the output terminals 1214 and 1216. Since the transistor 1234 of the differential amplifier is blocked, the voltage drop across the collector resistor 1246 is small, so that the gate electrodes of the field effect transistors 1248 and 1250 are supplied with a low voltage signal which blocks these transistors. As a result, the input signals W ₁ iy and ω _ε ί <-, which are fed to the input terminals 1204 and 1206, cannot be passed on to the output terminals 1214 and 1216.

When the control signal RTLL fed to the input terminal 1212 assumes a high value, the diode circuit 1218 is acted upon in the reverse direction, as a result of which the level of the signal at the lower end of the base current resistor 1220 rises. This signal is fed to transistor 1224 via threshold circuit 1222 in order to bring it into the conductive state. When transistor 1224 is on, the flow of current through the bias network comprising resistors 1228, 1230 and 1232 causes the voltage at the base of transistor 1234 to drop below + 12V, turning transistor 1234 on.
When transistor 1234 of the differential amplifier is on, the other transistor 1236 is off. As a result, the voltage drop across the collector resistor 1246 increases, while the voltage drop across the collector resistor 1240 decreases. As a result, the collector voltage 1234 is fed to the gate electrodes of the field effect transistors 1248 and 1250 in order to switch these transistors on, whereupon the input signals W ₁ t _f and m _e t _f are fed to the output terminals 1214 and 1216 respectively via the cathode-anode paths of the field effect transistors. The decrease in the voltage at the collector of transistor 1236 is fed to the gate electrodes of field effect transistors 1242 and 1244 to block these transistors, after which the signal at ground level fed to input terminals 1208 and 1210 can no longer be fed to the output terminals.

The output signal M of the operational amplifier 1162 then becomes B = η _ά + η _κ + η _Ιί . If, however, it is a two-circuit system, the switching contacts of a single-circuit switch 1178 are switched over so that the first kinematic lead angle signal CU ₁ ^ - is fed to an input of an operational amplifier 1180 and the signal M = η _ά + η » .

The attachment angle signal ε is also fed directly to an input of an operational amplifier 1252, to generate a signal L that is equal to ε (L = ε).

The output signals L and M of the operational amplifiers 1162 and 1252 can be fed directly to the inputs of operational amplifiers 1180 and 1257 by means of analog switches 1254a and 1254b of the type described above when a skew signal N is at the "off" level. When the tilt signal JV is "On", the L and M signals are passed through a tilt resolver 1256 before being fed to the inputs of the operational amplifiers 1180 and 1257 via the analog switches 1254a and 12546. While a tilt resolver has been specifically mentioned, it will be understood that other types of resolvers could be used instead.

Assuming that the inclined position resolver 1256 is in operation, the output signals L and M of the operational amplifiers are fed to a winding of the resolver and resolved into tower coordinates which take into account the angle of inclination of the tower with respect to the horizontal and produce output signals which in accordance with this Lean angles are resolved. One way to accomplish this is to have a first winding connected to a pendulum so that the first winding is moved relative to a stationary second winding and the inductive coupling between the legs of the windings changes with the position of the movable winding. The output of the lean resolver 1256 then becomes an uncorrected deflection signal D ' and an uncorrected elevation signal E' of the shape

D ' = McosN + LsiaN E' = LcosN + MsinN

These uncorrected deflection and elevation signals D ' and E' are then one of the summing inputs of the analog switches 1254 a and 12546

35 operational amplifiers 1180 and 1257, in which they are further corrected. Other inputs of operational amplifiers 1180 and 1257 receive jump signals Dj and Ej provided by variable attenuators 1258 and 1260 for each selected ammunition. Another input of the operational amplifiers 1180 and 1257 receives the parallax coefficient β = l / R - l / i? _c , which is generated by function generator 990 of Figure 1A. Another input signal of the operational amplifiers is the hanging of the gun barrel in the deflection D _d and the elevation E _d . The inputs of the operational amplifier contain an adjustable resistor for the gain, which multiplies the parallax coefficient β by the azimuth distance D ^ , which is determined for each tank. The input of the operational amplifier 1257 has an adjustment resistor for the gain, which multiplies the parallax coefficient β by the elevation distance D _pe , which is determined for each tank. If it is a two-gyro system, the signal m _e t _f from the product of the tower angular velocity in the elevation and the flight time is fed to another input of the operational amplifier 1257 in order to take into account the kinematic lead angle in the elevation. The output signals of the operational amplifiers 1180 and 1257 are passed through analog switches 1278 a and 1278 ft of the type described above, which only pass the signal on when the CALCULATE signal is received from them.
The operational amplifiers 1280 and 1282 receive the output signals of the analog switches 1278 a and 1278 b together with target line correction signals D _b and E _b in the deflection and elevation, respectively. These operational amplifiers 1280 and 1282 generate the deflection correction signal D and the elevation correction signal E, which signals are supplied to a servo amplifier.

For this purpose 2 sheets of drawings 309528/214

Claims (11)

Patent claims: 1. Ballistic computer for calculating fire control signals as a function of the target range and non-standard conditions, which are characteristic for the deviation of the environmental conditions and the ammunition properties from standard values, with a range multiplier for modifying the range data as a function of the non-standard conditions and for generating a standardized range signal, a time-of-flight function generator and an elevation angle function generator for multiplying the normalized distance signal with non-linear functions of the normalized distance and a time-of-flight multiplier and an elevation angle multiplier for modifying the output signals of the time-of-flight or elevation angle function generator as a function of the non-standard conditions and for generating a time-of-flight signal and a ballistic elevation angle signal, characterized in that at least one of the Mul multipliers (954, 964, 966, 968; 962, 964,980,982; 960, 964, 972, 974) an ammunition type multiplier (954; 962; 960) and a correction circuit (964, 966, 968; 964, 980, 982; 964, 972, 974) connected in series with the ammunition type multiplier which the ammunition type multiplier (954; 962; 960) multiplies the signal supplied to it by a transfer function characteristic of the ammunition type (l / i? "₀;t" ol "" o) and the correction circuit has a multiplier (964, 966; 964 , 980; 964, 972), which multiplies the signal fed to it by a function (B) of the deviations of the environmental conditions from the standard values, and comprises a summing circuit (968; 982; 974) for combining the output signals of the multiplier with the input signal fed to it . 2. Ballistic computer according to claim 1, characterized in that a further correction circuit (970, 984, 986 and 970, respectively) for the time of flight and the elevation angle multipliers (962, 964, 980, 982 and 960, 964, 972, 974) , 976, 978) is connected in series, which is used to compensate for the effects of changes in the muzzle velocity depending on the temperature and the pipe wear and again from a multiplier (970, 984 or 970.976) to multiply the supplied signal [t _fl (1 + B) or E ₁ (1 + BJ] with a function (Ji) of the deviation of the propellant charge temperature from the standard value and the pipe wear and a summing circuit (986 or 978) to combine the output signal of the derivative multiplier (970, 984 or 970, 976) with the input signal fed to it for generating an output signal (t _f or ε) corrected for non-standard conditions. 3. Ballistics computer according to claim 1, characterized in that the multipliers (964, 966; 964, 980; 964, 972) from the ammunition type multiplier (954; 962; 960) is multiplied by a signal which is for the The relationship Δ PfP ₀ - K _T Δ TfT _{0 is} characteristic, in which Δ T is the deviation of the air temperature from the standard air temperature T ₀ , ΔP is the deviation of the air pressure from the standard air pressure P ₀ and K _{T is} a preselected constant. 4. Ballistic computer according to one of the preceding claims, characterized in that it has a ballistic term multiplier (1160) for multiplying the corrected ballistic attachment angle signal (ε) with a selected constant (K _d ), which is assigned to a certain type of ammunition, for generating a drift signal (η _ά ) includes. 5. Ballistic computer according to one of the preceding claims, characterized in that it comprises a lead angle part (1264,1174,1170) which receives at least one angular velocity signal (W ₁ ; ω _ε ) and preferably also a cross wind signal (V _w ) and at least one lead angle signal (^) delivers, which is a function of the product of the angular velocity signal and possibly the crosswind signal with the time-of-flight signal (t _f ) . 6. Ballistic computer according to claim 5, characterized in that the lead angle part (1264, 1174, 1170) comprises a wind speed multiplier (1164, 1166, 1168) for modifying the received crosswind signal (V _w ) as a function of the non-standard conditions. 7. Ballistic computer according to claim 6, characterized in that the wind speed multiplier (1164, 1166, 1160) comprises a multiplier (1164) and a correction circuit (964, 1166, 1168) connected in series with the multiplier, of which the multiplier ( 1164) multiplies the crosswind signal (V _w ) supplied to it by a crosswind coefficient (K _w ) characteristic of the selected type of ammunition, and the correction circuit uses a multiplier (964, 1166) that converts the supplied signal with a function (B) of the deviations in the environmental conditions from the Standard values multiplied, and a summing circuit (1168) for combining the output of its multiplier with the input signals applied to it. 8. Ballistic computer according to claim 5, characterized in that a coordinate converter (1162, 1252, 1256, 1180, 1257) responds to the ballistic elevation angle signal (ε) and the at least one lead angle signal (%) and elevation and azimuth fire control signals (E ' or D ') in the coordinate system of a gun coupled to the ballistics computer. 9. Ballistic computer according to claims 4 and 8, characterized in that the coordinate converter (1162, 1252, 1256, 1180, 1257) also responds to the drift angle signal (η _ά ). 10. Ballistic computer according to claim 8 or 9, characterized in that means for receiving signals which are characteristic of the scanned parameters of the gun coupled to the ballistic computer, and a summing circuit (1257, 1180) for modifying the elevation and azimuth fire control signals ( E ' or D') are available as a function of the sampled parameters. 11. Ballistics computer according to one of claims 5 to 10, characterized in that the lead angle part (1264, 1174, 1170) contains a follow-up and clamping circuit (1172, 1262) which scans and stores the at least one angular velocity signal (W _{1), and} that these stored signals are multiplied by the time-of-flight signal (t _f ).