DE102022103550B4 - Control circuit for an electrical separator - Google Patents

Abstract

Device comprising:an electrical separator (3) with a plurality of emission electrodes (7) and a counter electrode (5); anda control circuit (120) which is set up to generate a high voltage from a low voltage of a direct voltage source (110) and to apply the high voltage to the emission electrodes (7) and the counter electrode (5) of the electrical separator (3) in order to create a direct current plasma between the emission electrodes (7) and the counter electrode (5); the control circuit (120) containing a control circuit (122) with a current measuring unit (256), the current measuring unit (256) passing through the emission electrodes (7) via the direct current plasma to the counter electrode ( 5) measures the direct current flowing from the electrical separator (3) to the common reference potential of the control circuit (120); and the control circuit (122) is set up to regulate the output power of the control circuit (120) to a reference current value based on the value of the measured flowing direct current, the control circuit (120) having a transformer (220) and a coil (1218) on the secondary side. has a high-voltage cascade (600) connected to the transformer (220) in order to convert the low voltage of the direct voltage source (110) or a direct voltage derived therefrom into the high voltage, the current measuring unit (256) measuring the direct current at a center tap of a voltage divider, the voltage divider being on is connected on one side to the secondary-side coil (1218) of the transformer (220) and on the other side is connected to the counter electrode (5) of the electrical separator (3) and the reference potential of the control circuit (120).

Description

The invention relates to a control circuit for generating a high voltage which is applied to the electrodes of an electrical separator. Another aspect is the use of such a control circuit in a device, for example a room air filter or a fine dust filter. The invention further relates to a circuit board on which a control circuit according to the invention is implemented using discrete components.

Electrical separators are used, for example, in room air filters, fine dust filters or similar devices. The electrodes of the electrostatic precipitator are conventionally driven with a pulsed positive high voltage in order to generate a plasma between the electrodes of the electrostatic precipitator. For this purpose, control circuits are used in the end devices that convert the low voltage on the input side into the high voltage. Transformers are often used that provide a correspondingly high multiplication factor by appropriately selecting the number of turns on the primary and secondary sides of the transformer.

The patent application WO 2021/224017 A1 describes a room air purifier in which an emission electrode surrounded by water lies opposite an arrangement of counter electrodes in order to form the desired electric field between the emission electrode and the counter electrode. The counter-emission electrodes are formed from needles.

The patent specification US 8,529,830 B2 describes an air purification device. The air purification device comprises a plasma reactor which is driven by a voltage supply circuit with voltage pulses, the width of the voltage pulses being able to be varied by the voltage supply circuit. The voltage supply circuit is supplied on the input side with an alternating voltage, which is first converted into a direct voltage using a rectifier circuit and a subsequent filter. A digital control circuit of the voltage supply circuit controls the primary side of a transformer using a switch with pulses of direct voltage, so that high-voltage pulses of 10-2 µs of a positive high voltage in the range from 12 kV to 16 kV (peak-valley value) are output on the secondary side of the transformer ) are generated, by means of which a plasma is generated. The control circuit is regulated based on the primary-side current flowing through the switch in the primary-side part of the voltage supply circuit. The pulse frequency of the power supply circuit is in the range 20-100 kHz.

The patent application US 2004/0033176 A1 describes an electrokinetic air conditioner that removes particles from the air to create a particle-free airflow. The air conditioner includes an ion generator with an electrode assembly including a first row of emitter electrodes, a second row of collector electrodes, and a high voltage generator. The high-voltage generator generates alternating voltage from the input side, which is first converted into a direct voltage using an oscillator that statically switches a switch at a frequency of 20 kHz to generate voltage pulses. The voltage pulses are increased into high-voltage pulses using a transformer. Another voltage multiplier is used on the secondary side of the transformer to generate high-voltage pulses.

Out of JP 2001-232241 A an electrical separator is known which includes an emission electrode and a counter electrode as well as a power source and a control device that controls the power source. The output voltage Vo of the power source is detected by a voltmeter, and the current of the power source I0 is detected by an ammeter. The control device inputs a preset voltage Voset and a preset current loset into the power source according to the detected values. The power source is then controlled so that the output voltage Vo or the output current I0 corresponds to the input preset voltage Voset or the preset current loset. The control device sets the preset voltage Voset close to the rated voltage Vb so that the power source maintains the constant value of the preset current loset. Thus, the increase of a current flow in generating a spark discharge is prevented and the development of a spark discharge is prevented.

Out of DE 10 2020 112 573 A1 a device for treating air is known, which comprises a flushing body surrounded by liquid, an air guide system which is set up to flow the air to be treated onto the flushing body, and an electrical separator which is assigned to the flushing body in such a way that solid and / or liquid particles are separated from the air to be treated flowing around the body and the separated particles get into the liquid.

Out of US 2008/0198563 A1 It is known to form insulating ribs in a housing that encloses circuits of an electrical device. Components mounted on a circuit board are separated based on whether they relate to the primary side of a power transformer or to the secondary side. Primary related components are typically mounted on the primary side of an isolation slot machined into a circuit board, and components related to the secondary side are mounted on the other side. The insulating ribs are strategically placed to protrude through the slots when the circuit board is mounted to the housing part. Thus, when the casing section is connected to another casing section, which may also include strategically placed insulating fins, the fins increase the breakdown voltage of the boundary.

This patent application is based, among other things, on the task of proposing an improved control circuit for generating a high voltage for driving an electrical separator. In improving the drive, one or more of the following considerations may be taken into account: the stability of the drive circuit over the desired temperature range in which it operates (e.g. -40°C to +160°C); a high response speed of the control loop to the ambient temperatures and parameters (e.g. rapid temperature changes, undesirable breakdowns of the plasma, etc.); high requirements for electromagnetic compatibility (EMC); Possibility of discrete construction of the circuit; Miniaturization of the individual components, especially the transformer and the control circuit. This task is solved by the subject matter of independent patent claim 1.

Another aspect and another task of this patent application lies in the design of a circuit board that implements the control circuit and can reliably prevent leakage currents and arcing on the circuit board. This task is solved by the subject matter of independent patent claim 22.

A significant further aspect and a further object of this invention is to provide a room air purifier, such as from the patent application WO 2021/224017 A1 known, to improve, in particular to specify a control and / or electronic control in order to ensure, on the one hand, sufficient cleaning results with regard to the use of a liquid that wets the counter electrode, and on the other hand to ensure the high requirements for the safety of the people in the room to be cleaned. This task is solved by the subject matter of independent patent claim 31.

One aspect of the invention relates to a drive circuit for generating a high voltage that is applied to the electrodes of an electrical precipitator. In contrast to conventional control circuits, the electrodes of the electrical precipitator are not controlled with a pulsed high voltage, but rather with a direct voltage in the high-voltage range, so that a direct current flows across the electrodes of the electrical precipitator. The control circuit of the control circuit includes a current measuring unit that measures the direct current flowing across the electrodes of the electrical separator. The control circuit regulates the output power of the control circuit based on the measured value of the flowing direct current and regulates the direct current to a reference current value. The control loop can, for example, have a control cycle of 5 ms or less, preferably 2 ms or less, more preferably 1 ms or less. The shorter the control cycle, the faster fluctuations in measured direct current can be detected and corrected.

One embodiment of the invention relates to a device with a control circuit which is set up to generate a high voltage from a low voltage of a direct voltage source and to apply the high voltage to the one or more emission electrodes and their counter electrode of the electrical separator in order to create a direct current plasma between the emission electrode(s). ) and the counter electrode. The control circuit includes a control circuit with a current measuring unit. The current measuring unit measures the direct current flowing through the emission electrode(s) via the direct current plasma to the counter electrode of the electrical separator to the common reference potential of the control circuit. The control circuit is set up to regulate the output power of the control circuit (or its output current) to a reference current value based on the value of the measured direct current flowing. The reference potential of the control circuit can be, for example, the reference potential that is used to determine all voltages in the control circuit. This reference potential is also referred to as internal ground. This may or may not correspond to the mass of the earth (common ground).

In a further embodiment, the control circuit is set up to set the output power of the control circuit (or its output current) based on the measured value of the direct current relative to the reference current value to an operating point at which a spark discharge of the direct current plasma between the emission electrodes and the counter electrode is prevented. It should be noted here that in the embodiments of the invention, the spark discharge/breakdown of the direct current plasma is viewed as a fault and the regulation by the control circuit prevents such breakdown of the direct current as far as possible.

Whether breakdowns occur depends, among other things, on the series resistances of the emission electrodes as well as their insulation and the conditions in the separation room. In the separation room, pressure fluctuations (e.g. slamming doors) or macroscopic particles (e.g. hair) can lead to breakdowns. While breakdowns due to macroscopic particles are difficult to control, pressure fluctuations in the separation chamber, for example, can be prevented with a high degree of reliability using the control according to the invention. For example, if one assumes the application of the invention in an air purifier with a Clean Air Delivery Rate (CADR) of 250 m ³ /h and tolerates a drop in the CADR of 10 m ³ /h, and sees breakdowns as the cause of this, which leads to restarts the board, a breakdown rate can be estimated as follows. If a restart takes 5 s (can be parameterized), then 10 m ³ /h corresponds to approximately 0.5 breakthroughs per minute. The breakdown rate of conventional electrical separators is often in the range of 40 or more breakdowns per minute.

In a further embodiment, the control circuit of the control circuit comprises a control unit that receives the measured value of the flowing direct current from the current measuring unit and generates a control signal. The control unit can be used, for example, by means of a microprocessor, a discrete circuit or an integrated circuit (e.g. Application Specific Integrated Circuit (ASIC) or programmable logic (e.g. Field Programmable Gate Array (FPGA), Programmable Logic Device (PLD), etc.) or combinations thereof The options mentioned can be designed as a possible microprocessor, for example a microcontroller from the S12ZVM family from NXP, such as the microprocessor S912ZVMBA6F0WLF from NXP. The control signal generated by the control unit is, for example, a transistor-transistor logic (TTL) control signal or CMOS control signal . The control unit can further be set up to vary the frequency of the control signal based on the value of the direct current and the reference current value in order to bring the direct current flowing through the emission electrodes via the direct current plasma to the counter electrode of the electrical separator to the common reference potential of the control circuit and the electrical separator towards the reference current value regulate. In an exemplary embodiment, the control signal has a frequency in the range from 120.0 kHz to 200.0 kHz inclusive, preferably in the range from 130.0 kHz to 170.0 kHz inclusive. The resonance frequency of the transformer is higher than the frequency of the control signal; it is preferably in the range including 200.0 kHz to 240.0 kHz.

According to the invention, the control circuit has a transformer and a high-voltage cascade connected to a secondary-side coil of the transformer in order to convert the low voltage of the low-voltage direct voltage source or a direct voltage derived therefrom into the high voltage. The current measuring unit measures the direct current at a center tap of a voltage divider, the voltage divider being connected on one side to the secondary-side coil of the transformer and on the other side being connected to the counter electrode of the electrical separator and the reference potential of the control circuit.

Another aspect of the invention relates to the structure of the transformer, which can be used in the different embodiments of the control circuit. The
Transformer contains either a primary side coil and a secondary side stage or alternatively, two primary side coils wound in the same direction and connected in series and a secondary side coil. In the latter case, the primary-side coils can optionally be pushed into one another in order to make the transformer more compact. In one embodiment of the invention, the windings of the secondary-side coil are divided into several partial coils. The transformer can further comprise a non-conductive housing (eg a winding body) which is divided into a plurality of housing sections which are arranged parallel to one another in one direction and are separated from one another by a non-conductive material and in which the respective partial coils are arranged. The transformer can be integrated in an EFD20 housing, for example. The resonance frequency of the transformer is preferably in the range 200.0 kHz to 240.0 kHz.

In an exemplary embodiment of the transformer or the control circuit, the ratio of the number of windings of the (one) primary-side coil or each of the two primary-side coils of the transformer and the number of windings of the secondary-side coil of the transformer is in the range from 0.015 to 0.025 inclusive. In an exemplary embodiment, this ratio is 0.02. For example, the number of windings of the primary-side coil(s) of the transformer is between 13 windings and 18 windings, and the number of windings of the secondary-side coil of the transformer is between 700 windings and 800 windings.

Further embodiments relate to the control of the transformer of the control circuit by power switches, such as Leis transistors (e.g. metal-oxide-semiconductor field effect transistors (MOSFET)) or thyristors.

In an exemplary embodiment, the transformer is controlled with two power switches. For example, the two power switches can be designed as n-channel metal-oxide-semiconductor field effect transistors (NMOS MOSFETs). The control circuit includes a transformer with two primary-side coils wound in the same direction and a secondary-side coil. Each primary-side coil has an end that is connected to the low voltage of the DC voltage source or a DC voltage derived therefrom. Furthermore, the control circuit comprises a first power switch (e.g. power transistor or thyristor), which is controlled with a frequency-variable pulse width modulated control signal (PWM control signal) and, depending on the control signal, connects a second end of a first of the two primary-side coils to the reference potential of the control circuit; and a second power switch (e.g. power transistor or thyristor), which is controlled with the inverted frequency-variable PWM control signal and, depending on the inverted frequency-variable PWM control signal, connects a second end of the second of the two primary-side coils to the reference potential of the control circuit. With a duty cycle of 50%, the "inverted frequency-variable PWM control signal" is a signal that is in phase with the frequency-variable PWM control signal but is inverted: A rising edge of the PWM control signal corresponds in time to a falling edge of the inverted PWM control signal and a Falling edge of the PWM control signal corresponds in time to a rising edge of the inverted PWM control signal.

In a further embodiment of the control circuit, several driver circuits, which adapt the TTL or CMOS control signal of the control unit of the control circuit to the signal levels of the power switches. Accordingly, the control circuit further comprises a first driver circuit, which is designed as a single or multi-stage level converter in order to adapt the signal level of the TTL/CMOS control signal to the signal level of the first power switch and to supply the adapted control signal to the first power switch as the frequency-variable PWM control signal; and a second driver circuit, which is designed as a single or multi-stage level converter, to adapt the signal level of the TTL/CMOS control signal to the signal level of the second power switch and to supply the adapted control signal to the first power switch as the inverted frequency-variable PWM control signal. Each of the first and second driver circuits can optionally have a plurality of resistors, diodes and transistors and can be set up to coordinate the charge and discharge of the gate capacitances of the transistors in the individual stages of the level converters in such a way that respective transistor pairs in each stage of the driver circuits, which are connected in series between a DC voltage corresponding to or derived from the low voltage and the reference potential, in a predetermined temperature range in which the control circuit is operated, preferably around range
-40° C to 160° C, are not conductive at the same time (ie a “shoot-through” is prevented). In particular, the driver circuits can be designed to be identical in construction, so that it can be ensured that the driver circuits show the same behavior in the event of temperature fluctuations, that is, for example, that there are changes in the rise times and fall times of the signal edges, which arise in particular due to the fluctuations in the resistance values of the resistors at different temperatures , turn out to be the same in the individual driver circuits.

In an alternative embodiment, the transformer is controlled with four power switches or alternatively by means of two power switches and two capacitors that form a full bridge. In this embodiment, the drive circuit comprises a transformer with a primary-side coil and a secondary-side coil. Optionally, the one primary-side coil can be formed by a series connection of coils wound in the same direction, in particular two coils. The control circuit further comprises a first power switch (e.g. power transistor or thyristor), which is controlled with a frequency-variable PWM control signal and, depending on the control signal of a control unit of the control circuit, connects a second end of the primary-side coil to a direct voltage corresponding to or derived from the low voltage; a second power switch (e.g. power transistor or thyristor), which is controlled with the inverted frequency-variable PWM control signal and, depending on the inverted control signal, connects a first end of the primary-side coil to the DC voltage corresponding to or derived from the low voltage; a third power switch (eg power transistor or thyristor), which is controlled with the inverted frequency-variable PWM control signal and, depending on the inverted control signal, connects the second end of the primary-side coil to the reference potential of the control circuit; and a fourth power switch (eg power transistor or thyristor), which is controlled with the frequency-variable PWM control signal and, depending on the control signal, the first end of the pri The market-side coil connects to the reference potential of the control circuit. Optionally, a capacitor can be connected between the gate connection and drain connection of the power switches designed as power transistors in order to suppress conducted interference. The first and second power switches can be designed, for example, as p-channel metal-oxide-semiconductor field effect transistors (PMOS MOSFETs). The third and fourth power switches can be designed, for example, as n-channel metal-oxide-semiconductor field effect transistors (NMOS MOSFETs). In the alternative embodiment, the first and third power switches are each replaced by a capacitor, so that a virtual ground point is formed between the two capacitors and is connected to the second end of the coil. In this case, no driver circuits need to be provided for the first and third power switches, ie in this embodiment only the remaining two (second and fourth) power switches are correspondingly controlled by driver circuits.

In a further embodiment, the control circuit comprises several driver circuits that adapt the TTL or CMOS control signal of the control unit of the control circuit to the signal levels of the power switches. The control circuit accordingly comprises a first driver circuit, which is designed as a single or multi-stage level converter in order to adapt the signal level of the TTL/CMOS control signal to the signal level of the first power switch and to supply the adapted control signal to the first power switch as the frequency-variable PWM control signal; a second driver circuit, which is designed as a single or multi-stage level converter, to adapt the signal level of the TTL/CMOS control signal to the signal level of the second power switch and to supply the adjusted control signal to the first power switch as the inverted frequency-variable PWM control signal; a third driver circuit, which is designed as a single or multi-stage level converter, to adapt the signal level of the TTL/CMOS control signal to the signal level of the third power switch and to supply the adjusted control signal to the first power switch as the frequency-variable PWM control signal; and a fourth driver circuit, which is designed as a single or multi-stage level converter, to adjust the signal level of the TTL/CMOS control signal to the signal level of the fourth power switch and to supply the adjusted control signal to the first power switch as the inverted frequency-variable PWM control signal.

Furthermore, according to a further embodiment, each of the first, second, third and fourth driver circuits can have a plurality of resistors, diodes and transistors and can be set up to coordinate the charge and discharge of the gate capacitances of the transistors in the individual stages of the level converters so that respective transistor pairs in each stage of the driver circuits, which are connected in series between a direct voltage corresponding to or derived from the low voltage and the reference potential, in a predetermined temperature range in which the control circuit is operated, preferably in the range -40 ° C to 160 ° C, not simultaneously are leading. In this embodiment too, the driver circuits can be designed to be identical in construction, so that it can be ensured that the driver circuits show the same behavior in the event of temperature fluctuations, i.e. that, for example, there are changes in the rise times and fall times of the signal edges, which are particularly due to the fluctuations in the resistance values of the resistors Different temperatures result in the same in the individual driver circuits.

In the above embodiments, the driver circuit(s) receiving the inverted PWM control signal may alternatively also receive the PWM control signal. In this case, the corresponding driver circuit(s) may comprise an inverter which first inverts the PWM control signal in order to obtain the inverted PWM control signal. In this case, the other driver circuit(s) comprise a delay loop for the PWM control signal, which compensates for the delay in the PWM control signal resulting from the inversion. Alternatively, in the above-mentioned embodiments, the driver circuit(s) that receive the PWM control signal can also receive the inverted PWM control signal and initially invert it accordingly. Here too, the other driver circuit(s) would compensate for the delay in the control signal by inverting it using a delay loop.

In the embodiments described above, the PWM control signal can have a duty cycle of 50%, so that the one or more primary-side coils of the transformer can be switched virtually continuously with the low voltage of the low-voltage source, or a direct voltage derived therefrom. If, when operating the control circuit in the desired temperature range, control errors occur due to temperature fluctuations (e.g. shoot-through of the series-connected transistor pairs of the driver circuits or the power switch pairs), the duty cycle of the PWM control signal could alternatively or additionally be reduced in order to prevent these control errors. However, this could be the ripple of the secondary output chip voltage of the transformer may increase, so that further measures for rectification on the secondary circuit side of the control circuit could be necessary.

The output voltage of the transformer may not be at the desired voltage level. Therefore, according to a further embodiment, the control circuit can have a multi-stage, in particular 3-stage, 4-stage, 5-stage, or 6-stage high-voltage cascade, which increases the output-side alternating voltage of the transformer into the direct current high voltage. In an exemplary embodiment, the high-voltage cascade is designed to convert the secondary-side output voltage of the transformer into a direct voltage in the high-voltage range, whose alternating component (AC component) compared to its direct voltage component (DC component) is 4% or less, preferably 3% or less is preferably 2% or less. The high-voltage cascade can be, for example, a Villard cascade or a high-voltage cascade with full-wave rectification, whereby the Villard cascade can be made significantly more compact. For example, each stage of the high-voltage cascade can include a series connection of several diodes and a series connection of several capacitors, which are designed as discrete components. In this case, the number of discrete components required for a Villard cascade is significantly lower than for a high-voltage cascade with full-wave rectification.

In a further embodiment, the control circuit further has an impedance matching circuit. The impedance matching circuit can be connected between the two outputs of the secondary-side coil of the transformer of the control circuit and the inputs of the high-voltage cascade. The impedance matching circuit serves to adapt the impedance between the power switches of the drive circuit, the transformer of the drive circuit and the high-voltage cascade of the drive circuit to one another.

In a further embodiment, the control circuit further has an input stage which is connected to the DC voltage source, the input stage comprising a common mode choke and/or a filter which are set up to attenuate conducted interference from the transformer and/or from the DC voltage source into the Control circuit to limit current flowing.

According to a further embodiment, the control circuit further comprises an input stage which comprises a filter, the output of the filter providing a DC voltage derived from the low voltage of the DC voltage source, which is applied to the primary-side coils of the transformer of the control circuit.

Another embodiment relates to the design of a circuit board that implements a control circuit according to the invention. In particular, this aspect concerns the discrete structure of at least the high-voltage part of the control circuit. Of course, the complete control circuit can also be constructed discretely. A further embodiment therefore relates to a circuit board on which a control circuit with discrete components is implemented. The control circuit has a low-voltage part and a high-voltage part, which is implemented on the circuit board. The board may have a substantially rectangular shape, but the invention is not limited to rectangular shapes of the board. However, without limiting the generality, it can be assumed that the board extends in a longitudinal direction and a transverse direction, and the board has an upper region in which the high-voltage part of the driving circuit is implemented and a lower region in which the low-voltage part of the driving circuit is implemented. having. A transformer of the control circuit can be installed at a point on the circuit board that defines the transition between the low-voltage part and the high-voltage part of the control circuit. Viewed in the longitudinal direction of the board, the transformer can be arranged on one side of the board in a border area between the upper (high-voltage) region and the lower (low-voltage) region of the board. The remaining part of the board has a longitudinally extending slot in the boundary region between the upper (high-voltage) region and the lower (low-voltage) region of the board, which separates the upper region of the board from the lower region of the board in order to avoid potential creep paths for leakage currents between the upper (high voltage) area and the lower (low voltage) area of the board. This makes it possible to prevent leakage currents and/or arcing from the high-voltage part of the control circuit into the low-voltage part of the control circuit.

In a further embodiment of the circuit board, the control circuit comprises a high-voltage cascade, which is implemented in the upper region of the circuit board using discrete components. The high-voltage cascade consists of a large number of capacitors and diodes, with some of the capacitors in a capacitor strip at an upper edge of the upper area of the board and another part of the capacitors are arranged in a longitudinally extending capacitor strip at a lower edge of the upper region of the board, which is located next to the longitudinally extending slot in the board. The diodes of the high-voltage cascade are arranged in a plurality of transversely extending diode strips between the lower edge and the upper edge of the upper region of the board. The board has recesses between the respective diode strips and the capacitor strips at the bottom edge and at the top edge of the upper area in order to prevent leakage currents and/or arcing between the diodes and capacitors of the high-voltage cascade of the control circuit.

In a further embodiment, groups of discrete diodes are connected in series in a zigzag-shaped arrangement in each diode strip, with slots in the board extending from the recesses extending between the individual discrete diodes of the groups in order to prevent leakage currents and/or arcing between the discrete ones To prevent diodes. In an advantageous development, the anodes and cathodes of the discrete diodes in the groups extend in the longitudinal direction. The slots starting from the recesses also extend in the longitudinal direction in the board between adjacent diodes.

Furthermore, in an advantageous development of the embodiment, the capacitors can be arranged in groups in the capacitor strips at the lower edge and at the upper edge of the upper area, the circuit board having a slot extending in the transverse direction between the two poles of each discrete capacitor in order to prevent leakage currents and/or to prevent arcing between the two poles of the discrete capacitors.

According to a further advantageous development of the embodiment, one of the diode strips is arranged in the longitudinal direction next to the transformer. The circuit board has a transversely extending slot which extends between the one diode strip and the transformer in order to prevent leakage currents and/or arcing between the diodes of the one diode strip and the secondary-side connections of the transformer in the high-voltage part of the drive circuit . Starting from the slot extending in the transverse direction, slots in the board can extend in the longitudinal direction between the adjacent discrete diodes of one diode strip in order to prevent leakage currents between the discrete diodes of the one diode strip.

In a further embodiment, the board has a further slot in the lower region of the board on an output side of the high-voltage cascade, which is essentially at least partially parallel to the longitudinally extending slot in the board, which separates the upper region of the board from the lower region of the Circuit board separates, runs. This further slot can start from an edge in the lower region of the board that delimits the board on the output side of the high-voltage cascade and can extend away from the edge. The additional slot can therefore interrupt the edge of the board.

As already stated, an essential aspect of this invention is room air purifiers, as shown in particular in the patent application WO 2021/224017 A1 are known to improve. In particular, an electronic control should be proposed in order to ensure sufficient cleaning results on the one hand and, on the other hand, to ensure the high demands on the safety of the people in the room to be cleaned with regard to the use of a liquid that wets the counter electrode. In particular, this aspect of the invention relates to a device that is independent and/or can be combined with a control circuit described herein, namely a room air purifier, and a method for treating, in particular humidifying, cleaning and/or washing, air. This device can be a humidifier, an air purifier, an air washer, or the like. These generic room air purifiers, also called air treatment devices, serve to process, in particular to humidify, clean and/or wash, air that is present in closed rooms and/or buildings. This air treatment device can have numerous areas of application, for example in medical technology or in the healthcare industry, in particular in doctor's offices, isolation rooms, hospital rooms, intensive care units or clean rooms, in private households, in particular in bedrooms, living rooms, kitchens or children's rooms, in public buildings and industrial buildings such as museums, theaters , government buildings, office spaces, classrooms, and/or universities, and/or in mobility, for example for vehicle interior cleaning, especially in taxis, rental cars, vehicle sharing concepts. For example, the so-called room air purifiers are free-standing devices and/or small electrical devices which can be placed on the floor in buildings or rooms or on shelves such as tables and which do not weigh more than 15 kilograms.

Particularly effective cleaning of indoor air is achieved through the use of so-called Air purifier achieved with the electrostatic separation technology, in which a liquid storage is additionally provided in order to at least partially, in particular completely, wet, in particular wash around, the counter electrode of the electrostatic precipitator with a flowable mass, such as a liquid. During operation of the room air purifier, the particles electrically charged by the electric separator are attracted to its counter electrode and can thus be caught and transported away in the liquid wetting on the counter electrode, which can in particular be designed as a continuously flowing liquid film, in particular while the air flow cleaned therefrom continues separately and finally released back into the environment. The liquid is generally a flowable rinsing and/or collector medium, for example water, in particular rainwater, a hygroscopic collecting material, such as sodium hydroxide dissolved in a liquid, a gel, which is heated to a certain temperature, for example , so that a liquid state of aggregation is achieved, such as a wax or the like, an ionic liquid, such as melted or dissolved salts, or even highly viscous oils that are mixed with electrically conductive particles, such as copper, are used. For example, the liquid may have a predetermined minimum electrical conductivity, for example of at least 0.005 S/m. The liquid storage can be designed as a local liquid storage. By local it is meant that the liquid storage is part of the room air purifier and/or is directly assigned to it, in contrast to a separate liquid storage or a separate liquid supply. For example, the liquid storage is arranged below the electrical separator. The liquid can then be pumped upwards, for example to the top of the counterelectrode, using a pump and then returned to the liquid reservoir via the counterelectrode in a structurally simple manner using the weight force. The particles separated by the electrostatic precipitator can be carried along by the liquid, transported to the liquid storage and collected there.

Further embodiments of the invention relate to a device and a method for treating, in particular humidifying, cleaning and/or washing, air. Such a device can in particular be a humidifier, a room air purifier, an air washer or the like. The device according to the invention comprises an electrical separator, as described in particular above. The electrostatic precipitator includes an emission electrode arrangement consisting of a plurality of emission needles and a counter electrode. The emission electrode arrangement and the counter electrode are electrically coupled to the control device according to the invention.

An embodiment of the invention accordingly provides a device, in particular a room air purifier, for treating air. This device comprises an electrical separator with an emission electrode, which is formed from a plurality of emission needles and one of the counter electrodes, which is formed at a distance from the emission needles and which is at least partially wetted, preferably surrounded by a liquid; and a control unit that operates the electrical separator. The control unit comprises a control circuit which is set up to apply a high voltage, in particular a direct voltage in the high voltage range, to the emission needles of the emission electrode and the counter electrode of the electrical precipitator in order to generate direct current plasma between the emission electrode and the counter electrode.

According to a further embodiment of the device, the control circuit comprises a control circuit which is set up to set the output power of the control circuit to an operating point relative to the reference current value based on a measured value of a direct current flowing through the emission electrode via the direct current plasma to the counter electrode of the electrical separator to the common reference potential of the control circuit set in which a spark discharge of the direct current plasma between the emission needles of the emission electrode and the counter electrode is prevented.

According to the invention, the control circuit comprises a current measuring unit, wherein the current measuring unit measures the direct current flowing through the emission electrode via the direct current plasma to the counter electrode of the electrical separator to the common reference potential of the control circuit.

According to the invention, the control circuit comprises a control unit which receives the measured value of the flowing direct current from the current measuring unit and generates a control signal, the control unit being set up to control the frequency of the control signal based on the measured value of the flow through the emission needles of the emission electrode via the direct current plasma to the counter electrode of the Electrical separator to vary the direct current flowing to the common reference potential of the control circuit and the reference current value in order to flow to the common reference potential of the control circuit and the electrical separator to regulate the direct current to the reference current value.

In a further embodiment of the device, the control circuit is arranged adjacent to the emission electrode and in particular distal to the counter electrode.

In a further embodiment of the device, the control unit and/or the control circuit drives a fan.

• 1 zeigt einen exemplarischen Systemaufbau gemäß einer Ausführungsform der Erfindung, in dem eine Ansteuerschaltung gemäß einer Ausführungsform der Erfindung von einer Gleichspannungsquelle mit einer Niederspannung gespeist wird, und diese in eine Hochspannung umsetzt, um die Emissionselektroden und die Gegenelektrode eines Elektroabscheiders mit einer Hochspannung zu beauftragen;
• 2 zeigt eine beispielhafte Ausführungsform der Ansteuerschaltung der 1;
• 3 zeigt eine erste beispielhafte Ausführungsform und Verschaltung von der Leistungsschaltern der 2, um die primärseitige Niederspannung mittels des Transformators hochzusetzen;
• 4a zeigt eine zweite beispielhafte Ausführungsform und Verschaltung von der Leistungsschaltern der 2, um die primärseitige Niederspannung mittels des Transformators hochzusetzen;
• 4b zeigt eine dritte beispielhafte Ausführungsform und Verschaltung von der Leistungsschaltern der 2, um die primärseitige Niederspannung mittels des Transformators hochzusetzen;
• 5 zeigt eine erste beispielhafte Ausführungsform einer Hochspannungskaskade in der 2;
• 6 zeigt eine zweite beispielhafte Ausführungsform einer Hochspannungskaskade in der 2;
• 7 zeigt eine erste beispielhafte Ausführungsform einer der Treiberschaltungen des Treibers in der 2;
• 8a zeigt eine zweite beispielhafte Ausführungsform einer der Treiberschaltungen des Treibers in der 2, wenn es sich bei dem zu treibenden Leistungsschalter um PMOS MOSFETs handelt;
• 8b zeigt eine zweite beispielhafte Ausführungsform einer der Treiberschaltungen des Treibers in der 2, wenn es sich bei dem zu treibenden Leistungsschalter um NMOS MOSFETs handelt;
• 9 zeigt eine erste beispielhafte Ausführungsform der Regelung der Ausgangsleistung der Ansteuerschaltung in 2 basierend auf dem über die Emissionselektroden zur Gegenelektrode des Elektroabscheiders zu einem Referenzpotenzial (GND) fließenden Gleichstrom;
• 10 zeigt eine zweite beispielhafte Ausführungsform der Regelung der Ausgangsleistung der Ansteuerschaltung in 2 basierend auf dem über die Emissionselektroden zur Gegenelektrode des Elektroabscheiders zu einem Referenzpotenzial (GND) fließenden Gleichstrom und einer gemessenen Temperatur der der Ansteuerschaltung;
• 11 zeigt ein exemplarisches Eingangsfilter zur Verwendung im Leistungsbegrenzer 212 der 2 gemäß einer beispielhaften Ausführungsform der Erfindung;
• 12 zeigt einen exemplarischen Aufbau eines Transformators gemäß einer Ausführungsform der Erfindung;
• 13 zeigt den Frequenzgang eines beispielhaften Transformators gemäß einer Ausführungsform der Erfindung;
• 14a zeigt eine Ausführungsform einer Platine, die eine Ansteuerschaltung gemäß einer Ausführungsform der Erfindung implementiert;
• 14b zeigt die Ausführungsform der Platine der 14a, wobei die in 2 gezeigten Komponenten der die eine Ansteuerschaltung gekennzeichnet sind;
• 15 zeigt Teilbereiche der Platine der 14a und die Anordnung von Schlitzen und Aussparungen in der Platine;
• 16 illustriert beispielhaft die Realisierung der Hochspannungskaskade, der Impedanzanpassung und des Transformators in 5 auf der Platine der 14a; und
• 17 zeigt eine vergrößerte Ansicht von Teilbereichen im Hochspannungsbereich der Platine der 14a und die Anordnung von Schlitzen und Aussparungen im Hochspannungsbereich der Platine; und
• 18 eine schematische, perspektivische Schnittansicht eines Ausschnitts eines beispielhaften Raumluftreinigers gemäß einer Ausführungsform der Erfindung, der eine Ansteuerschaltung gemäß einer Ausführungsform der Erfindung beinhaltet; und
• 19 eine schematische Schnittansicht von der Seite des Raumluftreinigers aus 18.

Further features and advantages of the invention emerge from the following description, in which preferred embodiments of the invention are explained using schematic drawings. Corresponding elements/functions are designated with identical reference numerals in the figures. This shows:

• 1 shows an exemplary system structure according to an embodiment of the invention, in which a drive circuit according to an embodiment of the invention is fed by a DC voltage source with a low voltage and converts this into a high voltage in order to charge the emission electrodes and the counter electrode of an electrical precipitator with a high voltage;
• 2 shows an exemplary embodiment of the control circuit 1 ;
• 3 shows a first exemplary embodiment and connection of the circuit breakers 2 to increase the primary side low voltage using the transformer;
• 4a shows a second exemplary embodiment and connection of the circuit breakers 2 to increase the primary side low voltage using the transformer;
• 4b shows a third exemplary embodiment and connection of the circuit breakers 2 to increase the primary side low voltage using the transformer;
• 5 shows a first exemplary embodiment of a high-voltage cascade in the 2 ;
• 6 shows a second exemplary embodiment of a high-voltage cascade in the 2 ;
• 7 shows a first exemplary embodiment of one of the driver circuits of the driver in the 2 ;
• 8a shows a second exemplary embodiment of one of the driver circuits of the driver in the 2 , if the power switch to be driven is PMOS MOSFETs;
• 8b shows a second exemplary embodiment of one of the driver circuits of the driver in the 2 , if the power switch to be driven is NMOS MOSFETs;
• 9 shows a first exemplary embodiment of the control of the output power of the control circuit in 2 based on the direct current flowing via the emission electrodes to the counter electrode of the electrostatic precipitator to a reference potential (GND);
• 10 shows a second exemplary embodiment of the control of the output power of the control circuit in 2 based on the direct current flowing via the emission electrodes to the counter electrode of the electrical separator to a reference potential (GND) and a measured temperature of the control circuit;
• 11 shows an exemplary input filter for use in the power limiter 212 of 2 according to an exemplary embodiment of the invention;
• 12 shows an exemplary structure of a transformer according to an embodiment of the invention;
• 13 shows the frequency response of an exemplary transformer according to an embodiment of the invention;
• 14a shows an embodiment of a circuit board that implements a drive circuit according to an embodiment of the invention;
• 14b shows the embodiment of the circuit board 14a , where the in 2 shown components which are marked with a control circuit;
• 15 shows parts of the circuit board 14a and the arrangement of slots and recesses in the board;
• 16 exemplifies the implementation of the high-voltage cascade, the impedance matching and the transformer in 5 on the circuit board of the 14a ; and
• 17 shows an enlarged view of partial areas in the high-voltage area of the circuit board 14a and the arrangement of slots and recesses in the high voltage area of the board; and
• 18 a schematic, perspective sectional view of a section of an example adhesive room air purifier according to an embodiment of the invention, which includes a control circuit according to an embodiment of the invention; and
• 19 a schematic sectional view from the side of the room air purifier 18 .

The invention relates to a control circuit for generating a high voltage which is applied to the electrodes of an electrical separator. Another aspect is the use of such a control circuit in a device with an electrical separator. This device can be, for example, an air treatment device, e.g. a room air filter, a humidifier, an air washer, an air sterilizer, an aerosol filter or a fine dust filter.

The control circuit can in principle be used in all end devices that separate particles or liquids or impurities from a gas stream using a low-energy plasma. An example is the PCT registration WO 2021/224017 A1 and the patent application DE 10 2021 128 345.0 of the applicant dated October 29, 2021, which describes an air treatment device with an electrical separator in which the control circuit according to the invention can be used. The disclosure of the PCT application WO 2021/224017 A1 and the disclosure of the patent application DE 10 2021 128 345.0 is hereby incorporated in its entirety by referencing.

An electrostatic precipitator essentially works on the following principle: release of electrical charges, especially electrons or positive ions; Charging particles that may be present in the air in an electric field; Transport of the electrically charged particles to an opposite pole; discharge of the charged particles at the opposite pole; and removal of the particles from the opposite pole. For example, the principle of charge generation underlying the electrostatic precipitator can be impact ionization. When a so-called corona field strength is exceeded, electrons can be released and interact with the surrounding gas molecules in the air, whereby a corona is formed. Whether this is a positive or negative corona depends on the high voltage applied to the electrodes. Free electrons present in the air are strongly accelerated in the electrostatic field of the corona, so that a gas discharge can occur. When it hits gas molecules in the air, additional electrons can be split off or attach to the gas molecules. The negative charges move within the air treatment device, in particular within the electrostatic precipitator. When a particle-laden air stream enters, the negatively charged charges attach to the particles. Due to the acting electrostatic force of the applied DC voltage field, which can be oriented transversely to the direction of flow of the air within the device, the negatively charged particles are deflected and can thus be separated from the air flow.

A further aspect and further embodiments of the invention, which are described herein, relate to the use of the control circuit according to the invention in devices, in particular air treatment devices with an electrical separator. In particular, the invention relates to the use of such a control circuit in a room air filter comprising an electrical separator with an emission electrode, which is formed from a plurality of emission needles and one of the counter electrodes, which is formed at a distance from the emission needles and which is preferably surrounded by a liquid, at least partially wetted . The embodiments of the invention can in particular air treatment devices with a volume flow rate of the air to be treated (at a CADR of more than 250 inclusive) up to and including 500 m ³ /h, preferably up to and including 350 m ³ /h and more preferably up to and including 300 m ³ /h regarding. Further embodiments may relate to air treatment devices in which the output power of the control circuit is in the range between 0.5 W and 20 W, preferably between 0.7 W and 17 W.

The control circuit can generate a high voltage, the magnitude of the direct component of which is in the range between 8 kV and 17 kV inclusive, preferably between 10 kV and 13 kV inclusive. The direct component of the direct current that the control circuit provides to an electrical separator is in the range between 0.100 µA and 1.000 µA, preferably in the range between 0.150 µA and 0.400 µA. However, the invention is not limited to these power ranges, voltage ranges and/or direct current ranges. The exact operating parameters with regard to power, output voltage and/or output current of the control circuit result from the area of application of the control circuit and may differ from the specified ranges. However, the specified ranges and parameters are typical for the use of the control circuit for cabin air filters or fine dust filters.

1 shows an exemplary system structure according to an embodiment of the invention. The system includes a control circuit 120 according to the invention, which is powered by a DC voltage Voltage source 110 is fed with a low voltage V _{DC_ext} . The DC voltage source 110 can be, for example, a commercially available 12 V direct current (DC) power supply or a battery. The control circuit 120 implements a DC/DC converter that increases the input-side low voltage V _{DC_ext} (or a voltage V' _{DC_ext} derived therefrom) into a high voltage V _plasma . The high voltage is applied to the emission electrodes 130 and the counter electrode 140 of an electrical precipitator on the output side to generate a direct current plasma between the respective emission electrodes 130 and the counter electrode 140. As mentioned at the beginning, the direct current plasma is used to separate particles/impurities in a gas stream, as is described in detail, for example, in the already mentioned PCT application WO 2021/224017A1 is described. The individual emission electrodes 130 can each be connected in series with a series resistor (not shown) in order to limit the current flow through the respective emission electrode 130.

By applying the high voltage V _plasma to the emission electrodes 130 and the counter electrode 140 of the electrical precipitator, a direct current I _plasma flows via the emission electrodes 130, the direct current plasma and the counter electrode 140. In the embodiment shown, it is assumed, for example, that the high voltage V _plasma generated by the control circuit 120 is a negative DC voltage. Accordingly, the arrow of the direct current I shows _plasma in the 1 and the other figures show the technical flow of electricity. In principle, however, it is also conceivable that the high voltage V _plasma is a positive direct voltage that is generated by the control circuit 120. However, applying a negative direct voltage has the advantage that it tends to achieve a higher separation performance and generates more ozone, which in turn actively supports air purification.

The control circuit 120 further includes a control circuit 122, which regulates the output power (P = I _plasma · V _plasma ) of the control circuit 120. The control circuit 122 regulates the direct current I _plasma flowing on the output side to the electrical separator to a desired reference current value (I _Ref ). The counter electrode 140 of the electrical separator as well as one end of the secondary-side coil of a transformer in the control circuit 120 can, as will be explained in more detail below, be connected to the reference potential (GND) of the control circuit 120, so that the direct current I _plasma flowing to the reference potential on the output side is detected and can be supplied to the control circuit 122 as an input variable. Based on the measured direct current I _plasma , the control circuit 120 varies the frequency f[n] of a control signal with which the power generator in the control circuit 120 controls the primary-side coil(s) of a transformer, so that the desired reference current value (I _Ref ) is set on the output side and the output power of the control circuit is regulated to the desired value. The cycle time of the control loop is chosen to be very short and is preferably in the range of 5 ms or less, preferably 2 ms or less, more preferably 1 ms or less. The shorter the control cycle, the faster fluctuations in measured direct current can be detected and corrected.

2 shows an exemplary embodiment of the control circuit 120 1 . The control circuit 120 is supplied on the input side by the DC voltage source 110 with a low voltage V _{DC_ext} . In the embodiments of the invention, the low voltage is, for example, in the range of a few 10 V, ie it is preferably including 50 V or less, particularly preferably including 20 V or less. The input voltage V _{DC_ext} is supplied to a power limiter 212 of the control circuit 120. The power limiter 212 contains, for example, a circuit that limits the power consumed by the control circuit 120, which limits the power flowing into the control circuit 120, particularly in the event of a short circuit on the output side (for example when the direct current plasma breaks down), and thus protects its components from destruction. The power limiter can also be used as described below in connection with 11 is discussed even more, have an input filter that, for example, suppresses conducted interference using an EMC filter. In the exemplary embodiment shown, the power limiter 212 provides a low voltage V' _{DC_ext} derived from the input voltage V _{DC_ext} . In the embodiments of the invention, the derived low voltage V' _{DC_ext} is, for example, in the range of a few 10 V, ie it is preferably 50 V or less, particularly preferably 20 V or less. It should be noted here that the derived low voltage V' _{DC_ext} is defined with respect to the reference potential GND of the control circuit 120, while the input-side low voltage V _{DC_ext} can also be defined with respect to another reference potential (eg the negative pole of a battery). In the embodiment shown, the power limiter 212 also provides a TTL or CMOS supply voltage V _TTL derived from the low voltage V _{DC_ext} on the input side for supplying the digital components (for example a microprocessor in the control unit 214) and/or components responsive to the TTL/CMOS level. Components in the driver 216 or the circuit breaker (e.g. NMOS-based parts/components). Typical voltage values for the TTL or CMOS supply voltage V _TTL are, for example, 5.0 V, 3.3 V, 2.5 V, 1.8 V, etc. In principle, it is also conceivable that the power limiter 212 has different TTL or CMOS as required -Supply voltages are available.

In the exemplary embodiment shown, the control circuit 122 consists of the "controlled system", which includes the control unit 214, the driver 216, the power switches 218, the transformer 220, the optional impedance matching 252 and the high-voltage cascade 254, so that between the electrodes 130, 140 of the Electrical separator a direct current plasma is formed. The control circuit also includes the current measuring unit 256, which measures the controlled variable, the plasma current I _Plasma , and returns it to the controller, the control unit 214.

The control unit 214 varies the frequency of the control signals (manipulated variable) that are supplied to the driver 216 by the control unit 214 based on the measured plasma current I _plasma . The control unit 214 can further be set up to adjust the frequency of the control signal based on the value of the plasma current I _Plasma and the reference current value I _Ref (not shown) to vary the direct current I _Plasma flowing to the common reference potential GND of the control circuit 120 and the electrical separator to the reference current value I _Ref . If the output plasma current I _Plasma increases in comparison to the reference current value I _Ref , the control unit 214 reduces the frequency of the control signals. If the output plasma current I _Plasma falls compared to the reference current value I _Ref , the control unit 214 increases the frequency of the control signals.

wobei das Steuersignal $\overline{V_{PWM}\left(f\right)}$

das invertierte Steuersignal V_PWM(f) ist. Die Steuersignale V_PWM(f) und V_PWM(f) der Regeleinheit 214 sind pulsweitenmodulierte Signale, die einen Tastgrad von ca. 50 %, bevorzugt 50 % aufweisen. Die Steuersignale V_PWM(f) und $\overline{V_{PWM}\left(f\right)}$

weisen einen TTL oder CMS Signalpegel auf, der der TTL oder CMOS-Versorgungsspannung V_TTL entspricht.The control unit 214 can be designed, for example, by means of a microprocessor, a discrete circuit or an integrated circuit (e.g. ASIC or programmable logic (e.g. FPGA, PLD, etc.) or combinations of the options mentioned. In the exemplary embodiment shown, the control unit 214 generates the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)},$

where the control signal $\overline{v_{PWM}\left(f\right)}$

is the inverted control signal V _PWM (f). The control signals V _PWM (f) and V _PWM (f) of the control unit 214 are pulse width modulated signals that have a duty cycle of approximately 50%, preferably 50%. The control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

have a TTL or CMS signal level that corresponds to the TTL or CMOS supply voltage V _TTL .

in den genannten Bereich regeln. Ein vorgegebener Frequenz-Referenzwert in diesem Bereich entspricht dem Referenzstromwert I_Ref. Entsprechend verringert oder erhöht die Regeleinheit 214 die Frequenz der Steuersignale V_PWM(f) und $\overline{V_{PWM}\left(f\right)}$

um diesen Frequenz-Referenzwert, je nachdem, ob der ausgangsseitige Plasmastrom I_Plasma sich relativ zum Referenzstromwert I_Ref erhöht oder abfällt. Die Resonanzfrequenz des Transformators 220 ist dabei höher ist als die Frequenz der Steuersignale V_PWM(f) und $\overline{V_{PWM}\left(f\right)}.$

Die Resonanzfrequenz des Transformators 220 liegt bevorzugt im Bereich einschließlich 200,0 kHz bis 240,0 kHz. Grundsätzlich ist es wünschenswert, die primärseitige Ansteuerfrequenz des Transformators 220 möglichst nahe an die Resonanzfrequenz des Transformators heranzuführen, da der Transformator bei der Resonanzfrequenz die höchste Effizienz bietet. Jedoch lässt sich die Resonanzfrequenz des Transformators 220 nur schwer exakt einstellen, da typische Frequenzgänge eines Transformators im Bereich um die Resonanzfrequenz stark ansteigen, bzw. abfallen. Dadurch würde in diesen „steilen“ Bereichen des Frequenzgangs eine kleine Änderung der Frequenz eine große Änderung der Ausgangsleistung der Anpassungsschaltung 120 bzw. des Plasmastroms bedingen. Diese großen Änderungen der Ausgangsleistung der Anpassungsschaltung 120 bzw. des Plasmastroms kann wiederum nur schwer vom Regelkreis ausgeregelt werden und kann unter Umständen zu einem instabilen Regelverhalten führen. Daher haben die Steuersignale V_PWM(f) und $\overline{V_{PWM}\left(f\right)}$

eine Frequenz, die in einem möglichst linearen Bereich des Frequenzgangs des Transformators 220 liegt, in dem gezeigten Beispiel, im Bereich einschließlich 130,0 kHz bis einschließlich 170,0 KHz.In an exemplary embodiment, the control signals V _PWM (f) and V _PWM (f) have a frequency in the range inclusive of 120.0 kHz up to and including 200.0 kHz, preferably in the range inclusive of 130.0 kHz up to and including 170.0 kHz. The control unit 214 can control the frequency of the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

in the specified area. A predetermined frequency reference value in this range corresponds to the reference current value I _Ref . The control unit 214 correspondingly reduces or increases the frequency of the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

by this frequency reference value, depending on whether the output plasma current I _Plasma increases or decreases relative to the reference current value I _Ref . The resonance frequency of the transformer 220 is higher than the frequency of the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}.$

The resonance frequency of the transformer 220 is preferably in the range 200.0 kHz to 240.0 kHz inclusive. In principle, it is desirable to bring the primary-side drive frequency of the transformer 220 as close as possible to the resonance frequency of the transformer, since the transformer offers the highest efficiency at the resonance frequency. However, the resonance frequency of the transformer 220 is difficult to set exactly, since typical frequency responses of a transformer increase or decrease sharply in the area around the resonance frequency. As a result, in these “steep” areas of the frequency response, a small change in frequency would cause a large change in the output power of the matching circuit 120 or the plasma current. These large changes in the output power of the matching circuit 120 or the plasma current can only be adjusted with difficulty by the control circuit and can, under certain circumstances, lead to unstable control behavior. Therefore, the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

a frequency that lies in the most linear possible range of the frequency response of the transformer 220, in the example shown, in the range including 130.0 kHz up to and including 170.0 KHz.

im Bereich vom mehr als 100 kHz und aufgrund der parasitären Impedanzen, wie Wicklungskapazitäten oder Streuinduktivitäten des Transformators 220, können dazu führen, dass in dem auf der primärseitigen Seite des Transformators 220 liegenden Schaltungsbereich der Ansteuerschaltung 120 (Niederspannungsteil 210) leitungsgebundene Störungen auftreten. Die Streuinduktivitäten des Transformators 220 werden mit steigendem Miniaturisierungsgrad des Transformators relevanter. Durch die geringere Querschnittsfläche steigt die Streuinduktivität und damit abgestrahlte Magnetfelder an. Due to the high operating frequency of the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

in the range of more than 100 kHz and due to the parasitic impedances, such as winding capacitances or leakage inductances of the transformer 220, can lead to line-related interference occurring in the circuit area of the control circuit 120 (low-voltage part 210) located on the primary side of the transformer 220. The leakage inductances of the transformer 220 become more relevant as the level of miniaturization of the transformer increases. Due to the smaller cross-sectional area, the leakage inductance and thus the radiated magnetic fields increase.

This can cause EMC problems and reduce the efficiency of the transformer 220. The additional heat dissipation problem therefore makes the miniaturization of the transistor 220 challenging. As the degree of miniaturization increases, the surface area available for dissipating heat from the transformer decreases squarely, ie it is more difficult to transport away the heat generated in the transformer 220. These line-bound interferences resulting from miniaturization can, as will be explained below, be compensated for by means of filtering in the power limiter 212.

werden in der 2 dem Treiber 216 der Ansteuerschaltung 120 zugeführt. Wie nachstehend in Bezug auf die 3 und 4 erläutert wird, dient der Treiber 216 dazu, die Signalpegel der Steuersignale V_PWM(f) und $\overline{V_{PWM}\left(f\right)}$

in Bezug auf Stromstärke und/oder Spannung an die Leistungsschalter 218, die den Transformator 220 antreiben, anzupassen. Dazu umfasst der Treiber 216 mehrere Treiberschaltungen. Exemplarisch sind in der 2 zwei solche Treiberschaltungen 216-1 und 216-2 gezeigt, die entsprechende zwei Leistungsschalter in der Stufe 218 ansteuern. Die Treiberschaltungen 216-1, 216-2 behalten dabei das relative Timing der Steuersignale V_PWM(f) und $\overline{V_{PWM}\left(f\right)}$

über einen vorgegebenen Temperaturbereich, in dem die Ansteuerschaltung betrieben werden soll, bei. Wie eingangs erwähnt, liegt dieser Temperaturbereich zwischen -40 °C und 160 °C, wobei die Erfindung nicht auf diesen Temperaturbereich beschränkt ist. Natürlich kann der Temperaturbereich kleiner sein und/oder mit dem angegebenen Temperaturbereich nur teilweise überlappen. Die Treiberschaltungen 216-1, 216-2 beinhalten eine oder mehrere Stufen von in Reihe geschaltete Pegelumsetzer, um die Steuersignale V_PWM(f) und $\overline{V_{PWM}\left(f\right)}$

auf die gewünschten Signalpegel zu bringen, und mit den angepassten Steuersignalen V'_PWM(f) und $\overline{V{\text{'}}_{PWM}\left(f\right)}$

die Leistungsschalter der Stufe 218 anzusteuern. Die Treiberschaltungen 216-1, 216-2 behalten die Frequenz Steuersignale V_PWM(f) und $\overline{V_{PWM}\left(f\right)}$

bei, sodass die angepassten Steuersignale V'_PWM(f) und $\overline{V{\text{'}}_{PWM}\left(f\right)}$

die gleiche Frequenz (und Tastgrad) wie die Steuersignale V_PWM(f) und $\overline{V_{PWM}\left(f\right)}$

aufweisen.The control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

will be in the 2 supplied to the driver 216 of the control circuit 120. As below in relation to the 3 and 4 As explained, the driver 216 serves to control the signal levels of the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

in terms of current and/or voltage to the power switches 218 that drive the transformer 220. For this purpose, the driver 216 includes several driver circuits. Examples are in the 2 two such driver circuits 216-1 and 216-2 are shown, which control corresponding two power switches in stage 218. The driver circuits 216-1, 216-2 maintain the relative timing of the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

over a predetermined temperature range in which the control circuit is to be operated. As mentioned at the beginning, this temperature range is between -40 ° C and 160 ° C, although the invention is not limited to this temperature range. Of course, the temperature range can be smaller and/or only partially overlap with the specified temperature range. The driver circuits 216-1, 216-2 include one or more stages of series-connected level shifters to drive the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

to bring it to the desired signal levels, and with the adapted control signals V' _PWM (f) and $\overline{v{\text{'}}_{PWM}\left(f\right)}$

to control the circuit breakers of level 218. The driver circuits 216-1, 216-2 maintain the frequency control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

so that the adapted control signals V ' _PWM (f) and $\overline{v{\text{'}}_{PWM}\left(f\right)}$

the same frequency (and duty cycle) as the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

exhibit.

The circuit breakers 218 alternately charge the primary-side coil(s) of the transformer 220 with the low voltage V' _{DC_ext} , so that an alternating voltage in the high-voltage range is output in the secondary-side coil of the transformer 220. For example, the output voltage V _transformer of the transformer 220 is in the range of 2 kV or more, preferably 2.5 kV or more. This output voltage on the secondary side of the transformer 220 is then passed through an impedance matching 252 and rectified by a high-voltage cascade 254 and at the same time converted to the desired output DC voltage V _plasma , which can be applied to the emission electrodes 130 and the counter electrode 140 of the electrical precipitator, as in 1 shown. In an exemplary embodiment, the high-voltage cascade 254 is designed to convert the secondary-side output voltage of the transformer 220 into a direct voltage V _plasma in the high-voltage range, whose alternating component (AC component) compared to its direct voltage component (DC component) is 4% or less, preferably 3%. or less, more preferably 2% or less. The DC voltage component of the DC voltage V _plasma at the output of the cascade 254 is, for example, in the range between 8 kV and 17 kV inclusive, preferably between 10 kV and 13 kV inclusive. In the example shown, the DC voltage V _Plasma at the output of the cascade 254 is a negative voltage, so the technical current direction of the DC I _Plasma is directed towards the cascade 254. The direct component of the plasma current I _Plasma that the control circuit 120 can provide to the electrical separator is, for example, in the range between 0.100 µA and 1.000 µA, preferably in the range between 0.150 µA and 0.400 µA. Exemplary embodiments of the high-voltage cascade 254 are discussed below with respect to 5 and 6 described.

Like in the 2 indicated, the control circuit 120 also includes a current measuring unit 256, which measures the flowing plasma current I _plasma on the output side and returns it to the control unit 214 as a controlled variable. The detection of the plasma current I _plasma is related to the 9 and 10 described in more detail.

The control circuit 120 can be divided into a low-voltage part 210 and a high-voltage part 250. The high-voltage part 250 includes the secondary-side coil of the transformer 220 and the downstream parts of the control circuit 120, in particular the impedance matching 252, the high-voltage cascade 254 and the current measuring unit 256. The low-voltage part 210 of the control circuit 120 includes the power limiter 212, the control unit 214, the driver 216, the circuit breakers 218 and the primary-side coil(s) of the transformer 220. The transformer 220 thus forms the transition from the low-voltage area 210 to the high-voltage area 250.

mit dem Referenzpotenzial GND verbunden. Steuersignale V'_PWM(f) und $\overline{V{\text{'}}_{PWM}\left(ƒ\right)}$

sind dazu mit dem Gate-Anschluss der Leistungsschalter 302 und 304 verbunden, die in dem gezeigten Ausführungsbeispiel durch NMOS MOSFETs gebildet werden. Wenn die Leistungsschalter 302 und 304 leiten, wird das jeweilige Ende, das mit dem Drain-Anschluss des jeweiligen NMOS MOSFETs 302, 304 verbunden ist, mit dem Referenzpotenzial GND verbunden. Die so erzeugte Wechselspannung an den primärseitigen Spulen des Transformators 220 induziert somit eine Wechselspannung in der sekundärseitigen Spule des Transformators 220. Wie in der 3 ebenfalls exemplarisch dargestellt, kann auf Seiten der sekundärseitigen Spule des Transformators 220 eine Impedanzanpassung 252 vorgesehen sein. Diese wird in dem gezeigten Ausführungsbeispiel exemplarisch durch die Kapazität 310 dargestellt, die zwischen die beiden Ausgänge der sekundärseitigen Spule des Transformators 220 geschalten werden. Diese Kapazität kann beispielsweise mittels eines oder mehrerer in Reihe geschalteter diskreter Kondensatoren gebildet werden. Die Impedanzanpassung 252 dient dazu, die Impedanz zwischen den Leistungsschaltern 218, dem Transformator 220 unter der nachgeschalteten Hochspannungskaskade 254 aneinander anzupassen. Dazu ist anzumerken, dass die parasitären Impedanzen, wie Wicklungskapazitäten oder Streuinduktivitäten, des Transformators 220 sich nur schwer korrekt simulieren lassen und in aller Regel analytisch bestimmt werden müssen. Der Aufbau des Transformators 220 ist zwar nicht sonderlich komplex, ist aber vorzugsweise in seiner Konstruktion auf das Gesamtsystem abgestimmt. 3 shows a first exemplary embodiment and connection of the circuit breakers 218 2 to increase the primary-side low voltage V' _{DC_ext} using the transformer 220. In the embodiment shown, the transformer 220 includes two primary-side coils that are wound in the same direction and connected in series. The center tap of the primary-side coils, which is formed at the connection point of the first coil and the second coil, is connected to the low voltage V' _{DC_ext} . The other ends of the two coils are alternately controlled based on the control signals V' _{and PWM} (f). $\overline{v{\text{'}}_{PWM}\left(ƒ\right)}$

connected to the reference potential GND. Control signals V' _PWM (f) and $\overline{v{\text{'}}_{PWM}\left(ƒ\right)}$

are connected to the gate connection of the power switches 302 and 304, which are formed by NMOS MOSFETs in the exemplary embodiment shown. When the power switches 302 and 304 conduct, the respective end connected to the drain of the respective NMOS MOSFET 302, 304 is connected to the reference potential GND. The alternating voltage thus generated on the primary-side coils of the transformer 220 thus induces an alternating voltage in the secondary-side coil of the transformer 220. As in the 3 Also shown as an example, an impedance matching 252 can be provided on the secondary side coil of the transformer 220. In the exemplary embodiment shown, this is exemplified by the capacitance 310, which is connected between the two outputs of the secondary-side coil of the transformer 220. This capacitance can be formed, for example, by means of one or more discrete capacitors connected in series. The impedance matching 252 serves to match the impedance between the power switches 218, the transformer 220 under the downstream high-voltage cascade 254. It should be noted that the parasitic impedances, such as winding capacitances or leakage inductances, of the transformer 220 are difficult to simulate correctly and generally have to be determined analytically. Although the structure of the transformer 220 is not particularly complex, its construction is preferably tailored to the overall system.

4a shows a second exemplary embodiment and the connection of the circuit breakers 218 2 to increase the primary-side low voltage V' _{DC_ext} using the transformer 220. In the embodiment shown, the transformer 220 includes a primary-side coil and a secondary-side coil. The primary-side coil can also be formed from two coils that are wound in the same direction and connected in series. The coils can be pushed into one another so that the size can be reduced. In the in 4a In the exemplary embodiment shown, the power switches 218 comprise a full bridge (H-bridge) of four power switches 402, 404, 406, 408. It has been shown that the primary-side coil(s) of the transformer 220 can be controlled with power switches in an H-bridge are connected, line-related interference compared to the in 3 shown solution can reduce.

angesteuert werden. Durch diese Ansteuerung wird abwechselnd die Niederspannung V'_{DC_ext} an ein Ende der primärseitigen Spule des Transformators 220 angelegt, während das jeweils andere Ende mit dem Referenzpotenzial GND verbunden wird. So wird beispielsweise in einem ersten Zyklus der Leistungstransistor 402 und der Leistungstransistor 408 geöffnet, sodass die Niederspannung V'_{DC_ext} über den geöffneten Leistungstransistor 402 am zweiten Ende der primärseitigen Spule anliegt und das erste Ende der primärseitigen Spule über den Leistungstransistor 408 mit dem Referenzpotenzial GND verbunden ist. Im nachfolgenden Zyklus sind der Leistungstransistor 404 und der Leistungstransistor 406 geöffnet, sodass die Niederspannung V'_{DC_ext} über den geöffneten Leistungstransistor 404 am ersten Ende der primärseitigen Spule anliegt und das zweite Ende der primärseitigen Spule über den Leistungstransistor 406 mit dem Referenzpotenzial GND verbunden ist.In the exemplary embodiment shown, the power switches 218 are designed as MOSFETs, the power switches 402 and 404 are PMOS MOSFETs, and the power switches 406 and 408 are NMOS MOSFETs. The power switches 402 and 408 are controlled by corresponding driver circuits 216 with the control signal V ' _PWM (f), while the power switches 404 and 406 are controlled by corresponding driver circuits 216 with the control signal $\overline{v{\text{'}}_{PWM}\left(ƒ\right)}$

be controlled. Through this control, the low voltage V' _{DC_ext} is alternately applied to one end of the primary-side coil of the transformer 220, while the other end is connected to the reference potential GND. For example, in a first cycle, the power transistor 402 and the power transistor 408 are opened, so that the low voltage V' _{DC_ext} is applied to the second end of the primary-side coil via the opened power transistor 402 and the first end of the primary-side coil is connected to the reference potential GND via the power transistor 408 is. In the subsequent cycle, the power transistor 404 and the power transistor 406 are opened, so that the low voltage V' _{DC_ext} is applied to the first end of the primary-side coil via the open power transistor 404 and the second end of the primary-side coil is connected to the reference potential GND via the power transistor 406.

In another embodiment, a capacitor may be connected between the gate and source of each of the power switches 402, ₄₀₄ , 406 and 408. However, this is in 4a Not shown. The use of capacitors between the gate terminal and source terminal of each of the power switches 402, ₄₀₄ , 406 and 408 has been shown to reduce conducted noise compared to that in 3 shown solution can further reduce.

Further optionally, the drain connection of the power switches 402 and ₄₀₄ can not be connected directly to the low voltage V' _{DC_ext} , but via a filter 412. The filter 412 can prevent line-related interference due to parasitic impedances and the high-frequency switching of the power switches 402, ₄₀₄ , Reduce 406 and 408.

The output of the secondary coil of the transformer 220 may be connected to a circuit 252 that matches the impedances of the power switches 218, the transformer 220 and the subsequent high-voltage cascade 254.

in etwa die halbe Niederspannung V'_{DC_ext}/2 ab. Der in die primärseitige Spule fließende Strom I_primär ist aufgrund der Schaffung des virtuellen Massepunkts 418 einer Dreieckspannung mit der halben Frequenz f der Ansteuerungssignale V'_PWM(f) und $\overline{V{\text{'}}_{PWM}\left(ƒ\right)}$

wie dies in der 4b exemplarisch dargestellt ist. Der Vorteil der in 4b gezeigten Ausführungsform liegt darin, dass sich durch die beiden Kondensatoren 414 und 416 und den geschaffenen virtuellen Massepunkt 418 leitungsgebundene Störungen durch die parasitären Impedanzen, insbesondere die Wicklungskapazitäten oder Streuinduktivitäten des Transformators 220, reduzieren lassen. 4b shows a third exemplary embodiment and the connection of the circuit breakers 218 2 to increase the primary-side low voltage V' _{DC_ext} using the transformer 220. The structure of the control of the primary-side coil(s) of the transformer 220 corresponds to that in 4a structure shown. In contrast to 4a are in the embodiment of 4b the two power switches 402 and 406 through the capacitors 414 and 416, which can have the same capacity. This creates a virtual ground point 418 between the two capacitors 414 and 416. Due to the high control frequency caused by the control signals V ' _PWM (f) and $\overline{v{\text{'}}_{PWM}\left(ƒ\right)}$

approximately half the low voltage V' _{DC_ext} /2. The current I flowing into the primary-side coil is _primary due to the creation of the virtual ground point 418 of a delta voltage with half the frequency f of the control signals V ' _PWM (f) and $\overline{v{\text{'}}_{PWM}\left(ƒ\right)}$

like this in the 4b is shown as an example. The advantage of in 4b The embodiment shown is that the two capacitors 414 and 416 and the created virtual ground point 418 can reduce line-related interference caused by the parasitic impedances, in particular the winding capacitances or leakage inductances of the transformer 220.

5 shows a first exemplary embodiment of a high-voltage cascade 254 in the 2 . An exemplary high-voltage cascade 254 is designed as a Villard cascade 500. The output voltage V _transformer of the transformer 220 is present at the input of the cascade 254. In the exemplary embodiment shown, the high-voltage cascade 254 comprises five stages 502, 504, 506, 508 and 510. Each stage 502, 504, 506, 508 and 510 doubles the AC voltage component of the output voltage V _transformer of the transformer 220. The cascade 254 can be noticeably lossy. The diodes have a relevant reverse current, especially at higher temperatures, and the amplitude of the alternating voltage component of the output voltage V _transformer is slightly smoothed. Therefore, a series of any number of cascade stages does not make sense. Although the number of cascade stages of the invention is not limited to five, it is advantageous to form no more than six cascade stages. Cascade steps 502, 504, 506, 508 and 510 form a “ladder”. Each of the cascade stages 502, 504, 506, 508 and 510 consists of two capacitors and two diodes. The second end 514 of the secondary-side coil of the transformer 220 is connected to the reference potential GND (as will be explained below). The first end 512 of the secondary-side coil of the transformer 220 is connected on the input side to the first capacitor of the first cascade stage 502. The other end of the capacitor is connected to the anode of the first diode, the cathode of which is in turn connected to the second end of the secondary coil of the transformer. The anode of the second diode of the first cascade stage 502 is connected to the cathode of the first diode of the first cascade stage 502 via the second capacitor of the first cascade stage 502. The cathode of the second diode is in turn connected to the anode of the first diode and the other end of the capacitor. The other stages 504, 506, 508 and 510 are also structured accordingly. The output voltage V _plasma of the cascade 254 is tapped at the anode of the second diode in the last stage 510 of the cascade 254. In principle, it is also possible to tap the output voltage V _plasma of the cascade 254 at the cathode of the second diode in the last stage 510 of the cascade 254. In this case, the unsmoothed high voltage is obtained - i.e. the DC component of the high voltage generated at this point in the cascade 254 plus an AC voltage component of approximately V _transformer . This option would be conceivable, for example, in applications in which breakdowns of the electrical separator are to be provoked

It should be noted that the arrangement of the cascade stages 502, 504, 506, 508 and 510 shown generates a negative DC voltage V _plasma on the output side. In order to generate a positive DC voltage, the anode and cathode of the individual diodes in stages 502, 504, 506, 508 and 510 simply have to be swapped with each other.

6 shows a second exemplary embodiment of a high-voltage cascade 254 in the 2 . In the exemplary embodiment shown, the high-voltage cascade 254 is designed as a high-voltage cascade 600 with full-wave rectification, which also consists of several stages. In the exemplary embodiment, as in the 5 Five cascade stages 602, 604, 606, 608 and 610 are shown, but here too the number of cascade stages is not limited to five. The structure of the high-voltage cascade 600 is based on a Villard cascade 500, which is mirrored on the signal path between the second end 514 of the secondary-side coil and the output voltage tap.

The two cascades 500 and 600 in the 5 and 6 differ essentially in the number of their elements, which are in the high voltage cascade 600 with full-wave rectification is almost twice as high as that in the Villard cascade 500. The individual diodes and capacitors in the two embodiments of the cascades 500 and 600 can be formed by discrete components. In this case, a single diode can be formed by connecting several discrete diodes in series. The capacitors can also be formed using a series connection of discrete capacitors. With the discrete structure of the cascade 254, the embodiment appears 5 advantageous if the miniaturization of the control circuit 120 is an important design aspect.

zur Verfügung, die jeweils einer Treiberschaltungen 216-1, 216-2 zugeführt werden. Die Anzahl der Treiberschaltungen entspricht dabei der Anzahl der Leistungsschalter 218. In dem in 3 gezeigten Ausführungsbeispiel mit zwei Leistungsschaltern 302, 304 sind somit zwei Treiberschaltungen 216-1, 216-2 vorgesehen, wobei eine der Treiberschaltungen mit dem Steuersignal V_PWM(f) und die andere Treiberschaltung mit dem Steuersignal $\overline{V_{PWM}\left(ƒ\right)}$

der Regeleinheit 214 angesteuert wird. In dem in 4 gezeigten Ausführungsbeispiel mit vier Leistungsschaltern 402, 404, 406 und 408 sind entsprechend vier Treiberschaltungen vorgesehen, wobei zwei der Treiberschaltungen mit dem Steuersignale $V_{PWM}\left(ƒ\right)$

und die anderen beiden Treiberschaltungen mit dem Steuersignal $\overline{V_{PWM}\left(ƒ\right)}$

der Regeleinheit 214 angesteuert werden. 7 shows a first exemplary embodiment of one of the driver circuits 216-1, 216-2 of the driver 216 in the 2 . As already in connection with the 2 executed, the control unit 214 provides the control signals V _PWM (f) and $\overline{v_{PWM}\left(ƒ\right)}$

available, each of which is fed to a driver circuit 216-1, 216-2. The number of driver circuits corresponds to the number of power switches 218. In the in 3 In the exemplary embodiment shown with two power switches 302, 304, two driver circuits 216-1, 216-2 are therefore provided, one of the driver circuits with the control signal V _PWM (f) and the other driver circuit with the control signal $\overline{v_{PWM}\left(ƒ\right)}$

the control unit 214 is controlled. In the in 4 In the exemplary embodiment shown with four power switches 402, 404, 406 and 408, four driver circuits are provided, with two of the driver circuits being connected to the control signals $v_{PWM}\left(ƒ\right)$

and the other two driver circuits with the control signal $\overline{v_{PWM}\left(ƒ\right)}$

the control unit 214 can be controlled.

However, the structure of the driver circuits is identical in the different embodiments. Only the (type of) transistors used in the individual stages of the driver circuits can be designed differently, for example depending on which voltage level is switched or which (type of) power switch controls the respective driver circuit. However, the corresponding resistors and diodes of the individual stages 700, 720 of the driver circuits can be designed to be identical in construction. This has the advantage that the effects of fluctuations in the operating temperature on the resistance values and thus the rise times and fall times of signal edges of the driver circuits 216-1, 216-2 are uniform in all driver circuits 216-1, 216-2 and therefore the switching timing is not negative influenced. In particular, this can ensure reliable operation of the control circuit 120 over the desired temperature range in which the control circuit 120 is to be operated. This makes it possible, in particular, to prevent a possible “shoot through” in the individual stages 700, 720 of the driver circuits 216-1, 216-2.

In the 7 Resistors that correspond to each other in their resistance value are designated with the same value RX. Thus, resistors 708, 712, 710 and 730 in stages 700, 720 have the same resistance value R1 (eg 100 Ω), while resistor 728 has resistance value R2 (eg 330 Ω). Diodes of identical construction are also identically labeled DX. The diodes 706, 714 and 726 can be designed to be identical in construction. However, the identical design of the resistors and the resistance values or the identical design of the diodes and transistors within a driver stage 700, 720 is not crucial; Rather, it is advantageous that the driver stages 700, 720 are designed to be identical across the different driver circuits 216-1 and 216-2. The same applies to those described below 8a and 8b .

The driver circuit 216, which is in the 7 is shown, exemplarily includes two driver stages 700 and 720. The two driver stages 700 and 720 in particular amplify the current that is supplied to the gate contact of the power switch 218 so that it switches faster. Alternatively or additionally, the driver stages 700 and 720 can also increase the voltage level to which the gate contact of the power switch 218 is applied, ie increase it to the voltage value of the rail voltage V _TTL or the low voltage V' _{DC_ext} .

wird dabei dem Mittelabgriff des Spannungsteilers zugeführt. Die erste Treiberstufe 700 umfasst ferner eine Reihenschaltung des PMOS MOSFETs 702 mit dem NMOS MOSFET 704 über den Widerstand 710, der der Strombegrenzung dient. Der Drain-Anschluss des PMOS MOSFETs 702 ist mit der TTL/CMOS Versorgungsspannung v_TTL oder der Niederspannung V'_{DC_ext} verbunden. Ein Abgriff zwischen dem des PMOS MOSFET 702 und dem NMOS MOSFET 704 liefert das Ausgangssignal der Treiberstufe 700.The first driver stage 700 includes a voltage divider on the input side consisting of two parallel circuits, each with a resistor and a diode. The first parallel connection is formed by resistor 708 and diode 706. The second parallel connection is formed by resistor 712 and diode 714. The control signal V _PWM (f) or $\overline{v_{PWM}\left(ƒ\right)}$

is fed to the center tap of the voltage divider. The first driver stage 700 further includes a series connection of the PMOS MOSFET 702 with the NMOS MOSFET 704 via the resistor 710, which serves to limit the current. The drain of the PMOS MOSFET 702 is connected to the TTL/CMOS supply voltage v _TTL or the low voltage V' _{DC_ext} . A tap between the PMOS MOSFET 702 and the NMOS MOSFET 704 provides the output signal of the driver stage 700.

an der Kathode des Transistor 706 anliegt. Die Anode der Diode 714 ist mit dem Mittelabgriff des Spannungsteilers verbunden, sodass das Steuersignal V_PWM (f) bzw. $\overline{V_{PWM}\left(ƒ\right)}$

an der Anode der Diode 714 anliegt. Die Kathode der Diode 714 ist mit dem Gate-Anschluss des PMOS MOSFETs 702 verbunden.The anode of diode 706 is connected to the gate of NMOS MOSFET 704. The cathode of diode 706 is center tapped of the voltage divider, so that the control signal V _PWM (f) or $\overline{v_{PWM}\left(ƒ\right)}$

applied to the cathode of the transistor 706. The anode of the diode 714 is connected to the center tap of the voltage divider, so that the control signal V _PWM (f) or $\overline{v_{PWM}\left(ƒ\right)}$

is applied to the anode of diode 714. The cathode of diode 714 is connected to the gate of PMOS MOSFET 702.

bereitstellt, mit dem ein entsprechender Leistungsschalter in der Stufe 218 der Ansteuerschaltung 120 angesteuert wird. PMOS MOSFET 722 ist mit seinem Drain-Anschluss mit der TTL/CMOS Versorgungsspannung V_TTL oder der Niederspannung V'_{DC_ext} verbunden.The second driver stage 720 receives the output signal provided at the tap between that of the PMOS MOSFET 702 and the NMOS MOSFET 704. This output signal is supplied to the gate terminal of the PMOS MOSFET 722. The money terminal of the NMOS MOSFET 724 is connected to the tap between the PMOS MOSFET 702 and the NMOS MOSFET 704 via a parallel connection of the diode 726 and the resistor 728. The output signal of the first driver stage 700 is present at the cathode of the diode 226, while its anode is connected to the gate terminal of the NMOS MOSFET 724. The PMOS MOSFET 722 and the NMOS MOSFET 724 are connected in series via the resistor 730, with a center tap between the PMOS MOSFET 722 and the NMOS MOSFET 724 providing the output signal V' _PWM (f) and $\overline{v{\text{'}}_{PWM}\left(ƒ\right)}$

provides with which a corresponding power switch in stage 218 of the control circuit 120 is controlled. PMOS MOSFET 722 has its drain connected to the TTL/CMOS supply voltage V _TTL or the low voltage V' _{DC_ext} .

an den Signalpegel des Leistungsschalters 218 anpassen und das angepasste Steuersignal V'_PWM (f) bzw. $\overline{V{\text{'}}_{PWM}\left(ƒ\right)}$

dem Leistungsschalter 218 als ein frequenzvariables PWM-Steuersignal zuzuführen. Ob die Transistoren 702 bzw. 722 mit der TTL/CMOS Versorgungsspannung v_TTL oder der Niederspannung V'_{DC_ext} verbunden sind, kann beispielsweise davon abhängen, ob der zu treibende Leistungsschalter 218 ein PMOS-basierter oder ein NMOS-basierter Leistungsschalter ist. Da PMOS-basierte Leistungsschalter üblicherweise aufgrund ihrer höheren Gate-Kapazität eine höhere Gate-Spannung benötigen, um eine kurze Schaltzeiten zu erreichen, ist es sinnvoll, teilweise sogar notwendig, zumindest die letzte Treiberstufe (hier Treiberstufe 720) mit der Niederspannung V'_{DC_ext} zu verbinden. Die anderen Treiberstufen können wahlweise mit TTL/CMOS Versorgungsspannung V_TTL oder der Niederspannung P'_{Dc_ext} verbunden sein, je nach Aufbau der Treiberstufe.The individual driver stages 700 and 720 each form a level converter, which adjusts the signal level of the TTL/CMOS control signal V _PWM (f) or $\overline{v_{PWM}\left(ƒ\right)}$

to the signal level of the power switch 218 and the adapted control signal V ' _PWM (f) or $\overline{v{\text{'}}_{PWM}\left(ƒ\right)}$

to the power switch 218 as a frequency-variable PWM control signal. Whether the transistors 702 or 722 are connected to the TTL/CMOS supply voltage v _TTL or the low voltage V' _{DC_ext} can depend, for example, on whether the power switch 218 to be driven is a PMOS-based or an NMOS-based power switch. Since PMOS-based power switches usually require a higher gate voltage due to their higher gate capacity in order to achieve short switching times, it makes sense, and in some cases even necessary, to set at least the last driver stage (here driver stage 720) with the low voltage V' _{DC_ext} connect. The other driver stages can be connected either to TTL/CMOS supply voltage V _TTL or the low voltage P' _{Dc_ext} , depending on the structure of the driver stage.

The diodes 706 and 714, or 726, enable the gate capacitance at the gate terminal of the NMOS transistors 704 and 724 to be discharged quickly and thus the gate capacitance of the PMOS transistors 702 and 722 and theirs to be charged Switching time is accelerated. Due to the blocking direction of the diodes, the charging of the money capacitance of the NMOS transistors 704 and 724 takes place via the resistors 708 and 712 or 728. This circuit arrangement allows the switching times of the PMOS transistors 702 and 722 and the NMOS transistors 704 and 724 be adjusted so that a “shoot through” of the transistor pairs in each of the driver stages 700, 720 can be prevented.

8a shows a second exemplary embodiment of one of the driver circuits 216-1, 216-2 of the driver 216 in the 2 , if the power switch 218 to be driven is PMOS MOSFETs. 8b shows a second exemplary embodiment of one of the driver circuits 216-1, 216-2 of the driver 216 in the 2 , if the power switch 218 to be driven is NMOS MOSFETs.

The two embodiments of the driver circuits 216-1, 216-2 of the driver 216 in the 8a and 8b can therefore further develop the driver circuit 7 to be viewed as. The driver circuits 216-1, 216-2 in the 8a and 8b Both contain a cascade of driver stages 800, 700, 720, compared to 7 Driver level 800 was added on the input side. The driver levels 700,720 in 8a and 8b essentially correspond to driver levels 700, 720 7 . In the 8a It is assumed that the power switch 218 is a PMOS-based switching element. For this reason, the transistors 702 and 722 in the driver stages 700, 720 are connected with their source connection to the low voltage V' _{DC_ext} . Optionally, the source connection of the PMOS transistor 722 can be connected to the reference potential GND via a filter capacitor (C2). Embodiment of the driver circuits 216-1, 216-2 in the 8b is basically identical to the embodiment of the embodiments of the driver circuits 216-1, 216-2 in FIG 8a , however, the PMOS-based transistors 702 and 722 in the driver stages 700, 720 are supplied with the TTL/CMOS supply voltage V _TTL , since it is assumed here that the power switch 218 is an NMOS-based power switch.

mithilfe der Operationsverstärker-Schaltung hochsetzt. Aufgrund des hochohmigen Eingangs des Operationsverstärker 802, mit dem das Steuersignal V_PWM (f) bzw. $\overline{V_{PWM}\left(f\right)}$

über den Widerstand 804 verbunden ist, kann verhindert werden, dass plötzlich auftretende Überspannungen bzw. Stromspitzen in den nachfolgenden Stufen 700, 720 die Regeleinheit 214 beschädigen können. Der Ausgang des Operationsverstärker 802 ist mit Mittelabgriff des eingangsseitigen Spannungsteilers der Treiberstufe 700, der aus einer Parallelschaltung der Diode 706 mit dem Widerstand 708 und einer Parallelschaltung der Diode 714 mit dem Widerstand 712 besteht, verbunden. Wie bereits in Zusammenhang mit 7 erläutert, ist es vorteilhaft, wenn alle Treiberschaltungen 216-1, 216-2 baugleich ausgeführt sind.The additional input stage 800 of the driver circuit 216-1, 216-2 in the 8a and 8b implements a level converter that changes the TTL/CMOS voltage of the control signal V _PWM (f) or $\overline{v_{PWM}\left(ƒ\right)}$

using the operational amplifier circuit. Due to the high-impedance input of the operational amplifier 802, with which the control signal V _PWM (f) or $\overline{v_{PWM}\left(f\right)}$

is connected via the resistor 804, it can be prevented that sudden overvoltages or current peaks in the subsequent stages 700, 720 can damage the control unit 214. The output of the operational amplifier 802 is connected to the center tap of the input-side voltage divider of the driver stage 700, which consists of a parallel connection of the diode 706 with the resistor 708 and a parallel connection of the diode 714 with the resistor 712. As already in connection with 7 explained, it is advantageous if all driver circuits 216-1, 216-2 are constructed identically.

9 shows a second exemplary embodiment of the current measuring unit 256 for determining a current measurement value, based on which the output power of the control circuit 120 in 2 is regulated. The current measurement value indicates the direct current I _plasma flowing via the emission electrodes 130 to the counter electrode 140 of the electrical separator to a reference potential GND. Like from the 9 As can be seen, the counter electrode 140 of the electrical separator is connected to the reference potential GND. In order to control the direct current I _plasma flowing into the electrical separator. To measure as precisely as possible, the secondary-side coil, more precisely the second end 514 of the secondary-side coil of the transformer 220, is connected to the reference potential GND via a voltage divider formed from the resistors 902 and 904. The direct current I _plasma . In the embodiment of the invention, it is therefore measured on the secondary side of the transformer 220. In the embodiment shown, resistors 902 and 904 have the same resistance R6. Due to Kirchhoff's rules, the plasma current I _plasma flows through the voltage divider due to this connection, so that a measuring voltage V _measurement , which represents the plasma current I _plasma , drops at the center tap between the resistors 902 and 904. By connecting the counter electrode 140 with the reference potential GND and the voltage divider 902/904 between the second end 514 of the secondary-side coil of the transformer 220 and the reference potential GND, all relevant factors that change the current flow can also be taken into account and the direct current I _plasma with very measured with high precision. The efficiency of the control circuit 122 can thus be significantly improved.

Since the measuring unit 256 is connected to the center tap of the voltage divider with a very high resistance (cf. operational amplifier 918), only a negligible current flows into the measuring unit 256. The measuring unit 256 here, for example, includes an input-side RC element (low-pass filter) made up of the resistors 910 and the Capacitor 912 and ESD protection formed by diodes 914 and 916. At the non-inverting input of the operational amplifier 918 - due to the negligible current in the measuring unit 256 - the potential is present at the center tap of the voltage divider between the resistors 902 and 904. The connection of the operational amplifier 918 creates a gain buffer or unity gain buffer. The inverting input of operational amplifier 918 is coupled to the output of operational amplifier 918. The output of the operational amplifier 918 is connected to another RC element (low-pass filter) made of the resistors 920 and the capacitor 922 and outputs the (analog) current measurement value I[t]. This analog current measurement value I[t] is converted into a digital current measurement value I[n] by an analog/digital (A/D) converter 924 and supplied to the control unit 214. Whether an A/D conversion of the analog current measurement value I[t] is necessary depends on whether the control unit works analog or digital. In one exemplary embodiment, the control unit 214 contains the A/D converter 924. As already mentioned, the control unit 214 can be designed as a microprocessor. The control cycle of the control loop 122 is preferably in the range of 5 ms or less, preferably 2 ms or less, more preferably 1 ms or less. The shorter the control cycle, the faster fluctuations in the measured plasma current I _plasma can be detected and corrected. In one exemplary embodiment, the control cycle is 1 ms. Accordingly, the control unit 214 adapts the frequency value f[n] to the respective current measurement value I[n] at intervals of 1 ms in order to adjust the output power of the control circuit 120, or its output current I _Plasma, to a reference output power or a reference current value I _Ref regulate.

10 shows a second exemplary embodiment of the current measuring unit 256 for determining a current measurement value, based on which the output power of the control circuit 120 in 2 is regulated. In addition, the control unit 214 also takes into account a temperature measurement value in this embodiment. The current measurement value indicates the direct current I _plasma flowing via the emission electrodes 130 to the counter electrode 140 of the electrical separator to a reference potential GND. The embodiment of the 10 is based on the embodiment of 9 . The current measuring unit 256 in 10 corresponds to the current measurement unit 256 in 9 . In addition, the control unit 214 receives a temperature measurement value T[t]. This can be done, for example, with another one in the 10 A/D converter, not shown, into one digital temperature measurement value T[n]. This additional AD converter can also be integrated into the control unit 214, for example. The temperature measurement value can, for example, measure the temperature in the area surrounding the circuit board on which the control circuit 120 is implemented. The temperature sensor can optionally also be part of the control circuit 120. However, it is also possible to connect an external temperature sensor to the control circuit 120, which is not provided on the same board as the control circuit 120. In principle, the temperature can be recorded at various points on the circuit board or the terminal in which the control circuit 120 is installed. Different temperatures can also be recorded at different points on the circuit board or the terminal in which the control circuit 120 is installed and fed to the control unit 214 in analog/digital form. The control unit 214 can use the one or more temperature measurements that it receives in different ways in controlling the output power of the drive circuit 120.

zu variieren, sodass der Tastgrad nicht 50 % sondern leicht mehr oder leicht weniger beträgt.Due to the temperature fluctuations, it is to be expected that the resistance value of the resistors in the driver circuits 216-1, 216-2 increases at higher temperatures. The control unit 214 could use this, for example, to adjust the duty cycle of the control signals V _PWM (f) or $\overline{v_{PWM}\left(f\right)}$

to vary so that the duty cycle is not 50% but slightly more or slightly less.

zusammen mit dem gemessenen Stromwert I[n] berücksichtigt. In einer Ausführungsform könnte die Regeleinheit 214 beispielsweise die Eingangsleistung des Transformators 220 reduzieren, wenn eine kritische Temperatur, beispielsweise des Transformators 220, gemessen wird, um die Schaltung 120 vor Überlastung zu schützen. Die Effizienz des Transformators 220 nimmt typischerweise ab, wenn das Kernmaterial sich seiner Curie-Temperatur nähert.Another possibility is that the control unit 214 uses the measured temperature(s) to adapt the frequency f[n] of the control signals V _PWM (f) or $\overline{v_{PWM}\left(f\right)}$

taken into account together with the measured current value I[n]. For example, in one embodiment, the control unit 214 could reduce the input power of the transformer 220 when a critical temperature, for example of the transformer 220, is measured to protect the circuit 120 from overload. The efficiency of the transformer 220 typically decreases as the core material approaches its Curie temperature.

und/oder durch Reduzierung der Frequenz f[n] der Steuersignale V_PWM(f) bzw. $\overline{V_{PWM}\left(f\right)},$

so dass ein thermisches Durchgehen verhindert werden kann.If the temperature of the transformer 220 or a correlating temperature is known, thermal runaway can be detected and prevented at an early stage. In one embodiment, the control unit 214 is configured to detect thermal runaway based on the temperature measurement value that represents the temperature of the transformer 220 (e.g., by comparison with an assigning threshold temperature selected based on the Curie temperature of the transformer 220 , and detecting that the threshold temperature has been exceeded) and reduce the input power of the transformer 220, for example by changing the duty cycle of the control signals V _PWM (f) or $\overline{v_{PWM}\left(f\right)}$

and/or by reducing the frequency f[n] of the control signals V _PWM (f) or $\overline{v_{PWM}\left(f\right)},$

so that thermal runaway can be prevented.

und/oder durch Reduzierung der Frequenz f[n] der Steuersignale V_PWM(f) bzw. $\overline{V_{PWM}\left(f\right)},$

so dass ein thermisches Durchgehen verhindert werden kannFurthermore, the reverse current of the diodes of the cascade 254 typically increases exponentially with the temperature, so that the efficiency drops and here too there is a risk of thermal runaway. In addition to the temperature of the transformer 220 or alternatively, one or more temperature sensors may detect the temperature of the diodes in the cascade 254 (in the case of multiple temperature sensors, at different locations in the cascade 254). The control unit 214 receives the temperature measurement values of the one or more temperature sensors and can detect an impending thermal runaway based on this (e.g. by comparing with an assigned threshold temperature and detecting that the threshold temperature has been exceeded). If there is a risk of thermal runaway, the control unit 214 can reduce the input power of the transformer 220, for example by changing the duty cycle of the control signals V _PWM (f) or $\overline{v_{PWM}\left(f\right)}$

and/or by reducing the frequency f[n] of the control signals V _PWM (f) or $\overline{v_{PWM}\left(f\right)},$

so that thermal runaway can be prevented

unterdrückt, wenn oder solange die Schwellentemperatur überschritten ist.Either additionally or alternatively, if a threshold temperature is exceeded, an emergency shutdown of the control circuit 120 could occur by the control unit 214 receiving the control signals V _PWM (f) or $\overline{v_{PWM}\left(f\right)}$

suppressed if or as long as the threshold temperature is exceeded.

11 shows an exemplary input filter 1100 for use in the power limiter 212 of the 2 according to an exemplary embodiment of the invention. The input filter 1100 includes a common mode choke with several windings wound in the same direction. Current from the DC voltage source 110 flows through the windings. The magnetic fields in the core cancel each other out, so that interference currents on the forward and return lines are attenuated. Furthermore, the input filter includes an EMC filter 1104 that conducts EMC interference due to the parasitic impedances of the transformer 220 are filtered. The EMC filter 1104 can be implemented, for example, using a Π filter or a T filter and adapted to the implementation-specific interference the. The input filter 1100 can, for example, provide the derived low voltage V' _{DC_ext} and the reference potential GND at its output. The reference potential GND is separated from the external ground by the filter circuit 1104 and the common mode choke 1102. Filter circuit 1104 and common mode choke 1102 are only effective for high frequency signals, so that the DC plasma current I _plasma is not influenced by filter circuit 1104 and common mode choke 1102. The direct current component of the low voltage V' _{DC_ext} can therefore also correspond to the input voltage V _{DC_ext} . Basically, in an alternative embodiment, it is provided that the power limiter 212 changes the level of the direct current component of the input voltage V _{DC_ext} , for example by changing the reference potential GND compared to the reference potential (eg external ground) of the direct voltage source 110 and/or changing the direct voltage level of the input voltage _{VDC_ext} .

bereit. Grundsätzlich ist es auch denkbar, dass die Regeleinheit 214 nur das Steuersignal V_PWM (f) bereitstellt. In diesem Falle könnten einzelne Treiberschaltungen 216-1, 216-2, die eine invertierte Version dieses Steuersignals benötigen, einen Inverter aufweisen, um das Steuersignal $\overline{V_{PWM}\left(f\right)}$

zu erzeugen. Die anderen Treiberschaltungen 216-1, 216-2, die das nicht-invertierte Steuersignal V_PWM(f) könnten ferner eine Verzögerungsschleife aufweisen, um die Verzögerung der Laufzeit in den Inverter der jeweils anderen Treiberschaltungen 216-1, 216-2 zu kompensieren. Eine weitere Möglichkeit besteht darin, dass der Operationsverstärker 802 in der Stufe 800 der Treiberschaltungen 216-1, 216-2 so verschaltet wird, dass der Operationsverstärker 802 das Steuersignal V_PWM (f) invertiert. Eine andere Möglichkeit ist, dass die Stufen 700 und 720 der Treiberschaltungen 216-1, 216-2 die in den beschriebenen Ausführungsformen das invertierte Steuersignal $\overline{V_{PWM}\left(f\right)}$

benötigen, Transistorpaare der einzelnen Stufen 700, 720 (z.B. Transistoren 702 und 704, und Transistoren 722 und 724) nicht zwischen die positive TTL/CMOS Versorgungsspannung V_TTL bzw. die abgeleitete positive Niederspannung V'_{DC_ext} und das Referenzpotential GND, sondern zwischen das Referenzpotential GND und eine negative TTL/CMOS Versorgungsspannung -V_TTL bzw. eine abgeleitete negative Niederspannung - V'_{DC_ext} geschaltet sind.In the described embodiments, the control unit 214 provided the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

ready. In principle, it is also conceivable that the control unit 214 only provides the control signal V _PWM (f). In this case, individual driver circuits 216-1, 216-2 that require an inverted version of this control signal could have an inverter to convert the control signal $\overline{v_{PWM}\left(f\right)}$

to create. The other driver circuits 216-1, 216-2, which provide the non-inverted control signal V _PWM (f), could further have a delay loop to compensate for the delay in the propagation time in the inverter of the respective other driver circuits 216-1, 216-2. Another possibility is that the operational amplifier 802 is connected in stage 800 of the driver circuits 216-1, 216-2 in such a way that the operational amplifier 802 inverts the control signal V _PWM (f). Another possibility is that the stages 700 and 720 of the driver circuits 216-1, 216-2 receive the inverted control signal in the described embodiments $\overline{v_{PWM}\left(f\right)}$

require, transistor pairs of the individual stages 700, 720 (e.g. transistors 702 and 704, and transistors 722 and 724) not between the positive TTL/CMOS supply voltage V _TTL or the derived positive low voltage V' _{DC_ext} and the reference potential GND, but between the reference potential GND and a negative TTL/CMOS supply voltage -V _TTL or a derived negative low voltage - V' _{DC_ext} are connected.

Another aspect of the invention relates to the design of the transformer 220 so that it can be implemented as compactly as possible. In one embodiment of the invention, the transformer 220 has a housing with a rectangular or square basic shape. The edge length of this basic shape is, for example, between 30 mm and 18 mm. For example, the transformer 220 may include an EFD20 housing. The secondary-side coil 1218 of the transformer 220 can be arranged in several chambers of the transformer housing 1228 that are insulated from one another by means of a dielectric material.

12 shows an exemplary structure of a transformer 220 according to an embodiment of the invention. The transformer can be divided into a primary side and a secondary side. The primary side includes, among other things and for example, one or more primary-side coil(s), here as an example two coils 1206, 1212. The primary side can be assigned to the low-voltage part of the control circuit 220. The secondary side in turn includes, among other things, the secondary-side coil 1218. The secondary side can be assigned to the high-voltage part of the control circuit 220.

The transformer 220 includes the transformer housing 1228, which can also be referred to as a winding body. This is made of an insulating material. In the exemplary embodiment shown, the transformer 220 comprises two primary-side coils 1206 and 1212. The first coil 1206 has the connections 1208 and 1210, the first connection 1208 also being the first end 1208 of the primary-side coil 1206 and the second connection 1210 also being the second End 1210 of the primary side coil 1206 is referred to. The second coil 1212 also has two connections 1214 and 1216. Connections 1214 and 1216 are also referred to as the first end 1214 and second end 1216 of the coil 1212. The second end 1210 of the coil 1206 and the first end 1214 of the second primary-side coil 1212 can be short-circuited with each other or short-circuited on the circuit board (see 3 , 4a and 4b) . The coils 1206 and 1212 are wound in the same direction. If the coils 1206 and 1212 are connected in series, the multiplication factor of the transformer 220 is halved. The secondary-side coil 1218 of the transformer 220 is in the exemplary embodiment shown 12 divided into three partial coils 1218-1, 1218-2 and 1218-3. Each of these partial coils is arranged in a chamber 1224-1, 1224-2, 1224-3 of the transformer housing 1228. The individual chambers 1224-1, 1224-2, 1224-3 are separated from each other or from a chamber 1230 containing the primary-side coils 1206 and 1212 by partitions 1226-1, 1226-2, 1226-3. It should be noted that there are only three examples here Chambers 1224-1, 1224-2, 1224-3 for receiving the partial coils 1218-1, 1218-2 and 1218-3 of the secondary-side windings of the coil 1218 are shown. However, the windings of the secondary-side coil 1218 can also be arranged in more or fewer chambers, for example two chambers, four chambers, five chambers, six chambers, seven chambers, eight chambers, etc. The chambers 1224-1, 1224-2, 1224-3 and 1230 may also be referred to as housing sections.

The division of the windings of the secondary-side coil 1218 into several chambers 1224-1,1224-2,1224-3, which are insulated from each other, makes it possible to reduce the potential differences between the winding layers. This makes it possible to reduce the thickness of the insulation around the individual conductors of the secondary-side coil 1218, which in turn enables a more compact design of the transformer 220. This also makes it possible to manufacture the transformer 220 using industrial processes. In previously used, hand-made transformers 220, the individual partial coils 1218-1, 1218-2 and 1218-3 of the secondary-side windings of the coil 1218 were insulated from each other using insulating foils drawn in by hand.

The outputs 1220 (512) and 1222 (514) of the secondary-side coil 1218 output the alternating voltage V _transformer of the transformer 220 induced in the secondary-side coil 1218. The outputs 1220 (512) and 1222 (514) of the secondary coil 1218 are connected to the impedance match 252 and the high voltage cascade 254, as shown in FIG 2 is shown.

In the in 12 In the embodiment shown, the transformer 220 further comprises two E-shaped interconnected cores made of a conductive material that transport the induced magnetic field. The cores can be made of iron, for example, but other materials, in particular soft magnetic materials, are also possible.

In an exemplary embodiment of the transformer 220, the ratio of the number of windings of each of the two primary-side coils 1208 and 1212 of the transformer 220 and the number of windings of the secondary-side coil 1218 of the transformer 220 is in the range from 0.015 to 0.025 inclusive. In an exemplary embodiment, this ratio is 0.02, which corresponds to a multiplication factor of 1:0.02=50. For example, the number of windings of the primary side coils 1206 and 1212 of the transformer 220 is between 13 windings and 18 windings inclusive, and the number of windings of the secondary side coil 1218 of the transformer 220 is between 700 windings and 800 windings inclusive. In an exemplary implementation, the number of turns of coils 1206 and 1212 is each 15 and the number of turns of secondary side study 1218 is 750.

liegt daher beispielweise in einem Bereich von einschließlich 130 kHz und einschließlich 170 kHz. 13 shows the frequency response of an exemplary transformer 220 according to an embodiment of the invention. The X-axis denotes the frequency, while the Y-axis denotes the amount of magnetic flux in the transformer 220. The resonance frequency of the transformer 220 is preferably in the range 200.0 kHz to 240.0 kHz inclusive. In an advantageous implementation, the resonance frequency of the transformer 220 is set to 230.0 kHz due to the design. As already explained, it makes sense to control the frequency f with which the primary side of the transformer 220 is controlled in a range of the frequency response of the transformer 220 that is as linear as possible. The control frequency of the control signals V _PWM (f) and $\overline{v_{PWM}\left(f\right)}$

is therefore, for example, in a range of 130 kHz and 170 kHz inclusive.

Another aspect of the invention relates to the design of a circuit board that implements a drive circuit according to the invention. In particular, this aspect relates to the discrete structure of at least the high-voltage part of the control circuit 120 on a circuit board and the use of such a circuit board in a room air purifier, the room air cleaner having an electrical separator with an emission electrode which is formed from a plurality of emission needles and one of the emission needles at a distance formed, counter electrode, which is at least partially wetted, preferably surrounded by a liquid. Of course, the complete control circuit can also be constructed discretely. 14a shows an embodiment of such a circuit board 1400, which implements a control circuit 120 according to an embodiment of the invention. The board shown extends by approx. 140 mm in the longitudinal direction and by approx. 70 mm in the transverse direction. Of course, this is just an example and the board size is not limited to it.

14b shows the embodiment of the circuit board 14a , where the in 2 Components shown which have a control circuit 120 are marked. The high-voltage cascade 254 is arranged in an upper region of the circuit board 1400 when viewed in the transverse direction. On the right side in the longitudinal direction is the transformer 220, the secondary-side coil 1218 of which is oriented upwards in the transverse direction. At the upper right edge of the circuit board 1400 several discrete capacitors connected in series are shown, which implement the impedance matching 252. The structure of the high-voltage cascade using the individual discrete components shown is in the 16 illustrated. The components of the low-voltage part 210 of the control circuit 120 can be found in the lower area of the circuit board 1400. The lower area of the board includes the power limiter 212 with the input filter. The connection (pins, plugs or the like) for the low-voltage source 110 is also provided near the input filter. The measuring unit 256, the control unit 214, as well as the driver 216 and the circuit breaker 218 can also be found in the lower area of the board.

15 shows parts of the circuit board 14a and the arrangement of slots 1510, 1512, 1522 and recesses 1524 in the circuit board 1400, which are intended to completely penetrate the circuit board 1400 and prevent sparking between the individual components of the high-voltage part 250, or from the high-voltage part 250 into the low-voltage part 210 and / or the leakage current paths (see arrows 1720,1722 in the 17 ) extended so that no harmful leakage currents that can form on the surface of the circuit board damage the components of the control circuit 120 and / or the direct current source 110 connected to the circuit board 1400. In particular, the control unit 214 can react sensitively to overvoltages.

Like in the 15 clarified, the circuit board 1400 can be divided into a high-voltage region 1502 in the upper region of the board and a low-voltage region 1504 in the lower region of the board. The transformer 220 is attached to one side, here the right side (in the longitudinal direction) of the circuit board 1400 and represents the transition between the low-voltage area 1504 and the high-voltage area 1502. A boundary area 1506 is defined between the high-voltage area 1502 and the low-voltage area 1504, which is extends essentially in the longitudinal direction of the board. A slot 1510 is formed in this boundary region 1506, which begins next to the transformer 220 and essentially extends over the remaining width of the circuit board 1400 in the longitudinal direction. In the exemplary embodiment shown, this slot 1510 does not break through the edge of the circuit board 1400 opposite the transformer 220, so that the circuit board 1400 retains its stability. The slot 1510 completely penetrates the board 1400. In the exemplary embodiment shown, the slot 1510 has a width of approximately 3 mm, so that it can reliably prevent the spread of leakage currents from the high-voltage area 1502 into the low-voltage area 1504. Furthermore, this slot 1510 extends possible leakage current paths (see, for example, arrow 1720 in 17 ) from the high voltage area 1502 to the low voltage area 1504.

In the upper area of the circuit board 1400, which defines the high-voltage area 1502, the high-voltage cascade 254 of the control circuit 120 is implemented using discrete diodes and capacitors. The output of the high-voltage cascade 254 is located in the left area 1520 of the high-voltage area 1502. The highest voltages are therefore to be expected there. The voltage level increases from the high voltage V _Trafo at the output of the transformer 220 in the high voltage range 1502 to the output value V _PLAsma at the end of the cascade 254. The high voltage cascade 254 corresponds in the exemplary embodiment to the embodiment in 5 . The capacitors are placed in a longitudinally extending, strip-shaped region 1516 at the upper edge of the high-voltage region 1502 and in a longitudinally extending, strip-shaped region 1514 at the lower edge of the high-voltage region 1502. Like in the 16 shown, the capacitors of the high-voltage cascade 254 are realized by a series connection of groups of capacitors. In the embodiment shown on board 1400, each of these groups consists of four capacitors, but the invention is not limited to this. The diodes 1704 of the cascade 254 are also formed by a series connection of individual discrete diodes 1704. Each diode of the cascade 254 is in the exemplary embodiment shown 16 realized by a series connection of five discrete diodes 1704, but the invention is not limited to this. These groups of diodes are also arranged in strips 1518 transversely on the board. The diodes 1704 of a diode strip 1518 are connected in a zigzag or meandering shape, as can be seen, for example, from the enlarged view in the 17 results. The diode strips 1518 extend transversely between the strip-shaped areas 1514 and 1516 in which the capacitors of the cascade 254 are mounted on the circuit board 1400. There are recesses 1524 in the circuit board 1400 between the individual adjacent diode strips 1518 (not all recesses have reference symbols in the 15 marked to promote clarity). This effectively prevents the spread of leakage currents between the different diode strips 1518. In addition to the diode strip 1518, which is formed immediately next to the transformer 220 on the circuit board 1400, a slot 1522 is also formed in the circuit board 1400. This essentially extends in the transverse direction. At the upper end of the slot 1522, viewed in the transverse direction, this can extend in the longitudinal direction, for example in the embodiment shown the slot 1522 is essentially T-shaped. The lengthening of the slot 1522 in the longitudinal direction serves to increase the length of the leakage current path (see, for example, arrow 1720 in 17 ) so that the risk of damage to components in the low-voltage area 1504 of the circuit board 1400 due to leakage currents can be reduced.

17 shows an enlarged view of partial areas in the high-voltage area 1502 of the circuit board 1400 14a and the arrangement of slots 1702, 1706 that break through the board and recesses 1524 in the high voltage area 1502 of the board 1400. Above the board 1400 is in 17 As an example, a group of capacitors connected in series from the high-voltage cascade 254 in the high-voltage area 1502 is shown. The capacitors extend in the longitudinal direction, ie the two connections 1710, 1712 of each capacitor as well as the conductor tracks for connecting the capacitors of the group extend in the longitudinal direction. As the 17 Illustrated, a slot 1702 is provided below each capacitor in the board 1400 between the two terminals 1710, 1712 with which each of the discrete capacitors is connected (eg soldered) to the board, so that the propagation of leakage currents between the respective terminals 1710 , 1712 of each capacitor is prevented as much as possible. The slots 1702 extend in the transverse direction and protrude in the transverse direction on both sides beyond the housing of the respective capacitor.

In the bottom right area 17 A detailed view of two adjacent diode strips 1518 is also shown. As can be seen better there, the diodes 1704 (not all diodes are provided with reference numbers in order not to undermine the clarity of the enlarged detail) on the circuit board 1400 are connected to one another in a zigzag shape, that is, the cathodes of a diode 1704 (except for the last diode in the diode strip 1518) is connected to the anode of the next diode 1704. The diodes 1704 are designed as discrete components. Their anode and cathode extend essentially in the longitudinal direction. The respective connections between the individual diodes 1704 extend substantially in the transverse direction, forming a zigzag pattern. As already mentioned, a recess 1524 is provided between adjacent diode strips 1518. Starting from these recesses 1524, further slots 1706 extend between adjacent diodes 1704 in the longitudinal direction of the circuit board 1400 in order to prevent the spread of leakage currents between the adjacent diodes 1704 within the respective diode strips 1518 as far as possible. Corresponding slots 1706 in the longitudinal direction of the board 1400 extend from the recesses 1524 also above or below the first or last diode of each diode strip 1518, as also shown in the enlarged detail in the 17 will be shown. It should also be noted that such slots 1706 also extend from the slot 1522 between the individual neighboring diodes 1704 of the diode strip 1518 lying next to the transformer 202.

Furthermore, in the 17 It can be seen that below the slot 1510, a further slot 1512 extends in the circuit board 1400, which is located at least in part in the area of the power limiter 212. In the exemplary embodiment shown, this slot 1512 extends from the edge of the board 1400 and breaks the edge. In the exemplary embodiment shown, the slot 1512 extends (approximately) in the area of the input filter 110 in which the connection of the DC voltage source 110 is also located. The slot 1512 is formed at least in part in the longitudinal direction of the board 1400. No components of the control circuit 120 are provided in the area between the slot 1510 and the slot 1512 or attached to the circuit board 1400. Like in the 17 indicated by the arrow 1722, the leakage current path between the output area 1520 of the cascade 254 and the connection of the DC voltage source 110 on the circuit board 1400 (in the area of the input filter 1100) is extended due to the slot 1512, so that the external DC voltage source 110, which is connected to the Circuit board 1400 or control circuit 120 is connected, can be better protected against leakage currents.

As mentioned at the beginning, a further aspect of the invention is the use of the control circuit according to the invention in an air treatment device, in particular a room air purifier with an electrical separator, such as that from the PCT application WO 2021/224017 A1 and the disclosure of the patent application DE 10 2021 128 345.0 . 18 shows an exemplary embodiment of a section of a room air purifier 1 according to the invention, shown in a schematic sectional view, which contains a control circuit 120 according to an embodiment of the invention. The control circuit 120 can be part of a control unit of the room air purifier 1, which controls and/or operates the room air cleaner 1. The control circuit 120 can be implemented on its own circuit board 1400. Alternatively, the control circuit 120 can also be part of a circuit board that implements further control functions of the room air purifier 1.

The room air purifier 1 is designed to be rotation-shaped and includes the following main components nen: A control circuit 120 according to one of the previously described embodiments of the invention, an electrical separator 3 for separating liquid and / or solid particles from the air to be treated with a rotation-shaped counter electrode 5 (which corresponds to the counter electrode 140 in the previously described embodiments of the invention) and an emission electrode 7, which in the exemplary embodiment is formed as an array of emission electrode needles 9 (which correspond to the emission electrodes 130 in the previously described embodiments of the invention) which is arranged above the rotational counter electrode 5. The control circuit 120 provides the electrical separator with a direct voltage V _plasma in the high-voltage range, which is applied and regulated to the counter electrode 5 and the emission electrode 7, in particular, the array of emission electrode needles 9, during operation of the room air purifier 1.

The main components of the room air purifier 1 may further include: a liquid reservoir 11; a liquid conveyor 13 connected to the liquid reservoir 11 for wetting the counter electrode 5 with liquid from the liquid reservoir 11; a rotational air duct 15 for supplying the air to be treated to the electrical separator 3 and for forwarding the air cleaned by the electrical separator 3 to a deflection body 17 arranged downstream of the electrical separator 3 in the center of rotation of the air cleaner 1 or the air duct 15, which directs the cleaned air against the Direction of gravity, i.e. upwards, is diverted in order to discharge the cleaned air back into the environment via a flow outlet 26; and a fan 27 for generating the air flow through the room air purifier 1. The fan 27 can be controlled by a control unit of the room air purifier 1.

When executed in 1 the liquid storage 11 is arranged below the other components of the room air purifier 1. The liquid delivery 13, the counter electrode 5, the deflection body 17, the emission electrode 7 and the fan 27 are arranged above this from bottom to top. The components are housed in a housing 67 made up of several parts. The outside of the air cleaner 1 is formed by a cylindrical housing part 69 and the top by a disk-shaped housing part. The housing part 69 and the further housing part can also be formed in one piece.

All components of the room air purifier 1 according to the invention are accommodated or accommodated within the housing 3. The housing 3 can also have a connection for an external DC voltage source 110, which can be connected to the control circuit 120 of the room air purifier 1 by means of the connection (e.g. a plug connector). In principle, the air to be treated, which is generally provided with the reference number 17 and which contains liquid and/or solid particles, is introduced laterally into the interior of the housing 67 via an air inlet 19 and fed to the electrical separator 5. After the electrodeposition process, the separated liquid and/or solid particles, which are generally marked with the reference number 20, are transported away into a collecting container 21, which is also arranged within the housing 67, while the cleaned fresh air, which is provided with the reference number 22, is transported away, in particular by means of the deflection body 17 is deflected. The air can pass through an air aftertreatment system, which can include, for example, an ozone filter, and finally the cleaned air, which may have had its ozone content reduced, which is provided with the reference number 24, leaves via the air outlet 26, which can have, for example, grid-shaped or lamella-shaped outlet openings 29, the housing 67 or the room air purifier 1 towards the environment.

The liquid for wetting the counter electrode 5 is basically pumped from the liquid reservoir 11 to an upper side 25 of the counter electrode 5 using a pump (not shown) via a line 23 connected to the liquid reservoir 11. The liquid storage 11 and the collecting container 21 can be the same component or include different liquid basins.

The operation of the room air purifier 1 according to the invention is based on 19 described in detail and takes place in relation to the air flow 17 to be cleaned as follows. Via the air supply 15, the air 17 to be treated reaches an air duct structure 31 and finally into the interior of the room air purifier 1. The inflow direction E is in 19 indicated by an arrow at the entrance. However, it should be clear that not every air particle enters the room air purifier 1 parallel to the inflow direction E, but can reach the air inlet 19 somewhat obliquely with respect to the inflow direction E. In the exemplary embodiment, the air supply 15, like the room air purifier as a whole, is designed to be rotational, so that there is a circumferential air inflow and the air supply 15 essentially has a ring shape with the same cross-section distributed over the circumference.

The air supply 15 defines an air duct structure 31, which has a curved passage channel for the air to be treated into the interior of the room air purifier 1 limited. The air supply channel comprises an upstream channel wall 43, on which the inflowing air undergoes a first deflection of at least 30° with respect to the inflow direction E. As in 19 can be seen, the upstream channel wall 43 is concave in relation to the inflow direction E, so that the inflowing air can flow against the channel wall 43 with as little pressure loss as possible and, guided along it, can continue to flow towards the interior of the room air purifier 1. The air supply channel also includes one opposite the upstream channel wall 43 downstream channel wall 45, which is also shaped in such a way that the air flow deflected from the upstream channel wall 43 into an intermediate flow direction Z can flow against the channel wall 45 with as little pressure loss as possible and, guided thereon, can continue to flow towards the interior of the room air purifier 1 along an outflow direction A. As in 19 can be seen, the channel wall 45 is also designed and shaped to be concave with respect to the intermediate flow direction Z. At the downstream channel wall 45, the air flow undergoes a further deflection of at least 30° relative to the intermediate flow direction Z and is finally discharged in the direction of the separation space formed between emission electrode needles 9 and counter electrode 5.

Both the downstream channel wall 45 and the upstream channel wall 43 each include a convexly curved flow separation edge 47, 49, at which the air flow leaves the channel walls 43, 45 as a free jet into the interior of the room air purifier, i.e. without further structural guidance and / or support in the Course of the flow. The convex curvature of the flow separation edges 49, 47 also results in the lowest possible flow losses/pressure losses in the area of the flow outlet 6. A laminar flow can form throughout the entire course of the air supply duct, which can spread without turbulence and/or without pressure loss.

The air supply 15 in the area of the flow outlet 6 has a flow outlet surface. This flow outlet area is smaller than a deposition space cross-sectional area limited by the counter electrode 5 and the emission electrode 7, which represents the height or the distance between the emission electrode needles 9 and the counter electrode 5, is indicated.

At the flow separation edge 47, 49, at which the air is released in the direction of the electrical separator, in particular in a free jet, an imaginary extension T of the air guide surface beyond the flow separation edge is drawn, which does not cross the emission electrode needles 9, but in the direction of the counter electrode 5 is oriented and crosses it. The flow separation edge 47, 49 has a diffuser and/or spoiler-like effect on the air flow and causes a targeted introduction of the air flow into the separation space between the emission electrode needles 9 and the counter electrode 5, because the orientation of the air guide surface according to the invention can be reliably ensured that the air flow to be treated and Most of the air to be cleaned, in particular exclusively, reaches the electrode cloud below the emission electrode needles 9, forming a so-called plasma cone 39.

As already described above, the counter electrode 5 is wetted with a liquid in order to collect and transport away the particles separated from the air. The removal of the particles is indicated by arrow 20. The liquid flows in particular uniformly and/or as a calm liquid film on the surface 25 of the counter electrode 5, which according to the preferred embodiment has a funnel shape, into its center of rotation and is finally collected in the collecting container 21. After leaving the air duct structure 31 at the flow outlet 6, the air to be treated flows without obstacles, first through the electrical separator 3 and finally through the capacitor 33, the functioning and structure of which are explained below.

The emission electrode 7 comprises an array of emission electrode needles 9, which are placed on a rear side of an air guide wall 35 that delimits the air flow path and is located away from the deposition space between the emission electrode needles 9 and the counter electrode 5. The control circuit 120 is conductively connected to the emission electrode 7 and the counter electrode 5 in order to apply a high voltage to the emission electrode 7 and the counter electrode 5. The counter electrode 5 is connected to the reference potential GND of the control circuit 120. As explained above, the control circuit 120 can regulate the plasma current flowing from the emission electrode 7 via the direct current plasma to the counter electrode 5, so that the desired plasma current is established. The air duct wall 35 is formed essentially parallel to the counterelectrode contour and extends rotationally from radially outside to radially inward in the direction of the central deflection body 17. The air duct wall 35 is electrically conductive and has a capacitor plate 37 forming a capacitor plate downstream of the electrical separator 3, in particular the emission electron needles 9 Section that is connected to the control circuit 120 and, together with the counter electrode, builds up an electric high-voltage field F ( 19 ).

In the area of the electrical separator 3, the emission electrode needles generate dense electron clouds in the form of so-called plasma cones 39, in which the particles present in the air are electrically charged in order to separate the charged particles 41 from the air. The deposition occurs because the charged particles are attracted to the grounded counter electrode 5 in accordance with the technical current direction T _R. Because it was found according to the invention that the length of the electrical separator 3 viewed in the flow direction is not sufficient to reliably and effectively separate enough particles from the air to be converted, the downstream capacitor plate arrangement 37 and the built-up high-voltage electric field F present therein are used to give a negative to impose an attractive force F _C on charged particles 41, which causes the electrically charged particle 41 to be deflected or deflected in the direction of the counter electrode 5 and finally to be carried away there by the liquid wetting and transported away into the collecting container 21. The cleaned air 22 is deflected vertically upwards via the deflection body 17 and finally fed to the environment (reference number 24).

The features disclosed in the above description, the figures and the claims can be important both individually and in any combination for the implementation of the invention in the various embodiments.

Claims (39)

Device comprising: an electrical separator (3) with a plurality of emission electrodes (7) and a counter electrode (5); and a control circuit (120) which is set up to generate a high voltage from a low voltage of a direct voltage source (110) and to apply the high voltage to the emission electrodes (7) and the counter electrode (5) of the electrical separator (3) in order to create a direct current plasma between the emission electrodes (7) and the counter electrode (5); wherein the control circuit (120) contains a control circuit (122) with a current measuring unit (256), the current measuring unit (256) passing through the emission electrodes (7) via the direct current plasma to the counter electrode (5) of the electrical separator (3) to the common reference potential of the control circuit (120) measures flowing direct current; and the control circuit (122) is set up to regulate the output power of the control circuit (120) to a reference current value based on the value of the measured flowing direct current, wherein the control circuit (120) has a transformer (220) and a high-voltage cascade (600) connected to a secondary-side coil (1218) of the transformer (220) in order to convert the low voltage of the DC voltage source (110) or a DC voltage derived therefrom into the high voltage , wherein the current measuring unit (256) measures the direct current at a center tap of a voltage divider, the voltage divider being connected on one side to the secondary-side coil (1218) of the transformer (220) and on the other side to the counter electrode (5) of the electrical separator (3 ) and the reference potential of the control circuit (120). Device according to Claim 1 , wherein the control circuit (122) is set up to set the output power of the control circuit (120) based on the measured value of the direct current relative to the reference current value to an operating point at which a spark discharge of the direct current plasma between the emission electrodes (7) and the counter electrode (5 ) is prevented. Device according to Claim 1 or 2 , wherein the control circuit (122) further comprises: a control unit (214) which receives the measured value of the flowing direct current from the current measuring unit (256) and generates a control signal, the control unit (214) being set up to determine the frequency of the control signal based on the measured value of the direct current flowing through the emission electrodes (7) via the direct current plasma to the counter electrode (5) of the electrical separator (3) to the common reference potential of the control circuit (120) and the reference current value to vary the to the common reference potential of the control circuit (120) and to regulate the direct current flowing through the electrical separator (3) to the reference current value. Device according to one of the Claims 1 until 3 , wherein the windings of the secondary-side coil (1218) are divided into a plurality of partial coils and the transformer (220) comprises a non-conductive housing (1228) which is divided into a plurality of housing sections arranged parallel to one another in one direction and separated from one another by a non-conductive material, in in which the respective partial coils are arranged. Device according to one of the Claims 1 until 4 , wherein the ratio of the number of windings of each of the two primary-side coils (1206, 1212) and the number of windings of the secondary-side coil (1218) of the transformer (220) is in the range from 0.015 to 0.025 inclusive, preferably 0.02. Device according to Claim 5 , wherein the number of windings of the primary-side coil (1206, 1212) of the transformer (220) is between 13 windings and 18 windings inclusive and the number of windings of the secondary-side coil (1218) of the transformer (220) is between 700 windings and inclusive 800 windings. Device according to one of the Claims 3 until 6 , wherein the control signal has a frequency in the range from 120.0 kHz to 200.0 kHz inclusive, preferably in the range from 130.0 kHz to 170.0 kHz inclusive, and the resonance frequency of the transformer (220) is higher than the frequency of the control signal, preferably in the range including 200.0 kHz to 240.0 kHz. Device according to one of the Claims 1 until 7 , wherein the control circuit (120) comprises: a transformer (220) with two primary-side coils (1206, 1212) wound in the same direction and a secondary-side coil (1218), each primary-side coil (1206, 1212) having an end that is connected to the low voltage the direct voltage source (110) or a direct voltage derived therefrom is connected; a first power transistor (402, 404, 406, 408), which is controlled with a frequency-variable PWM control signal and, depending on the control signal, a second end of a first of the two primary-side coils (1206, 1212) with the reference potential of the control circuit (120) connects; and a second power transistor (402, 404, 406, 408), which is controlled with the inverted frequency-variable PWM control signal and, depending on the inverted control signal, a second end of the second of the two primary-side coils (1206, 1212) with the reference potential of the control circuit (120) connects. Device according to Claim 8 , wherein the control signal generated by the control unit (214) of the control circuit (122) is a transistor-transistor logic (TTL) or CMOS control signal; and the control circuit (120) further comprises: a first driver circuit (216), which is designed as a single-stage or multi-stage level converter in order to adapt the signal level of the TTL/CMOS control signal to the signal level of the first power transistor (402, 404, 406, 408). and supplying the matched control signal to the first power transistor (402, 404, 406, 408) as the variable frequency PWM control signal; and a second driver circuit (216), which is designed as a single-stage or multi-stage level converter in order to adapt the signal level of the TTL/CMOS control signal to the signal level of the first power transistor (402, 404, 406, 408) and the adapted control signal to the first power transistor ( 402, 404, 406, 408) as the inverted frequency variable PWM control signal. Device according to Claim 9 , wherein each of the first and second driver circuits (216) has a plurality of resistors, diodes and transistors and is set up to coordinate the charge and discharge of the gate capacitances of the transistors in the individual stages of the level converter so that respective transistor pairs in each stage of the Driver circuits (216), which are connected in series between a direct voltage corresponding to or derived from the low voltage and the reference potential, in a predetermined temperature range in which the control circuit (120) is operated, preferably around the range -40 ° C to 160 ° C, are not simultaneously leading. Device according to one of the Claims 1 until 7 , wherein the drive circuit (120) further comprises: a transformer (220) with a primary-side coil (1206, 1212) and a secondary-side coil (1218); a first power transistor (402, 404, 406, 408), which is controlled with a frequency-variable PWM control signal and, depending on the control signal of a control unit (214) of the control circuit (122), a second end of the primary-side coil (1206, 1212). connects to a direct voltage corresponding to or derived from the low voltage; a second power transistor (402, 404, 406, 408), which is controlled with the inverted frequency-variable PWM control signal and, depending on the inverted control signal, a first end of the primary-side coil (1206, 1212) with the DC voltage corresponding to or derived from the low voltage connects; a third power transistor (402, 404, 406, 408), which is controlled with the inverted frequency-variable PWM control signal and, depending on the inverted control signal, connects the second end of the primary-side coil (1206, 1212) to the reference potential of the control circuit (120). ; and a fourth power transistor (402, 404, 406, 408), which is controlled with the frequency-variable PWM control signal and, depending on the control signal, connects the first end of the primary-side coil (1206, 1212) to the reference potential of the control circuit (120); wherein, optionally, a capacitor (414, 416) is connected between the gate connection and drain connection of the power transistors (402, 404, 406, 408) in order to suppress conducted interference. Device according to Claim 11 , wherein the control circuit (120) further comprises: a first driver circuit (216), which is designed as a single-stage or multi-stage level converter in order to adapt the signal level of the TTL/CMOS control signal to the signal level of the first power transistor (402, 404, 406, 408). to adapt and supply the adapted control signal to the first power transistor (402, 404, 406, 408) as the variable frequency PWM control signal; a second driver circuit (216), which is designed as a single-stage or multi-stage level converter in order to adapt the signal level of the TTL/CMOS control signal to the signal level of the second power transistor (402, 404, 406, 408) and the adapted control signal to the first power transistor (402 , 404, 406, 408) as the inverted frequency variable PWM control signal; a third driver circuit (216), which is designed as a single-stage or multi-stage level converter in order to adapt the signal level of the TTL/CMOS control signal to the signal level of the third power transistor (402, 404, 406, 408) and the adapted control signal to the first power transistor (402 , 404, 406, 408) as the variable frequency PWM control signal; and a fourth driver circuit (216), which is designed as a single-stage or multi-stage level converter in order to adapt the signal level of the TTL/CMOS control signal to the signal level of the fourth power transistor (402, 404, 406, 408) and the adapted control signal to the first power transistor ( 402, 404, 406, 408) as the inverted frequency variable PWM control signal. Device according to one of the Claims 1 until 7 , wherein the drive circuit (120) further comprises: a transformer (220) with a primary-side coil (1206, 1212) and a secondary-side coil (1218); a first capacitor (414, 416), one end of which is connected to a second end of the primary-side coil (1206, 1212) and at the other end of which a direct voltage corresponding to or derived from the low voltage is present; a second power transistor (402, 404, 406, 408), which is controlled with a frequency-variable PWM control signal and, depending on the control signal, connects the first end of the primary-side coil (1206, 1212) to the reference potential of the control circuit (120); a first power transistor (402, 404, 406, 408), which is controlled with the inverted frequency-variable PWM control signal and, depending on the inverted control signal, a first end of the primary-side coil (1206, 1212) with the DC voltage corresponding to or derived from the low voltage connects; a second capacitor (414, 416), one end of which is connected to the second end of the primary-side coil (1206, 1212) and the other end of which is connected to the reference potential of the control circuit (120). Device according to Claim 12 or 13 , wherein each of the driver circuits (216) has a plurality of resistors, diodes and transistors and is set up to coordinate the charge and discharge of the gate capacitances of the transistors in the individual stages of the level converters so that respective transistor pairs in each stage of the driver circuits (216) , which are connected in series between a direct voltage corresponding to or derived from the low voltage and the reference potential, are not simultaneously conductive in a predetermined temperature range in which the control circuit (120) is operated, preferably in the range -40 ° C to 160 ° C . Device according to one of the Claims 1 until 14 , wherein the control circuit (120) has a multi-stage, in particular 3-stage, 4-stage, 5-stage, or 6-stage, high-voltage cascade (600) in order to increase an output-side alternating voltage of a transformer (220) into the direct current high voltage. Device according to Claim 15 , wherein the high-voltage cascade (600) is a Villard cascade or a high-voltage cascade with full-wave rectification. Device according to Claim 15 or 16 , wherein each stage of the high-voltage cascade (600) comprises a series connection of several diodes and a series connection of several capacitors, which are designed as discrete components. Device according to one of the Claims 14 until 17 , wherein the control circuit (120) further includes an impedance matching circuit (252) which is connected between the two outputs of the secondary-side coil (1218) of the transformer (220) of the control circuit (120) and the inputs of the high-voltage cascade (600), the impedance matching circuit (252) is set up to adapt the impedance between power switches of the control circuit (120), the transformer (220) and the high-voltage cascade (600) to one another. Device according to one of the Claims 1 until 18 , wherein the control circuit (120) further comprises an input stage which is connected to the DC voltage source (110), the input stage being a common mode choke (1102) and/or a filter (1104) which is designed to attenuate conducted interference from the transformer (220) and/or to limit the current flowing from the DC voltage source (110) into the control circuit (120). Device according to one of the Claims 1 until 19 , wherein the control circuit (120) further comprises an input stage which comprises a filter, the output of the filter providing a direct voltage derived from the low voltage of the direct voltage source (110), with which the primary-side coils (1206, 1212) of the transformer ( 220) of the control circuit (120). Device according to one of the Claims 1 until 20 comprising a circuit board (1400) on which the control circuit (120) is implemented with discrete components, the control circuit (129) having a low-voltage part (210) and a high-voltage part (250), the circuit board (1400) extending in a longitudinal direction and Transverse direction extends, and the circuit board (1400) has an upper region in which the high-voltage part (250) of the control circuit (120) is implemented and a lower region in which the low-voltage part (250) of the control circuit (120) is implemented; wherein a transformer (220) of the control circuit (120), which defines the transition between the low-voltage part (210) and high-voltage part (250) of the control circuit (120), in the longitudinal direction on one side of the circuit board (1400) in a border area (1506) between the is arranged in the upper region and the lower region of the board (1400), and in the remaining part of the board (1400), the board (1400) has a longitudinally extending slot (1510) in the board (1400) in the boundary region (1506), which separates the upper area of the circuit board (1400) from the lower area of the circuit board (1400) in order to extend the path for leakage currents between the high-voltage part (250) of the control circuit (120) and the low-voltage part (210) of the control circuit (120) and / or to prevent arcing from the high-voltage part (250) of the control circuit (120) into the low-voltage part (210) of the control circuit (120). Device according to Claim 21 , wherein the control circuit (120) comprises a high-voltage cascade (600), which is implemented in the upper region of the circuit board (1400) by means of discrete components, the high-voltage cascade (600) consisting of a large number of capacitors and diodes, some of the capacitors being in a capacitor strip (1516) arranged at an upper edge of the upper region of the circuit board (1400) and another part of the capacitors in a longitudinally extending capacitor strip (1516) at a lower edge of the upper region of the circuit board (1400) , which is located next to the longitudinally extending slot (1510) in the circuit board (1400); and wherein the diodes of the high voltage cascade (600) are arranged in a plurality of transversely extending diode strips (1518) between the lower edge and the upper edge of the upper region of the circuit board (1400), the circuit board (1400) being between the respective diodes -Strip (1518) and the capacitor strip (1516) has recesses (1524) at the bottom edge and at the top edge of the upper area in order to prevent leakage currents between the diodes and capacitors of the high-voltage cascade (600) of the control circuit (120). Device according to Claim 22 , wherein in each strip groups of discrete diodes are connected in series in a zigzag arrangement, with slots in the circuit board (1400) extending from the recesses (1524) extending between the individual discrete diodes of the groups in order to prevent leakage currents between the discrete diodes impede. Device according to Claim 23 , wherein the anodes and cathodes of the discrete diodes in the groups extend longitudinally and the slots extending from the recesses (1524) also extend longitudinally in the circuit board (1400). Device according to one of the Claims 22 until 24 , wherein the capacitors are arranged in groups in the capacitor strips (1516) at the lower edge and at the upper edge of the upper region, the circuit board (1400) extending in the transverse direction between the two poles of each discrete capacitor (414, 416). Has slot to prevent leakage currents between the two poles of the discrete capacitors. Device according to one of the Claims 22 until 25 , wherein one of the diode strips (1518) is arranged in the longitudinal direction next to the transformer (220), and the circuit board (1400) has a transversely extending slot which is between the one diode strip (1518) and the transformer ( 220) extends to prevent leakage currents between the diodes of one diode strip (1518) and the secondary-side connections of the transformer (220) in the high-voltage part (250) of the control circuit (120). Device according to Claim 26 , wherein starting from the transversely extending slot between the adjacent discrete diodes of the one diode strip (1518), slots in the circuit board (1400) extend in the longitudinal direction in order to prevent leakage currents between the discrete diodes of the one diode strip (1518). impede. Device according to one of the Claims 22 until 27 , wherein in the lower region of the circuit board (1400), the circuit board (1400) has a further slot on an output side of the high-voltage cascade (600), which is essentially at least partially parallel to the longitudinally extending slot (1510) in the circuit board (1400 ), which separates the upper area of the board (1400) from the lower area of the board (1400). Device according to Claim 28 , wherein the further slot extends away from an edge in the lower region of the circuit board (1400) that delimits the circuit board (1400) on the output side of the high-voltage cascade (600). Control circuit (120) for use in a device, in particular a room air purifier (1), for treating air, the control circuit (120) being set up to supply a direct voltage in the high-voltage range to the emission needles (9) of an emission electrode (7) and a counter electrode ( 5) to create an electrical separator (3) of the device in order to generate a direct current plasma between the emission electrode (7) and the counter electrode (5), wherein the control circuit (120) has a control circuit (122) which is set up to control the output power of the control circuit (120) based on a measured value of a signal transmitted through the emission electrode (7) via the direct current plasma to the counter electrode (5) of the electrical separator (3). to set the common reference potential of the control circuit (120) flowing direct current relative to the reference current value to an operating point at which a spark discharge of the direct current plasma between the emission needles (9) of the emission electrode (7) and the counter electrode (5) is prevented, wherein the control circuit (122) contains a current measuring unit (256), the current measuring unit (256) measuring the direct current flowing through the emission electrode (7) via the direct current plasma to the counter electrode (5) of the electrical separator (3) to the common reference potential of the control circuit (120). , and the control circuit (122) contains a control unit (214) which receives the measured value of the flowing direct current from the current measuring unit (256) and generates a control signal, the control unit (214) being set up to control the frequency of the control signal based on the measured value of the to vary the direct current flowing through the emission needles (9) of the emission electrode (7) via the direct current plasma to the counter electrode (5) of the electrical separator (3) to the common reference potential of the control circuit (120) and the reference current value in order to vary the direct current flowing to the common reference potential of the control circuit (120) and the electric separator (3) to regulate the direct current flowing to the reference current value. Control circuit (120). Claim 30 , wherein the control circuit (120) has a transformer (220) and a high-voltage cascade (600) connected to a secondary-side coil (1218) of the transformer (220) in order to convert the low voltage of the DC voltage source (110) or a DC voltage derived therefrom into the high voltage convert, the current measuring unit (256) measuring the direct current at a center tap of a voltage divider, the voltage divider being connected on one side to the secondary-side coil (1218) of the transformer (220) and on the other side to the counter electrode (5) of the electrical separator (3) and the reference potential of the control circuit (120). Control circuit (120) according to one of the Claims 30 until 31 , wherein the control circuit (120) is also like the control circuit of the device according to one of Patent claims 1 until 20 is trained. Circuit board (1400) on which a control circuit (120) according to one of the Claims 30 until 32 is implemented with discrete components. Use of a control circuit (120) according to one of the Claims 30 until 32 or a circuit board (1400). Claim 33 in a room air purifier (1), the room air cleaner (1) having an electrical separator (3) with an emission electrode (7) which is formed from a plurality of emission needles (9) and a counter electrode (5) which is formed at a distance from the emission needles (9). ), which is at least partially wetted, preferably surrounded by a liquid. Device, in particular room air cleaner (1), for treating air, comprising: an electrical separator (3) with an emission electrode (7), which is formed from a plurality of emission needles (9), and one which is spaced apart from the emission needles (9). Counter electrode (5), which is at least partially wetted, preferably surrounded by a liquid; and a control unit which operates the electrical separator (3); wherein the control unit is a control circuit (120), which is set up to apply a direct voltage in the high-voltage range to the emission needles (9) of the emission electrode (7) and the counter electrode (5) of the electrostatic precipitator (3) in order to create a direct current plasma between the emission electrode (7) and the counter electrode ( 5), wherein the control circuit (120) has a control circuit (122) which is set up to control the output power of the control circuit (120) based on a measured value of a through the emission electrode (7) via the direct current plasma to the counter electrode (5) of the Electrical separator (3) to set the direct current flowing to the common reference potential of the control circuit (120) relative to the reference current value to an operating point at which a spark discharge of the direct current plasma between the emission needles (9) of the emission electrode (7) and the counter electrode (5) is prevented, whereby the control circuit (122) further comprises: a control unit (214) which receives the measured value of the flowing direct current from the current measuring unit (256) and generates a control signal, the control unit (214) being set up to determine the frequency of the control signal based on the measured To vary the value of the direct current flowing through the emission needles (9) of the emission electrode (7) via the direct current plasma to the counter electrode (5) of the electrical separator (3) to the common reference potential of the control circuit (120) and the reference current value in order to vary the value of the direct current flowing to the common reference potential of the control circuit (120). 120) and the electric separator (3) to regulate the direct current flowing to the reference current value. Device according to Claim 35 , wherein the control circuit (122) contains a current measuring unit (256), the current measuring unit (256) measuring the direct current flowing through the emission electrode (7) via the direct current plasma to the counter electrode (5) of the electrical separator (3) to the common reference potential of the control circuit (120). measures. Device according to one of the Claims 35 until 36 , wherein the control circuit (120) is arranged adjacent to the emission electrode (7) and in particular distal to the counter electrode (5). Device according to one of the Claims 35 until 37 , wherein the control unit and / or the control circuit (120) drives a fan (27). Device according to one of the Claims 35 until 38 , wherein the control circuit (120) is also like the control circuit of the device according to one of Patent claims 1 until 29 is designed and / or the control circuit (120) by means of a circuit board (1400). Claim 33 is realized.

Images