CN201374643Y - Control device applied to input unbalanced three-phase boost type AC-DC converter - Google Patents

Abstract

The utility model provides a be applied to unbalanced three-phase boost type AC/DC converter's of input controlling means which characterized in that: the whole control device is formed by integrating a controller of a three-phase boost type AC-DC converter with an input voltage, a current sensor and an output voltage sensor, wherein the controller of the three-phase boost type AC-DC converter consists of a converter voltage controller, a pulse width modulation module, a positive sequence q-axis current command, a negative sequence d-axis current command, a negative sequence q-axis current command, a positive sequence current controller and a negative sequence current controller, and can operate under three-phase balanced and unbalanced power input to achieve input-side unit power factor and reduce output-side secondary waves so as to minimize a voltage stabilizing capacitor, thereby not only reducing the cost of a system, but also improving the efficiency of the system.

Description

Be applied to import the control device of uneven three-phase voltage increasing type AC/DC changeover switch

Technical field

The utility model belongs to electric and electronic technical field, particularly a kind of input side power factor correction function and the control device of eliminating Er Ripple ripple of outlet side voltage that is applied to uneven three-phase AC/DC changeover switch.

Background technology

By existing document [1] Hong-Seok Song, Kwanghee Nam, " Dual currentcontrol scheme for PWM converter under unbalanced conditions; " IEEETrans.Ind.Electron., vol.46, No.5, pp.953-959, Oct.1999.; [2] Y.Suh, V.Tijeras and T.A.Lipo, " A nonlinear control of the instantaneous power indq synchronous frame for pwm ac/dc converter under generalizedunbalanced operating conditions; " in Conf.Rec.IEEE-IACON ' 02, vol.2, pp.1189-1196, Oct.2002.; [3] Peng Xiao, Keith A.Corzine, Ganesh K.Venayagamoorthy, " Cancellation Predictive Control for Three-Phase PWMRectifiers under Harmonic and Unbalanced Input Conditions; " IECON2006-32nd Annual Conference on, pp.1816-1821, Nov.2006.; And [4] I.Etxeberria-Otadui, U.Viscarret; M.Caballero, A.Rufer, S.Bacha, " NewOptimized PWM VSC Control Structures and Strategies Under UnbalancedVoltage Transients; " IEEE Trans.Ind.Electron., vol.54, No.5, pp.2902-2914, Oct.2007. can find out in, import the ripple that unbalanced voltage can cause 120Hz really, and document [1] proposes the double-current controller in the operation of three-phase imbalance pulse-width modulation transducer, document [2] then proposes the instantaneous power nonlinear Control rule of the synchronous frame of d-q, applies to the power supply of three-phase imbalance input widely.

In addition, document [5] Hong-Seok Song, In-Won Joo, Kwanghee Nam, " Sourcevoltage sensorless estimation scheme for PWM rectifiers under unbalancedconditions, " IEEE Trans.Ind.Electron., vol.50, No.6, pp.1238-1245, Dec.2003.; [6] S.Hansen, M.Malinowski, F.Blaabjerg, and M.P.Kazmierkowski, " Sensorless control strategies for PWM rectifier, " in Proc.IEEE APEC, pp.832-838,2000.; [7] T.Ohnuki, O.Miyashita, P.Lataire, and G.Maggetto, " Control of a three-phase PWM rectifier using estimatedAC-side and DC-side voltages, " IEEE Trans.Power Electron., vol.14, pp.222-226, Mar.1999.; [8] D.-C.Lee and D.-S.Lim, " AC voltage andcurrent sensorless control of three-phase PWM rectifiers; " in Proc.IEEEPESC, pp.588-593, the 2000. control rules that propose no sensing operate in the situation of three-phase imbalance.

Yet up to the present, have only the existing document of few Number to consider circuit and switch switch cost, to operating in the three-phase AC/DC changeover switch of uneven input, transducer outlet side voltage is subjected to the situation of 120Hz Ripple wave action, therefore how to improve present existing method, and consider and operate in the three-phase imbalance AC/DC changeover switch, because of being subjected to the influence of uneven power supply and line loss, and the current control device that proposes, to reduce outlet side voltage 120Hz De Ripple ripple, make output voltage more clean steadily, be the primary problem that the utility model institute desire solves.

Summary of the invention

Main purpose of the present utility model, be to provide a kind of control device that is applied to import uneven three-phase voltage increasing type AC/DC changeover switch, it can be at the three-phase voltage increasing type AC/DC changeover switch of input unbalance voltage, in order to be suppressed at two times of lines Dian Ya Ripple ripple frequently on the output direct current chain, make output voltage for the direct current offered load is required more stably, and reach the input side specific work because of purpose.

Main purpose of the present utility model is achieved by following technical proposals:

A kind of control device that is applied to import uneven three-phase voltage increasing type AC/DC changeover switch is made up of a three-phase AC/DC changeover switch and its controller, it is characterized in that:

This three-phase voltage increasing type AC/DC changeover switch controller comprises AC/DC transducer voltage controller, a pulse-width modulation module and a forward-order current controller, a negative-sequence current controller, a positive sequence q shaft current order, a negative phase-sequence d shaft current order and a negative phase-sequence q shaft current order of an individual operation; And this AC/DC transducer voltage controller obtains required positive sequence direct-axis current order with voltage commands and two input signals of output voltage measuring value after as calculated, and export this forward-order current controller to the q shaft voltage with positive sequence q shaft current order and current sensor actual positive sequence d axle that measures and the actual positive sequence d axle that q shaft current and voltage-sensor measure, and actual negative phase-sequence d axle that this current sensor measures and q shaft current and the order of this negative phase-sequence d shaft current, the actual negative phase-sequence d axle that the order of this negative phase-sequence q shaft current and this voltage-sensor measure exports this negative-sequence current controller to the q shaft voltage, then this forward-order current controller and this negative-sequence current controller are calculated the value addition that is produced, and deliver to six switch signals of this pulse-width modulation module generation to drive six switches of this AC/DC transducer.

This three-phase AC/DC changeover switch input side is a three phase mains, outlet side then is a DC load, and this three-phase AC/DC changeover switch arrange in pairs or groups this three-phase AC/DC changeover switch controller and an input side three pole reactor, an outlet side electric capacity of voltage regulation, and an input voltage sensor, an output voltage sensor and a current sensor, this three-phase AC/DC changeover switch controller then for six semiconductor power switch control signals of output controlling this three-phase AC/DC changeover switch, and in order to control transformation device input side electric current and outlet side voltage.

The three-phase voltage of this three-phase AC/DC changeover switch input side and outlet side voltage are surveyed by this input voltage sensor and this output voltage sensibility reciprocal respectively, and the input side three-phase current is then measured by current sensor and obtains.

The beneficial effects of the utility model are: this control device, no matter the variation of line loss why, little and the output voltage that comparable single controller of the ripple that transducer produced or dual Control device come is comparatively near bid value, and the influence that is not subject to converter efficiency, compared with multiple advantages such as traditional dual Control device is more strong.

Description of drawings

Fig. 1 is a control block schematic diagram of the present utility model;

Fig. 2 is a control flow chart of the present utility model;

Fig. 3 is the utility model three-phase AC/DC changeover switch power circuit Organization Chart;

Fig. 4 is the Organization Chart that the utility model three-phase AC/DC changeover switch controller uses the digital signals processor;

Fig. 5 imports the wherein circuit framework figure of a phase of three-phase voltage sensor for the utility model;

Fig. 6 is the circuit framework figure of the utility model output voltage sensor;

Fig. 7 is the wherein circuit framework figure of a phase of the utility model three-phase current sensor;

Fig. 8 is the inside block schematic diagram of the utility model forward-order current controller;

Fig. 9 is the inside block schematic diagram of the utility model negative-sequence current controller;

Figure 10 is the signal order schematic diagram of the utility model positive sequence q shaft current order;

Figure 11 is the signal order schematic diagram of the utility model negative phase-sequence d shaft current order;

Figure 12 is the signal order schematic diagram of the utility model negative phase-sequence q shaft current order;

Figure 13 is a datagram of adopting the single control device method simulation of voltage guiding;

Figure 14 is the datagram of traditional dual Control device method simulation;

The datagram of the controller simulation that Figure 15 carries for the utility model;

Figure 16 is the utility model simulation test figure, and wherein (a) is in order to show negative phase-sequence direct-axis current I _DN(b) in order to show output voltage V _o(c) in order to show the negative phase-sequence d-axis and the negative phase-sequence quadrature-axis voltage V of controlling signal stator frame _DsNAnd V _QsN

Embodiment

Below in conjunction with drawings and Examples the utility model is further described.

Referring to Fig. 1 to Fig. 4, shown in the figure the selected embodiment of the utility model, this only for the usefulness of explanation, is not subjected to the restriction of this structure in patent application.

A kind of control device that is applied to import uneven three-phase voltage increasing type AC/DC changeover switch provided by the utility model mainly is made up of with its controller 1 a three-phase AC/DC changeover switch 6, and its input side is a three phase mains 2, and outlet side then is a DC load 8.Wherein this three-phase AC/DC changeover switch 6 these three-phase AC/DC changeover switch controllers 1 of collocation, an input side three pole reactor 5, an outlet side electric capacity of voltage regulation 7, and an input voltage sensor 31, an output voltage sensor 32 and a current sensor 4; This three-phase AC/DC changeover switch controller 1 is then controlled this three-phase AC/DC changeover switch 6 for six semiconductor power switch control signals of output.As shown in Figure 3, in order to control transformation device input side electric current and outlet side voltage.(Ea, Eb Ec) can measure the circuit framework figure of this voltage- sensor 31,32 such as Fig. 5 and shown in Figure 6 by this voltage- sensor 31,32 respectively with outlet side voltage (Vo) to the three-phase voltage of input side in addition; And the input side three-phase current (ia, ib ic) then can be by current sensor 4 measurement acquisitions, and wherein the circuit framework figure of this current sensor 4 is then as shown in Figure 7.

Referring to Fig. 8 to Figure 12, the built-in function square of three-phase voltage increasing type AC/DC changeover switch controller 1 of the present utility model includes an individual operation AC/DC transducer voltage controller 16, a pulse-width modulation module 17 and a forward-order current controller 11, a negative-sequence current controller 12, a positive sequence q shaft current order 13, a negative phase-sequence d shaft current order 14 and a negative phase-sequence q shaft current order 15.Wherein this AC/DC transducer voltage controller 16 obtains voltage commands (Vo*) after as calculated required positive sequence d-axis IdP* current order and exports this forward-order current controller 11 with the actual positive sequence d axle that voltage-sensor 31 measures with q shaft voltage EdP, EqP with q shaft current IdP, IqP with actual positive sequence d axle that this positive sequence q shaft current order 13 and current sensor 4 measure with two input signals of output voltage (Vo) measuring value.In like manner, the actual negative phase-sequence d axle that measures of this current sensor 4 exports this negative-sequence current controller 12 with the actual negative phase-sequence d axle that this negative phase-sequence d shaft current order 14, negative phase-sequence q shaft current order 15 and this voltage-sensor 31 measure with q shaft voltage EdN, EqN with q shaft current IdN, IqN; Then this forward-order current controller 11 and this negative-sequence current controller 12 are calculated the value addition that is produced, deliver to this pulse-width modulation module 17 and produce six switch signals, in order to driving six switches of AC/DC transducer, with reach the input side specific work because of and reduce the generation of Er Ripple ripple of outlet side voltage.

Fig. 3 is the circuit framework figure of three-phase pulse-width modulation transducer, wherein is input as three phase mains, connects boost inductance and substitutional connection series resistance and six controllable switch, and outlet side is made up of the load in parallel of an electric capacity of voltage regulation.At first, do a modeling for convenient behaviour is done in three-phase imbalance pulse-width modulation transducer, the input voltage of three-phase imbalance and input current can be described as by positive sequence and negative phase-sequence composition to be formed, being described below of mathematics:

$E_{\mathrm{dqs}}=e^{\mathrm{j\ωt}}{E}_{\mathrm{dq}}^p+e^{-\mathrm{j\ωt}}{E}_{\mathrm{dq}}^n$

$=e^{\mathrm{j\ωt}}(E_d^p+j{E}_q^p)+e^{-\mathrm{j\ωt}}(E_d^n+j{E}_q^n)---\left(1\right)$

$I_{\mathrm{dqs}}=e^{\mathrm{j\ωt}}{I}_{\mathrm{dq}}^p+e^{-\mathrm{j\ωt}}{I}_{\mathrm{dq}}^n$

$=e^{\mathrm{j\ωt}}(I_d^p+j{I}_q^p)+e^{-\mathrm{j\ωt}}(I_d^n+j{I}_q^n)---\left(2\right)$

Wherein, ω is the power supply angular frequency, ω=377rad/sec, frequency f=60Hz.Consideration by Fig. 2 A side see into the complex power of transducer, can be expressed as:

$S_{\mathrm{in}}=\frac{3}{2}{E}_{\mathrm{dqs}}{I}_{\mathrm{dqs}}^{*}=P_{\mathrm{in}}\left(t\right)+j{Q}_{\mathrm{in}}\left(t\right)---\left(3\right)$

Wherein, Pin (t) and Qin (t) are respectively the compositions of instantaneous real merit and virtual work, and in addition, Pin (t) and Qin (t) also can be expressed as following form:

P _in(t)＝P _in+P _inc2cos2ωt+P _ins2sin2ωt (4)

Q _in(t)＝Q _in+Q _inc2cos2ωt+Q _ins2sin2ωt (5)

Consider equation (5), suppose that the instantaneous virtual work Qin (t) of input pulse-width modulation transducer is zero, the input side electric current of transducer just can reach specific work because of.With approximate way, also can make that average input virtual work is zero to be Qin=0, reach the input side electric current reach specific work because of purpose, and do not influence output voltage, because output voltage is only relevant with real merit.Secondly consider to operate in the output voltage of three-phase imbalance pulse-width modulation transducer, from equation (4) as can be known, import instantaneous real merit and have two times of lines composition frequently, wherein instantaneous real merit comprises average real merit Pin and high-order term Pinc2, Pins2.The instantaneous real merit of the input of transducer transfers to outlet side, and the size of decision direct current chain bus bar voltage, yet Pinc2 and Pins2 can cause 120Hz De Ripple ripple on the accurate position of output direct current.Traditional way under the situation of not considering line loss and switch switch cost, can make directly that Pinc2 and Pins2 are zero, to keep the clean accurate position of transducer output direct current.

And in the utility model, consider line loss and line loss is discussed and Shu goes out the relation of Dian Ya Ripple ripple; In traditional method, the input specific work because of analysis with output two multiple scale analysis all observe from the A side of Fig. 3, the utility model then propose specific work because of analysis do observation with two multiple scale analysis in different places, specific work because of analysis observe in the A side, and the analysis of two frequencys multiplication is observed in the B side, at this moment, the switch switch cost also can equivalence be done observation from the B side.

Consider the sum total equivalent resistance Rs of switch switch cost and line loss, the power loss of Rs can be write as shown in the formula:

$S_{\mathrm{Line}}=\frac{3}{2}{V}_{\mathrm{dqs}\_\mathrm{Rs}}{I}_{\mathrm{dqs}}^{*}$

$=P_{\mathrm{Rs}}\left(t\right)+j{Q}_{\mathrm{Rs}}\left(t\right)---\left(6\right)$

Wherein

$\left\{\begin{array}{c}{P}_{\mathrm{Rs}}\left(t\right)=P_{\mathrm{dc}\_\mathrm{Rs}}+{\mathrm{<figure-callout\; id="2"\; label="P\; c"\; filenames="Y2009200063230011H1.png,Y2009200063230013H1.png"\; state="\{\{state\}\}">P</figure-callout>}}_c\cos 2\mathrm{\&omega;t}+{\mathrm{<figure-callout\; id="2"\; label="P\; s"\; filenames="Y2009200063230011H1.png,Y2009200063230013H1.png"\; state="\{\{state\}\}">P</figure-callout>}}_s\sin 2\mathrm{\&omega;t}\\ Q_{\mathrm{Rs}}\left(t\right)=Q_{\mathrm{dc}\_\mathrm{Rs}}+{\mathrm{<figure-callout\; id="2"\; label="Q\; c"\; filenames="Y2009200063230011H1.png,Y2009200063230013H1.png"\; state="\{\{state\}\}">Q</figure-callout>}}_c\cos 2\mathrm{\&omega;t}+{\mathrm{<figure-callout\; id="2"\; label="Q\; s"\; filenames="Y2009200063230011H1.png,Y2009200063230013H1.png"\; state="\{\{state\}\}">Q</figure-callout>}}_s\sin 2\mathrm{\&omega;t}\end{array}\right.$

$P_{\mathrm{dc}\_\mathrm{Rs}}=\frac{3}{2}(V_{d\_\mathrm{Rs}}^p{I}_d^p+V_{q\_\mathrm{Rs}}^p{I}_q^p+V_{d\_\mathrm{Rs}}^n{I}_d^n+V_{q\_\mathrm{Rs}}^n{I}_q^n)---\left(7\right)$

${\mathrm{<figure-callout\; id="2"\; label="P\; c"\; filenames="Y2009200063230011H1.png,Y2009200063230013H1.png"\; state="\{\{state\}\}">P</figure-callout>}}_c=\frac{3}{2}(V_{d\_\mathrm{Rs}}^p{I}_d^n+V_{q\_\mathrm{Rs}}^p{I}_q^n+V_{d\_\mathrm{Rs}}^n{I}_d^p+V_{q\_\mathrm{Rs}}^n{I}_q^p)---\left(8\right)$

${\mathrm{<figure-callout\; id="2"\; label="P\; s"\; filenames="Y2009200063230011H1.png,Y2009200063230013H1.png"\; state="\{\{state\}\}">P</figure-callout>}}_s=\frac{3}{2}(V_{q\_\mathrm{Rs}}^n{I}_d^p-V_{d\_\mathrm{Rs}}^n{I}_q^p-V_{q\_\mathrm{Rs}}^p{I}_d^n+V_{d\_\mathrm{Rs}}^p{I}_q^n)---\left(9\right)$

P _o＝P _in-P _{dc_Rs} (10)

P _oc2＝P _{in_c2}-P _{Rs_c2} (11)

P _os2＝P _{in_s2}-P _{Rs_s2} (12)

By above derivation, consider the input side specific work because of not having the analysis of circuit two frequencys multiplication, and the substitutional connection loss is taken into account with outlet side, integrate (5), (10), (11) and (12) formula, can get (13) formula:

$\left[\begin{array}{c}\frac{2}{3}{\mathrm{<figure-callout\; id="2"\; label="P\; o"\; filenames="Y2009200063230011H1.png,Y2009200063230013H1.png"\; state="\{\{state\}\}">P</figure-callout>}}_o\\ \frac{2}{3}{Q}_{\mathrm{in}}\\ \frac{2}{3}{\mathrm{<figure-callout\; id="2"\; label="P\; os"\; filenames="Y2009200063230011H1.png,Y2009200063230013H1.png"\; state="\{\{state\}\}">P</figure-callout>}}_{\mathrm{os}}\\ \frac{2}{3}{\mathrm{<figure-callout\; id="2"\; label="P\; oc"\; filenames="Y2009200063230011H1.png,Y2009200063230013H1.png"\; state="\{\{state\}\}">P</figure-callout>}}_{\mathrm{oc}}\end{array}\right]=\left[\begin{array}{cccc}(E_d^p-R_s{I}_d^p)& (E_q^p-R_s{I}_q^p)& (E_d^n-R_s{I}_d^n)& (E_q^n-R_s{I}_q^n)\\ E_q^p& {-E}_d^p& E_q^n& {-E}_d^n\\ (E_q^n-R_s{I}_q^n)& -(E_d^n-R_s{I}_d^n)& -(E_q^p-R_s{I}_q^p)& (E_d^p-R_s{I}_d^p)\\ (E_d^n-R_s{I}_d^n)& (E_q^n-R_s{I}_q^n)& (E_d^p-R_s{I}_d^p)& (E_q^p-R_s{I}_q^p)\end{array}\right]\left[\begin{array}{c}{I}_d^p\left(t\right)\\ I_q^p\left(t\right)\\ I_d^n\left(t\right)\\ I_q^n\left(t\right)\end{array}\right]---\left(13\right)$

Observe (13) formula, for reach input side current unit merit because of, must make that the value of importing average fictitious power is zero, i.e. Qin=0.Secondly, for obtaining the accurate position of clean output voltage, by the B side see into real merit Pos2 of two frequencys multiplication and Poc2 need be set at zero.For simplifying control strategy, adopt voltage guiding control rule, equation (13) can be simplified as follows:

$\left[\begin{array}{c}\frac{2}{3}{\mathrm{<figure-callout\; id="0"\; label="P\; o"\; filenames="Y2009200063230019H1.png,Y2009200063230019H3.png"\; state="\{\{state\}\}">P</figure-callout>}}_o\\ 0\\ 0\\ 0\end{array}\right]=\left[\begin{array}{cc}(E_d^p-R_s{I}_d^p)& (E_d^n-R_s{I}_d^n)\\ E_q^p& E_q^n\\ E_q^n& {-E}_q^p\\ (E_d^n-R_s{I}_d^n)& (E_d^p-R_s{I}_d^p)\end{array}\right]\left[\begin{array}{c}{I}_d^p\left(t\right)\\ I_d^n\left(t\right)\end{array}\right]---\left(14\right)$

When the input side three-phase voltage can precisely be obtained by the phase-locked loop, the q axle composition that can make positive sequence and negative phase-sequence was zero, and equation (14) can be simplified as follows:

$\left[\begin{array}{c}\frac{2}{3}{P}_o\\ 0\end{array}\right]\left[\begin{array}{cc}(E_d^p-R_s{I}_d^p)& (E_d^n-R_s{I}_d^n)\\ (E_d^n-R_s{I}_d^n)& (E_d^p-R_s{I}_d^p)\end{array}\right]\left[\begin{array}{c}{I}_d^p\left(t\right)\\ I_d^n\left(t\right)\end{array}\right]---\left(15\right)$

Equation (15) is one group of nonlinear equation, and it separates as follows:

$I_{\mathrm{dp}}=\frac{1}{6}\frac{3E_{\mathrm{dp}}+{(9E_{\mathrm{dp}}^2-24P_o{R}_s+36R_s{E}_{\mathrm{dn}}{I}_{\mathrm{dn}}-36R_s^2{I}_{\mathrm{dn}}^2)}^{\frac{1}{2}}}{R_s}---\left(16\right)$

$I_{\mathrm{dn}}=\frac{-E_{\mathrm{dn}}}{E_{\mathrm{dp}}-2R_s{I}_{\mathrm{dp}}}{I}_{\mathrm{dp}}---\left(17\right)$

By aforementioned to importing the modeling process of uneven pulse wave width modulation transducer, as can be known, for reach input side current unit merit because of, and obtain clean output voltage, the negative-sequence current I of d axle _DnBe forward-order current I by the d axle _DpDetermine, and with the switch and the line loss resistance R of equivalence _STake into account, its relational expression is shown in (17), if do not consider the switching losses and the line loss resistance R of equivalence _S, then relational expression can be rewritten into as shown in the formula:

$I_{\mathrm{dn}}=\frac{-E_{\mathrm{dn}}}{E_{\mathrm{dp}}}{I}_{\mathrm{dp}}---\left(18\right)$

Operate in the unbalanced modeling of input according to the pulse wave width modulation transducer, and can learn that from (16) formula and (17) formula negative-sequence current has the coupling terms of a forward-order current, forward-order current also has the coupling terms of a negative-sequence current.Yet the coupling terms of negative phase-sequence is much smaller than the coupling terms of positive sequence, so negative phase-sequence can be left in the basket to the coupling terms of positive sequence.Therefore based on the analysis of front theory, the control device that the utility model is carried mainly is to eliminate outlet side voltage secondary ripple, and finish input side current unit merit because of.Fig. 1 is control device calcspar of the present utility model, and the control loop of forward-order current adopts voltage guiding control rule, and equation (17), (18) formula then are applied to the control loop of negative-sequence current.For the feasibility of the checking control strategy of carrying, utilize breadboardin software PSPICE to simulate, below parameter for simulating:

L＝1.6mH，R _s＝0.2ohm，C _o＝2200μF，

e _a(t)＝100cosωt+10cosωt?V，ω＝120πrad/s

e _b(t)＝100cos(ωt-2π/3)+10cos(ωt+2π/3)V

e _c(t)＝100cos(ωt+2π/3)+10cos(ωt-2π/3)V

Compare for convenience, three kinds of control strategies are discussed.First method is to adopt the single controller of voltage guiding, and second method is the dual Control device, and the third method is the control strategy that the utility model is carried.The line loss R of equivalence _SBe changed to 0.2 Ω from 0.001 Ω, can observe the amplitude size of direct current chain voltage secondary ripple, Simulation result, collating is distinguished corresponding first method, second method and the third method at Figure 13, Figure 14 and Figure 15.From Simulation result, can be observed, work as R _SDuring=0.001 Ω, the voltage ripple that the dual Control device is produced is less than the single controller method of adopting voltage guiding, yet, work as R _SDuring=0.2 Ω, the voltage ripple that the dual Control device is produced then than the single controller of adopting voltage guiding come serious.From this example as can be seen, the method for traditional dual Control device is not considered the loss of line impedance integral body.Obviously, the voltage ripple size that produced of traditional dual Control device is vulnerable to the influence of converter efficiency.Analog waveform such as Figure 16 of the control strategy of carrying, R at this moment _S=0.2 Ω, Figure 16 (a) are negative phase-sequence direct-axis current I _DNWaveform, Figure 16 (b) is an output voltage V _oWaveform, Figure 16 (c) then is the negative phase-sequence d-axis and the negative phase-sequence quadrature-axis voltage V of controlling signal stator frame _DsN, V _QsNWaveform.

And as can be known by above analog result, control device provided by the utility model, no matter the variation of line loss why, little and the output voltage that comparable single controller of the ripple that transducer produced or dual Control device come is comparatively near bid value, and the influence that is not subject to converter efficiency, compared with multiple advantages such as traditional dual Control device is more strong.

Claims (3)

1. A control device applied to an input unbalanced three-phase step-up AC-DC converter, consisting of a three-phase AC-DC converter and its controller, characterized in that: The three-phase step-up AC-DC converter controller includes a single-operating AC/DC converter voltage controller, a pulse width modulation module, a positive-sequence current controller, a negative-sequence current controller, a positive sequence q-axis current command, a negative-sequence d-axis current command, and a negative-sequence q-axis current command; and the AC/DC converter voltage controller calculates the two input signals of the voltage command and the output voltage measurement value to obtain the The desired positive-sequence direct-axis current command is combined with the positive-sequence q-axis current command and the actual positive-sequence d-axis measured by the current sensor and the actual positive-sequence d measured by the q-axis current and voltage sensor. axis and q axis voltages are output to the positive sequence current controller together, and the actual negative sequence d axis and q axis currents measured by the current sensor are related to the negative sequence d axis current command, the negative sequence q axis current The command and the actual negative-sequence d-axis and q-axis voltages measured by the voltage sensor are output to the negative-sequence current controller, and then the positive-sequence current controller and the negative-sequence current controller calculate the resulting The values are summed and sent to the pulse width modulation module to generate six switching signals to drive the six switches of the AC/DC converter. 2. The control device applied to the input unbalanced three-phase step-up AC-DC converter according to claim 1, characterized in that: the input side of the three-phase AC-DC converter is a three-phase power supply, and the output side is a constant and the three-phase AC-DC converter is equipped with the three-phase AC-DC converter controller and an input-side three-phase inductor, an output-side stabilizing capacitor, an input voltage sensor, an output voltage sensor and an A current sensor, and the three-phase AC-DC converter controller is used to output six semiconductor power switch control signals to control the three-phase AC-DC converter, and to control the input side current and output side voltage of the converter. 3. The control device applied to the input unbalanced three-phase step-up AC-DC converter according to claim 2, characterized in that: the three-phase voltage on the input side of the three-phase AC-DC converter and the voltage on the output side are respectively determined by the The input voltage sensor senses and measures the output voltage, and the three-phase current at the input side is measured by the current sensor.

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