CN117254698B - CLLC circuit bidirectional switching control method outside limit gain - Google Patents
CLLC circuit bidirectional switching control method outside limit gain Download PDFInfo
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- CN117254698B CN117254698B CN202311515440.6A CN202311515440A CN117254698B CN 117254698 B CN117254698 B CN 117254698B CN 202311515440 A CN202311515440 A CN 202311515440A CN 117254698 B CN117254698 B CN 117254698B
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- 230000002457 bidirectional effect Effects 0.000 title claims abstract description 29
- 230000010363 phase shift Effects 0.000 claims abstract description 24
- 230000009467 reduction Effects 0.000 claims description 20
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- 230000001360 synchronised effect Effects 0.000 claims description 18
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- 238000004364 calculation method Methods 0.000 description 8
- 238000004146 energy storage Methods 0.000 description 5
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- 238000010586 diagram Methods 0.000 description 3
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Classifications
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
- H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
- H02M3/33584—Bidirectional converters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/01—Resonant DC/DC converters
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
- H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
- H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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- Engineering & Computer Science (AREA)
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Abstract
The invention relates to the field of DC-DC converters, in particular to a CLLC circuit bidirectional switching control method outside limit gain. Comprising the following steps: calculating the limit voltage gain of the CLLC circuit variable frequency control in an off-line mode, and judging the current working direction and the current gain state; if the circuit gain is smaller than the limit voltage gain，For a primary side driving signal and a secondary side driving signal, performing a two-degree-of-freedom switching control based on a switching frequency and a phase shift angle of a circuit; and generating a corresponding driving signal of each switching tube according to the primary side driving signal and the secondary side driving signal. The invention realizes stable and accurate control of power outside the frequency conversion control limit gain based on the double degrees of freedom of frequency conversion and phase shift, can realize smooth switching of the working direction, and avoids power impact and bus voltage fluctuation in the traditional switching process; because the voltage gain is optimized only for the scene that the voltage gain exceeds the frequency conversion control limit gain, the method is not in conflict with the traditional frequency conversion control scheme, and can be directly applied to the existing CLLC circuit system as a supplementary strategy.
Description
Technical Field
The invention relates to a control or regulation technology for conversion equipment between alternating current and direct current, belongs to the field of DC-DC converters, and particularly relates to a CLLC circuit bidirectional switching control method outside limit gain. The operation mode control of the bidirectional CLLC circuit can be used for a plurality of application occasions such as micro-grid operation, electric automobile charging and discharging, energy storage systems and the like.
Background
In order to achieve the aim of carbon neutralization and carbon peak, more and more new energy power generation equipment is connected into a power grid, but photovoltaic and wind power have the problems of output fluctuation and intermittence, and hidden danger is brought to the stable operation of the power grid. In order to stabilize the fluctuation of new energy power generation, energy storage equipment needs to be added in a power grid. The bidirectional isolation DC-DC converter is a power interface for the energy storage device to access the power grid, wherein the bidirectional CLLC circuit topology is widely adopted by virtue of high frequency, high efficiency and high power density.
The bidirectional CLLC circuit is developed from a traditional unidirectional LLC circuit, inherits the fundamental wave analysis and frequency conversion control scheme of the unidirectional LLC circuit, and can realize the soft switching and boosting operation in the positive and negative directions. To adapt energy storage systems of different voltage classes, a bi-directional CLLC circuit needs to have a wide voltage gain regulation capability. However, when the CLLC circuit adopting the traditional variable frequency control has limit voltage gain, namely the gain is smaller than the limit voltage gain, the variable frequency control cannot cover the whole power range, and the circuit has a dead zone of power control. This dead zone limits the accuracy and stability of the power control of the CLLC circuit under low gain, low load conditions. In the energy storage application, the bidirectional converter needs to have smooth and rapid charge-discharge switching capability, and the CLLC circuit adopting the traditional frequency conversion control can only realize smooth switching in a smaller gain range. When the gain requirement exceeds the limit gain of the circuit, a dead zone of power control can cause power fluctuation in the working direction switching process, so that bus voltage oscillation is caused, and the safe and stable working of the connected converter and the power grid is threatened.
Meanwhile, when the voltage gain is not 1, the change of the circuit buck-boost working mode can occur simultaneously along with the switching of the working direction; the equivalent voltage gain of the circuit is suddenly changed, which makes the analysis of the working state of the circuit difficult. When only variable frequency control is used, the frequency cannot be mutated with the gain. The phase shift control is added on the basis of the variable frequency control, so that two sections of variable frequency control intervals which cannot be suddenly changed can be connected; however, the conventional methods do not consider the switching conditions of multiple control modes, and also cannot realize smooth change of the driving signal and continuous change of the control parameters in the forward and reverse directions, so that smooth switching of the running state of the circuit is difficult to ensure.
It can be seen that the conventional variable frequency control of the bidirectional CLLC converter has problems of dead zones of limiting gain and power control and uneven switching of the working direction. The disclosed traditional optimization method combines frequency conversion and phase shift control, expands the voltage gain range, but fails to realize smooth switching in the forward and reverse directions. Thus, improvements in the art are desirable.
Disclosure of Invention
The invention aims to solve the technical problem of overcoming the defects in the prior art and providing a CLLC circuit bidirectional switching control method outside the limit gain.
In order to solve the technical problems, the invention adopts the following solutions:
providing a CLLC circuit bidirectional switching control method outside limit gain, wherein the primary side and the secondary side of the CLLC circuit are in bidirectional symmetry; the control method comprises the following steps:
(1) In an off-line mode, the limit voltage gain of the variable frequency control of the CLLC circuit is calculated according to the following modeM limit ：
（1）
Resonant frequencyf r Is determined by circuit parameters and is calculated according to the following formula:
（2）
in the above-mentioned formulae, L rp is a primary side resonance inductance;L m exciting an inductor for the transformer;f s is the switching frequency;C rp is a primary side resonance capacitor;
(2) Implementing bidirectional switching control, comprising:
(2.1) judging the current working direction and the current gain state of the CLLC circuit based on circuit state detection; if the circuit gain is smaller than the limit voltage gainM limit Step (2.2) is performed;
(2.2) Primary side drive Signal for CLLC CircuitG p And a secondary side drive signalG s Performing a circuit-based switching frequencyf s And phase shift angleφIs controlled by switching the two degrees of freedom;
(2.3) according to the Primary side drive SignalG p And a secondary side drive signalG s A corresponding drive signal for each switching tube is generated.
As a preferable scheme of the invention, in the CLLC circuit with two-way symmetry of primary side and secondary side, a switching tube S is adopted 1 ~S 4 Forms a primary side switch bridge and is composed of a switch tube S 5 ~S 8 Forms a secondary side switch bridge, and the driving signals respectively corresponding to the secondary side switch bridge are respectivelyG 1 ~ G 4 AndG 5 ~G 8 the method comprises the steps of carrying out a first treatment on the surface of the According to the primary drive signalG p And a secondary side drive signalG s Generating a corresponding driving signal for each switching tube, whereinG 1 AndG 4 and (3) withG p The same as,G 2 AndG 3 and (3) withG p Complementary to each other,G 5 AndG 8 and (3) withG s The same as,G 6 AndG 7 and (3) withG s Complementary.
In the CLLC circuit, the circuit parameters of the CLLC circuit are as followsL rp =n 2 L rs AndC rp =C rs /n 2 is a relationship of (2); wherein,L rp is a primary side resonance inductance;nis the transformation ratio of the transformer;L rs the secondary resonance inductance;C rp is a primary side resonance capacitor;C rs is the secondary resonance capacitance.
As a preferred embodiment of the present invention, the output current is based oni s Current input to primary sidei p Primary side input powerP p Or minor edgeOutput powerP s The current working direction of the CLLC circuit is judged according to any circuit state.
As a preferred scheme of the present invention, the current working direction of the CLLC circuit is specifically determined according to any one of the following modes: to output currenti s As a basis for the purposes of this,i s >0 is a positive direction,i s <0 is reverse direction,i s =0 is the switching threshold state where the transmission power is 0; by primary side input currenti p As a basis for the purposes of this,i p >0 is a positive direction,i p <0 is reverse direction,i p =0 is the switching threshold state where the transmission power is 0; with primary input powerP p As a basis for the purposes of this,P p >0 is a positive direction,P p <0 is reverse direction,P p =0 is the switching threshold state where the transmission power is 0; with secondary side output powerP s As a basis for the purposes of this,P s >0 is a positive direction,P s <0 is reverse direction,P s =0 is the switching threshold state where the transmission power is 0.
In the step (2.1), the input voltage of the CLLC circuit is detectedV p And output voltageV s To determine the current gain state:V s /V p ≥M limit when the circuit gain is larger than or equal to the limit gain, the subsequent switching control is not needed;V s /V p <M limit when the circuit gain is smaller than the limit gain, the subsequent switching control is required.
In the step (2.2), the phase shift angle is preferably setφRepresenting secondary side drive signals in buck modeG s Relative to the primary drive signalG p Phase angle with lag of on and off time;φrepresentative of boost modeG s Relative toG p Phase angle of leading off timeG s And (3) withG p The opening time is the same;
(2.2-1) upon switching from the Forward Buck state to the reverse boost stateG p AndG s the switching control of (a) includes the steps of:
(a) The circuit works in a forward voltage-reducing state, and when receiving the switching signal, the circuit controls the primary side driving signalG p Switching frequencyf s Increase of secondary side driving signalG s Following upG p Changing, phase shifting angleφPhase shift angle for synchronous rectificationφ SR I.e.φ=φ SR ；
(b) Determining whether the switching frequency has reached the maximum switching frequency, i.e.f s =f smax Returning to the step (a) if the step (a) is not established, and entering the next step if the step (a) is established;
(c) The working is kept at the maximum switching frequency, the driving signal of the secondary side is actively controlled, and the phase shift angle is reducedφ；
(d) Judging whether the phase shift angle is 0, namelyφ=0, and return to step (c) is not true; if true, the circuit transmission power is 0, the forward voltage reduction state is finished, and the next step is carried out;
(e) Controlling secondary side drive signalsG s Switching frequencyf s Reduction of primary drive signalG p Following upG s Changing, phase shifting angleφPhase shift angle for synchronous rectificationφ SR I.e.φ=φ SR The circuit operates in a reverse boost state;
(2.2-2) upon switching from the reverse boost state to the forward buck stateG p AndG s the switching control operation content of (a) is reversed to that of the step (2.2-1), namely, the control is performed in reverse order from (e) to (a).
As a preferable mode of the invention, the phase shift angle of the synchronous rectificationφ SR Is obtained offline by adopting any one method of fundamental wave analysis, time domain analysis and simulation analysis; then build up by making a tableφ SR And mapping with the circuit state, realizing synchronous rectification control by using a table look-up mode, and avoiding complex operation in the controller directly.
As a preferred embodiment of the present invention, the input voltage of the CLLC circuit is detected in step (2.1) and step (2.2)V p And output voltageV s To determine the current gain state and describe that the forward gain is less than the limit gain (i.eV s /V p <M limit ) A control scheme at that time; the scheme comprises two conditions of switching from a forward voltage-reducing state to a reverse voltage-increasing state shown in the step (2.2-1) and switching from the reverse voltage-increasing state to the forward voltage-reducing state shown in the step (2.2-2), and belongs to the forward and reverse sequences of the same control scheme;
according to the characteristics of the circuit parameters and the gain characteristics which are symmetrical in the positive and negative working directions, when the reverse gain is smaller than the limit gain (i.eV p /V s <M limit ) In this case, the same logic as that of the control scheme is true, and only the difference is that the logic concepts of the primary side and the secondary side need to be exchanged, and the voltage reduction direction is set to be the forward direction.
Compared with the prior art, the invention has the beneficial effects that:
1. the invention breaks through the minimum voltage gain limit existing when the traditional CLLC circuit adopts variable frequency control, and realizes stable and accurate control of power beyond the limit gain of variable frequency control based on the double degrees of freedom of variable frequency and phase shift. Therefore, the invention can realize smooth switching of the working direction, and avoid power impact and bus voltage fluctuation in the traditional switching process.
2. The switching control strategy of the invention only optimizes the scenes of the voltage gain exceeding the frequency conversion control limit gain, and does not conflict with the traditional frequency conversion control scheme; therefore, the control method can be directly applied to the existing CLLC circuit system as a supplementary strategy.
3. Compared with other power regulation and switching control strategies adopting multiple degrees of freedom, the invention is applied to the condition of smaller power transmission, and the characteristics and the high-efficiency topological advantages of the traditional variable frequency control soft switch and low off current are reserved under the condition of unidirectional heavy load.
4. The technical scheme of the invention not only realizes smooth change of power and current from the aspect of external electrical characteristics, but also is complete and continuous in control, and is convenient to be fused with other CLLC optimization control strategies.
5. The invention does not need to add an additional auxiliary circuit and a detection element, and only performs control optimization on the basis of traditional variable frequency control, thereby avoiding efficiency reduction and cost increase.
Drawings
Fig. 1 is a block diagram of a method of the present invention for exemplary master and CLLC circuit bi-directional switching control.
Fig. 2 is a control flow chart of the CLLC circuit bidirectional switching control method.
Fig. 3 is a circuit waveform and state diagram based on bidirectional switching control.
Detailed Description
The invention will be further described with reference to specific examples, but the scope of the invention is not limited thereto.
Firstly, it should be noted that the control method of the present invention is only suitable for the primary-secondary side bidirectional symmetrical CLLC converter circuit, and the circuit parameters thereof satisfyL rp =n 2 L rs AndC rp =C rs /n 2 is a relationship of (3).
Wherein,L rp is a primary side resonance inductance;nis the transformation ratio of the transformer;L rs the secondary resonance inductance;C rp is a primary side resonance capacitor;C rs is the secondary resonance capacitance.
The control method mainly comprises two parts of calculation of the limiting voltage gain and double-degree-of-freedom switching control.
1. Calculation of limiting gain
In the present invention, when the CLLC circuit parameters (primary side resonant inductanceL rp Secondary side resonant inductorL rs Primary side resonance capacitorC rp Secondary side resonance capacitorC rs Excitation inductor of transformerL m Sum-to-transformation rationEtc.) that has limited voltage gain for variable frequency controlM limit I.e. the voltage gain of the circuit cannot be made by variable frequency controlMLess thanM limit 。
M limit Calculated in an off-line manner, the calculation formula is as follows:
（1）
wherein,f s for switching frequency, resonant frequencyf r The calculation formula is determined by circuit parameters as follows:
（2）
that is, when the circuit parameters are determined, the variable frequency control limit gain of any CLLC circuit can be obtained through the calculationM limit 。
Because the CLLC circuit structure is bilaterally symmetrical with the primary side and the secondary side, the gain of the control scheme applied in a certain forward and reverse direction is smaller than that of the control schemeM limit In the following, the bidirectional switching control method will be described with respect to the forward direction of the step-down operation only.
2. Bidirectional switching control
In the present invention, bidirectional switching for CLLC circuits is achieved by a two-degree-of-freedom switching control. The bidirectional switching control includes: circuit state judgment, double-degree-of-freedom switching control and generation of a driving signal:
(1) Based on the circuit state detection, judging the current working direction and the current gain state of the CLLC circuit; if the circuit gain is smaller than the limit voltage gainM limit And (3) executing the step (2).
Optionally, according to the output currenti s Current input to primary sidei p Primary side input powerP p Or secondary side output powerP s The current working direction of the CLLC circuit is judged according to any circuit state. The method comprises the following steps:
to output currenti s As a basis for the purposes of this,i s >0 is a positive direction,i s <0 is reverse direction,i s =0 is the switching threshold state where the transmission power is 0. By primary side input currenti p As a basis for the purposes of this,i p >0 is a positive direction,i p <0 is reverse direction,i p =0 is the switching threshold state where the transmission power is 0. With primary input powerP p As a basis for the purposes of this,P p >0 is a positive direction,P p <0 is reverse direction,P p =0 is the switching threshold state where the transmission power is 0. With secondary side output powerP s As a basis for the purposes of this,P s >0 is a positive direction,P s <0 is reverse direction,P s =0 is the switching threshold state where the transmission power is 0.
By detecting input voltageV p And output voltageV s To determine the current gain state, the following is specific: when (when)V s /V p ≥M limit When the circuit gain is larger than or equal to the limit gain, the double-degree-of-freedom switching control is not needed;V s /V p <M limit at the time of electricityThe path gain is smaller than the limit gain, and two-degree-of-freedom switching control is required.
(2) Dual degree of freedom switching control
The double degrees of freedom of the invention is that the pointer drives signals to the primary side of the CLLC circuitG p And a secondary side drive signalG s Performing a circuit-based switching frequencyf s And phase shift angleφIs controlled by switching in two degrees of freedom.
Wherein the phase shift angleφRepresenting secondary side drive signals in buck modeG s Relative to the primary drive signalG p Phase angle with lag of on and off time;φrepresentative of boost modeG s Relative toG p Phase angle of leading off timeG s And (3) withG p The turn-on times are the same.
(2.1) for the example of switching from the forward buck state to the reverse boost stateG p AndG s the two-degree-of-freedom switching control of (a) is explained.
The control process specifically comprises the following steps:
step 1: the circuit works in a forward voltage-reducing state, and when receiving the switching signal, the circuit controls the primary side driving signalG p Switching frequencyf s Increase of secondary side driving signalG s Following upG p The phase-shifting angle being that of synchronous rectification, i.eφ=φ SR ；
Step 2: determining whether the switching frequency has reached the maximum switching frequency, i.e.f s =f smax Returning to the step 1 if the step is not established, and entering the next step if the step is established;
step 3: the working is kept at the maximum switching frequency, the driving signal of the secondary side is actively controlled, and the phase shift angle is reducedφ；
Step 4: judging whether the phase shift angle is 0, namelyφ=0, and return to step 3 is not true; description of the electricity if trueThe transmission power of the path is 0, the forward voltage reduction state is finished, and the next step is carried out;
step 5: controlling secondary side drive signalsG s Switching frequencyf s Reduction of primary drive signalG p Following upG s The phase-shifting angle being that of synchronous rectification, i.eφ=φ SR The circuit operates in a reverse boost state.
(2.2) upon switching from the reverse boost State to the Forward Buck State, forG p AndG s the step of the switching control of (2.1) is reversed, i.e., the control is performed in the order from step 5 to step 1.
It should be noted that the forward gain is less than the limit gain (i.e.V s /V p <M limit ) The control scheme comprises a step 1-5 of switching from a forward voltage reduction state to a reverse voltage reduction state and a step 5-1 of switching from the reverse voltage reduction state to the forward voltage reduction state, and belongs to the forward and reverse sequences of the same control scheme.
According to the characteristics of the circuit parameters and the gain characteristics which are symmetrical in the positive and negative working directions, when the reverse gain is smaller than the limit gain (i.eV p /V s <M limit ) In this case, the same logic as that of the control scheme is true, and only the difference is that the logic concepts of the primary side and the secondary side need to be exchanged, and the voltage reduction direction is set to be the forward direction. In particular, the method can be subdivided into the following two schemes:
(2.3) when switching from the reverse buck state to the forward boost state, the scheme is described only with respect to the buck operation direction as the forward direction according to the symmetry of the circuit, so that the state only requires the exchange of the logic concepts of the primary side and the secondary side, and the buck direction is set as the forward direction, and the specific steps are similar to those of (2.1).
(2.4) when switching from the forward boost state to the reverse buck state, the scheme is described only with respect to the buck operation direction as the forward direction according to the symmetry of the circuit, so that the state only requires the exchange of the logic concepts of the primary side and the secondary side, and the buck direction is set as the forward direction, and the specific steps are similar to those of (2.2).
In the control of the two-degree-of-freedom switching, the synchronous rectification phase shift angleφ SR Can be obtained offline by adopting fundamental wave analysis, time domain analysis, simulation analysis and other methods, and can be used for making a table and establishingφ SR And mapping with the circuit state, and realizing synchronous rectification control in a table look-up mode, thereby avoiding direct complex operation in the controller.
(3) Generating a drive signal
The driving signal generating module generates a driving signal according to the primary sideG p And a secondary side drive signalG s A corresponding drive signal for each switching tube is generated.
One specific application example:
the CLLC circuit and the bidirectional switching control system outside the limit gain proposed in this example are divided into a power section and a control section as shown in fig. 1.
1. For the power portion, with the left-to-right power transfer being forward,V p andC p respectively input bus voltage and capacitance, S 1 ~S 4 Constitutes a primary side switch bridge, A and B are the midpoints of two bridge arms,L rp AndC rp is primary side resonance inductance and capacitance,L m Andnis a high-frequency transformerTExciting inductance and turn ratio,L rs AndC rs is secondary side resonance inductance and capacitance S 5 ~S 8 The secondary side switch bridge is formed, C and D are the midpoints of two bridge arms,V s AndC s the output bus voltage and the capacitance, respectively. The CLLC circuit has symmetric primary and secondary sides, and circuit parameters satisfyingL rp =n 2 L rs AndC rp =C rs /n 2 is a relationship of (3).
2. The control part of the circuit system comprises two parts of calculation of limit gain and control of bidirectional switching.
When the CLLC circuit parameters are determined, the CLLC circuit parameters have the limit voltage gain of variable frequency controlM limit I.e. the voltage gain of the circuit cannot be made by variable frequency controlMLess thanM limit In this case, it is necessary to perform a two-degree-of-freedom switching control based on phase shifting and frequency conversion.
The calculation of the limiting gain may be accomplished off-line based on circuit parameters, with specific calculations being as described above.
Due to the symmetrical circuit structure, the gain is smaller than that of the forward voltageM limit The description of the bidirectional switching control method is given.
3. The CLLC circuit bidirectional switching control consists of three parts, namely circuit state detection, double-freedom switching control and driving signal generation:
(1) Circuit state detection
By detecting output currenti s To judge the current working direction of the machine,i s >0 is a positive direction,i s <0 is reverse direction,i s =0 is the switching threshold state where the transmission power is 0. By detecting input voltageV p And output voltageV s To determine the current gain state of the gain,V s /V p ≥M limit when the circuit gain is larger than or equal to the limit gain, the double-degree-of-freedom switching control is not needed;V s /V p <M limit when the circuit gain is smaller than the limit gain, the two-degree-of-freedom switching control is required.
(2) Dual degree of freedom switching control
The two degrees of freedom refer to the switching frequency of the circuitf s And phase shift angleφ. Wherein,φrepresenting secondary side drive signals in buck modeG s Relative toPrimary side drive signalG p Phase angle with lag of on and off time;φrepresentative of boost modeG s Relative toG p Phase angle of leading off timeG s And (3) withG p The turn-on times are the same.
Taking the switching of the forward buck state to the reverse boost state as illustrated in FIG. 2, for exampleG p AndG s the two-degree-of-freedom switching control of (2) includes the steps of:
step 1: the circuit works in a forward voltage-reducing state, and when receiving the switching signal, the circuit controls the primary side driving signalG p Switching frequencyf s Increase of secondary side driving signalG s Following upG p The phase-shifting angle being that of synchronous rectification, i.eφ=φ SR ；
Step 2: determining whether the switching frequency has reached the maximum switching frequency, i.e.f s =f smax Returning to the step 1 if the step is not established, and entering the next step if the step is established;
step 3: the working is kept at the maximum switching frequency, the driving signal of the secondary side is actively controlled, and the phase shift angle is reducedφ；
Step 4: judging whether the phase shift angle is 0, namelyφ=0, and return to step 3 is not true; if true, the circuit transmission power is 0, the forward voltage reduction state is finished, and the next step is carried out;
step 5: controlling secondary side drive signalsG s Switching frequencyf s Reduction of primary drive signalG p Following upG s The phase-shifting angle being that of synchronous rectification, i.eφ=φ SR The circuit operates in a reverse boost state.
(3) Drive signal generation
Taking the CLLC circuit shown in FIG. 1 as an example, the primary side switching bridge is formed by a switching tube S 1 ~S 4 Combined and auxiliary side switchBridge switch tube S 5 ~S 8 The corresponding driving signals are respectivelyG 1 ~ G 4 AndG 5 ~ G 8 . According to the primary drive signalG p And a secondary side drive signalG s Generating a corresponding driving signal for each switching tube, whereinG 1 AndG 4 and (3) withG p The same as,G 2 AndG 3 and (3) withG p Complementary to each other,G 5 AndG 8 and (3) withG s The same as,G 6 AndG 7 and (3) withG s Complementary.
Fig. 3 is a circuit waveform and a state diagram of the CLLC circuit bidirectional switching control method beyond the limit gain proposed in the present invention, showing the complete 6 stages of the switching process from forward buck to reverse boost:
stage 1: after receiving the switching signal of the working direction, the controller increases the drivingG p AndG s is of the switching frequency of (a)f s Phase angle of bothφ=φ SR At this time, the first and second electrodes are connected,G p in the state of active control, the control system is in an active control state,G s in a synchronous rectification passive control state, phase followsG p Varying, transmission powerP>0. Wherein,φ SR the phase angle difference in the synchronous rectification state is determined by the circuit state and can be obtained through various schemes such as a fundamental wave analysis method, a time domain analysis method, a simulation analysis method and the like.
Stage 2: when (when)f s To increase tof smax The limit switching frequency, subject to hardware conditions, cannot continue to rise,P>0 at this timeφ=φ SR For a pair ofPFurther control of (a) requires a changeφStage 3 is entered.
Stage 3: holdingf s =f smax Will beφAs a degree of freedom of the control,G s slave synchronization integerThe passive control state of the flow becomes active control,φfrom the slaveφ SR The reduction is started and the process is started,Pfurther reducing.
Stage 4: when (when)φWhen the value of the sum is =0,P=0, the circuit switching process completes the forward part, at which timeG p AndG s are in active control state.
Stage 5: in reverse boost operation, toG s In order to be actively controlled,G p for passive control of synchronous rectificationφ=φ SR . In the reduction off s In the course of (a) the process,Phold at 0 at this timeφ=φ SR =0。
Stage 6: reduction off s Up toP<0 at this timeG p Synchronous rectification passive control is still maintained, butφ=φ SR >And 0, the circuit starts to transmit power to the reverse direction, and the switching process is completed.
Based on the above, the invention breaks through the limit of the minimum voltage gain existing when the traditional CLLC circuit adopts variable frequency control, realizes the stable control of the power beyond the limit gain of variable frequency control based on the double degrees of freedom of variable frequency and phase shift, further realizes the smooth switching of the working direction, and avoids the power impact and bus voltage fluctuation in the traditional switching process.
The method provided by the invention is only optimized in control under the condition that the traditional variable frequency control cannot be implemented, and the advantages of soft switching and low turn-off current are maintained in the heavy load and limit gain range. In addition, the invention does not add an additional auxiliary circuit and a detection element, thereby avoiding efficiency reduction and cost increase.
Finally, it should also be noted that the above list is merely a few specific embodiments of the present invention. It will be apparent that the invention is not limited to the above embodiments, but that many variations of hardware circuit arrangements and control program arrangements are possible. All modifications directly derived or suggested to one skilled in the art from the present disclosure should be considered as being within the scope of the present invention.
Claims (8)
1. A CLLC circuit bidirectional switching control method outside limit gain is characterized in that the primary side and the secondary side of the CLLC circuit are in bidirectional symmetry; the control method comprises the following steps:
(1) In an off-line mode, the limit voltage gain of the variable frequency control of the CLLC circuit is calculated according to the following modeM limit ：
（1）
Resonant frequencyf r Is determined by circuit parameters and is calculated according to the following formula:
（2）
in the above-mentioned formulae, L rp is a primary side resonance inductance;L m exciting an inductor for the transformer;f s is the switching frequency;C rp is a primary side resonance capacitor;
(2) Implementing bidirectional switching control, comprising:
(2.1) judging the current working direction and the current gain state of the CLLC circuit based on circuit state detection; if the circuit gain is smaller than the limit voltage gainM limit Step (2.2) is performed;
(2.2) Primary side drive Signal for CLLC CircuitG p And a secondary side drive signalG s Performing a circuit-based switching frequencyf s And phase shift angleφIs controlled by switching the two degrees of freedom; wherein the method comprises the steps of
Phase shift angleφRepresenting secondary side drive signals in buck modeG s Relative to the primary drive signalG p Phase angle with lag of on and off time;φrepresentative of boost modeG s Relative toG p Phase angle of leading off timeG s And (3) withG p The opening time is the same;
(2.2-1) upon switching from the Forward Buck state to the reverse boost stateG p AndG s the switching control of (a) includes the steps of:
(a) The circuit works in a forward voltage-reducing state, and when receiving the switching signal, the circuit controls the primary side driving signalG p Switching frequencyf s Increase of secondary side driving signalG s Following upG p Changing, phase shifting angleφPhase shift angle for synchronous rectificationφ SR I.e.φ=φ SR ；
(b) Determining whether the switching frequency has reached the maximum switching frequency, i.e.f s =f smax Returning to the step (a) if the step (a) is not established, and entering the next step if the step (a) is established;
(c) The working is kept at the maximum switching frequency, the driving signal of the secondary side is actively controlled, and the phase shift angle is reducedφ；
(d) Judging whether the phase shift angle is 0, namelyφ=0, and return to step (c) is not true; if true, the circuit transmission power is 0, the forward voltage reduction state is finished, and the next step is carried out;
(e) Controlling secondary side drive signalsG s Switching frequencyf s Reduction of primary drive signalG p Following upG s Changing, phase shifting angleφPhase shift angle for synchronous rectificationφ SR I.e.φ=φ SR The circuit operates in a reverse boost state;
(2.2-2) upon switching from the reverse boost state to the forward buck stateG p AndG s the switching control operation content of (a) is opposite to that of the step (2.2-1), namely, the switching control operation content is controlled in the reverse order from (e) to (a);
(2.3) according to the Primary side drive SignalG p And a secondary side drive signalG s A corresponding drive signal for each switching tube is generated.
2. The method according to claim 1, wherein the switching tube S is used in the CLLC circuit with two-way symmetry of primary and secondary sides 1 ~S 4 Forms a primary side switch bridge and is composed of a switch tube S 5 ~S 8 Forms a secondary side switch bridge, and the driving signals respectively corresponding to the secondary side switch bridge are respectivelyG 1 ~ G 4 AndG 5 ~ G 8 the method comprises the steps of carrying out a first treatment on the surface of the According to the primary drive signalG p And a secondary side drive signalG s Generating a corresponding driving signal for each switching tube, whereinG 1 AndG 4 and (3) withG p The same as,G 2 AndG 3 and (3) withG p Complementary to each other,G 5 AndG 8 and (3) withG s The same as,G 6 AndG 7 and (3) withG s Complementary.
3. The method according to claim 1, wherein the CLLC circuit has circuit parameters satisfying the following conditionsL rp =n 2 L rs AndC rp =C rs /n 2 is a relationship of (2); wherein,L rp is a primary side resonance inductance;nis the transformation ratio of the transformer;L rs the secondary resonance inductance;C rp is a primary side resonance capacitor;C rs is the secondary resonance capacitance.
4. The method according to claim 1, wherein in the step (2.1), the output current is used as a referencei s Current input to primary sidei p Primary side input powerP p Or secondary side output powerP s The current working direction of the CLLC circuit is judged according to any circuit state.
5. The method of claim 4, wherein the current operating direction of the CLLC circuit is determined in accordance with any one of the following manners: to output currenti s As a basis for the purposes of this,i s >0 is a positive direction,i s <0 is reverse direction,i s =0 is the switching threshold state where the transmission power is 0; by primary side input currenti p As a basis for the purposes of this,i p >0 is a positive direction,i p <0 is reverse direction,i p =0 is the switching threshold state where the transmission power is 0; with primary input powerP p As a basis for the purposes of this,P p >0 is a positive direction,P p <0 is reverse direction,P p =0 is the switching threshold state where the transmission power is 0; with secondary side output powerP s As a basis for the purposes of this,P s >0 is a positive direction,P s <0 is reverse direction,P s =0 is the switching threshold state where the transmission power is 0.
6. The method according to claim 1, wherein in the step (2.1), the input voltage of the CLLC circuit is detected by detecting the input voltage of the CLLC circuitV p And output voltageV s To determine the current gain state:V s /V p ≥M limit when the circuit gain is larger than or equal to the limit gain, the subsequent switching control is not needed;V s /V p <M limit when the circuit gain is smaller than the limit gain, the subsequent switching control is required.
7. The method of claim 1, wherein the phase shift angle of the synchronous rectificationφ SR Is based on fundamental wave analysisAny one method of time domain analysis and simulation analysis is obtained off-line; then build up by making a tableφ SR And mapping with the circuit state, realizing synchronous rectification control by using a table look-up mode, and avoiding complex operation in the controller directly.
8. The method according to claim 1, wherein the step (2.1) and the step (2.2) are performed by detecting an input voltage of the CLLC circuitV p And output voltageV s Judging the current gain state, and describing a control scheme that the forward gain is smaller than the limit gain; the scheme comprises two conditions of switching from a forward voltage-reducing state to a reverse voltage-increasing state shown in the step (2.2-1) and switching from the reverse voltage-increasing state to the forward voltage-reducing state shown in the step (2.2-2), and belongs to the forward and reverse sequences of the same control scheme;
according to the characteristic that the circuit parameters and the gain characteristics are symmetrical in the positive working direction and the negative working direction, when the reverse gain is smaller than the limit gain, the logic of the control scheme is also established, and the difference is that the logic concepts of the primary side and the secondary side are only needed to be exchanged, and the voltage reduction direction is set to be the positive direction.
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