CN116683750A - IPOS-DC/DC converter cascading single-phase inverter and method - Google Patents

IPOS-DC/DC converter cascading single-phase inverter and method Download PDF

Info

Publication number
CN116683750A
CN116683750A CN202310694136.6A CN202310694136A CN116683750A CN 116683750 A CN116683750 A CN 116683750A CN 202310694136 A CN202310694136 A CN 202310694136A CN 116683750 A CN116683750 A CN 116683750A
Authority
CN
China
Prior art keywords
output
converter module
power
voltage
power tube
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Pending
Application number
CN202310694136.6A
Other languages
Chinese (zh)
Inventor
张岩
王子铟
李震朝
刘进军
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Xian Jiaotong University
Original Assignee
Xian Jiaotong University
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Xian Jiaotong University filed Critical Xian Jiaotong University
Priority to CN202310694136.6A priority Critical patent/CN116683750A/en
Publication of CN116683750A publication Critical patent/CN116683750A/en
Pending legal-status Critical Current

Links

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/007—Plural converter units in cascade
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H02M1/0077—Plural converter units whose outputs are connected in series
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/01—Resonant DC/DC converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/157—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators with digital control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53873—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with digital control
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention discloses an IPOS-DC/DC converter cascading single-phase inverter and a method, which belong to the technical field of electric power, and an LLC resonant converter module and a bidirectional Buck converter module adopt an input parallel output series structure; input DC power supply V in Is of (3)The pole is connected with the input positive pole of the LLC resonant converter module and the input positive pole of the bidirectional Buck converter module, and is input with a direct current power supply V in The negative electrode of the LLC resonant converter module is connected with the input negative electrode of the LLC resonant converter module and the input negative electrode of the bidirectional Buck converter module, the output positive electrode of the LLC resonant converter module is connected with the positive electrode of the full-control inverter bridge INV direct current bus, the output negative electrode of the LLC resonant converter module is connected with the output positive electrode of the bidirectional Buck converter module, the output negative electrode of the bidirectional Buck converter module is connected with the negative electrode of the full-control inverter bridge INV direct current bus, the input side and the output side of the bidirectional Buck converter module are grounded together, and the output of the full-control inverter bridge INV is connected with a single-phase alternating current power grid or an alternating current load. The single-phase inverter adopts asymmetric design of module power and output capacitance, so that double frequency ripple waves can be eliminated.

Description

IPOS-DC/DC converter cascading single-phase inverter and method
Technical Field
The invention belongs to the technical field of electric power, and particularly relates to an IPOS-DC/DC converter cascade single-phase inverter and a method.
Background
The two-stage single-phase inverter system is widely applied to high-quality electric energy conversion occasions such as new energy power generation, energy storage systems, rail transit, electric automobile interaction with a power grid and the like, the front stage of the two-stage single-phase inverter system is used for processing input direct-current electric energy through a direct-current converter, for example, tracking the maximum power point of a photovoltaic array, lifting a direct-current bus to a required voltage level through a boost converter and the like, and the rear stage of the two-stage single-phase inverter system is responsible for outputting alternating-current power for single-phase grid connection or alternating-current load. There is a significant difference in the frequency doubling ripple power between the input dc power and the output ac power, and the presence of this fluctuating power increases the requirements and size of passive components in the system, further leading to increased power losses. Therefore, intensive research on single-phase converters with active energy storage and power decoupling functions has become one of hot spot directions in recent years.
In order to avoid a series of interference generated in the power treatment process of the double frequency ripple oscillation, a passive element is usually introduced on a direct current bus, and particularly, an electrolytic capacitor with a large capacity value and an energy density far higher than that of an inductor is used for buffering a source load difference, but the electrolytic capacitor is short in service life, large in size, easy to influence the capacity value by factors such as ambient temperature and ripple current, and is not suitable for occasions with certain requirements on equipment service life.
The influence of the double frequency ripple power on the system is reduced mainly by the following methods: firstly, an active power decoupling technology is that an auxiliary branch is added in a system, and the buffer capacitor can compensate ripple power by controlling an auxiliary branch switching tube, so that the function of ripple suppression is realized, but the mode increases the number of circuit switching tubes and is complex to control; secondly, virtual impedance is introduced into a control loop in a control mode, so that the output impedance of the front-stage DC/DC converter tends to be infinite at the ripple frequency, and the method reduces the secondary ripple current on the inductor to a certain extent, but also has the problem of complex control.
However, none of the above solutions solves the inherent contradiction between the small capacitance of the dc bus and the low voltage ripple and increases the complexity of the circuit more or less.
Disclosure of Invention
The invention aims to provide an IPOS-DC/DC converter cascading single-phase inverter and a method thereof, wherein the single-phase inverter adopts an asymmetric design of module power and output capacitance, and can eliminate double frequency ripple.
In order to achieve the above purpose, the invention adopts the following technical means:
A first object of the present invention is to provide an IPOS-DC/DC converter cascade single-phase inverter comprising: input DC power supply V in The system comprises an LLC resonant converter module, a bidirectional Buck converter module and a full-control inverter bridge INV; the LLC resonant converter module and the bidirectional Buck converter module adopt an input-parallel output series structure;
the input DC power supply V in The positive electrode of the LLC resonant converter module is connected with the input positive electrode of the bidirectional Buck converter module, and the direct current power supply V is input in The negative electrode of the LLC resonant converter module is connected with the input negative electrode of the LLC resonant converter module and the input negative electrode of the bidirectional Buck converter module, the output positive electrode of the LLC resonant converter module is connected with the positive electrode of the full-control inverter bridge INV direct current bus, the output negative electrode of the LLC resonant converter module is connected with the output positive electrode of the bidirectional Buck converter module, the output negative electrode of the bidirectional Buck converter module is connected with the negative electrode of the full-control inverter bridge INV direct current bus, the input side and the output side of the bidirectional Buck converter module are grounded together, and the output of the full-control inverter bridge INV is connected with a single-phase alternating current power grid or an alternating current load.
As a further improvement of the embodiment of the invention, the LLC resonant converter module includes a primary full bridge, an isolated resonant cavity, and a secondary full bridge;
The primary full bridge comprises 4N-channel MOSFET power tubes Q with reverse diodes 1 MOSFET power tube Q 2 MOSFET power tube Q 3 MOSFET power tube Q 4 The isolation resonant cavity comprises a resonant capacitor C r1 Resonant inductance L r1 Transformer T1, transformer excitation inductance L m1 The secondary full bridge comprises an output capacitor C 1 N-channel MOSFET power tube Q with reverse diode 5 MOSFET power tube Q 6 MOSFET power tube Q 7 MOSFET power tube Q 8 The method comprises the steps of carrying out a first treatment on the surface of the MOSFET power tube Q 1 Drain electrode of (d) and MOSFET power tube Q 2 The drain electrode of (C) is connected with an input DC power supply V in Positive electrode of MOSFET power tube Q 1 Source of (d) and MOSFET power tube Q 2 The source electrodes of (a) are respectively connected with the MOSFET power tube Q 3 Drain electrode of (d) and MOSFET power tube Q 4 Drain electrode of MOSFET power tube Q 3 Source electrode of (C) is connected with MOSFET power tube Q 4 Source of (d) and input dc power V in Negative electrode of (C), resonance capacitance C r1 Is connected with MOSFET power tube Q 1 Source of (C), resonance capacitance C r1 Is connected with the resonant inductance L by the negative electrode r1 Positive electrode of (a), resonant inductance L r1 The negative electrode of the transformer T1 is connected with the positive electrode of the primary side and the exciting inductance L m1 The positive pole of the transformer T1 is connected with the exciting inductance L by the negative pole of the primary side m1 Is of the negative pole of (1) and MOSFET power tube Q 2 A source of (a); transformer T1 secondary side positive electrode is connected with MOSFET power tube Q 5 Source of (d) and MOSFET power tube Q 7 Drain electrode of MOSFET power tube Q 5 Is connected with MOSFET power tube Q 6 Drain electrode of MOSFET power tube Q 6 Source electrode of (C) is connected with MOSFET power tube Q 8 And the drain electrode of the transformer T1 and the secondary side cathode of the transformer T1, the MOSFET power tube Q 7 Source electrode of (C) is connected with MOSFET power tube Q 8 A source of (a); output capacitor C 1 The positive electrode of (a) is connected with the output positive electrode of the LLC resonant converter module, and the capacitor C is output 1 The negative electrode of the positive pole LLC resonant converter module is connected with the output negative electrode of the bidirectional Buck converter module.
As an implementation of the inventionFurther improvement of the embodiment, the bidirectional Buck converter module comprises an N-channel MOSFET power tube Q with a reverse diode 9 MOSFET power tube Q 10 Power inductance L, output capacitance C 2 The method comprises the steps of carrying out a first treatment on the surface of the MOSFET power tube Q 9 The drain electrode of (2) is connected with the input voltage V in Positive electrode of MOSFET power tube Q 9 Source electrode of (C) is connected with MOSFET power tube Q 10 And the drain electrode of the power inductance L, and the MOSFET power tube Q 10 Is connected with the input voltage V in Negative electrode of (C) and output capacitance C 2 The negative electrode of the power inductance L is connected with the output capacitor C 2 Is a positive electrode of (a); output capacitor C 2 The positive electrode of the (C) is connected with the output negative electrode of the LLC resonant converter module, and the capacitor C is output 2 The negative electrode of the capacitor is connected with the ground of the direct current bus.
As a further improvement of the embodiment of the present invention, the fully-controlled inverter bridge INV includes a MOSFET power tube S ap MOSFET power tube S an MOSFET power tube S bp MOSFET power tube S bn MOSFET power tube S ap MOSFET power tube S an MOSFET power tube S bp MOSFET power tube S bn Two pairs of the two pairs are connected in series and then in parallel.
A second object of the present invention is to provide a control method for an IPOS-DC/DC converter cascade single-phase inverter, characterized by comprising: the LLC resonant converter module flows in a one-way power mode in a full frequency doubling interval, and the bidirectional Buck converter module flows in a forward and reverse power mode in the full frequency doubling interval; the LLC resonant converter module adopts variable frequency control, adjusts the switching frequency of the system through the PR controller, and changes the voltage gain of the system so as to enable the output voltage to fluctuate in a double frequency manner on a given direct current point voltage; the bidirectional Buck converter module adopts PWM control and works in an intermittent mode, and the output voltage is regulated by regulating the duty ratio of a switching tube.
As a further improvement of the embodiment of the invention, the difference between the output alternating current power and the input direct current power is the double frequency ripple power, and the double frequency ripple power is formed by an output capacitor C 1 And output capacitance C 2 Complete buffering, its relationship satisfies:
p C1+C2 ＝p o -P in
wherein p is C1+C2 Is a capacitor C 1 And capacitor C 2 Total power absorbed, p o Is the output power of the inverter, P in For inputting dc power.
As a further improvement of the embodiment of the invention, the capacitor C 1 Capacitance C 2 DC working point voltage is different, capacitance is different, and V C1 >V C2 ,C 1 >C 2 The total capacitance voltage expression is:
v C1 ＝V C1 +v C_2ω
v C2 ＝V C2 -v C_2ω
V C1 is the output capacitance C 1 Direct current operating point of V C2 Is the output capacitance C 2 Direct current operating point v of (2) C_2ω Is the capacitance voltage doubling ripple quantity.
As a further improvement of the embodiment of the present invention, the expression of the output capacitor voltage is:
wherein v is C1 (t)、v C2 (t) represents the output capacitance C 1 、C 2 V of (V) bus Representing the DC component, P, of the DC bus voltage dc Represents the average power of the system, C 1 、C 2 Representing the output capacitance C 1 、C 2 ω represents the power frequency angular frequency of the ac output of the ac side inverter.
As a further improvement of the embodiment of the invention, the capacitive voltage ripple amplitude should not exceed the minimum dc operating point of the module:
thereby forming the constraint condition between the capacitance parameter and the DC working point voltage as follows:
wherein P is dc Represents the average power of the system, C 1 、C 2 Representing the output capacitance C 1 、C 2 Capacitance of V C1 Representing the output capacitance C 1 Direct current operating point of V C2 Representing the output capacitance C 2 ω represents the power frequency angular frequency of the ac output of the ac side inverter.
As a further improvement of the embodiment of the present invention, the bidirectional Buck converter module, which is controlled by PWM, operates in an intermittent mode, and adjusts an output voltage by adjusting a duty cycle of a switching transistor, includes:
The LLC resonant converter module and the bidirectional Buck converter module are in an asymmetric mode in power and output capacitance, direct-current bus voltage is controlled by the LLC resonant converter module and the bidirectional Buck converter module together, and alternating-current side output voltage is controlled by the full-control inverter bridge INV; the LLC resonant converter adopts a variable frequency control method, and the switching frequency is obtained by closed-loop control;
sampling output capacitance voltage v C1 As a control quantity, with a target capacitance voltage v C1 * Comparing, wherein the error obtains a switching frequency fS1 through a PR controller, and the PR controller simultaneously controls a direct-current working point and a ripple component of the capacitor voltage; the secondary side adopts synchronous rectification data, and a synchronous rectification chip is utilized to control the on and off of a switching tube in an under-resonance mode;
for the bidirectional Buck converter module, in an intermittent mode, the main switching tube and the synchronous rectifying tube do not generate complementary waves any more, and the main switching tube and the synchronous rectifying tube are switched along with the switching of the power flow direction; the duty ratio D of the main switching tube is obtained by a PR controller in a closed loop manner;
sampling input voltage V in And a DC component V of the output voltage C2 The on duty ratio D of the synchronous rectifying tube is obtained after the DSP operation 1 Realizing the intermittent synchronization of the synchronous rectifying tube by changing the rising comparison value and the falling comparison value of the triangular carrier wave in the EPWM module Accurate turn-off in mode.
Compared with the prior art, the invention has the following advantages:
the invention adopts LLC resonant converter module and bidirectional Buck converter module with input parallel output series structure, the power and output capacitance of the modules adopt asymmetric design, and the waveform of the active control capacitance is controlled by closed loop to complementarily fluctuate by frequency doubling, so as to maintain the constant bus voltage. The essence of the power distribution is that due to the capacitance C 1 Greater than the capacitance C 2 And the corresponding power is larger, so the capacitor C 1 When the voltage is increased, the absorbed double frequency power is larger than that of the system, and the capacitor C 2 Only a small amount of power is needed to compensate for the portion of the power absorbed by capacitor C1 that exceeds the system doubling power. Through asymmetric design, the distribution of the double frequency ripple power between the two capacitors is improved, so that the requirement of the direct current bus output capacitor is reduced while the low ripple of the bus is maintained. In the traditional design, the capacitors are the same in size and the direct current working point is the same, the double frequency ripple power flows into the direct current bus capacitor, the synchronous fluctuation of the low frequency ripple voltage on the capacitor is shown, the capacitor synchronously absorbs power or releases power, the voltage ripple on the direct current bus is the total ripple sum of the output voltage of the module, and in order to reduce the voltage ripple, an electrolytic capacitor with a large capacitance value is generally adopted, so that the capacitor has the advantages of huge volume, short service life and low stability, and is difficult to apply to the application occasions with high power density and long service time. The invention skillfully utilizes the design of module power difference and output capacitance value of the capacitor, under the condition of actively controlling voltage double frequency fluctuation of the capacitor, not only eliminates low-frequency ripple of bus voltage, but also reduces the requirement of bus capacitance value, and fundamentally solves the inherent contradiction between low ripple and bus small capacitance value, so that the film capacitor can replace electrolytic capacitor to become a feasible way. The two-stage single-phase inversion system designed based on the method basically eliminates the ripple of the bus voltage and obviously reduces the THD of the inverter output voltage.
Drawings
The drawings described herein are for illustration purposes only and are not intended to limit the scope of the present disclosure in any way. In addition, the shapes, proportional sizes, and the like of the respective components in the drawings are merely illustrative for aiding in understanding the present invention, and are not particularly limited. In the drawings:
FIG. 1 is a circuit topology employed by the present invention;
FIG. 2 is a closed-loop control block diagram of an LLC resonant converter module and a bi-directional Buck converter module employed in the present invention;
FIG. 3a shows the voltage v across two output capacitors without the asymmetric design of the module power and output capacitors according to the present invention C1 、v C2 Dc bus voltage v bus A waveform of the waveform;
FIG. 3b shows the AC output voltage v without the asymmetric design of the module power and output capacitance according to the present invention ac Is a waveform of (a);
FIG. 3c is a waveform of an AC output voltage without the asymmetric design of module power and output capacitance values proposed by the present invention;
FIG. 4a shows the voltage v across two output capacitors in an asymmetric design of the power and output capacitors of a module according to the present invention C1 、v C2 And dc bus voltage v bus Is a waveform of (a);
FIG. 4b shows the output voltage v under an asymmetric design of the module power and the output capacitance according to the present invention ac A waveform;
FIG. 4c shows the AC output voltage v using the complementary control algorithm according to the present invention ac Waveform.
Detailed Description
In order to make the technical solution of the present invention better understood by those skilled in the art, the technical solution of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention, and it is apparent that the described embodiments are only some embodiments of the present invention, not all embodiments. All other embodiments, which can be made by those skilled in the art based on the embodiments of the invention without making any inventive effort, shall fall within the scope of the invention.
It will be understood that when an element is referred to as being "disposed on" another element, it can be directly on the other element or intervening elements may also be present. When an element is referred to as being "connected" to another element, it can be directly connected to the other element or intervening elements may also be present. The terms "vertical," "horizontal," "left," "right," and the like are used herein for illustrative purposes only and are not meant to be the only embodiment.
Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this invention belongs. The terminology used herein in the description of the invention is for the purpose of describing particular embodiments only and is not intended to be limiting of the invention. The term "and/or" as used herein includes any and all combinations of one or more of the associated listed items.
In order to better understand the present solution, the following description will clearly and completely describe the solution in the embodiments of the present invention with reference to the accompanying drawings in the embodiments of the present invention, and it is apparent that the embodiments are only some embodiments of the present invention, not all embodiments, and are not intended to limit the scope of the disclosure of the present invention. In addition, in the following description, descriptions of well-known structures and techniques are omitted so as not to unnecessarily obscure the concepts of the present disclosure. All other embodiments, which can be made by those skilled in the art based on the embodiments of the present invention without making any inventive effort, shall fall within the scope of the present invention.
The invention is suitable for application occasions of isolated two-stage single-phase grid-connected inverters, and particularly relates to a low-frequency ripple eliminating method of a DC/DC converter cascade single-phase inverter with asymmetric module power and output capacitance design and input and output connected in series.
As shown in FIG. 1, a circuit diagram for cascading IPOS-DC/DC converters with single-phase inverter according to the present invention comprises 1 input DC power supply V in 1 LLC resonant converter module, 1 bidirectional Buck converter module and 1 full-control inverter bridgeThe INV, LLC resonant converter module and the bidirectional Buck converter module adopt an input-parallel output series (IPOS) structure;
input DC power supply V in The positive electrode of the LLC resonant converter module is connected with the input positive electrode of the bidirectional Buck converter module, and the direct current power supply V is input in The negative electrode of the LLC resonant converter module is connected with the input negative electrode of the LLC resonant converter module and the input negative electrode of the bidirectional Buck converter module, the output positive electrode of the LLC resonant converter module is connected with the positive electrode of the full-control inverter bridge INV direct current bus, the output negative electrode of the LLC resonant converter module is connected with the output positive electrode of the bidirectional Buck converter module, the output negative electrode of the bidirectional Buck converter module is connected with the negative electrode of the full-control inverter bridge INV direct current bus, the input side and the output side of the bidirectional Buck converter module are grounded together, and the output of the full-control inverter bridge INV is connected with a single-phase alternating current power grid or an alternating current load.
According to the embodiment of the invention, the output serial capacitors of the two DC-DC modules are utilized, and on the premise of not changing the existing topology, the double frequency ripple power buffering and the capacitor voltage ripple complementation are realized by the difference of the capacitance values of the output capacitors and the non-voltage equalizing design of the direct current working points of the output capacitors, namely, the rated powers of different modules are different, so that the total capacitance value of the electrolytic capacitor required by a bus is reduced, and the power density of the system is improved. In the scheme, the output capacitance corresponding to the high-power module is larger, and the output capacitance corresponding to the low-power module is smaller, so that the compensation of ripple power can be realized only by a smaller fluctuation value of the output capacitance. The high-power module in the proposal can be applied to a unidirectional DC-DC module and a bidirectional DC-DC module through reasonable circuit parameter design, has flexibility, and adopts a bidirectional DC/DC converter. Compared with the traditional scheme, the scheme greatly reduces the demand on the direct current bus capacitor, basically eliminates the double frequency ripple of the direct current bus voltage, avoids the double frequency ripple from being fed forward to the input end of the DC/DC converter, and improves the output power quality of the rear-stage inverter.
The LLC resonant converter module comprises a primary full bridge, an isolation resonant cavity and a secondary full bridge, wherein the primary full bridge comprises 4 diodes with reverse directionsN-channel MOSFET power tube Q of tube 1 MOSFET power tube Q 2 MOSFET power tube Q 3 MOSFET power tube Q 4 The isolation resonant cavity comprises a resonant capacitor C r1 Resonant inductance L r1 Transformer T1, transformer excitation inductance L m1 The secondary full bridge comprises 4N-channel MOSFET power tubes Q with reverse diodes 5 MOSFET power tube Q 6 MOSFET power tube Q 7 MOSFET power tube Q 8 Output capacitance C 1 . MOSFET power tube Q 1 Drain electrode of (d) and MOSFET power tube Q 2 The drain electrode of (C) is connected with an input DC power supply V in Positive electrode of MOSFET power tube Q 1 Source of (d) and MOSFET power tube Q 2 The source electrodes of (a) are respectively connected with the MOSFET power tube Q 3 Drain electrode of (d) and MOSFET power tube Q 4 Drain electrode of MOSFET power tube Q 3 Source electrode of (C) is connected with MOSFET power tube Q 4 Source of (d) and input dc power V in Negative electrode of (C), resonance capacitance C r1 Is connected with MOSFET power tube Q 1 Source of (C), resonance capacitance C r1 Is connected with the resonant inductance L by the negative electrode r1 Positive electrode of (a), resonant inductance L r1 The negative electrode of the transformer T1 is connected with the positive electrode of the primary side and the exciting inductance L m1 The positive pole of the transformer T1 is connected with the exciting inductance L by the negative pole of the primary side m1 Is of the negative pole of (1) and MOSFET power tube Q 2 Is a source of (c). Transformer T1 secondary side positive electrode is connected with MOSFET power tube Q 5 Source of (d) and MOSFET power tube Q 7 Drain electrode of MOSFET power tube Q 5 Is connected with MOSFET power tube Q 6 Drain electrode of MOSFET power tube Q 6 Source electrode of (C) is connected with MOSFET power tube Q 8 And the drain electrode of the transformer T1 and the secondary side cathode of the transformer T1, the MOSFET power tube Q 7 Source electrode of (C) is connected with MOSFET power tube Q 8 Is a source of (c). Output capacitor C 1 The positive electrode of (a) is connected with the output positive electrode of the LLC resonant converter module, and the capacitor C is output 1 The negative electrode of the positive pole LLC resonant converter module is connected with the output negative electrode of the bidirectional Buck converter module.
The bidirectional Buck converter module is composed of an N-channel MOSFET power tube Q with a reverse diode 9 MOSFET power tube Q 10 Power inductance L, output capacitance C 2 Composition is prepared. MOSFET workRate tube Q 9 The drain electrode of (2) is connected with the input voltage V in Positive electrode of MOSFET power tube Q 9 Source electrode of (C) is connected with MOSFET power tube Q 10 And the drain electrode of the power inductance L, and the MOSFET power tube Q 10 Is connected with the input voltage V in Negative electrode of (C) and output capacitance C 2 The negative electrode of the power inductance L is connected with the output capacitor C 2 Is a positive electrode of (a). Output capacitor C 2 The positive electrode of the (C) is connected with the output negative electrode of the LLC resonant converter module, and the capacitor C is output 2 The negative electrode of the capacitor is connected with the ground of the direct current bus.
Output capacitor C 1 And output capacitance C 2 The input end of the series connection full control inverter bridge INV; positive pole output of full-control inverter bridge INV is connected with filter inductance L f The negative output of the full-control inverter bridge INV is connected with the public ground of the alternating current side; filter capacitor C f And load resistance R L Parallel connection, one end is connected with an inductance L f The other end is connected with the common ground of the alternating current side.
Preferably, the fully-controlled inverter bridge consists of four MOSFET power tubes S ap 、S an 、S bp 、S bn Composition is prepared. The full-control inverter bridge INV comprises a MOSFET power tube S ap MOSFET power tube S an MOSFET power tube S bp MOSFET power tube S bn MOSFET power tube S ap MOSFET power tube S an MOSFET power tube S bp MOSFET power tube S bn Two pairs of the two pairs are connected in series and then in parallel.
According to the transfer relation of the double frequency power of the two-stage IPOS-DCDC module cascading single-phase inverter, under the conditions of difference module power and capacitance value of the capacitor, the capacitor voltage ripple is designed to be in a complementary state, so that the double frequency ripple power difference between the system source load and the output capacitor is completely born, the redistribution of the double frequency power between the two capacitors is optimized, the low-frequency ripple of the direct current bus voltage is eliminated, and the required total capacitance value of the direct current bus is reduced.
Wherein the difference between the output AC power and the input DC power is a double frequency ripple power, and the power is obtained by an output capacitor C 1 And output capacitance C 2 Complete buffering, its relationship satisfies:
p C1+C2 ＝p o -P in
Wherein p is C1+C2 Is a capacitor C 1 And capacitor C 2 Total power absorbed, p o Is the output power of the inverter, P in For inputting dc power.
Preferably, the control objective of the IPOS-DCDC converter cascade single-phase inverter with low-frequency ripple suppression capability is to transfer all the differences between the inverter output power and the input DC power to the DC bus capacitor C 1 And C 2 And the output capacitor voltage ripple complementarily fluctuates in double frequency by control, so that the double frequency ripple on the direct current bus voltage is reduced.
The power and output capacitance of the two DC-DC modules are both designed asymmetrically, and the capacitance C 1 Capacitance C 2 DC working point voltage is different, capacitance is different, and V C1 >V C2 ,C 1 >C 2 By manually controlling the voltage difference of the direct-current working points of the capacitors, and the ripple voltage complementarily fluctuates in a double frequency mode, the power absorption of one capacitor at any moment is realized, the power is released by the other capacitor, and the total capacitor voltage expression is as follows:
v C1 ＝V C1 +v C_2ω
v C2 ＝V C2 -v C_2ω
V C1 is the output capacitance C 1 Direct current operating point of V C2 Is the output capacitance C 2 Direct current operating point v of (2) C_2ω Is the capacitance voltage doubling ripple quantity.
The high-capacity electrolytic capacitor is used for buffering most of the double-frequency ripple power, the low-power module is used for buffering a small part of the double-frequency ripple power, and the power variation is kept equal to the power difference between the input and the output of the system while the bus voltage is kept constant.
As a further improvement of the invention, the IPOS-DC/DC converter is cascaded with a single-phase inverter, and a design flow of module power and output capacitance is given so as to realize different application occasions of power unidirectional and power bidirectional or power bidirectional and power bidirectional. When the difference between the power of the module and the product of the output capacitance is larger than a specific condition, the high-power module works in a power unidirectional mode, and when the difference is smaller than the specific condition, the high-power module works in a power bidirectional mode. The low-power modules all need to work in a power bidirectional mode.
As a further improvement, under the asymmetric design scheme of the module power and the output capacitance, when the power and the output capacitance meet certain design conditions, the LLC resonant converter module flows in one direction in the full frequency doubling interval, and the bidirectional Buck module requires forward and reverse power flow in the full frequency doubling interval. The LLC resonant converter module adopts variable frequency control, adjusts the switching frequency of the system through the PR controller, and changes the voltage gain of the system so as to realize the double frequency fluctuation of the output voltage on a given direct current point voltage; the bidirectional Buck module adopts PWM control, works in an intermittent mode, and adjusts output voltage by adjusting the duty ratio of a switching tube.
The LLC resonant converter module adopts synchronous rectification technology, and works in an under-resonance area through reasonable parameter design, so that ZVS of a primary side switching tube and ZCS of a secondary side switching tube are realized, and system loss is reduced. The bidirectional Buck module adopts synchronous rectification technology, the main switching tube and the synchronous rectifying tube do not generate complementary waves in the intermittent mode, and the turn-off signal of the synchronous rectifying tube is calculated by the digital control chip and then is given.
When the power and the output capacitance meet certain design conditions, the LLC resonant converter module flows in one direction in a full frequency doubling interval, and the bidirectional Buck module flows in the forward and reverse directions in the full frequency doubling interval. The LLC resonant converter module adopts variable frequency control, adjusts the switching frequency of the system through the PR controller, and changes the voltage gain of the system so as to realize the double frequency fluctuation of the output voltage on a given direct current point voltage; the bidirectional Buck module adopts PWM control, works in an intermittent mode, and adjusts output voltage by adjusting the duty ratio of a switching tube.
The expression of the output capacitance voltage is:
wherein v is C1 (t)、v C2 (t) represents the output capacitance C 1 、C 2 V of (V) bus Representing the DC component, P, of the DC bus voltage dc Represents the average power of the system, C 1 、C 2 Representing the output capacitance C 1 、C 2 ω represents the power frequency angular frequency of the ac output of the ac side inverter.
The ripple amplitude of the capacitance voltage should not exceed the minimum dc operating point of the module:
thereby forming a constraint condition between the capacitance parameter and the DC working point voltage:
the output current expression of the DC-DC module is:
wherein i is out1 (t)、i out2 (t) represents the output currents of LLC resonant converter and bidirectional Buck resonant converter, respectively, V bus Representing the DC component, P, of the DC bus voltage dc Represents the average power of the system, C 1 、C 2 Representing the output capacitance C 1 、C 2 ω represents the power frequency angular frequency of the ac output of the ac side inverter.
Output current i to module 1 out1 (t) sorting the discussion, when the module employed is a power unidirectional module, outputting a current i out1 And (t) is constantly greater than 0, and the amplitude of the frequency doubling component and the direct current component meet the relation:
thereby obtaining the following steps:
wherein P is dc Represents the average power of the system, C 1 、C 2 Representing the output capacitance C 1 、C 2 Capacitance of V C1 Representing the output capacitance C 1 Direct current operating point of V C2 Representing the output capacitance C 2 ω represents the power frequency angular frequency of the ac output of the ac side inverter.
When the capacitance of the system and the voltage design and distribution of the direct current working point meet the above relation, the module 1 only needs to work in a power unidirectional mode, and when the relation is not met, the module 1 needs to work in a power forward and reverse switching mode. The greater the difference between the dc operating point voltage and the capacitance, the easier it is to satisfy this relationship, and the easier it is for the module 1 to operate in the power unidirectional mode.
Output current i to module 2 out2 (t) analysis due to V C1 >V C2 ，C 1 >C 2 Thus i out2 The amplitude of the frequency-doubled component of (t)Is constantly greater than its DC component +.>Within a doubling interval, i out2 (t) there must be a section less than 0, at which time the module 2 is required to have power bidirectional flow and seamless switching capability.
Under the asymmetric design scheme of module power and output capacitance, the direct current bus voltage is commonly controlled by the two DC-DC modules at the front stage, and the output voltage at the alternating current side is controlled by the full-control inverter bridge. The LLC resonant converter adopts a variable frequency control method, and the primary side switching tube is 50%The square wave signals of the upper and lower switching tubes of the same bridge arm are complementary, and the switching frequency is obtained by closed-loop control. Sampling output capacitance voltage v C1 As a control quantity, with a target capacitance voltage v C1 * Comparing, and obtaining the switching frequency f by the error through the PR controller S1 The PR controller simultaneously controls the DC operating point and the ripple component of the capacitor voltage. The secondary side adopts synchronous rectification technology, and accurate on and off of the switching tube in the under-resonance mode is realized by utilizing a synchronous rectification chip.
For the bidirectional Buck module, in the intermittent mode, the main switching tube and the synchronous rectifying tube do not generate complementary waves, and the main switching tube and the synchronous rectifying tube are switched along with the switching of the power flow direction. The duty ratio D of the main switching tube is obtained by the PR controller in a closed loop manner, and the duty ratio D of the synchronous rectifying tube 1 From the law of volt-second equilibrium of inductance:
sampling input voltage V in And a DC component V of the output voltage C2 The conducting duty ratio D of the synchronous rectifying tube can be obtained after the DSP operation 1 The accurate turn-off of the synchronous rectifier tube in the intermittent mode can be realized by changing the rising comparison value and the falling comparison value of the triangular carrier wave in the EPWM module.
The control method of the present invention is described in detail below with reference to specific examples:
under the asymmetric design of module power and output capacity, the analysis process of the low-frequency ripple suppression scheme of the IPOS-LLC resonant converter and bidirectional Buck converter module cascading single-phase inverter system is as follows:
when the single-phase inverter is in grid-connected operation or connected with an alternating current load, the input direct current is generally converted into power frequency alternating current by adopting a PWM (pulse width modulation) technology, and the alternating current output side of the single-phase inverter is connected with a single-phase alternating current power grid or an alternating current load after the switching frequency harmonic wave is filtered by an LC (inductance-capacitance) filter.
The specific implementation of the PWM pulse width modulation technique is: selecting modulation wave as single-phase power grid frequency sine wave v con ＝Msin(ωt), wherein M is modulation degree, and the carrier wave is bipolar high-frequency triangular wave v tri When v tri <v con At the time S ap And S is bn Conduction, S an And S is bp Turn-off, the midpoint voltage of two bridge arms of the inverter is v bus . When v tri >v con At the time S an And S is bp Conduction, S ap And S is bn Turn-off, the midpoint voltage of two bridge arms of the inverter is-v bus . Grid-connected voltage v of inverter ac (t) is the average value of the bridge arm midpoint voltage in one carrier period:
v ac (t)＝MV bus sin(ωt)＝V ac sin(ωt) (1)
wherein M is modulation degree, V bus Is the DC bus voltage, omega is the angular frequency of the output AC voltage, V ac Is the amplitude of the ac voltage.
Assuming that the ac side voltage and current phases are the same, the inverter ac side current is obtained as:
i ac (t)＝I ac sin(ωt) (2)
wherein ω is the angular frequency of the output AC current, I ac Is the amplitude of the alternating current.
The available AC side power p ac (t) is:
wherein ω is the angular frequency of the output AC voltage, V ac Is the amplitude of the alternating voltage, I ac Is the amplitude of the alternating current.
Since the front stage IPOS-DCDC module only provides DC power P dc The partial dc power is the same as the dc component in the ac side power, namely:
thus, the input DC power P can be obtained dc And output AC side power p ac (t) Power difference p of (2) e (t)：
Assuming a DC bus voltage V bus Constant, the DC bus current i can be obtained by conservation of power bus (t)
Wherein I is bus Is the direct current bus current i bus The direct current component of (t), i bus_2ω Is the direct current bus current i bus A frequency-doubled component of (t).
Assuming that the output capacitor voltage ripple component fluctuates by a frequency doubling, the capacitor voltage is expressed as:
wherein V is C1 、V C2 To output DC components of capacitor voltage, the magnitudes of which are different, commonly support bus DC voltage, i.e. V C1 +V C2 ＝V bus (V C1 >V C2 )，v C1_2ω For the amplitude of the ripple quantity of the output capacitor voltage, θ is the output capacitor voltage v C1 (t) phase difference from the output side AC voltage, when the capacitance voltage satisfies the ripple complement, the bus voltage V bus Constant.
The voltage-current relationship of the capacitor can be obtained, and the expression of the capacitor current is as follows:
the total power absorbed by the available output capacitance is:
assuming a frequency doubling power difference p of the system e (t) is entirely composed of the output capacitance C 1 Output capacitance C 2 Buffering:
2ω(V C1 C 1 -V C2 C 2 )v C1_2ω cos(2ωt+θ)＝P dc cos(2ωt) (10)
by solving the equation, the amplitude and phase angle of the required capacitor voltage ripple can be solved as follows:
the final expression for the target capacitance voltage is:
substituting (11) into (8), and similarly obtaining the final expression of the capacitance current as follows:
as known from kirchhoff's current law, the output current expression of the DC-DC module is:
output current i to module 1 out1 (t) sorting the discussion, when the module employed is a power unidirectional module, outputting a current i out1 And (t) is constantly greater than 0, and the amplitude of the frequency doubling component and the direct current component meet the relation:
thereby obtaining the following steps:
when the capacitance value and the direct current working point voltage design distribution of the system meet the relation of the formula (16), the module 1 only needs to work in a power unidirectional mode, and when the relation of the formula (16) is not met, the module 1 needs to work in a power forward and reverse switching mode. The greater the difference between the dc operating point voltage and the capacitance, the easier it is to satisfy this relationship, and the easier it is for the module 1 to operate in the power unidirectional mode.
Output current i to module 2 out2 (t) analysis due to V C1 >V C2 ，C 1 >C 2 Thus i out2 The amplitude of the frequency-doubled component of (t)Is constantly greater than its DC component +.>Within a doubling interval, i out2 (t) there must be a section less than 0, at which time the module 2 is required to have power bidirectional flow and seamless switching capability.
As can be obtained from the condition that the capacitance voltage is constantly greater than 0, the ripple amplitude of the capacitance voltage should not exceed the minimum dc operating point of the module:
thereby forming a constraint condition between the capacitance parameter and the DC working point voltage:
the embodiment of the invention provides a control method of a double frequency ripple suppression circuit of an LLC resonant converter with parallel input and serial output and a bidirectional Buck converter module cascading full-control inverter bridge INV, which specifically comprises the following steps:
the direct current bus voltage is commonly controlled by the two DC-DC modules of the front stage, and the output voltage of the alternating current side is controlled by the fully-controlled inverter bridge INV. The LLC resonant converter adopts a variable frequency control method, the primary side switching tube is a 50% square wave signal, the upper switching tube signal and the lower switching tube signal of the same bridge arm are complementary, and the switching frequency is obtained by closed loop control. Sampling output capacitance voltage v C1 As a control quantity, with a target capacitance voltage v C1 * Comparing, and obtaining the switching frequency f by the error through the PR controller S1 The PR controller simultaneously controls the DC operating point and the ripple component of the capacitor voltage. The secondary side adopts synchronous rectification technology, and accurate on and off of the switching tube in the under-resonance mode is realized by utilizing a synchronous rectification chip.
For the bidirectional Buck converter module, in the intermittent mode, the main switching tube and the synchronous rectifying tube do not generate complementary waves any more, and the main switching tube and the synchronous rectifying tube are switched along with the switching of the power flow direction. The duty ratio D of the main switching tube is obtained by the PR controller in a closed loop manner, and the duty ratio D of the synchronous rectifying tube 1 From the law of volt-second equilibrium of inductance:
sampling input voltage V in And a DC component V of the output voltage C2 The conducting duty ratio D of the synchronous rectifying tube can be obtained after the DSP operation 1 The accurate turn-off of the synchronous rectifier tube in the intermittent mode can be realized by changing the rising comparison value and the falling comparison value of the triangular carrier wave in the EPWM module.
A control block diagram of the converter is thus established as shown in fig. 2.
Through direct control to the DC bus voltage ripple, the voltage on the capacitor is stabilized at a DC working point, and the ripple is controlled to be in a complementary state, so that the requirements of the DC bus voltage ripple and the DC bus capacitor are reduced while the power conversion of the converter is realized.
To verify the theoretical analysis of the converter, the invention gives a design example.
The converter parameters are as follows: v (V) bus ＝400V、I bus ＝1A、V in ＝60V、V C1 ＝350V、V C2 ＝50V、f S_buck ＝150kHz、f inv ＝60kHz、f ac ＝50Hz、C 1 ＝600μF、C 2 ＝200μF、n＝0.2、L r ＝1.1μH、L m1 ＝3.3μH、L＝2μH、C r ＝93nF、V ac_rms ＝220V、L f ＝2mH、C f ＝0.43μF、R L ＝121Ω、M＝0.777。
FIG. 3a shows the voltage v across the output capacitance of two DC-DC modules (each employing LLC resonant converter modules) without the asymmetric design of module power and output capacitance values as proposed by the present invention C1 、v C2 Dc bus voltage v bus V of (v) C1 、v C2 The direct current working points are the same, the voltage ripple wave synchronously fluctuates in a double frequency mode, the peak value of the fluctuation voltage is 11V, and the voltage is superimposed on the direct current bus voltage, the direct current bus voltage V bus The amount of double frequency fluctuation appears, and the peak value of the ripple peak is 22V.
FIG. 3b shows the input side current i of two DC-DC modules designed asymmetrically without the proposed module power and output capacitance in1 、i in2 And total input current i in The waveform of (i.e., the power supply current) the double frequency ripple current flows into the input side such that the direct current power supply current is no longer constant, and the peak-to-peak value of the double frequency ripple current is 8.59A. If devices sensitive to current ripple, such as an energy storage unit and a battery bank, are connected to the input side, the ripple current will cause serious heat generation, accelerate aging of the device, and are difficult to apply to electric energy conversion occasions with long service life.
FIG. 3c shows an AC output voltage v without the asymmetric design of module power and output capacitance ac Is a waveform of (a). At the input DC bus voltage v bus Under the condition of double frequency ripple, the output voltage waveform of the rear-stage full-bridge inverter generates certain distortion, and THD is 1.89%.
FIG. 4a shows the power and power output of a module according to the invention Output capacitance value asymmetric design, voltage v on output capacitance of two DC-DC modules C1 、v C2 And dc bus voltage v bus As can be seen from the waveforms of (c), v C1 And v C2 Direct current working point voltages are different, average power of modules is not equally divided, V C1 ＝350V、V C2 =50v, and the capacitance is different, C 1 ＝600μF、C 2 =200 μf. By closed loop control, the ripple voltage waveform forms a complementary state as desired. The voltage ripple of the direct current bus is reduced to about 0.4V, which is the ripple of the high-frequency component of the system, and the low-frequency ripple component is basically eliminated, so that compared with 22V which is not controlled by an algorithm, the ripple suppression effect is quite obvious.
FIG. 4b shows the input side current i of two DC-DC modules designed asymmetrically with the module power and output capacitance values proposed by the present invention in1 、i in2 And total input current i in The output current of the single module fluctuates in a double frequency ripple waveform mode, the power is different, the direct current working points are different, but the fluctuation amplitude of the double frequency ripple is the same, the fluctuation directions are opposite, the total input current is kept basically constant after superposition, namely, the input side power supply only needs to provide direct current power, and compared with the traditional design, the influence of the secondary ripple current on input side equipment is obviously reduced.
FIG. 4c shows the AC output voltage v using the complementary control algorithm of the present invention ac Waveform. Under the condition that the ripple of the input direct current bus voltage is improved, the waveform distortion of the output voltage is reduced, and the THD of the output voltage is reduced to 1.08%.
Many embodiments and many applications other than the examples provided will be apparent to those of skill in the art upon reading the above description. The scope of the present teachings should, therefore, be determined not with reference to the above description, but instead should be determined with reference to the appended claims, along with the full scope of equivalents to which such claims are entitled. The disclosures of all articles and references, including patent applications and publications, are incorporated herein by reference for the purpose of completeness. The omission of any aspect of the subject matter disclosed herein in the preceding claims is not intended to forego such subject matter, nor should the applicant be deemed to have such subject matter not considered to be part of the disclosed subject matter.
The foregoing is a further elaboration of the present invention, and it is not intended that the invention be limited to the specific embodiments shown, but rather that a number of simple deductions or substitutions be made by one of ordinary skill in the art without departing from the spirit of the invention, all shall be deemed to fall within the scope of the invention as defined by the claims which are filed herewith.

Claims (10)

1. An IPOS-DC/DC converter cascaded single-phase inverter comprising: input DC power supply V in The system comprises an LLC resonant converter module, a bidirectional Buck converter module and a full-control inverter bridge INV; the LLC resonant converter module and the bidirectional Buck converter module adopt an input-parallel output series structure;
the input DC power supply V in The positive electrode of the LLC resonant converter module is connected with the input positive electrode of the bidirectional Buck converter module, and the direct current power supply V is input in The negative electrode of the LLC resonant converter module is connected with the input negative electrode of the LLC resonant converter module and the input negative electrode of the bidirectional Buck converter module, the output positive electrode of the LLC resonant converter module is connected with the positive electrode of the full-control inverter bridge INV direct current bus, the output negative electrode of the LLC resonant converter module is connected with the output positive electrode of the bidirectional Buck converter module, the output negative electrode of the bidirectional Buck converter module is connected with the negative electrode of the full-control inverter bridge INV direct current bus, the input side and the output side of the bidirectional Buck converter module are grounded together, and the output of the full-control inverter bridge INV is connected with a single-phase alternating current power grid or an alternating current load.
2. The IPOS-DC/DC converter cascade single-phase inverter of claim 1, wherein the LLC resonant converter module includes a primary full bridge, an isolated resonant cavity, a secondary full bridge;
The primary full bridge comprises 4N-channel MOSFET power tubes Q with reverse diodes 1 MOSFET power tube Q 2 MOSFET power tube Q 3 MOSFET power tube Q 4 The isolation resonant cavity comprisesResonance capacitor C r1 Resonant inductance L r1 Transformer T1, transformer excitation inductance L m1 The secondary full bridge comprises an output capacitor C 1 N-channel MOSFET power tube Q with reverse diode 5 MOSFET power tube Q 6 MOSFET power tube Q 7 MOSFET power tube Q 8 The method comprises the steps of carrying out a first treatment on the surface of the MOSFET power tube Q 1 Drain electrode of (d) and MOSFET power tube Q 2 The drain electrode of (C) is connected with an input DC power supply V in Positive electrode of MOSFET power tube Q 1 Source of (d) and MOSFET power tube Q 2 The source electrodes of (a) are respectively connected with the MOSFET power tube Q 3 Drain electrode of (d) and MOSFET power tube Q 4 Drain electrode of MOSFET power tube Q 3 Source electrode of (C) is connected with MOSFET power tube Q 4 Source of (d) and input dc power V in Negative electrode of (C), resonance capacitance C r1 Is connected with MOSFET power tube Q 1 Source of (C), resonance capacitance C r1 Is connected with the resonant inductance L by the negative electrode r1 Positive electrode of (a), resonant inductance L r1 The negative electrode of the transformer T1 is connected with the positive electrode of the primary side and the exciting inductance L m1 The positive pole of the transformer T1 is connected with the exciting inductance L by the negative pole of the primary side m1 Is of the negative pole of (1) and MOSFET power tube Q 2 A source of (a); transformer T1 secondary side positive electrode is connected with MOSFET power tube Q 5 Source of (d) and MOSFET power tube Q 7 Drain electrode of MOSFET power tube Q 5 Is connected with MOSFET power tube Q 6 Drain electrode of MOSFET power tube Q 6 Source electrode of (C) is connected with MOSFET power tube Q 8 And the drain electrode of the transformer T1 and the secondary side cathode of the transformer T1, the MOSFET power tube Q 7 Source electrode of (C) is connected with MOSFET power tube Q 8 A source of (a); output capacitor C 1 The positive electrode of (a) is connected with the output positive electrode of the LLC resonant converter module, and the capacitor C is output 1 The negative electrode of the positive pole LLC resonant converter module is connected with the output negative electrode of the bidirectional Buck converter module.
3. The IPOS-DC/DC converter cascade single-phase inverter of claim 1, wherein the bi-directional Buck converter module includes an N-channel MOSFET power tube Q with a reverse diode 9 MOSFET power tube Q 10 Power inductance L, output capacitance C 2 The method comprises the steps of carrying out a first treatment on the surface of the MOSFET power tube Q 9 The drain electrode of (2) is connected with the input voltage V in Positive electrode of MOSFET power tube Q 9 Source electrode of (C) is connected with MOSFET power tube Q 10 And the drain electrode of the power inductance L, and the MOSFET power tube Q 10 Is connected with the input voltage V in Negative electrode of (C) and output capacitance C 2 The negative electrode of the power inductance L is connected with the output capacitor C 2 Is a positive electrode of (a); output capacitor C 2 The positive electrode of the (C) is connected with the output negative electrode of the LLC resonant converter module, and the capacitor C is output 2 The negative electrode of the capacitor is connected with the ground of the direct current bus.
4. The IPOS-DC/DC converter cascade single-phase inverter according to claim 1, wherein the fully controlled inverter bridge INV comprises MOSFET power tubes S ap MOSFET power tube S an MOSFET power tube S bp MOSFET power tube S bn MOSFET power tube S ap MOSFET power tube S an MOSFET power tube S bp MOSFET power tube S bn Two pairs of the two pairs are connected in series and then in parallel.
5. The control method of the IPOS-DC/DC converter cascade single-phase inverter according to any one of claims 1 to 4, comprising: the LLC resonant converter module flows in a one-way power mode in a full frequency doubling interval, and the bidirectional Buck converter module flows in a forward and reverse power mode in the full frequency doubling interval; the LLC resonant converter module adopts variable frequency control, adjusts the switching frequency of the system through the PR controller, and changes the voltage gain of the system so as to enable the output voltage to fluctuate in a double frequency manner on a given direct current point voltage; the bidirectional Buck converter module adopts PWM control and works in an intermittent mode, and the output voltage is regulated by regulating the duty ratio of a switching tube.
6. The method for controlling a cascaded single-phase inverter of an IPOS-DC/DC converter as claimed in claim 5, wherein the difference between the output AC power and the input DC power is a double frequency ripple power, the double frequency ripple power being formed by an output capacitor C 1 And output capacitance C 2 Complete buffering, its relationship satisfies:
p C1+C2 ＝p o -P in
wherein p is C1+C2 Is a capacitor C 1 And capacitor C 2 Total power absorbed, p o Is the output power of the inverter, P in For inputting dc power.
7. The method for controlling a cascaded single-phase inverter of an IPOS-DC/DC converter as claimed in claim 5,
capacitor C 1 Capacitance C 2 DC working point voltage is different, capacitance is different, and V C1 >V C2 ,C 1 >C 2 The total capacitance voltage expression is:
v C1 ＝V C1 +v C_2ω
v C2 ＝V C2 -v C_2ω
V C1 is the output capacitance C 1 Direct current operating point of V C2 Is the output capacitance C 2 Direct current operating point v of (2) C_2ω Is the capacitance voltage doubling ripple quantity.
8. The method for controlling a cascaded single-phase inverter of an IPOS-DC/DC converter as claimed in claim 5,
the expression of the output capacitance voltage is:
wherein v is C1 (t)、v C2 (t) represents the output capacitance C 1 、C 2 V of (V) bus Representing the DC component, P, of the DC bus voltage dc Represents the average power of the system, C 1 、C 2 Representing the output capacitance C 1 、C 2 ω represents the power frequency angular frequency of the ac output of the ac side inverter.
9. The method for controlling a cascaded single-phase inverter of an IPOS-DC/DC converter as claimed in claim 5,
the capacitor voltage ripple amplitude should not exceed the minimum dc operating point of the module:
thereby forming the constraint condition between the capacitance parameter and the DC working point voltage as follows:
Wherein P is dc Represents the average power of the system, C 1 、C 2 Representing the output capacitance C 1 、C 2 Capacitance of V C1 Representing the output capacitance C 1 Direct current operating point of V C2 Representing the output capacitance C 2 ω represents the power frequency angular frequency of the ac output of the ac side inverter.
10. The method for controlling the IPOS-DC/DC converter cascade single-phase inverter according to claim 5, wherein the bidirectional Buck converter module is controlled by PWM, and operates in the discontinuous mode to regulate the output voltage by adjusting the duty ratio of the switching transistor, comprising:
the LLC resonant converter module and the bidirectional Buck converter module are in an asymmetric mode in power and output capacitance, direct-current bus voltage is controlled by the LLC resonant converter module and the bidirectional Buck converter module together, and alternating-current side output voltage is controlled by the full-control inverter bridge INV; the LLC resonant converter adopts a variable frequency control method, and the switching frequency is obtained by closed-loop control;
sampling output capacitance voltage v C1 As a control quantity, with a target capacitance voltage v C1 * Comparing, and obtaining the switching frequency f by the error through the PR controller S1 The PR controller simultaneously controls the direct-current working point and the ripple component of the capacitor voltage; the secondary side adopts synchronous rectification data and uses synchronization The rectifying chip controls the on and off of the switching tube in the under-resonance mode;
for the bidirectional Buck converter module, in an intermittent mode, the main switching tube and the synchronous rectifying tube do not generate complementary waves any more, and the main switching tube and the synchronous rectifying tube are switched along with the switching of the power flow direction; the duty ratio D of the main switching tube is obtained by a PR controller in a closed loop manner;
sampling input voltage V in And a DC component V of the output voltage C2 The on duty ratio D of the synchronous rectifying tube is obtained after the DSP operation 1 The accurate turn-off of the synchronous rectifier tube in the intermittent mode is realized by changing the rising comparison value and the falling comparison value of the triangular carrier wave in the EPWM module.
CN202310694136.6A 2023-06-12 2023-06-12 IPOS-DC/DC converter cascading single-phase inverter and method Pending CN116683750A (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN202310694136.6A CN116683750A (en) 2023-06-12 2023-06-12 IPOS-DC/DC converter cascading single-phase inverter and method

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN202310694136.6A CN116683750A (en) 2023-06-12 2023-06-12 IPOS-DC/DC converter cascading single-phase inverter and method

Publications (1)

Publication Number Publication Date
CN116683750A true CN116683750A (en) 2023-09-01

Family

ID=87786969

Family Applications (1)

Application Number Title Priority Date Filing Date
CN202310694136.6A Pending CN116683750A (en) 2023-06-12 2023-06-12 IPOS-DC/DC converter cascading single-phase inverter and method

Country Status (1)

Country Link
CN (1) CN116683750A (en)

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117277821A (en) * 2023-11-17 2023-12-22 湖南北顺源智能科技有限公司 Mixed type high-voltage direct-current electric energy converter system of submarine observation network and control method
CN117353764A (en) * 2023-12-04 2024-01-05 湖南北顺源智能科技有限公司 High-power cascade power amplifier system for underwater acoustic communication and control method thereof

Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117277821A (en) * 2023-11-17 2023-12-22 湖南北顺源智能科技有限公司 Mixed type high-voltage direct-current electric energy converter system of submarine observation network and control method
CN117277821B (en) * 2023-11-17 2024-02-06 湖南北顺源智能科技有限公司 Mixed type high-voltage direct-current electric energy converter system of submarine observation network and control method
CN117353764A (en) * 2023-12-04 2024-01-05 湖南北顺源智能科技有限公司 High-power cascade power amplifier system for underwater acoustic communication and control method thereof
CN117353764B (en) * 2023-12-04 2024-02-23 湖南北顺源智能科技有限公司 High-power cascade power amplifier system for underwater acoustic communication and control method thereof

Similar Documents

Publication Publication Date Title
CN107834886B (en) A kind of single-stage boost inverter and its control method
CN112234808B (en) Double-frequency ripple suppression circuit and suppression method of single-phase inverter
CN116683750A (en) IPOS-DC/DC converter cascading single-phase inverter and method
Shi et al. A single-phase grid-connected PV converter with minimal DC-link capacitor and low-frequency ripple-free maximum power point tracking
CN103490653B (en) Grid-connected electric current and DC voltage secondary Ripple Suppression control system and control method
CN111355397B (en) Single-phase high-gain photovoltaic grid-connected inverter with continuous input current and control method
CN209030101U (en) A kind of energy back feed device based on Boost flyback booster circuit
CN103401267A (en) Grid-connected circuit for small-sized wind-driven generator
Tian et al. A single-phase transformerless common-ground type PV inverter with active power decoupling
US20220077818A1 (en) Photovoltaic module
CN115051565A (en) Bidirectional half-bridge direct-current converter grid-connected inverter and ripple wave control method
CN102545681B (en) Step wave synthesis three-phase inverter capable of eliminating low frequency harmonic waves and control method
JP2017511103A (en) Power conversion electronics
CN205249078U (en) Three inverter in Z source and air conditioning system
CN104124885A (en) H6 photovoltaic grid-connected inverter of interleaved parallelly-connected BOOST structure
CN117039976A (en) CLLC bidirectional resonant converter cascading grid-connected inverter and inhibition method thereof
CN103227474B (en) A kind of photovoltaic power generation grid-connecting system based on solid-state transformer
CN110071652B (en) Low-leakage-current five-switch non-isolated single-phase photovoltaic grid-connected inverter and grid-connected system
US20200169216A1 (en) Photovoltaic module
CN103326602A (en) Inverter
CN113726210B (en) Low-frequency ripple suppression circuit and method for direct-current bus of two-stage double-active-bridge grid-connected inverter
CN203278685U (en) Inverter
CN116317499A (en) Single-phase inverter based on flying capacitor type three-level boost and control method
CN113224964B (en) Control method of single-phase single-stage boost inverter
CN112117925B (en) DCM single-bridge-arm integrated split-source inverter control method for photovoltaic grid-connected occasions

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination