CN116599242A - Synchronous control method of bidirectional wireless charging system - Google Patents

Synchronous control method of bidirectional wireless charging system Download PDF

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Publication number
CN116599242A
CN116599242A CN202310430597.2A CN202310430597A CN116599242A CN 116599242 A CN116599242 A CN 116599242A CN 202310430597 A CN202310430597 A CN 202310430597A CN 116599242 A CN116599242 A CN 116599242A
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current
wireless charging
charging system
secondary side
output
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钟文兴
郑康兴
刘恒
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Zhejiang University ZJU
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Zhejiang University ZJU
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/10—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling
    • H02J50/12—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling of the resonant type
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
    • H02J50/80—Circuit arrangements or systems for wireless supply or distribution of electric power involving the exchange of data, concerning supply or distribution of electric power, between transmitting devices and receiving devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J2207/00—Indexing scheme relating to details of circuit arrangements for charging or depolarising batteries or for supplying loads from batteries
    • H02J2207/20—Charging or discharging characterised by the power electronics converter
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/70—Energy storage systems for electromobility, e.g. batteries
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/7072—Electromobility specific charging systems or methods for batteries, ultracapacitors, supercapacitors or double-layer capacitors

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Charge And Discharge Circuits For Batteries Or The Like (AREA)

Abstract

The invention discloses a synchronous control method of a bidirectional wireless charging system, which comprises the following steps: after the bidirectional wireless charging system is started, calculating the mutual inductance of the bidirectional wireless charging system, and calculating the reference bypass duty ratio of the secondary side rectifier in synchronization according to the reference value of the output voltage or current; when the output voltage or current rises to a reference value, starting the power ring and the synchronous ring, and initializing the output bypass duty ratio of the power ring; the power loop performs closed-loop control of the output bypass duty ratio; the synchronous ring performs closed-loop phase-shifting control of the synchronous signal, and further controls phase synchronization of the synchronous signal and the zero crossing point of the input current of the secondary side rectifier. The invention can realize zero-voltage turn-on of the secondary side rectifier Mos, does not need an alternating current sensor, reduces the system cost, improves the reliability, and solves the problem of phase disturbance caused by the frequency deviation of the primary secondary side controller crystal oscillator in the bidirectional wireless charging system.

Description

Synchronous control method of bidirectional wireless charging system
Technical Field
The invention relates to the technical field of wireless charging, in particular to a synchronous control method of a bidirectional wireless charging system.
Background
The wireless charging technology can transmit electric energy from a power supply to a load in a non-contact mode, and has the advantages of convenience, rapidness, safety and reliability. In the field of electric automobile charging, the bidirectional wireless charging technology can realize bidirectional energy allocation between a vehicle end and a power grid, can improve the utilization efficiency of energy and the power supply flexibility of the power grid, and has wide application prospect.
In the bidirectional wireless charging system, the primary side and the secondary side respectively adopt independent controllers to output control signals with the same frequency, but the phase difference of the primary side control signals and the secondary side control signals can generate periodic change in the control process due to the small deviation of the crystal oscillator frequencies of different controllers, so that the output power oscillates. Even if the primary and secondary controllers perform wireless communication, the existing wireless communication means such as bluetooth and wifi have ms-level communication delay, the control signal period of the wireless charging system is generally of mu s level, the real-time synchronization requirement of the control signal is difficult to meet, and the wireless communication signal is easily affected by electromagnetic interference, so that the safety and stability of the system are not good.
Therefore, in order to ensure the stability of the power transmission of the bidirectional wireless charging system, a proper phase synchronization method is required to control the phase difference of the primary and secondary side control signals. The prior art can be divided into two types, and the first type indirectly acquires the phase information of the primary side for synchronization by adding additional hardware equipment on the secondary side, for example, an auxiliary coil or an alternating current sensor is added for high-frequency current sampling, so that the phase difference of the primary side and the secondary side is controlled. The method has higher hardware precision requirement on the sampling part, and when the current harmonic content is higher, the phase information obtained by sampling can generate offset, and compensation is needed, so that the cost and design complexity of the system are increased. And secondly, no additional hardware circuit is needed, only the direct current information of the load is needed, and the phase synchronization is carried out by tracking the output current extremum or the phase angle extremum of the closed loop bypass of the active rectifier bridge through a disturbance method. The method is not influenced by drift and detuning of system parameters, but the system performance is influenced by disturbance period and disturbance quantity, when the frequency difference of the primary side controller and the secondary side controller is large, the defect of large output ripple exists, and the disturbance method is only suitable for symmetrical bypass control (current and voltage fundamental wave in phase of a rectifying circuit) of an active rectifying bridge, and for soft switch bypass control (current lead voltage fundamental wave of the rectifying circuit) capable of realizing Mos soft switch of the rectifying bridge, the extremum of the bypass phase angle of the output current or the rectifying circuit does not correspond to a synchronous point, and at the moment, the disturbance method cannot realize phase synchronization.
Disclosure of Invention
The invention aims to provide a synchronous control method of a bidirectional wireless charging system. The invention can realize zero-voltage turn-on and synchronous control of the secondary side rectifier, reduce the system cost and improve the reliability, and solve the problem of phase disturbance caused by the frequency deviation of the primary secondary side controller crystal oscillator in the bidirectional wireless charging system.
The technical scheme of the invention is as follows: a synchronous control method of a bidirectional wireless charging system comprises a primary side circuit and a secondary side circuit; the primary side circuit comprises an inverter circuit connected with a direct-current voltage source, and the output end of the inverter circuit is connected with a primary side series compensation capacitor and a transmitting coil; the secondary side circuit comprises a secondary side rectifier connected with a load, and the input end of the secondary side rectifier is connected with a secondary side series compensation capacitor and a receiving coil; the bidirectional wireless charging system realizes the control of the output voltage or current and the output power direction of the bidirectional wireless charging system by controlling the output bypass duty ratio of the secondary rectifier and the phase duty ratio of the synchronous signal, and the synchronous control method comprises the following steps:
step 1, a primary side circuit performs low-power input after starting up, a secondary side rectifier enters an uncontrolled rectification mode, mutual inductance of the bidirectional wireless charging system is calculated, and a reference bypass duty ratio of the secondary side rectifier during synchronization is calculated according to a reference value of output voltage or current;
step 2, when the output voltage or current rises to a reference value, starting a power ring and a synchronous ring, and giving an initial value to the duty ratio of an output bypass; the power loop performs closed-loop control of the duty ratio of the output bypass, and reduces the error between the output voltage or current and a reference value; the synchronous ring performs closed-loop phase-shifting control of the synchronous signal, reduces the error of the duty ratio of the output bypass and the duty ratio of the reference bypass, and further controls the phase synchronization of the synchronous signal and the zero crossing point of the input current of the secondary rectifier.
In the step 1, the mutual inductance calculation is that the primary circuit inputs the primary dc bus voltage U with low power at this time through low-speed non-real-time wireless communication in1 Transmitting the mutual inductance M to a secondary side circuit, and calculating the mutual inductance M by the secondary side circuit according to the following formula:
wherein: i O The current is the direct current bus current in the secondary circuit; ω is the angular frequency of the two-way wireless charging system, ω=2pi f, f is the operating frequency of the two-way wireless charging system.
In the synchronous control method of the bidirectional wireless charging system, the calculation of the reference bypass duty ratio of the secondary side rectifier includes the reference bypass duty ratio d when the secondary side rectifier performs constant current control ref The calculation formula is as follows:
wherein I is O_ref U is the reference value of the current of the direct current bus of the secondary circuit in1_rated The voltage of a primary circuit direct current bus under rated power is M is mutual inductance, omega is the angular frequency of a bidirectional wireless charging system, and d a_ref Is a reference value for the phase duty cycle of the synchronous signal of the secondary circuit.
In the synchronous control method of the bidirectional wireless charging system, the reference bypass duty ratio of the secondary side rectifierThe calculation includes referencing the bypass duty cycle d when the secondary side is performing constant voltage control ref The calculation formula is as follows:
wherein U is o_ref U is the reference value of the DC bus voltage of the secondary side circuit in1_rated The voltage of a primary circuit direct current bus under rated power is M is mutual inductance, omega is the angular frequency of a bidirectional wireless charging system, and d a_ref A reference value for the phase duty ratio of the synchronous signal of the secondary circuit; r is R L In order to be a load equivalent resistance,U o and I o The sampling values of the output direct current voltage and the output direct current are respectively.
In the synchronous control method of the bidirectional wireless charging system, the output bypass duty ratio initialization value is as follows:
setting a reference current threshold I o_ref1 When the reference current I o_ref Less than the reference current threshold I o_ref1 Initial value d initial For a reference bypass duty cycle d ref ；
When the reference current I o_ref Greater than a reference current threshold I o_ref1 Initial value d initial For bypassing the duty cycle threshold d 1 The initial phase duty cycle is calculated as follows:
in the synchronous control method of the bidirectional wireless charging system, the reference current threshold value I o_ref1 The solution is obtained by the following equation:
the synchronous control method of the bidirectional wireless charging system further comprises the step of reversely flowing output power, wherein the output bypass duty ratio of the power loop is kept unchanged, the synchronous loop controls the synchronous signal to shift 180 degrees step by step, and then the power loop is enabled to resume closed loop control, so that the reverse flowing of power is realized.
Compared with the prior art, the invention has the following beneficial effects:
1. the invention only needs to collect the secondary side direct current side output information of the bidirectional wireless charging system, does not need an alternating current sensor to collect the current high-frequency phase information, further does not need a complex high-frequency filter circuit, reduces the design complexity of the system and reduces the system cost.
2. The invention can realize the phase synchronization of the primary and secondary sides only by low-speed non-real-time wireless communication at the starting-up stage without real-time rapid wireless communication of the primary and secondary sides, solves the phase disturbance problem caused by the crystal oscillator frequency deviation of the primary and secondary side controller in the bidirectional wireless charging system, and improves the reliability of the system.
3. Compared with the synchronous control method of the WPT system adopting the direct current information synchronization by adopting the disturbance method, the synchronous control method of the invention adopts closed loop feedback control, has higher response speed and good dynamic performance, does not have the problem that the system performance of the disturbance method is influenced by disturbance period and disturbance quantity, and is designed for the soft switch bypass control (current leading voltage fundamental wave of the rectifier circuit) capable of realizing the soft switch of the rectifier bridge Mos, thereby reducing the switching loss of the rectifier bridge and being beneficial to improving the system efficiency.
Drawings
FIG. 1 is a basic circuit topology of a two-way wireless charging system of the present invention;
FIG. 2 shows the driving control waveforms and the Sync waveforms of the 4 Mos transistors on the secondary side;
FIG. 3 is the current i at stage a e Is a schematic diagram of the flow path of (a);
FIG. 4 is the current i at stage b e Is a schematic diagram of the flow path of (a);
FIG. 5 is the current i at stage c e Is a schematic diagram of the flow path of (a);
FIG. 6 is the current i at stage d e Is a schematic diagram of the flow path of (a);
FIG. 7 is the current i at stage e e Is a schematic diagram of the flow path of (a);
FIG. 8 is the current i at stage f e Is a schematic diagram of the flow path of (a);
FIG. 9 is a schematic diagram of a control method of the present invention;
FIG. 10 shows the output DC current I O Sync phase duty cycle d of synchronization signal a Is a graph of the relationship of (2);
FIG. 11 is a reference bypass duty cycle d ref Sync phase duty cycle d of synchronization signal a Is a relationship of (2);
FIG. 12 shows the output DC current I O Output bypass duty cycle d and Sync phase duty cycle d a Is a relationship diagram of (1);
FIG. 13 is a phase duty cycle d of the Sync signal a When the output power is smaller than 0, the direct current I of the output power loop O A variation relationship curve according to the duty ratio d of the output bypass;
FIG. 14 is a phase duty cycle d of the Sync signal a When the output power is larger than 0, the direct current I of the output power loop O A variation relationship curve according to the duty ratio d of the output bypass;
FIG. 15 is a reference bypass duty cycle d ref Sync phase duty cycle d of synchronization signal a Is a relationship of (2);
fig. 16 is an initial phase duty |d a_initial I, reference bypass duty cycle d ref With reference current I O_ref Is a relationship of (2);
FIG. 17 is a 3A constant current control simulation waveform of a resistive load;
FIG. 18 is a waveform of a resistive load constant voltage 60V control simulation;
fig. 19 is a 3A constant current control simulation waveform of the battery load.
Fig. 20 is an experimental waveform of a battery load constant current of 3.5A to a constant voltage of 52.9V;
fig. 21 is an experimental waveform of battery load constant currents 3.5A and 2.5A.
Detailed Description
The invention is further described in connection with the accompanying drawings and examples which are not to be construed as limiting the invention, but are intended to cover the full scope of the claims and will become more fully apparent to those of ordinary skill in the art from the following examples.
Examples: a synchronous control method of a bidirectional wireless charging system, wherein the basic circuit topology of the bidirectional wireless charging system is shown in figure 1, and the bidirectional wireless charging system comprises a primary side circuit and a secondary side circuit; the primary side circuit comprises an inverter circuit connected with a direct-current voltage source, and the output end of the inverter circuit is connected with a primary side series compensation capacitor and a transmitting coil; the secondary side circuit comprises a secondary side rectifier connected with a load, and the input end of the secondary side rectifier is connected with a secondary side series compensation capacitor and a receiving coil;
in this embodiment, the inverter circuit is formed by an H-bridge composed of four Mos tubes S1, S2, S3, S4, and is configured to convert dc power into ac power. The transmitting coil is used for transmitting the electric energy transmitted by the inverter circuit through a magnetic field, and the receiving coil is used for receiving the energy transmitted by the transmitting coil.
The primary side is connected in series with a compensation capacitor C p The configuration is according to the formula:
wherein L is p Is self-inductance of the transmitting coil, ω is the angular frequency of the bidirectional wireless charging system, ω=2pi f, f is the operating frequency of the bidirectional wireless charging system.
The secondary side is connected in series with the compensation capacitor C S Is configured according to the formula:
wherein L is s Is the self-inductance of the receiving coil.
The bidirectional wireless charging system configured according to the above formula has constant current source output characteristic, rectifier input current I e Is not affected by load variation, and the formula is as follows.
Wherein I is e Input current amplitude for secondary rectifier, U in The voltage is input to the primary side direct current bus (namely, the voltage of the direct current voltage source).
In this embodiment, the primary dc bus input voltage, the secondary dc bus output current and the voltage are collected by a dc current-voltage sensor.
The rectification circuit consists of 4 Mos tubes S5, S6, S7 and S8 and is used for converting alternating current into direct current. The driving control waveforms of the 4 Mos tubes of the rectifying circuit and the waveform of the synchronous signal Sync are shown in fig. 2, wherein beta is the phase angle of the secondary side rectifying voltage 0 level,for the phase difference between the secondary side synchronous signal Sync and the zero crossing point of the input current of the rectifier, when the synchronous signal Sync lags behind the zero crossing point of the input current of the secondary side rectifier +.>For convenience of the following description, d is defined as the secondary rectifier bypass duty cycle,definition d a For the phase duty cycle of the secondary synchronization signal Sync and the zero crossing of the rectifier input current,/->
The Mos tubes S5, S6, S7, S8 and the synchronous signal Sync are digital square waves with the duty ratio of 0.5 in the secondary side controller, and the frequency is preset as the system working frequency. The control logic is as follows: the rising edge of the driving control waveform of S5 is triggered and synchronized by the rising edge of the synchronizing signal Sync, the pulse widths of S5 and S7 are (pi/2+beta)/(2pi f), the phase angle of S7 lagging S5 is fixed to pi/2, the driving control waveform of S6 is complementary to S5, and the driving control waveform of S7 is complementary to S8.
According to fig. 2, the secondary control waveform can be specifically divided into 6 phases:
stage a: input current i at rectifier e In the positive half period of (1), due to the delay of the synchronization signal Sync, the current i e Zero crossing point, S6 and S7 are conducted, and current i e As shown in fig. 3, the system instantaneous power flows from the dc side back to the ac side.
Stage b: s6 is turned off, S5 is turned on, at the moment, S5 and S7 are turned on, the secondary side is in a bypass state, the bypass duty ratio d of the secondary side rectifier is determined by a secondary side controller, and the current i e As shown in fig. 4, the system instantaneous power is not transferred to the dc side.
Stage c: after the bypass is finished, S7 is turned off, S8 is turned on, at the moment, S5 and S8 are turned on, and current i e As shown in fig. 5, the system instantaneous power flows forward from the ac side to the dc side.
Stage d: input current i at rectifier e Due to the lag of the synchronization signal Sync, the current i e Zero crossing point, at this time, S5 and S8 are still conducted, and current i e As shown in fig. 6, the system instantaneous power flows from the direct current side to the alternating current side.
Stage e: s5 is turned off, S6 is turned on, at the moment, S6 and S8 are turned on, the secondary side is in a bypass state, the bypass duty ratio d of the secondary side rectifier is determined by a secondary side controller, and the current i e As shown in fig. 7, the system instantaneous power is not transferred to the dc side.
Stage f: after the bypass is finished, S8 is turned off, S7 is turned on, at the moment, S6 and S7 are turned on, and current i e As shown in fig. 8, the system instantaneous power flows forward from the ac side to the dc side.
From FIG. 2, the secondary DC output current I in steady state can be deduced O With the output bypass duty cycle d of the secondary rectifier and the Sync phase duty cycle d of the synchronous signal Sync a The relationship between the two is shown in the following formula:
it can be seen that the bidirectional wireless charging system of the invention is realized by the auxiliary pairThe output bypass duty cycle d of the side rectifier and the phase duty cycle d of the synchronization signal a To realize the control of the output voltage or current and the output power direction of the bidirectional wireless charging system and due to the current i e Zero-crossing point advanced synchronous signal Sync can realize zero-voltage turn-on of 4 Mos tubes on the secondary side, and reduces switching loss.
Based on the above two-way wireless charging system, as shown in fig. 9, the synchronization control method in this embodiment includes the following steps:
step 1, a primary side circuit performs low-power input after starting up, a secondary side rectifier enters an uncontrolled rectification mode, mutual inductance of the bidirectional wireless charging system is calculated, and a reference bypass duty ratio of the secondary side rectifier during synchronization is calculated according to a reference value of output voltage or current;
the mutual inductance calculation is that the primary circuit inputs the primary DC bus voltage U with low power at the moment through low-speed non-real-time wireless communication in1 Primary side DC bus voltage U of rated power input in_rated Transmitting the mutual inductance M to a secondary side circuit, and calculating the mutual inductance M by the secondary side circuit according to the following formula:
wherein: i o The current is the direct current bus current in the secondary circuit; ω is the angular frequency of the two-way wireless charging system, ω=2pi f, f is the operating frequency of the two-way wireless charging system. In other embodiments, the present invention may also employ other commonly used mutual inductance identification methods.
After the mutual inductance is calculated, the reference bypass duty ratio calculation of the secondary side rectifier comprises the reference bypass duty ratio d when the secondary side rectifier performs constant current control ref The calculation formula is as follows:
wherein I is o_ref U is the reference value of the current of the direct current bus of the secondary circuit in1_rated The voltage of a primary circuit direct current bus under rated power is M is mutual inductance, omega is the angular frequency of a bidirectional wireless charging system, and d a_ref Is a reference value for the phase duty cycle of the synchronous signal of the secondary circuit.
And when the secondary side performs constant voltage control, the reference bypass duty ratio d ref The calculation formula is as follows:
wherein U is o_ref U is the reference value of the DC bus voltage of the secondary side circuit in1_rated The voltage of a primary circuit direct current bus under rated power is M is mutual inductance, omega is the angular frequency of a bidirectional wireless charging system, and d a_ref A reference value for the phase duty ratio of the synchronous signal of the secondary circuit; r is R L In order to be a load equivalent resistance,U o and I o The sampling values of the output direct current voltage and the output direct current are respectively.
Step 2, the primary side carries out rated direct current bus voltage input, when the output voltage or current rises to a reference value, a power ring and a synchronous ring are opened, and an initial value d is given to the output bypass duty ratio d initial The method comprises the steps of carrying out a first treatment on the surface of the The power loop performs closed-loop control of the duty ratio of the output bypass, and reduces the error between the output voltage or current and a reference value; the synchronous ring performs closed-loop phase-shifting control of the synchronous signal Sync, and reduces the duty ratio d of the output bypass and the duty ratio d of the reference bypass ref Thereby controlling the phase synchronization of the synchronous signal and the zero crossing point of the input current of the secondary side rectifier, namely controlling d a Is a proper constant value d a_ref 。
When the power reverse flow is needed, the output bypass duty ratio of the power loop is kept unchanged, the synchronous loop controls the synchronous signal to shift 180 degrees step by step, and then the power loop is enabled to resume closed loop control, so that the power reverse flow is realized.
In this step, the output bypass duty cycle initialization is performed as follows:
setting a reference current threshold I o_ref1 The reference current threshold I o_ref1 The solution is obtained by the following equation:
for simplicity of calculation, the present embodiment uses an approximate numerical solution formula as a simplification of the above formula, the approximate numerical solution formula being as follows:
in this case, d a_ref ＝0.1；
When the reference current I o_ref Less than the reference current threshold I o_ref1 Initial value d initial For a reference bypass duty cycle d ref ；
When the reference current I o_ref Greater than a reference current threshold I o_ref1 Initial value d initial For bypassing the duty cycle threshold d 1 ，d 1 Is calculated as follows:
therefore, the invention realizes the control of the output voltage/current, the output power direction and the synchronous state of the bidirectional wireless charging system through the closed-loop control of the power loop and the synchronous loop, and is applicable to the load which is a resistor or a battery.
In order to further explain the beneficial effects of the invention, the embodiment is used for explaining the power disturbance caused by the frequency difference of the primary side controller and the secondary side controller of the bidirectional wireless charging system, and is not suitable for the synchronous control of the soft switch bypass in the case of an extremum disturbance method in the background art.
First, in the bidirectional wireless charging system, a scheme of independent controllers is adopted for a primary side and a secondary side. Wherein the control signal period of the primary side controller determines the input current i of the secondary side rectifier e The period of the secondary synchronization signal Sync is determined by the period of the control signal of the secondary controller.The control signal period of the primary and secondary side controllers also has a small period difference delta T due to the small deviation of the crystal oscillator frequencies of different controllers. The period difference DeltaT will result in i e Phase duty cycle d with synchronization signal Sync a Generating periodic variation, d is known from the formula a Will cause the output DC current I to be generated by periodic variation of (a) O As shown in fig. 10.
Taking the primary and secondary side controller crystal oscillator accuracy as 50ppm as an example (1 ppm=10) -6 ) When the working frequency of the system is set to be 85kHz and the primary frequency and the secondary frequency are selected to be 72MHz, the control signal period difference delta T of the primary side and the secondary side is 0.59ns, and the method is as follows:
can calculate the fluctuation period T of the system output current power 0.24s. Therefore, a proper synchronization control method is required to maintain d a Constant.
The description of the extremum disturbance method in the background art is not applicable to soft switch bypass synchronous control. As can be seen from fig. 2, the secondary side control method of the present invention is soft switch bypass control, and requires Sync lag current i e ZVS (zero voltage on), i.e. d, of 4 Mos on the secondary side can be achieved a It is required to be 0 or more. If the current extremum disturbance method is used for synchronization, it can be seen from FIG. 10 that when the power is transmitted in the forward direction, the output bypass duty cycle d is greater than 0, the output current I o D corresponding to maximum value a When the current is smaller than 0, ZVS of 4 Mos on the secondary side cannot be realized, namely the synchronous method of the traditional current extremum disturbance method is not suitable for soft switch bypass control.
When the bypass duty ratio extremum disturbance method is adopted for synchronization, the reference current I is adopted O_ref Taking 3A as an example, a reference bypass duty cycle d can be obtained from the formula ref Sync phase duty cycle d of synchronization signal a As shown in fig. 11.
From fig. 11 it can be seen that the reference bypass duty cycle d ref The extreme point of (2) also corresponds to d a Intervals of less than 0, and d a The same reference current I also appears in the interval smaller than 0 o_ref Corresponding to two bypass duty cycles d ref The bypass duty cycle extremum perturbation method is also not applicable to the synchronization of soft switch bypass control in the present invention.
Therefore, aiming at soft switch bypass control, the invention provides a synchronous control method for closed-loop feedback control according to the reference value of the bypass duty ratio of the secondary side rectifier during synchronization. The power ring and the synchronizing ring in the scheme of the invention are described as follows:
according to the formula hereinbeforeThe obtained secondary side direct current side output current I under the steady state o With the output bypass duty cycle d of the secondary rectifier and the Sync phase duty cycle d of the synchronous signal Sync a The relationship between them is shown in fig. 12.
Taking forward charging into consideration, the Sync phase duty ratio d of the synchronous signal a The direct current I of the output power loop can be obtained unchanged o The relationship of the change with the bypass duty cycle d is shown in fig. 13 and 14.
As can be seen from FIG. 13, when d a Less than 0, i.e. synchronous signal Sync leading rectification circuit i e When I o The dependence of the change with the bypass duty cycle d is non-monotonic. Wherein, the liquid crystal display device comprises a liquid crystal display device,is determined by the following formula:
order theD corresponding to the maximum value of the output current can be obtained, and the calculation formula is as follows:
d 1 ＝|d a |；
suppose when d is equal to 0, I o Equal to I o_ref D is then 1 Can furtherExpressed as:
if d is more than 0 and less than d 1 Then output DC current I o Increases with the increase of the duty ratio d of the output bypass, has monotonicity, and the control monotonicity at the moment is thatIf d 1 D is less than 1, then output DC current I O Decreasing with increasing output bypass duty ratio d, and controlling monotonicity to be +.>Visible direct current I o The overall control logic with the output bypass duty cycle d is non-monotonic.
As can be seen from FIG. 14, when d a Greater than 0, i.e. synchronous signal Sync hysteretic rectification circuit i e At the time, output DC current I o Decreasing with increasing of the duty ratio d of the output bypass, and controlling monotonicity at the momentThe overall control logic is monotonic.
D when the rectifier Mos fully realizes soft switching and needs to be stable a 0 or more, so that the control monotonicity of the monotonicity power ring is selected asI.e. output DC current I o The control logic as a power loop decreases with increasing output bypass duty cycle d. Consider d a When the voltage is too large, the reactive component of the rectifier impedance increases, resulting in reduced ACAC efficiency, thereby setting the synchronous signal Sync phase reference duty ratio d a_ref 0.1. The steady-state operating point of the power loop is shown as point C in FIG. 14, and the output DC current I corresponding to point A, B, C in FIG. 13 and FIG. 14 o Output direct current I corresponding to 3A and D points o Is d a Maximum current at = -0.22.
For synchronous rings, with reference current I o_ref For the example of 3A, define d a_initial To keep the duty ratio d of the output bypass at 0 when starting up, when outputting the DC current I o Rising to reference current I o_ref Reference bypass duty cycle d ref Sync phase duty cycle d of synchronization signal a The relationship of (2) is shown in FIG. 15, monotonicityIs determined by the following formula
Order theA reference bypass duty cycle d equal to 0 can be obtained ref Phase duty cycle d corresponding to maximum value a The method comprises the following steps:
from the above formula, d a1 And d 1 The absolute values of (a) are the same.
From fig. 15 it can be seen that the reference bypass duty cycle d ref Duty cycle d with phase a There are also two monotonicities, and when d a ＜d a1 When d ref There are two solutions. Due to the synchronous signal Sync phase reference duty cycle d a_ref Is 0.1, thus selecting control monotonicityI.e. reference bypass duty cycle d ref Duty cycle d with phase a Is reduced as a function of the control logic of the synchronizer ring. Since the synchronizer ring bypasses the duty cycle d according to the output of the power ring a The closed-loop phase-shifting control is carried out, so that the speed of the synchronous ring is required to be slower than that of the power ring, the synchronous ring is set to carry out phase-shifting control once every N control periods of the power ring, and the control period of the power ring and the system resonance period are setThe same applies. Fig. 13 and 14 satisfy the same conditions as A, B, C in fig. 15.
Still further, in the scheme of the invention, the power loop outputs the bypass duty cycle initial value d initial The description of (2) is as follows:
as can be seen from fig. 13, 14 and 15, when d a At < 0, the control monotonicity of the synchronization loop and the power loop may be opposite to design, which may result in the output of the controller failing to converge. Therefore, it is necessary to design appropriate control variable initial values and controller parameters to ensure that the control monotonicity of the power loop and the synchronization loop always conforms to the design.
As can be seen from fig. 10, the dc current I is output at the start-up o Rising to reference current I o_ref When d a May be greater than 0 or may be less than 0. When the primary control period is greater than the secondary control period, the secondary synchronous signal Sync continues to lead the rectifying circuit i e ，d a Will continue to decrease; output DC current I o Rising to reference current I o_ref When d a Greater than 0, the control monotonicity of the power ring and the synchronous ring are respectively as followsAnd->The design is met. When the primary control period is smaller than the secondary control period, the secondary synchronous signal Sync continuously lags behind the rectifying circuit i e ，d a Will continue to increase when outputting DC current I o Rising to reference current I o_ref When d a Is smaller than 0, and in the synchronous ring, the working point moves rightwards from the point A, and the control monotonicity is +.>The design is met. But the control monotonicity of the power loop is +.>It is therefore necessary to design the output bypass duty cycle d with a suitable initial value to ensure that its control monotonicity is +.>As can be seen from FIG. 13, when the DC current I is outputted o Rising to reference current I o_ref After that, when the initial value of the output bypass duty ratio d satisfies d initial ≥d 1 When the operating point of the power loop jumps from the point A to the right side of the point D, the control monotonicity is +.>
Bypass duty cycle threshold d 1 Reference bypass duty cycle d ref With reference current I o_ref The relationship of (2) is shown in FIG. 16. To ensure that the output of the system follows the reference value quickly, an initial value d initial As close as possible to the reference bypass duty cycle d ref . Thus, the selection principle of the initial value can be described as:
in this embodiment, the reference current threshold I of the system o_ref1 Calculated 2.72A.
When the reference current I is o_ref Greater than a reference current threshold I o_ref1 If the initial value d is selected initial For a reference bypass duty cycle d ref The system can be kept stable, but the overshoot of the starting current is larger.
In order to verify the effect of the invention, a simulation platform is built in placs by taking a two-way wireless power transmission system as an example, and system parameters are shown in table 1.
Parameter name Parameter value Parameter name Parameter value
M 72.17μH Cp 6.83nF
Lp 514.9μH Cs 42.92nF
Ls 81.72μH Co 95nF
R Lp 0.98Ω R Ls 0.11Ω
TABLE 1
Setting primary side DC bus input voltage U in 380V (S3 of the primary side inverter is kept off, S4 is kept on, and the primary side inverter works in a half-bridge mode, and then the actual equivalent input DC voltage U in 190V), when the load is a resistor, the load resistor is set to 17.8Ω, when the load is a battery, the battery is equivalent to a constant voltage source string resistor, the constant voltage source is set to 50.84V, and the resistor is set to 0.1Ω. Wherein the reference current is set to I o_ref Is 3A, case1 is the initial value d initial For a reference bypass duty cycle d ref Case2 is the initial value d initial For bypassing the duty cycle threshold d 1 . The 3A constant current control simulation waveform of the resistance load is shown as 17, and the constant voltage is controlled at 60VThe simulated waveform is shown at 18 and the 3A constant current control simulated waveform of the battery load is shown at 19. Fig. 20 and 21 are experimental waveforms of a load as a battery. In fig. 21, the output after the system is started is controlled to be a constant current output of 3.5A, and when the battery voltage rises to 52.9V, the system is converted to a constant voltage output of 52.9V. In fig. 21, the output current reference value of the system is switched from 3.5A to 2.5A, and the output is still stable.
The control of the output voltage/current, the output power direction and the synchronous state of the bidirectional wireless charging system is realized through the closed-loop control of the power loop and the synchronous loop, and the bidirectional wireless charging system is applicable to a load which is a resistor or a battery.
In summary, the invention only needs to collect the secondary side direct current side output information of the bidirectional wireless charging system, does not need to collect the current high-frequency phase information by the alternating current sensor and the real-time wireless communication of the primary and secondary sides, can realize zero voltage switching on and synchronous control of the secondary side rectifier Mos element, reduces the system cost and improves the reliability, and solves the phase disturbance problem caused by the primary and secondary side controller crystal oscillator frequency deviation in the bidirectional wireless charging system.

Claims (7)

1. A synchronous control method of a bidirectional wireless charging system comprises a primary side circuit and a secondary side circuit; the primary side circuit comprises an inverter circuit connected with a direct-current voltage source, and the output end of the inverter circuit is connected with a primary side series compensation capacitor and a transmitting coil; the secondary side circuit comprises a secondary side rectifier connected with a load, and the input end of the secondary side rectifier is connected with a secondary side series compensation capacitor and a receiving coil; the bidirectional wireless charging system realizes the control of the output voltage or current and the output power direction of the bidirectional wireless charging system by controlling the output bypass duty ratio of the secondary rectifier and the phase duty ratio of the synchronous signal, and is characterized in that: the synchronous control method comprises the following steps:
step 1, a primary side circuit performs low-power input after starting up, a secondary side rectifier enters an uncontrolled rectification mode, mutual inductance of the bidirectional wireless charging system is calculated, and a reference bypass duty ratio of the secondary side rectifier during synchronization is calculated according to a reference value of output voltage or current;
step 2, when the output voltage or current rises to a reference value, starting a power ring and a synchronous ring, and giving an initial value to the duty ratio of an output bypass; the power loop performs closed-loop control of the duty ratio of the output bypass, and reduces the error between the output voltage or current and a reference value; the synchronous ring performs closed-loop phase-shifting control of the synchronous signal, reduces the error of the duty ratio of the output bypass and the duty ratio of the reference bypass, and further controls the phase synchronization of the synchronous signal and the zero crossing point of the input current of the secondary rectifier.
2. The synchronization control method of a bidirectional wireless charging system according to claim 1, wherein: in step 1, the mutual inductance calculation is that the primary side circuit inputs the primary side DC bus voltage U with low power at the moment through low-speed non-real-time wireless communication in1 Transmitting the mutual inductance M to a secondary side circuit, and calculating the mutual inductance M by the secondary side circuit according to the following formula:
wherein: i o The current is the direct current bus current in the secondary circuit; ω is the angular frequency of the two-way wireless charging system, ω=2pi f, f is the operating frequency of the two-way wireless charging system.
3. The synchronization control method of a bidirectional wireless charging system according to claim 2, wherein: the reference bypass duty cycle calculation of the secondary side rectifier comprises the reference bypass duty cycle d when the secondary side rectifier performs constant current control ref The calculation formula is as follows:
wherein I is o_ref U is the reference value of the current of the direct current bus of the secondary circuit in1_rated Is rated as workPrimary side circuit direct current bus voltage under the rate, M is mutual inductance, omega is angular frequency of a bidirectional wireless charging system, and d a_ref Is a reference value for the phase duty cycle of the synchronous signal of the secondary circuit.
4. The synchronization control method of a bidirectional wireless charging system according to claim 2, wherein: the reference bypass duty cycle calculation of the secondary side rectifier comprises the reference bypass duty cycle d when the secondary side performs constant voltage control ref The calculation formula is as follows:
wherein U is o_ref U is the reference value of the DC bus voltage of the secondary side circuit in1_rated The voltage of a primary circuit direct current bus under rated power is M is mutual inductance, omega is the angular frequency of a bidirectional wireless charging system, and d a_ref A reference value for the phase duty ratio of the synchronous signal of the secondary circuit; r is R L In order to be a load equivalent resistance,U o and I o The sampling values of the output direct current voltage and the output direct current are respectively.
5. The synchronous control method of a bidirectional wireless charging system according to claim 3 or 4, wherein: the output bypass duty cycle initialization is as follows:
setting a reference current threshold I o_ref1 When the reference current I o_ref Less than the reference current threshold I o_ref1 Initial value d initial For a reference bypass duty cycle d ref ；
When the reference current I o_ref Greater than a reference current threshold I o_ref1 Initial value d initial For bypassing the duty cycle threshold d 1 The initial phase duty cycle is calculated as follows:
6. the synchronous control method of a bidirectional wireless charging system according to claim 5, wherein: the reference current threshold I o_ref1 The solution is obtained by the following equation:
7. the synchronization control method of a bidirectional wireless charging system according to claim 1, wherein: the synchronous control method further comprises the reverse flow of output power, at the moment, the output bypass duty ratio of the power loop is kept unchanged, the synchronous loop is controlled to gradually shift the synchronous signal by 180 degrees, and then the power loop is restored to be in closed loop control, so that the reverse flow of power is realized.
CN202310430597.2A 2023-04-21 2023-04-21 Synchronous control method of bidirectional wireless charging system Pending CN116599242A (en)

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