WO2019076365A1 - Power conversion system and applications thereof - Google Patents

Power conversion system and applications thereof Download PDF

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Publication number
WO2019076365A1
WO2019076365A1 PCT/CN2018/110996 CN2018110996W WO2019076365A1 WO 2019076365 A1 WO2019076365 A1 WO 2019076365A1 CN 2018110996 W CN2018110996 W CN 2018110996W WO 2019076365 A1 WO2019076365 A1 WO 2019076365A1
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Prior art keywords
voltage
resonant
capacitor
controller
current
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PCT/CN2018/110996
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French (fr)
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Jianlong TIAN
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Tian Jianlong
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Publication of WO2019076365A1 publication Critical patent/WO2019076365A1/en

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • H02M7/2195—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration the switches being synchronously commutated at the same frequency of the AC input voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4815—Resonant converters
    • H02M7/4818—Resonant converters with means for adaptation of resonance frequency, e.g. by modification of capacitance or inductance of resonance circuits
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02P—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN THE PRODUCTION OR PROCESSING OF GOODS
    • Y02P80/00—Climate change mitigation technologies for sector-wide applications
    • Y02P80/10—Efficient use of energy, e.g. using compressed air or pressurized fluid as energy carrier

Definitions

  • This invention relates generally to power electronic systems. Specifically, it’s a series of techniques to converter AC mains to a required DC voltage and their applications.
  • the techniques include a power factor correction (PFC) technique, a technique to dynamically detect the system resonant frequency by timed sampling, a technique to dynamically adjust or stabilize the output voltage or power of the system by tuning/detuning and a synchronous rectification technique.
  • PFC power factor correction
  • the applications are a new type electric motor driven by resonant converters and a new type of resonant switch mode power supply. These are only two typical examples of the extensive applications of the techniques.
  • any power electronic systems wherever there is an electromagnetic oscillation there is a potential for the use of these techniques, such as wireless power transfer (WPT) systems, switch mode power supplies, DC-DC converters, electric motors, electromagnetic induction heating systems, high voltage direct current transmission (HVDC) , some kind of instruments, etc.
  • WPT wireless power transfer
  • HVDC high voltage direct current transmission
  • the system driving frequency and innate resonant frequency can be made always equal to each other, making the system always work on square wave driving, soft-switching and resonant conditions, which will increase the system efficiency and lower the Electro Magnetic Interference (EMI) greatly.
  • EMI Electro Magnetic Interference
  • the innate resonant frequency of a WPT or any other electromagnetic oscillation systems in power electronic fields is a very important factor, which influences almost every important aspect of the system such as resonance, soft-switching, power transfer ability, efficiency, EMI, etc.
  • the innate resonant frequency of the system needs to be known first so that the system can be driven with this frequency to realize soft-switching and resonance, which makes the power transfer ability and efficiency of the system optimized.
  • the innate resonant frequency of a WPT system is not constant but changes with many factors such as the coupling coefficient between the primary and secondary side, the variation of the load and many other parameters of the system.
  • This invention proposes the following four techniques, i.e. a power factor correction (PFC) technique, a technique to dynamically detect the system resonant frequency by timed sampling, a technique to dynamically adjust or stabilize the output voltage or power of the system through the effect of tuning/detuning of a soft-switching capacitor, a synchronous rectification technique, and two of the applications of these techniques, i.e. a new type electric motor driven by resonant converters and a new type of resonant switch mode power supply.
  • PFC power factor correction
  • Fig. 1 shows a block diagram of the overall structure of the power conversion system proposed in this invention.
  • Fig. 2 shows a circuit diagram of the power factor correction technique proposed in this invention.
  • Fig. 3 shows the waveforms of key signals of the power factor correction technique proposed in this invention.
  • Fig. 4 shows a circuit diagram of detecting the innate system resonant frequency by timed sampling in a variable frequency system.
  • Fig. 5 shows a circuit diagram of detecting the innate system resonant frequency by timed sampling in a fixed frequency system.
  • Fig. 6 shows a circuit diagram of the technique to stabilize the output voltage or power of electromagnetic oscillation systems in the way of parallel tuning.
  • Fig. 7 shows a circuit diagram of the technique to stabilize the output voltage or power of electromagnetic oscillation systems in the way of serial tuning.
  • Fig. 8 shows a circuit diagram of the synchronous rectification technique proposed in this invention.
  • Fig. 9 shows the waveforms of key signals of the synchronous rectification technique.
  • Fig. 10 shows a circuit diagram of a new type electric motor driven by resonant converters.
  • Fig. 11 shows a circuit diagram of the new type resonant switch mode power supplies.
  • Fig. 12 shows a circuit diagram of the new type resonant switch mode power supplies when the tuning capacitor is shorted.
  • This part includes the following six sections:
  • Fig. 1 is the representative graph of this invention which shows a power conversion system to converter the AC main into a required DC voltage, which is divided into three parts, i.e. the power factor correction (PFC) part 2, the DC-AC conversion part 3 and the AC-DC conversion part 4.
  • the input of this power conversion system is an AC main 1 and the output a DC voltage 5.
  • This section introduces the first part of the power conversion system, i.e. the PFC part 2.
  • the circuit structure of this part is shown in Fig. 2, where v s 6 represents the voltage of the AC main, i s 7 the current of the AC main.
  • C f 10 is the filter capacitance for the voltage across the load of the system R load 11.
  • the basic operating principle is to control the two switches S1 and S2 turn on and off alternately at the zero voltage crossing points of v s 6. This guarantees the soft-switching of the two switches S1 and S2.
  • S1 is turned on and S2 turned off, when one branch of the current starts from the positive terminal of v s , through L1, L2, and back to the negative terminal of v s while another branch of the current starts from the positive terminal of v s , through L1, C f , R load , S1 and back to the negative terminal of v s .
  • S2 is turned on and S1 turned off, when one branch of the current starts from the negative terminal of v s , through L2, L1 and back to the positive terminal of v s while another branch of the current starts from the negative terminal of v s , through L2, C f , R load , S2 and back to the positive terminal of v s .
  • Fig. 3 shows the waveforms of the gate driving signal Gate_S1 for the switch S1, the voltage v s and current i s of the AC main input, from which it can be seen that v s and i s is in phase, meaning that the power factor is one.
  • the two switches S1 and S2 are turned on and off at the zero voltage crossing points of v s which means that S1 and S2 are soft-switching. So the proposed method can realize both of the two goals, i.e. making the power factor of the system to be one and realizing soft-switching for the switches of the system, which is very difficult for conventional power factor correction methods.
  • Section 2.1 the basic operating principle of the method to detect the innate system resonant frequency by timed sampling is described in detail in Section 2.1 with an example when this method is applied in a variable frequency system. Secondly, the situation when this method is applied in a fixed frequency system is described in Section 2.2.
  • the circuit topology of the method is shown in Fig. 4, where the part in the dashed block, i.e. the controller 18 is the essence of the method. It includes a timed sampling module 16, a voltage conversion module 17, a PI (Proportional-Integral) controller 19 and a voltage controlled oscillator (VCO) 20.
  • the outside part of the dashed block is mainly a switch mode DC-AC converter 14 with a resonant tank 15.
  • the gate driving signal 21 of the DC-AC converter 14 is generated by the VCO 20, which can be regarded as the actuator of the controller 18.
  • the most important part of the controller 18 is the timed sampling module 16, which samples the resonant voltage or current of the resonant tank 15 at exactly the moment when the switches in the DC-AC converter 14 are turned on or off.
  • this sampled voltage or current is zero, it is zero voltage switching (ZVS) or zero current switching (ZCS) and the innate system resonant frequency equals to the gate driving frequency 21.
  • ZVS zero voltage switching
  • ZCS zero current switching
  • the gate driving frequency 21 does not equal to the innate system resonant frequency.
  • the function of the voltage conversion module 17 is to generate a voltage signal according to the sign and value of the sampled voltage or current.
  • the voltage generate by 17 equals to the reference voltage of the PI controller 19, when the output voltage of the PI controller 19 and the output frequency of the VCO 20 remains constant.
  • the absolute value of the output voltage of the voltage conversion module 17 is proportional to the value of the sampled voltage or current, when the output voltage of 17 does not equal to the reference voltage of the PI controller 19 which will lead to the output voltage of the PI controller 19 and the output frequency of the VCO 20 changing continuously until the driving frequency 21 of the system equals to the innate system resonant frequency, when the system regains its resonant and soft-switching condition and the value of the sampled voltage or current becomes zero again.
  • the function of the PI controller 19 is to generate a control voltage for the VCO 20 to control its output frequency according to the output voltage from the voltage conversion module 17.
  • the general operating principle is that when the output voltage from the voltage conversion module 17 equals to the reference voltage of the PI controller 19, the output voltage of the PI controller 19 as well as the output frequency 21 of the VCO 20 remains constant, which means that the driving frequency of the system equals to the innate system resonant frequency and soft-switching is achieved at the moment.
  • the output voltage from the voltage conversion module 17 does not equal to the reference voltage of the PI controller 19
  • the output voltage of the PI controller 19 and therefore the output frequency 21 of the VCO 20 will vary continuously until the driving frequency 21 of the system equals to the innate system resonant frequency and soft-switching is achieved.
  • the situation when the value of the sampled voltage or current is zero is roughly the same, i.e. in this situation the output voltage of the voltage conversion module 17 equals to the reference voltage of the PI controller 19; the output voltage of the PI controller 19 and the output frequency 21 of the VCO 20 remains constant; the driving frequency of the system equals to the innate system resonant frequency meaning the system is working on resonant and soft-switching condition at the moment.
  • the detailed operation may be a little different according to the type of the DC-AC converter used and the type of sampling, i.e. what kind of signal (voltage or current) is sampled and at which moment (the rising or falling edge of the system gate driving signal) , etc.
  • one situation may be that the value of the sampled voltage or current being positive means that the driving frequency of the system is higher than the innate system resonant frequency; the output voltage of the voltage conversion module 17 is larger than the reference voltage of the PI controller 19; the output voltage of the PI controller 19 and the output frequency 21 of the VCO 20 will decrease continuously until the driving frequency 21 of the system equals to the innate resonant frequency of the system.
  • the value of the sampled voltage or current is negative, everything will be the inverse, i.e.
  • the driving frequency of the system is lower than the innate system resonant frequency; the output voltage of the voltage conversion module 17 is smaller than the reference voltage of the PI controller 19; the output voltage of the PI controller 19 and the output frequency 21 of the VCO 20 will increase continuously until the driving frequency 21 of the system equals to the innate resonant frequency of the system.
  • Fig. 5 shows the situation when the above method is applied to realize a fixed frequency system.
  • the main difference between the fixed frequency system of this section and the variable frequency system in the last section is the “actuator” of the control loop.
  • the actuator is a VCO 20, while for a fixed frequency system as described in this section, the actuator is a voltage controlled variable capacitor (VCVC) 24 or 26 as shown in Fig. 5.
  • VCVC voltage controlled variable capacitor
  • the length of time C1 and C2 are connected into the circuit is controlled by the output pulse width of the mono-stable flip flop 30, while the output pulse width of the mono-stable flip flop 30 is controlled by the output voltage of the PI controller in the controller 29. So the length of time the two capacitors C1 and C2 are connected into the circuit, i.e. the average equivalent capacitance of C1 and C2, is controlled by the output voltage of the controller 29.
  • the innate resonant frequency of the system can be adjusted to follow the driving frequency of the system.
  • the driving frequency of the system is fixed, a fixed frequency system will be realized which always works on soft-switching and resonant conditions meaning that the driving and innate resonant frequency of the system always equal to each other.
  • the general operating principle is that when the controller 29 finds that the driving frequency of the system does not equal to the innate system resonant frequency, the average equivalent capacitance of the VCVC will be controlled to adjust the innate system frequency making it follow and finally equal to the driving frequency of the system. Specifically, for example, when the driving frequency of the system is larger than the innate system resonant frequency, the VCVC will be controlled to connect into the circuit for a shorter period of time to increase the system innate resonant frequency and vice versa. As the VCVC has been presented in another patent, its detailed working principles will not be repeated here.
  • the circuit structure and operating principle of the other parts of the controller 29 in Fig. 5 is identical to their counterparts in Fig. 4. In short, in a variable frequency system, the driving frequency of the system is changed to follow the innate system resonant frequency while in a fixed frequency system, the innate system resonant frequency is adjusted or compensated to follow the system driving frequency.
  • This section presents a method to adjust the output voltage or power of oscillation systems through the effect of tuning/detuning of switch mode capacitors.
  • the output voltage or power of the system is adjusted by how many cycles the capacitor is connected into the circuit.
  • the switch connected to the tuning capacitor can be controlled to turn on and off at the zero voltage or current crossing points to realize soft-switching, which guarantees the stable operation of the system, and greatly increases the system efficiency and reduces EMI.
  • the section is divided into two parts. The basic operating principle of the method is introduced in detail in the first section with an example when the method is applied in a parallel tuning situation. In section 3.2, the serial tuning situation of the method is introduced.
  • capacitor C 32 is used as a parallel tuning capacitor, which is connected with a switch S 42 in serial.
  • the conduction period of the switch S 42 By adjusting the conduction period of the switch S 42, the length of time the capacitor C 32 is connected into the circuit can be adjusted, which influences the output voltage V out 34 of the system through the tuning/detuning effect of the capacitor C 32.
  • Soft-switching for the switch S 42 is realized by detecting the Zero Voltage Crossing (ZVC) points of the resonant voltage v res 33 when it changes from its positive half cycle to the negative half cycle. It should be noted that during the whole negative half cycle of v res 33, the body diode of the switch S 42 conducts. So it is soft-switching if S 42 is turned on or off during this period, i.e. the whole negative half cycle of v res 33. This is guaranteed by detecting the ZVC points of the resonant voltage v res 33 when it changes from its positive half cycle to the negative half cycle and feeding this information into a micro-controller 38.
  • ZVC Zero Voltage Crossing
  • the micro-controller can generate the gate driving signal for the switch S 42 according to this information to guarantee the soft-switching of the switch.
  • the ZVC points information can be input into a pulse counter 41 in the micro-controller 38 and the micro-controller can generate a rising edge for the switch S 42 when the counter overflows, which guarantees the capacitor C 32 is connected into the circuit after a fixed number of cycles of the resonant voltage v res 33.
  • both the ZVC points information from the ZVC points detection module 35 and the information from the mono-stable flip flop 37 need to be considered in the programming of the micro-controller 38 to guarantee that the falling edge happens at a moment when it is both soft-switching for the switch S 42 and after some proper cycles the capacitor C 32 is connected into the circuit.
  • the proper number of cycles the capacitor C 32 is connected into the circuit is determined by the output pulse width of the mono-stable flip flop 37, which reflects the error or difference between the actual output voltage of the system and the set value.
  • the output pulse width of the mono-stable flip flop 37 is controlled by the output voltage of the PI controller 36 which monitors the output voltage V out 34 of the system.
  • the PI controller 36 According to the fluctuation of the output voltage V out 34, the PI controller 36 generates an output voltage to control the output pulse width of the mono-stable flip flop 37 and finally the length of time or how many cycles the capacitor C 32 is connected into the circuit through generating the falling edge for the switch S 42 at a proper time by the micro-controller 38.
  • the output voltage V out 34 of the system can be adjusted or stabilized finally.
  • the above operation can be realized through the two interrupts 39 and 40 in the micro-controller 38. Specifically, set a flag in one of the interrupts meaning that this interrupt has happened and check this flag in another interrupt.
  • the microcontroller can generate a falling edge for the switch S 42 to disconnect the capacitor C 32 from the circuit.
  • both of the two tasks i.e. the task to realize soft-switching for the switch S 42 and the task to guarantee the capacitor C 32 has been connected into the circuit for a proper period of time, can be fulfilled.
  • the method to control the output pulse width of a mono-stable flip flop has been proposed in another patent, so will not be repeated here.
  • Fig. 7 The situation when the method is applied as a serial tuning capacitor is shown in Fig. 7. As can be seen, the only difference between the parallel and serial tuning situation is that there are two capacitors C1 54 and C2 53 connected with the switch S 52. This is to guarantee that the circuit will not be open when the switch S 52 is turned off. All the other parts of the circuit in Fig. 7 are of common knowledge or have been presented in the last section, and the basic operating principle of the method is the same as in the parallel tuning situation, so is repeated here.
  • This section proposes a method to realize synchronous rectification by current instead of voltage detection.
  • Mosfets are often used to replace diodes in rectification circuits as shown in Fig. 8.
  • One key issue in rectification circuits where diodes are replaced by Mosfets is how to control the on and off of the Mosfets to guarantee the on and off of the Mosfets are in phase with the rectified voltages and/or currents.
  • this function is finished mainly by the two modules, Module1 (56) and Module2 (65) as shown in Fig. 8.
  • the basic operating principle is that when the amplitude of the input AC voltage v s 66 is higher than the output voltage V out 60 of the system, the body diodes of the Mosfets in the related branch begin to conduct, which leads to the current in the related branch is higher than zero.
  • the related Module Mode1 or Module2
  • the output of the Module turns from low to high to control the Mosfets in the related branch to conduct, making the current in the related branch flow mainly through the Mosfets instead of their body diodes.
  • the current in related branches drops to zero, it will be detected by the related Module and the output of the related Module will turn from high to low to shut down the Mosfets in related branches.
  • the functions of the two Modules (Module1 and Module2) in Fig. 8 are as below:
  • Module1 Detects whether the current i 1 is zero or not and outputs a voltage to control the two switches S1 and S4 to be on when i 1 is not zero and off when i 1 is zero.
  • Module2 Detects whether the current i 2 is zero or not and outputs a voltage to control the two switches S2 and S3 to be on when i 2 is not zero and off when i 2 is zero.
  • Fig. 9 shows the waveforms of the AC input of the system v s 66, the DC output of the system V out 60, the current i 1 and the output signal of the Module1 GATE S1S4 55, where GATE S1S4 55 is the gate driving signal for the two switches S1 and S4. It can be seen from Fig. 9 that driven by GATE S1S4 55, the two switches S1 and S4 are on when the current i 1 is not zero and off when i 1 is zero, meaning synchronous rectification is realized. The situation of the other two switches S2 and S3 is similar, i.e.
  • Fig. 9 Another feature which can be seen from Fig. 9 is that the two switches S1 and S4 do not turn off when v s 66 becomes lower than V out 60, but turn off only when the current i 1 drops to zero, which avoids the drawback of the traditional rectification method which turns off S1 and S4 immediately after v s 66 becomes lower than V out 60 instead of after i 1 drops to zero, which makes i 1 flow through the body diodes of S1 and S2 so that the power loss is increased and the efficiency of the system lowered.
  • the basic difference between the proposed method and traditional ones is that synchronization is realized by detecting the current instead of the voltage of the system.
  • Traditional method is to compare the voltage in the two nodes A and B with the output voltage of the system V out 60. When the voltage in node A is higher than V out 60, the two switches S1 and S4 are controlled to turn on; otherwise they are controlled to turn off. Similarly, the two switches S2 and S3 are controlled to turn on when the voltage in node B is higher than V out 60 and off when the voltage in node B is lower than V out 60.
  • Another feature of this invention is that different from the traditional way of using only one filter inductor between the two nodes C and D, two different inductors L1 and L2 are used in the two branches as shown in Fig. 8. This is because that if only one inductor is used between the two nodes C and D, when the current in one branch drops to zero and the Mosfets in the related branch is shut down, a current glitch may be induced in another branch. So when this current glitch is detected by the current detection Module in another branch, the output of this current detection Module may turns from low to high to turn on the Mosfets in this branch wrongly leading to the malfunction of the system. By using two different inductors in the two different branches, this phenomenon can be avoided effectively.
  • Fig. 8 is only one embodiment of this invention. Those skilled in the art can find any number of variations without departure from the spirit or scope of the Applicant’s general inventive concept. For example, apply the method in half bridge, half wave; half bridge full wave, three phase or multiphase rectification circuits, etc. It is not the intention of the Applicant to restrict or in any way to limit the invention to the specific details.
  • Fig. 10 shows an example of the applications of the technique described above, i.e. firstly, adding capacitors to the coils in electric motors to form resonant tanks 68, and secondly driving the resonant tanks with switch mode DC-AC resonant converters 69.
  • Rotating magnetic field can be generated by driving the switch mode DC-AC converters with square waves in properly separated phases.
  • the system can be regarded as composed of the DC-AC converter, the resonant tank and the load.
  • the innate resonant frequency of the system can be changed.
  • the rotation rate and torque of the motor can be adjusted and controlled.
  • Fig. 10 is just one embodiment of the new type electric motors driven by resonant converters. Those skilled in the art can find any number of variations. It is not the intention of the Applicant to restrict or in any way limit the invention to the specific details shown in Fig. 10.
  • Fig. 11 shows the circuit structure of a new type of resonant switch mode power supply, where the function of Controller 1 (82) , to detect the innate system resonant frequency, can be realized with the method introduced in Section 2 of this invention or any other innate system frequency detection methods.
  • Controller 1 82
  • the switch mode DC-AC converter 70 can be driven with this frequency so that soft-switching and resonance can be realized to increase the efficiency of the system and lower the EMI.
  • the output voltage of the system V out 79 is not adjusted in the primary side by changing the duty cycle of the gate driving signal for the switch mode DC-AC converter 70, but in the secondary side by changing the duty cycle of the gate driving signal for the switch S1 (81) , which controls the length of time the capacitor C3 (75) is connected into the circuit.
  • the output voltage of the system V out 79 changes with the change of the length of time the capacitor capacitor C3 (75) is connected into the circuit because of its tuning/detuning effect.
  • the value of the capacitor C3 (75) is infinity meaning that it can be shorted so that the output voltage V out 79 is only adjusted by the duty cycle of the switch S1 (81) as shown in Fig.
  • controller 2 (80) can be realized by the method introduced in Section 3 of this invention or any other similar ones to guarantee the soft-switching of the switch S1 (81) and the capacitor C3 (75) having been connected into the circuit for a proper length of time or cycles. In this way, all the switches in this new type of switch mode power supply work in soft-switching condition so that the efficiency of this type of power supply can be increased greatly and EMI lowered dramatically.

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Abstract

A Power conversion system which always works on square wave driving, soft-switching and resonant conditions supported by the following four techniques and two of their applications are disclosed. ① A power factor correction technique; ② A technique to detect the innate resonant frequency of switch mode DC-AC converter driven oscillation systems by timed sampling; ③ A technique to dynamically adjust or stabilize the output voltage or power of electromagnetic oscillation systems by the tuning/detuning effect of soft-switching capacitors; ④ A synchronous rectification technique. The two applications are a new type electric motor driven by resonant converters and a new type of resonant switch mode power supply.

Description

[Title established by the ISA under Rule 37.2] POWER CONVERSION SYSTEM AND APPLICATIONS THEREOF
CROSS REFERENCE TO RELATED APPLICATIONS
The present application claims the priority of the New Zealand Patent Application No. 736612, entitled “A Power Conversion System and its Applications” , filed with the New Zealand Intellectual Property Office on October 20, 2017, the entity of which is incorporated herein by reference.
FIELD
This invention relates generally to power electronic systems. Specifically, it’s a series of techniques to converter AC mains to a required DC voltage and their applications. The techniques include a power factor correction (PFC) technique, a technique to dynamically detect the system resonant frequency by timed sampling, a technique to dynamically adjust or stabilize the output voltage or power of the system by tuning/detuning and a synchronous rectification technique. The applications are a new type electric motor driven by resonant converters and a new type of resonant switch mode power supply. These are only two typical examples of the extensive applications of the techniques. In fact, any power electronic systems wherever there is an electromagnetic oscillation there is a potential for the use of these techniques, such as wireless power transfer (WPT) systems, switch mode power supplies, DC-DC converters, electric motors, electromagnetic induction heating systems, high voltage direct current transmission (HVDC) , some kind of instruments, etc. By using these techniques, the system driving frequency and innate resonant frequency can be made always equal to each other, making the system always work on square wave driving, soft-switching and resonant conditions, which will increase the system efficiency and lower the Electro Magnetic Interference (EMI) greatly.
BACKGROUND
The innate resonant frequency of a WPT or any other electromagnetic oscillation systems in power electronic fields is a very important factor, which influences almost every important aspect of the system such as resonance, soft-switching, power transfer ability, efficiency, EMI, etc. Once the frequency of the system is under control, every important aspect of the system will be under control. To control the frequency of the system, the innate resonant frequency of the system needs to be known first so that the system can be driven with this frequency to realize soft-switching and resonance, which makes the power transfer ability and efficiency of the system optimized. However, the innate resonant frequency of a WPT system is not constant but changes with many factors such as the coupling coefficient between the primary and secondary side, the variation of the load and many other parameters of the system. So a technique to detect the ever-changing system resonant frequency in real time is very important. All the techniques proposed in this invention are based on the concept of innate system resonant frequency and making the innate system resonant frequency equal to the system driving frequency. The details of these techniques are described in the “DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION” section  through the following six parts:
1) A power factor correction technique;
2) A technique to detect the innate system resonant frequency by timed sampling;
3) A technique to dynamically adjust or stabilize the output voltage or power of electromagnetic oscillation systems by tuning/detuning;
4) A synchronous rectification technique;
5) A new type electric motor driven by resonant converters;
6) A new type of resonant switch mode power supply.
SUMMARY
This invention proposes the following four techniques, i.e. a power factor correction (PFC) technique, a technique to dynamically detect the system resonant frequency by timed sampling, a technique to dynamically adjust or stabilize the output voltage or power of the system through the effect of tuning/detuning of a soft-switching capacitor, a synchronous rectification technique, and two of the applications of these techniques, i.e. a new type electric motor driven by resonant converters and a new type of resonant switch mode power supply. These techniques are based on the concept of resonance and soft-switching realized by making the driving frequency of the system equal to the innate resonant frequency of the system, which are completely different from traditional concepts of hard-switched conversion for the design of power electronic systems. The efficiency and power transfer ability of power electronic systems will be increased greatly by the proposed techniques and the EMI of the system decreased dramatically.
BRIEF DESCRIPTION OF THE DRAWINGS
The accompanying drawings which are incorporated in and constitute part of the specification, illustrate embodiments of the invention and, together with the general and detailed descriptions of the invention given above and below, serve to explain the principles of the invention.
Fig. 1 shows a block diagram of the overall structure of the power conversion system proposed in this invention.
Fig. 2 shows a circuit diagram of the power factor correction technique proposed in this invention.
Fig. 3 shows the waveforms of key signals of the power factor correction technique proposed in this invention.
Fig. 4 shows a circuit diagram of detecting the innate system resonant frequency by timed sampling in a variable frequency system.
Fig. 5 shows a circuit diagram of detecting the innate system resonant frequency by timed sampling in a fixed frequency system.
Fig. 6 shows a circuit diagram of the technique to stabilize the output voltage or power of electromagnetic oscillation systems in the way of parallel tuning.
Fig. 7 shows a circuit diagram of the technique to stabilize the output voltage or power of electromagnetic oscillation systems in the way of serial tuning.
Fig. 8 shows a circuit diagram of the synchronous rectification technique proposed in this invention.
Fig. 9 shows the waveforms of key signals of the synchronous rectification technique.
Fig. 10 shows a circuit diagram of a new type electric motor driven by resonant converters.
Fig. 11 shows a circuit diagram of the new type resonant switch mode power supplies.
Fig. 12 shows a circuit diagram of the new type resonant switch mode power supplies when the tuning capacitor is shorted.
DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION
This part includes the following six sections:
1) A power factor correction technique;
2) A technique to detect the innate system resonant frequency by timed sampling;
3) A technique to dynamically adjust or stabilize the output voltage or power of electromagnetic oscillation systems by tuning/detuning;
4) A synchronous rectification technique;
5) A new type electric motor driven by resonant converters;
6) A new type of resonant switch mode power supply.
1. A power factor correction technique
Fig. 1 is the representative graph of this invention which shows a power conversion system to converter the AC main into a required DC voltage, which is divided into three parts, i.e. the power factor correction (PFC) part 2, the DC-AC conversion part 3 and the AC-DC conversion part 4. The input of this power conversion system is an AC main 1 and the output a DC voltage 5.
This section introduces the first part of the power conversion system, i.e. the PFC part 2. The circuit structure of this part is shown in Fig. 2, where v s 6 represents the voltage of the AC main, i s 7 the current of the AC main. C f 10 is the filter capacitance for the voltage across the load of the system R load 11. The basic operating principle is to control the two switches S1 and S2 turn on and off alternately at the zero voltage crossing points of v s 6. This guarantees the soft-switching of the two switches S1 and S2. Specifically, at the positive half cycle of v s, S1 is turned on and S2 turned off, when one branch of the current starts from  the positive terminal of v s, through L1, L2, and back to the negative terminal of v s while another branch of the current starts from the positive terminal of v s, through L1, C f, R load, S1 and back to the negative terminal of v s. At the negative half cycle of v s, S2 is turned on and S1 turned off, when one branch of the current starts from the negative terminal of v s, through L2, L1 and back to the positive terminal of v s while another branch of the current starts from the negative terminal of v s, through L2, C f, R load, S2 and back to the positive terminal of v s.
Fig. 3 shows the waveforms of the gate driving signal Gate_S1 for the switch S1, the voltage v s and current i s of the AC main input, from which it can be seen that v s and i s is in phase, meaning that the power factor is one. As mentioned above, the two switches S1 and S2 are turned on and off at the zero voltage crossing points of v s which means that S1 and S2 are soft-switching. So the proposed method can realize both of the two goals, i.e. making the power factor of the system to be one and realizing soft-switching for the switches of the system, which is very difficult for conventional power factor correction methods.
2. A technique to detect the innate system resonant frequency by timed sampling
This section is divided into two parts. Firstly, the basic operating principle of the method to detect the innate system resonant frequency by timed sampling is described in detail in Section 2.1 with an example when this method is applied in a variable frequency system. Secondly, the situation when this method is applied in a fixed frequency system is described in Section 2.2.
2.1 The basic operating principle of the method when it is applied in a variable frequency system
The circuit topology of the method is shown in Fig. 4, where the part in the dashed block, i.e. the controller 18 is the essence of the method. It includes a timed sampling module 16, a voltage conversion module 17, a PI (Proportional-Integral) controller 19 and a voltage controlled oscillator (VCO) 20. The outside part of the dashed block is mainly a switch mode DC-AC converter 14 with a resonant tank 15. The gate driving signal 21 of the DC-AC converter 14 is generated by the VCO 20, which can be regarded as the actuator of the controller 18.
The most important part of the controller 18 is the timed sampling module 16, which samples the resonant voltage or current of the resonant tank 15 at exactly the moment when the switches in the DC-AC converter 14 are turned on or off. When this sampled voltage or current is zero, it is zero voltage switching (ZVS) or zero current switching (ZCS) and the innate system resonant frequency equals to the gate driving frequency 21. When this sampled voltage or current is not zero, the gate driving frequency 21 does not equal to the innate system resonant frequency.
The function of the voltage conversion module 17 is to generate a voltage signal according to the sign and value of the sampled voltage or current. When the value of the sampled voltage or current is zero, the voltage generate by 17 equals to the reference voltage of the PI controller 19, when the output voltage of the PI controller 19 and the output frequency of the VCO 20 remains constant. Otherwise, if the value of  the sampled voltage or current is not zero, the absolute value of the output voltage of the voltage conversion module 17 is proportional to the value of the sampled voltage or current, when the output voltage of 17 does not equal to the reference voltage of the PI controller 19 which will lead to the output voltage of the PI controller 19 and the output frequency of the VCO 20 changing continuously until the driving frequency 21 of the system equals to the innate system resonant frequency, when the system regains its resonant and soft-switching condition and the value of the sampled voltage or current becomes zero again.
The function of the PI controller 19 is to generate a control voltage for the VCO 20 to control its output frequency according to the output voltage from the voltage conversion module 17. The general operating principle is that when the output voltage from the voltage conversion module 17 equals to the reference voltage of the PI controller 19, the output voltage of the PI controller 19 as well as the output frequency 21 of the VCO 20 remains constant, which means that the driving frequency of the system equals to the innate system resonant frequency and soft-switching is achieved at the moment. When the output voltage from the voltage conversion module 17 does not equal to the reference voltage of the PI controller 19, the output voltage of the PI controller 19 and therefore the output frequency 21 of the VCO 20 will vary continuously until the driving frequency 21 of the system equals to the innate system resonant frequency and soft-switching is achieved.
Generally, the situation when the value of the sampled voltage or current is zero is roughly the same, i.e. in this situation the output voltage of the voltage conversion module 17 equals to the reference voltage of the PI controller 19; the output voltage of the PI controller 19 and the output frequency 21 of the VCO 20 remains constant; the driving frequency of the system equals to the innate system resonant frequency meaning the system is working on resonant and soft-switching condition at the moment. The detailed operation may be a little different according to the type of the DC-AC converter used and the type of sampling, i.e. what kind of signal (voltage or current) is sampled and at which moment (the rising or falling edge of the system gate driving signal) , etc. For example, one situation may be that the value of the sampled voltage or current being positive means that the driving frequency of the system is higher than the innate system resonant frequency; the output voltage of the voltage conversion module 17 is larger than the reference voltage of the PI controller 19; the output voltage of the PI controller 19 and the output frequency 21 of the VCO 20 will decrease continuously until the driving frequency 21 of the system equals to the innate resonant frequency of the system. On the contrary, if the value of the sampled voltage or current is negative, everything will be the inverse, i.e. it means that the driving frequency of the system is lower than the innate system resonant frequency; the output voltage of the voltage conversion module 17 is smaller than the reference voltage of the PI controller 19; the output voltage of the PI controller 19 and the output frequency 21 of the VCO 20 will increase continuously until the driving frequency 21 of the system equals to the innate resonant frequency of the system.
It should be noted that above is just one embodiment of the invention. Those skilled in the art can find any number of variations. For example, all the functions of the three modules in the controller 18, i.e. the  timed sampling module 16, the voltage conversion module 17 and the PI controller 19, can be realized in digital manners within a micro-controller. It is not the intention of the Applicant to restrict or in any way to limit the invention to the specific details.
2.2 Application of the method in a fixed frequency system
Fig. 5 shows the situation when the above method is applied to realize a fixed frequency system. The main difference between the fixed frequency system of this section and the variable frequency system in the last section is the “actuator” of the control loop. For a variable frequency system as described in the last section, the actuator is a VCO 20, while for a fixed frequency system as described in this section, the actuator is a voltage controlled variable capacitor (VCVC) 24 or 26 as shown in Fig. 5. There are actually two of such VCVCs in Fig. 5, one consists of C1 and S1 while the other consists of C2 and S2. The length of time C1 and C2 are connected into the circuit is controlled by the output pulse width of the mono-stable flip flop 30, while the output pulse width of the mono-stable flip flop 30 is controlled by the output voltage of the PI controller in the controller 29. So the length of time the two capacitors C1 and C2 are connected into the circuit, i.e. the average equivalent capacitance of C1 and C2, is controlled by the output voltage of the controller 29. By adjusting the average equivalent capacitance of C1 and C2, the innate resonant frequency of the system can be adjusted to follow the driving frequency of the system. When the driving frequency of the system is fixed, a fixed frequency system will be realized which always works on soft-switching and resonant conditions meaning that the driving and innate resonant frequency of the system always equal to each other.
The general operating principle is that when the controller 29 finds that the driving frequency of the system does not equal to the innate system resonant frequency, the average equivalent capacitance of the VCVC will be controlled to adjust the innate system frequency making it follow and finally equal to the driving frequency of the system. Specifically, for example, when the driving frequency of the system is larger than the innate system resonant frequency, the VCVC will be controlled to connect into the circuit for a shorter period of time to increase the system innate resonant frequency and vice versa. As the VCVC has been presented in another patent, its detailed working principles will not be repeated here. The circuit structure and operating principle of the other parts of the controller 29 in Fig. 5 is identical to their counterparts in Fig. 4. In short, in a variable frequency system, the driving frequency of the system is changed to follow the innate system resonant frequency while in a fixed frequency system, the innate system resonant frequency is adjusted or compensated to follow the system driving frequency.
3. A technique to adjust or stabilize the output voltage or power of electromagnetic oscillation systems by tuning/detuning
This section presents a method to adjust the output voltage or power of oscillation systems through the effect of tuning/detuning of switch mode capacitors. The output voltage or power of the system is adjusted by how many cycles the capacitor is connected into the circuit. By detecting the resonant voltage or current of the system, the switch connected to the tuning capacitor can be controlled to turn on and off at the zero  voltage or current crossing points to realize soft-switching, which guarantees the stable operation of the system, and greatly increases the system efficiency and reduces EMI. The section is divided into two parts. The basic operating principle of the method is introduced in detail in the first section with an example when the method is applied in a parallel tuning situation. In section 3.2, the serial tuning situation of the method is introduced.
3.1 The basic operating principle of the method when it is applied as a parallel tuning capacitor
The circuit topology of the method is shown in Fig. 6. As can be seen, capacitor C 32 is used as a parallel tuning capacitor, which is connected with a switch S 42 in serial. By adjusting the conduction period of the switch S 42, the length of time the capacitor C 32 is connected into the circuit can be adjusted, which influences the output voltage V out 34 of the system through the tuning/detuning effect of the capacitor C 32. There are mainly two tasks which need to be finished for this method to work smoothly. The first is how to realize soft-switching for the switch S 42. The second is how to determine the length of time the capacitor C 32 is connected into the circuit so that the output voltage of the system can be stabilized.
Soft-switching for the switch S 42 is realized by detecting the Zero Voltage Crossing (ZVC) points of the resonant voltage v res 33 when it changes from its positive half cycle to the negative half cycle. It should be noted that during the whole negative half cycle of v res 33, the body diode of the switch S 42 conducts. So it is soft-switching if S 42 is turned on or off during this period, i.e. the whole negative half cycle of v res 33. This is guaranteed by detecting the ZVC points of the resonant voltage v res 33 when it changes from its positive half cycle to the negative half cycle and feeding this information into a micro-controller 38. The micro-controller can generate the gate driving signal for the switch S 42 according to this information to guarantee the soft-switching of the switch. For example, the ZVC points information can be input into a pulse counter 41 in the micro-controller 38 and the micro-controller can generate a rising edge for the switch S 42 when the counter overflows, which guarantees the capacitor C 32 is connected into the circuit after a fixed number of cycles of the resonant voltage v res 33.
To generate the falling edge for the switch S 42, both the ZVC points information from the ZVC points detection module 35 and the information from the mono-stable flip flop 37 need to be considered in the programming of the micro-controller 38 to guarantee that the falling edge happens at a moment when it is both soft-switching for the switch S 42 and after some proper cycles the capacitor C 32 is connected into the circuit. The proper number of cycles the capacitor C 32 is connected into the circuit is determined by the output pulse width of the mono-stable flip flop 37, which reflects the error or difference between the actual output voltage of the system and the set value. The output pulse width of the mono-stable flip flop 37 is controlled by the output voltage of the PI controller 36 which monitors the output voltage V out 34 of the system. According to the fluctuation of the output voltage V out 34, the PI controller 36 generates an  output voltage to control the output pulse width of the mono-stable flip flop 37 and finally the length of time or how many cycles the capacitor C 32 is connected into the circuit through generating the falling edge for the switch S 42 at a proper time by the micro-controller 38. Through the tuning/detuning effect of the capacitor C 32, the output voltage V out 34 of the system can be adjusted or stabilized finally. For the purpose of quick response, the above operation can be realized through the two interrupts 39 and 40 in the micro-controller 38. Specifically, set a flag in one of the interrupts meaning that this interrupt has happened and check this flag in another interrupt. If the flag shows that the other interrupt has happened, the microcontroller can generate a falling edge for the switch S 42 to disconnect the capacitor C 32 from the circuit. In this way, both of the two tasks, i.e. the task to realize soft-switching for the switch S 42 and the task to guarantee the capacitor C 32 has been connected into the circuit for a proper period of time, can be fulfilled. The method to control the output pulse width of a mono-stable flip flop has been proposed in another patent, so will not be repeated here.
It should be noted that above is just one embodiment of the invention. Those skilled in the art can find any number of variations without departure from the spirit or scope of the Applicant’s general inventive concept. For example, in special situations, it can be assumed that the value of the tuning capacitor C 32 is infinity meaning that it can be shorted so that the output voltage of the system is adjusted simply with the duty cycle of the switch S 42 as shown in Fig. 12; or some of the functions such as those of the PI controller 36 and the mono-stable flip flop 37 can be realized in a micro-controller with proper programming; or the functions of the micro-controller 38 can also be realized through properly designed hardware digital logic. Those skilled in the art can find any number of variations. It is not the intention of the Applicant to restrict or in any way limit the invention to the specific details.
3.2 The situation when the method is applied as a serial tuning capacitor
The situation when the method is applied as a serial tuning capacitor is shown in Fig. 7. As can be seen, the only difference between the parallel and serial tuning situation is that there are two capacitors C1 54 and C2 53 connected with the switch S 52. This is to guarantee that the circuit will not be open when the switch S 52 is turned off. All the other parts of the circuit in Fig. 7 are of common knowledge or have been presented in the last section, and the basic operating principle of the method is the same as in the parallel tuning situation, so is repeated here.
4. A synchronous rectification technique
This section proposes a method to realize synchronous rectification by current instead of voltage detection. To further increase efficiency, Mosfets are often used to replace diodes in rectification circuits as shown in Fig. 8. One key issue in rectification circuits where diodes are replaced by Mosfets is how to control the on and off of the Mosfets to guarantee the on and off of the Mosfets are in phase with the rectified voltages and/or currents. In this invention, this function is finished mainly by the two modules, Module1 (56) and Module2 (65) as shown in Fig. 8. The basic operating principle is that when the amplitude of the input AC voltage v s 66 is higher than the output voltage V out 60 of the system, the body  diodes of the Mosfets in the related branch begin to conduct, which leads to the current in the related branch is higher than zero. When a non-zero current is detected by the related Module (Module1 or Module2) , the output of the Module turns from low to high to control the Mosfets in the related branch to conduct, making the current in the related branch flow mainly through the Mosfets instead of their body diodes. When the current in related branches drops to zero, it will be detected by the related Module and the output of the related Module will turn from high to low to shut down the Mosfets in related branches. The functions of the two Modules (Module1 and Module2) in Fig. 8 are as below:
(1) Module1: Detects whether the current i 1 is zero or not and outputs a voltage to control the two switches S1 and S4 to be on when i 1 is not zero and off when i 1 is zero.
(2) Module2: Detects whether the current i 2 is zero or not and outputs a voltage to control the two switches S2 and S3 to be on when i 2 is not zero and off when i 2 is zero.
The above functions of the two Modules can be realized through voltage comparators after converting the two current signals i 1and i 2 into voltage signals. Fig. 9 shows the waveforms of the AC input of the system v s 66, the DC output of the system V out 60, the current i 1 and the output signal of the Module1 GATE S1S4 55, where GATE S1S4 55 is the gate driving signal for the two switches S1 and S4. It can be seen from Fig. 9 that driven by GATE S1S4 55, the two switches S1 and S4 are on when the current i 1 is not zero and off when i 1 is zero, meaning synchronous rectification is realized. The situation of the other two switches S2 and S3 is similar, i.e. they are on when i 2 is not zero and off when i 2 is zero. Another feature which can be seen from Fig. 9 is that the two switches S1 and S4 do not turn off when v s 66 becomes lower than V out 60, but turn off only when the current i 1 drops to zero, which avoids the drawback of the traditional rectification method which turns off S1 and S4 immediately after v s 66 becomes lower than V out 60 instead of after i 1 drops to zero, which makes i 1 flow through the body diodes of S1 and S2 so that the power loss is increased and the efficiency of the system lowered.
The basic difference between the proposed method and traditional ones is that synchronization is realized by detecting the current instead of the voltage of the system. Traditional method is to compare the voltage in the two nodes A and B with the output voltage of the system V out 60. When the voltage in node A is higher than V out 60, the two switches S1 and S4 are controlled to turn on; otherwise they are controlled to turn off. Similarly, the two switches S2 and S3 are controlled to turn on when the voltage in node B is higher than V out 60 and off when the voltage in node B is lower than V out 60. There are mainly two problems with this traditional method. Firstly, if there are no filter inductors before the filter capacitor C f 58, when S1, S4 or S2, S3 conduct, the voltages in node A or B will roughly equal to the output voltage of the system V out 60, which may cause oscillation for the outputs of the voltage comparator which compares the voltages between the two nodes A or B and the output voltage of the system V out 60, which makes the operation of the system unstable. Secondly, if there are filter inductors before the filter capacitor C f 58, when S1, S4 or S2, S3 turn off, the current in the filter inductors usually does not drop to zero, so this current will flow through the body diodes of related Mosfets increasing the power loss and lowering the  efficiency of the system. None of the above two problems exist for the proposed method in this invention which detects directly the current instead of the voltage of the system.
Another feature of this invention is that different from the traditional way of using only one filter inductor between the two nodes C and D, two different inductors L1 and L2 are used in the two branches as shown in Fig. 8. This is because that if only one inductor is used between the two nodes C and D, when the current in one branch drops to zero and the Mosfets in the related branch is shut down, a current glitch may be induced in another branch. So when this current glitch is detected by the current detection Module in another branch, the output of this current detection Module may turns from low to high to turn on the Mosfets in this branch wrongly leading to the malfunction of the system. By using two different inductors in the two different branches, this phenomenon can be avoided effectively.
It should be noted that Fig. 8 is only one embodiment of this invention. Those skilled in the art can find any number of variations without departure from the spirit or scope of the Applicant’s general inventive concept. For example, apply the method in half bridge, half wave; half bridge full wave, three phase or multiphase rectification circuits, etc. It is not the intention of the Applicant to restrict or in any way to limit the invention to the specific details.
5. A new type electric motor driven by resonant converters
Fig. 10 shows an example of the applications of the technique described above, i.e. firstly, adding capacitors to the coils in electric motors to form resonant tanks 68, and secondly driving the resonant tanks with switch mode DC-AC resonant converters 69. Rotating magnetic field can be generated by driving the switch mode DC-AC converters with square waves in properly separated phases. By detecting the innate system resonant frequency in the way presented in Section 2 or any other proper methods and driving the system with the detected system resonant frequency, the system can be made always work on soft-switching and resonant conditions so that the efficiency of the system can be increased and EMI lowered. The system can be regarded as composed of the DC-AC converter, the resonant tank and the load. By connecting different numbers and values of capacitors into the resonant tanks dynamically, the innate resonant frequency of the system can be changed. Combined with adjusting the source voltage of the switch mode DC-AC resonant converters, which influences the current in the resonant tank or in the coils of the motor, the rotation rate and torque of the motor can be adjusted and controlled.
It should be noted however that Fig. 10 is just one embodiment of the new type electric motors driven by resonant converters. Those skilled in the art can find any number of variations. It is not the intention of the Applicant to restrict or in any way limit the invention to the specific details shown in Fig. 10.
6. A new type of resonant switch mode power supply
Fig. 11 shows the circuit structure of a new type of resonant switch mode power supply, where the function of Controller 1 (82) , to detect the innate system resonant frequency, can be realized with the  method introduced in Section 2 of this invention or any other innate system frequency detection methods. With the innate system resonant frequency known, the switch mode DC-AC converter 70 can be driven with this frequency so that soft-switching and resonance can be realized to increase the efficiency of the system and lower the EMI. Different from traditional switch mode power supplies, the output voltage of the system V out 79 is not adjusted in the primary side by changing the duty cycle of the gate driving signal for the switch mode DC-AC converter 70, but in the secondary side by changing the duty cycle of the gate driving signal for the switch S1 (81) , which controls the length of time the capacitor C3 (75) is connected into the circuit. The output voltage of the system V out 79 changes with the change of the length of time the capacitor capacitor C3 (75) is connected into the circuit because of its tuning/detuning effect. In special situations, it can also be regarded that the value of the capacitor C3 (75) is infinity meaning that it can be shorted so that the output voltage V out 79 is only adjusted by the duty cycle of the switch S1 (81) as shown in Fig. 12. The function of controller 2 (80) can be realized by the method introduced in Section 3 of this invention or any other similar ones to guarantee the soft-switching of the switch S1 (81) and the capacitor C3 (75) having been connected into the circuit for a proper length of time or cycles. In this way, all the switches in this new type of switch mode power supply work in soft-switching condition so that the efficiency of this type of power supply can be increased greatly and EMI lowered dramatically.
What Fig. 11 and Fig. 12 show is only the parallel tuning situation. It should be noted that this is just one embodiment of this invention. Those skilled in the art can find any number of variations without departure from the spirit or scope of the Applicant’s general inventive concept. It is not the intention of the Applicant to restrict or in any way limit the invention to the specific details.
While the present inventions have been illustrated by the descriptions of the embodiments thereof, and while the embodiments have been described in detail, it is not the intention of the Applicant to restrict or in any way limit the scope of the appended claims to such details. Additional advantages and modifications will readily appear to those skilled in the art. Therefore, the invention in its broader aspects is not limited to the specific details, representative apparatus and methods, and illustrative examples shown and described. Accordingly, departures may be made from such details without departure from the spirit or scope of the Applicant’s general inventive concept. Reference to any prior art in this specification does not constitute an admission that such prior art forms part of the common general knowledge.

Claims (7)

  1. A Power conversion system, comprising:
    a power factor correction part, a DC-AC conversion part and an AC-DC conversion part;
    wherein:
    the DC-AC conversion part comprises a switch mode DC-AC converter, a resonant tank and a controller for detecting an innate system resonant frequency and drive the switch mode DC-AC converter with the detected innate system resonant frequency;
    the AC-DC conversion part comprises a resonant tank, a switch mode capacitor tuning/detuning circuit, a Proportional-Integral (PI) controller, a rectifier, a filter capacitor and a load;
    the resonant tanks of the DC-AC and AC-DC conversion parts comprise capacitors and inductors connected together in parallel and/or serial;
    the DC-AC and AC-DC conversion parts are coupled together via the capacitors or inductors in their resonant tanks.
  2. The power factor conversion part in Claim 1, comprising:
    a sinusoidal input voltage v s, two inductors L1 and L2, two switches S1 and S2, a filter capacitor and a load;
    wherein:
    the two inductors L1, L2, and the two switches S1, S2 are configured as a push pull structure;
    the filter capacitor is configured to provide a relatively stable DC voltage for the load;
    the two switches S1 and S2 are turned on and off alternately when v s is zero, so they are soft-switching;
    specifically, at the positive half cycle of v s, S1 is on and S2 is off while at the negative half cycle of v s, S2 is on and S1 is off;
    as such, the voltage and current of the sinusoidal input v s and i s are roughly in phase meaning that the power factor is one.
  3. The controller of the DC-AC conversion part in Claim 1, comprising:
    a timed sampling module, a voltage conversion module, a PI controller and a voltage controlled oscillator (VCO) ;
    wherein:
    the function of the timed sampling module is to sample the resonant voltage or current at exactly the moment when the switches in the switch mode DC-AC converter are turned on or off;
    if the value of the sampled voltage or current is zero, it means that the system is working on soft-switching and resonant conditions, and a driving frequency equals to the innate system resonant frequency at the moment;
    otherwise, if the value of the sampled voltage or current is not zero, it means that the driving frequency of the system does not equal to the innate system resonant frequency at the moment;
    the function of the voltage conversion module is to change the sampled voltage or current from a digital form into the form of an analog voltage;
    when the sampled voltage or current is zero, the voltage from the voltage conversion module equals to the reference voltage of the PI controller;
    when the sampled voltage or current is not zero, the voltage from the voltage conversion module does not equal to the reference voltage of the PI controller, which makes the output voltage of the PI controller change continuously until the voltage from the voltage conversion module becomes equal to the reference voltage of the PI controller;
    the changing output voltage from the PI controller changes the output frequency of the VCO;
    the output frequency of the VCO is used as the driving frequency of the system to change the system driving frequency until it equals to the innate system resonant frequency;
    at the above condition, the system is working on soft-switching and resonant condition, the sampled voltage or current is zero, the output voltage of the voltage conversion module equals the reference voltage of the PI controller, the output voltage of the PI controller and the output frequency of the VCO remains constant until the innate system resonant frequency deviates from the system driving frequency again;
    instead of the controlling the output frequency of a VCO, the changing output voltage from the PI controller can also be used to change the equivalent average capacitance of a VCVC to compensate the innate system resonant frequency making it equal to or follow the driving frequency of the system;
    all the functions of the three modules, i.e. the timed sampling module 16, the voltage conversion module 17 and the PI controller 19, can also be realized within a micro-controller or any other programmable components in the form of a software.
  4. The switch mode capacitor tuning/detuning circuit of the AC-DC conversion part in Claim 1, comprising:
    a switch mode capacitor, a zero voltage crossing (ZVC) points detection module, a module to determine how long the capacitor should be connected into the circuit named module1 and a module to generate a driving signal for a switch named module2;
    wherein:
    the switch mode capacitor comprises a capacitor and a switch, located before the rectifier and used as a parallel or serial tuning capacitor in the resonant tank of the AC-DC conversion part;
    in special situations, the capacitance of the capacitor can be regarded as infinity meaning that the capacitor can be shorted;
    the ZVC points detection module monitors the zero voltage crossing points of the resonant voltage across the capacitor of the switch mode capacitor;
    module1 comprises a PI controller and a mono-stable flip flop;
    the PI controller monitors an output voltage of the system and outputs a voltage to control an output pulse width of the mono-stable flip flop;
    how long or how many cycles the capacitor is connected into the circuit is determined by the output pulse width of the mono-stable flip flop;
    module2 generates the driving signal for the switch according to two signals, one is a ZVC signal from the ZVC points detection module, the other is a output pulse width signal from the mono-stable flip flop, specifically:
    module2 generates a high signal for the switch of the switch mode capacitor to connect the capacitor into the circuit periodically after a fixed number of cycles of the resonant voltage across the capacitor according to the signal from the ZVC points detection module;
    module2 generates a low signal for the switch of the switch mode capacitor to disconnect the capacitor from the circuit at a ZVC point of the resonant voltage across the capacitor after receiving an end edge (rising or falling edge) of the output pulse width signal from the mono-stable flip flop;
    as such, the output voltage or power of the system can be adjusted or stabilized by how long or how many cycles the capacitor is connected into the circuit;
    another feature of such a system is that the output voltage of the system can also be adjusted by a reference voltage of the PI controller;
    the functions of the PI controller and the mono-stable flip flop can also be realized in a micro-controller or any other programmable components with proper programming.
  5. The rectifier of the AC-DC conversion part in Claim 1, comprising:
    a rectification bridge comprising four switches S1, S2, S3, S4, two inductors L1, L2, and two current detection modules: module1 and module2;
    wherein:
    a first branch comprises S1, S4 in series with L1 wherein S1 and S2 are controlled to turn on and off concurrently;
    a second branch comprises S2, S3 in series with L2 wherein S2 and S3 are controlled to turn on and off concurrently;
    different from traditional methods which control the on and off of the switches by detecting the voltage of the system, this invention controls the on and off of the switches by detecting the currents in the two branches;
    module1 detects the current in the first branch, turn on S1 and S4 when the current is not zero and turn off S1 and S4 when the current is zero;
    module2 detects the current in the second branch, turn on S2 and S3 when the current is not zero and turn off S2 and S3 when the current is zero;
    when the amplitude of the AC input of the system is higher than the DC output of the system, the body diodes of the switches in a related branch conduct transitorily first making the current in the related branch becomes not to be zero; after detecting that the current in the related branch is not zero, the current detection module in the related branch outputs a high signal to turn on the switches in the related branch to replace the conducting body diodes;
    when the current in a related branch drops to zero, the current detection module in the related branch outputs a low signal to turn off the switches in the related branch;
    one feature of this invention is that two different inductors L1 and L2 are used respectively for the two different branches instead of using only one filter inductor for both of the two branches,  this is because that if a common filter inductor is used for both of the two branches, when the current in one branch drops to zero and the switches in this branch are turned off, a current glitch may be induced in another branch, which when detected by the current detection module in another branch, may cause the output of this current detection module turn from low to high to turn on the switches in this branch wrongly leading to the malfunction of the system; by using two different inductors in the two different branches, this phenomenon can be avoided effectively;
    by detecting current instead of voltage to determine the on and off of the switches, the following two problems can be avoided:
    firstly, if there is no filter inductor between the rectification bridge and the output voltage of the system, when the switches in the rectification bridge conduct, the amplitude of the AC input of the system equals roughly to the DC output of the system, which may cause the output of the voltage comparator which compares the voltage of the AC input of the system and the voltage of the DC output of the system to oscillate leading to unstableness for the operation of the system;
    secondly, if there is filter inductor between the rectification bridge and the output voltage of the system, when the voltage of the AC input of the system drops lower than the DC output of the system so that the switches in related branches are turned off, the current in the related branch is still not zero because of the existence of the filter inductor, which causes the remaining current flow through the body diodes of the switches leading to increased power losses.
  6. One application of the techniques in any one of the claims from 1 to 5 -a new type electric motor driven by switch mode DC-AC resonant converters, comprising:
    resonant tanks, switch mode DC-AC resonant converters, a controller;
    wherein:
    the resonant tanks are formed by adding capacitors to the coils in the electric motor;
    the resonant tanks are driven by switch mode DC-AC resonant converters;
    the controller can be realized by the technique in Claim 3 or any other ones which can detect the innate system resonant frequency;
    driving the switch mode DC-AC resonant converters with the detected innate system resonant frequency to realize resonance and soft-switching;
    by driving different switch mode DC-AC resonant converters with square waves in properly separated phases, rotating magnetic field can be generated;
    the innate system resonant frequency can be adjusted or changed dynamically by adding different numbers and/or values of capacitors to the coils of the electric motor in real time;
    the source voltage of the switch mode DC-AC resonant converters influences the current in the resonant tanks or the coils of the motor;
    the rotation rate and torque of the motor can be adjusted by changing the innate system resonant frequency in combination with adjusting the source voltage of the switch mode DC-AC resonant converters.
  7. Another application of the techniques in any one of the claims from 1 to 5 –a new type of resonant switch mode power supply, comprising:
    a primary side circuit, a high frequency transformer and a secondary side circuit;
    wherein:
    the primary side circuit comprises a switch mode DC-AC converter, a primary side controller and a primary side resonant capacitor;
    the secondary side circuit comprises a secondary side resonant capacitor, a switch mode capacitor, a rectifier, a secondary side controller and a common filter capacitor and load wherein a voltage across the filter capacitor and load is the output voltage of the system;
    the primary and secondary side circuits are coupled together through the high frequency transformer;
    the primary side controller is realized by the techniques in Claim 3 or any other ones with similar functions to guarantee soft-switching for the switch mode DC-AC converter;
    the output voltage of the system is not adjusted in the primary side by changing the duty cycle of the switches in the switch mode DC-AC converter, but adjusted in the secondary side through the tuning/detuning effect of the switch mode capacitor;
    the on and off of the switch mode capacitor is controlled by the secondary side controller;
    the secondary side controller is realized by the techniques in Claim 4 or any other ones with similar functions to guarantee soft-switching for the switch mode capacitor and stabilization of the output voltage of the system through the tuning/detuning effect of the switch mode capacitor;
    the rectifier in the secondary side circuit is realized by the techniques in Claim 5 or any other  ones with similar functions to guarantee the switches in the rectifier are soft-switching;
    as such, all the switches in this new type resonant switch mode power supply are soft-switching so that the efficiency of the system is increased greatly and EMI lowered dramatically.
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