CN116169917A - Model predictive control algorithm for low common-mode voltage without weight coefficient - Google Patents

Model predictive control algorithm for low common-mode voltage without weight coefficient Download PDF

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CN116169917A
CN116169917A CN202310459719.0A CN202310459719A CN116169917A CN 116169917 A CN116169917 A CN 116169917A CN 202310459719 A CN202310459719 A CN 202310459719A CN 116169917 A CN116169917 A CN 116169917A
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mode voltage
current
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张树林
张正松
宋玉明
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Hope Senlan Science & Technology Corp ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H02M1/123—Suppression of common mode voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/483—Converters with outputs that each can have more than two voltages levels
    • H02M7/487—Neutral point clamped inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53875—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output
    • H02M7/53876—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with analogue control of three-phase output based on synthesising a desired voltage vector via the selection of appropriate fundamental voltage vectors, and corresponding dwelling times
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H02P21/0017—Model reference adaptation, e.g. MRAS or MRAC, useful for control or parameter estimation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

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Abstract

The invention discloses a model predictive control algorithm of a common-mode voltage with low weight coefficient, which aims at a three-level diode clamping type inverter and a permanent magnet synchronous motor driving system. Compared with the traditional algorithm, the algorithm solves the problem that large leakage current, large common-mode current and large shaft current can be generated due to overlarge common-mode voltage, ensures the neutral-point potential balance, omits weight coefficient adjustment, and has lower output current harmonic content and smaller torque pulsation.

Description

Model predictive control algorithm for low common-mode voltage without weight coefficient
Technical Field
The invention relates to the field of control of permanent magnet three-level driving systems, in particular to a model predictive control algorithm for a low common-mode voltage without a weight coefficient.
Background
The permanent magnet synchronous motor is widely applied to a transmission device due to the advantages of simple structure, small noise, large starting moment, high system efficiency and the like; when the voltage level of the two-level inverter of the traditional permanent magnet synchronous motor is higher, the defects of high output harmonic content, higher dv/dt, higher switch stress caused by di/dt and the like exist; the cascade topology structure can expand the voltage class and the power class range, but has the defects of more switching tubes, low reliability, higher cost and the like; the three-level inverter of the permanent magnet synchronous motor is widely applied to medium and high voltage occasions due to the advantages of fewer required switches, lower output harmonic content, lower dv/dt and the like.
However, the three-level inverter of the permanent magnet synchronous motor has some defects due to the characteristics of the topological structure of the three-level inverter: firstly, in order to achieve optimal control performance of the motor, precisely controlling the output current of the motor is one of the problems that we have to solve; secondly, due to the existence of the distributed capacitance to the ground, a larger common-mode voltage is generated on the output side, electromagnetic interference is caused by the generated high-frequency leakage current, the surrounding electrical equipment cannot work normally when serious, the common-mode voltage also damages the electrical insulation of the motor, and the service life of the motor is shortened; third, the neutral point potential fluctuation is too large or even serious pull-out is caused when the inverter works, and further the output voltage is distorted, and the harmonic content of the output current is increased.
Disclosure of Invention
In order to solve the problems, the invention provides a model predictive control algorithm with low common mode voltage and no weight coefficient, which has lower output voltage harmonic wave and high-precision output current control compared with the traditional model predictive algorithm, realizes cooperative suppression of midpoint potential and common mode voltage, omits calculation of weight coefficient of an objective function, and has higher practicality.
The invention provides a model predictive control algorithm for a low common-mode voltage without a weight coefficient, which comprises the following steps:
s1, according to a three-level diode clamping inverter topological structure, 27 groups of switching states can be obtained, and 27 basic voltage vectors corresponding to the switching states under an alpha beta coordinate system can be obtained; 27 basic voltage vectors areV ooo 、V ppp 、V nnn 、V poo 、V onn 、V ppo 、V oon 、V opo 、V non 、V opp 、V noo 、V oop 、V nno 、V pop 、V ono 、V pnn 、V pon 、V ppn 、V opn 、V npn 、V npo 、V npp 、V nop 、V nnp 、V onp 、V pnp 、V pno 。
S2, discarding the common-mode voltage amplitude value to beU dc/2 and U dc 8 basic voltage vectors of/3 and the rest 19 basic voltage vectors form a low common-mode voltage vector control set under an alpha beta coordinate system;
wherein the basic voltage vector with the common-mode voltage amplitude of 0 isV ooo 、V pon 、V opn 、V npo 、V nop 、V onp 、V pno The method comprises the steps of carrying out a first treatment on the surface of the Common-mode voltage amplitude isU dc The basic voltage vector of/6 isV poo 、V oon 、V opo 、V noo 、V oop 、V ono 、V pnn 、V ppn 、V npn 、V npp 、V nnp 、V pnp The method comprises the steps of carrying out a first treatment on the surface of the Common-mode voltage amplitude isU dc The basic voltage vector of/3 isV onn 、V non 、V nno 、V ppo 、V opp 、V pop The method comprises the steps of carrying out a first treatment on the surface of the Common-mode voltage amplitude isU dc The basic voltage vector of/2 isV ppp 、V nnn The method comprises the steps of carrying out a first treatment on the surface of the Reject common mode voltage amplitude ofU dc/2 and U dc 8 basic voltage vectors of/3V ppp 、V nnn 、V onn 、V non 、V nno 、V ppo 、V opp 、V pop The remaining 19 basic voltage vectors areV ooo 、V pon 、V opn 、V npo 、V nop 、V onp 、V pno 、V poo 、V oon 、V opo 、V noo 、V oop 、V ono 、V pnn 、V ppn 、V npn 、V npp 、V nnp 、V pnp 。
S3, performing Park coordinate transformation on the low common-mode voltage vector set in the alpha beta coordinate system to obtain the low common-mode voltage vector set in the dq coordinate system; the Park coordinate transformation formula is:
Figure SMS_1
，
wherein ,
Figure SMS_2
、/>
Figure SMS_3
for the quantity in the alpha beta coordinate system,f d 、f q for the quantity in the dq coordinate system, +.>
Figure SMS_4
The electric angle of the rotor of the permanent magnet synchronous motor is obtained.
S4, establishing an output current prediction model, whereintSubstituting the moments into 19 basic voltage vectors in the low common-mode voltage vector set under the dq coordinate system respectively to calculate 19 basic voltage vectorstPredicted output current at +1;
the output current prediction formula is as follows:
Figure SMS_5
，
wherein ,I d (t+1)、I q (t+1) are respectivelytThe d-axis and q-axis output currents at +1 moment,I d (t)、I q (t) Respectively istThe current is output by the d axis and the q axis at the moment,U dGroup 、U qGroup the basic voltage vectors in the low common mode voltage vector set in the dq coordinate system,Rfor the resistance of the stator,T c the control period of the output current is indicated,L d for d-axis inductance、L qFor the q-axis inductance,
Figure SMS_6
for rotor angular velocity>
Figure SMS_7
Is a permanent magnet flux linkage.
S5, respectively taking 19 predicted output currents into an output current evaluation functionJ a Obtaining 19 different valuesJ a The 19 different values are comparedJ a Ordered from small to large and respectively namedJ a1 ~J a19 ；
Wherein the output current evaluation functionJ a The formula is:
Figure SMS_8
。
s6, selectingnIndividual output current evaluation functionsJ a1 ~J na （nNot less than 2), corresponding to itnSubstituting the values of the switch states of the basic voltage vectors into a neutral point potential prediction formula to obtainnPersonal (S)tPredicted value of midpoint potential at +1 timeU np (t+1)；
The neutral potential prediction model formula is as follows:
Figure SMS_9
，
wherein the method comprises the steps ofU np (t+1) istThe midpoint potential at the time +1,T S for the sampling period of the output current,Cis the capacitance value of the direct current bus capacitor,I a (t) Is thattThe current is output by the phase a at the moment,I b (t) Is thattThe current is output by the phase b at the moment,I c (t) Is thattOutputting current at the moment c;K x (t) Is thattTime of dayxThe value of the switching state of the phase bridge armx=a, b, c), whenxPhase switching tube K x1 And K x2 Conduction, K x3 And K x4 K at turn-off x When =1xPhase switching tube K x2 And K x3 Conduction, K x1 And K x4 K at turn-off x When=0, whenxPhase switching tube K x3 And K x4 Conduction, K x1 And K x2 K at turn-off x =-1； K x1 、K x2 、K x3 、K x4 Sequentially representx4 switching tubes from top to bottom.
S7, willnPersonal (S)tPredicted value of midpoint potential at +1 timeU np (t+1) carry-in midpoint potential evaluation functionJ b In (1) to obtainnOf different valuesJ b ；
Wherein the midpoint potential evaluation functionJ b The formula is:
Figure SMS_10
，/>
wherein,J b as a function of the evaluation of the midpoint potential,U np （t+1) istMidpoint potential at +1.
S8, selecting the minimum valueJ b The corresponding basic voltage vector is used as the optimal wave generating vector to act on the switching device.
Compared with the prior art, the invention has the following advantages: the method has the advantages of low output voltage harmonic wave and high-precision output current control, simultaneously realizes cooperative inhibition of midpoint potential and common-mode voltage, omits calculation of objective function weight coefficients, and has high practicality.
Drawings
FIG. 1 is a three-level topology diagram provided by the present invention;
FIG. 2 is a schematic diagram of 19 basic voltage vectors provided by the present invention in a two-phase stationary αβ coordinate system;
FIG. 3 is a flowchart illustrating the steps of an algorithm provided by the present invention;
FIG. 4 is a graph showing the experimental results of the algorithm of the present invention, wherein a is the output phase voltage of the present invention, b is the output current of the present invention, and c is the output common mode voltage of the present invention;
FIG. 5 is a graph showing the experimental results of a conventional algorithm, wherein a is the output current, and b is a graph of harmonic analysis of the output current;
fig. 6 is a graph showing experimental results of the algorithm of the present invention, wherein a is an output current, and b is a harmonic analysis graph of the output current.
Detailed Description
The present invention will be described in further detail with reference to the drawings and detailed description below to facilitate understanding of the present invention by those skilled in the art, and it should be noted that all the inventions which make use of the inventive concept are protected by the present invention insofar as the various changes are within the spirit and scope of the present invention as defined and defined by the appended claims without departing from the principle of the present invention.
The invention provides a model predictive control algorithm for a low common-mode voltage without a weight coefficient, which comprises the following steps:
step 1: fig. 1 is a three-level topology structure diagram provided by the present invention, fig. 3 is a flowchart showing algorithm steps provided by the present invention, and as shown in fig. 1 and fig. 3, a basic voltage vector is defined in a synchronous rotation αβ coordinate system as:
Figure SMS_11
，
wherein the method comprises the steps ofU bvv As a basic voltage vector of the power supply,U dc is a sampled value of the dc bus voltage,jin imaginary units, K a 、K b 、K c Is a switching function, defined as K x =p、o、n，x=a, b, c represent three-phase legs, K of a three-level NPC inverter, respectively x1 、K x2 、K x3 、K x4 Sequentially representx4 switching tubes from top to bottom of the phase bridge arm; let p represent K x State when=1, represented by o as K x State when=0, K is represented by n x State at = -1; p represents a switching tube K x1 And K x2 Conduction, K x3 And K x4 Turning off; o represents a switch tube K x2 And K x3 Conduction, K x1 And K x4 Turning off; n represents a switching tube K x3 And K x4 Conduction, K x1 And K x2 Turning off; any combination of three of p, o and n corresponds to a basic voltage vector, and is 3 3 The 27 combinations correspond to 27 basic voltage vectors.
Step 2: fig. 2 is a schematic diagram of 19 basic voltage vectors provided by the present invention in a two-phase stationary αβ coordinate system, fig. 3 is a flowchart of algorithm steps provided by the present invention, and as shown in fig. 2 and fig. 3, a common-mode voltage amplitude calculation formula is as follows:
Figure SMS_12
，
wherein the method comprises the steps ofU cmv For the common mode voltage amplitude,U dc k is the sampling value of the DC bus voltage a 、K b 、K c Is a switching function; wherein the basic voltage vector with the common-mode voltage amplitude of 0 isV ooo 、V pon 、V opn 、V npo 、V nop 、V onp 、V pno The method comprises the steps of carrying out a first treatment on the surface of the Amplitude of common-mode voltageIs thatU dc The basic voltage vector of/6 isV poo 、V oon 、V opo 、V noo 、V oop 、V ono 、V pnn 、V ppn 、V npn 、V npp 、V nnp 、V pnp The method comprises the steps of carrying out a first treatment on the surface of the Common-mode voltage amplitude isU dc The basic voltage vector of/3 isV onn 、V non 、V nno 、V ppo 、V opp 、V pop The method comprises the steps of carrying out a first treatment on the surface of the Common-mode voltage amplitude isU dc The basic voltage vector of/2 isV ppp 、V nnn The method comprises the steps of carrying out a first treatment on the surface of the Reject common mode voltage amplitude ofU dc 2 sumU dc 8 basic voltage vectors of/3V ppp 、V nnn 、V onn 、V non 、V nno 、V ppo 、V opp 、V pop The remaining 19 basic voltage vectors areV ooo 、V pon 、V opn 、V npo 、V nop 、V onp 、V pno 、V poo 、V oon 、V opo 、V noo 、V oop 、V ono 、V pnn 、V ppn 、V npn 、V npp 、V nnp 、V pnp 。
Step 3: performing Park coordinate transformation on the low common-mode voltage vector set in the alpha beta coordinate system to obtain the low common-mode voltage vector set in the dq coordinate system; the Park coordinate transformation formula is:
Figure SMS_13
，
wherein,
Figure SMS_14
、/>
Figure SMS_15
for the quantity in the alpha beta coordinate system,f d 、f q for the quantity in the dq coordinate system, +.>
Figure SMS_16
The electric angle of the rotor of the permanent magnet synchronous motor is obtained.
Step 4: FIG. 3 is a flowchart showing the steps of the algorithm provided by the present invention, wherein as shown in FIG. 3, a system prediction model formula is built, an output current prediction model is built, and then, the algorithm is implemented in the following stepstSubstituting the moments into 19 basic voltage vectors in the low common-mode voltage vector set under the dq coordinate system respectively to calculate 19 basic voltage vectorstPredicted output current at +1; the output current prediction formula is as follows:
Figure SMS_17
，
wherein,I d (t+1)、I q (t+1) are respectivelytThe d-axis and q-axis output currents at +1 moment,I d (t)、I q (t) Respectively istThe current is output by the d axis and the q axis at the moment,U dGroup 、U qGroup the basic voltage vectors in the low common mode voltage vector set in the dq coordinate system,Rfor the resistance of the stator,T c the control period of the output current is indicated,L d for d-axis inductance、L qFor the q-axis inductance,
Figure SMS_18
for rotor angular velocity>
Figure SMS_19
Is a permanent magnet flux linkage.
Step 5: FIG. 3 is a flowchart showing the algorithm steps provided by the present invention, wherein 19 predicted output currents are respectively brought into the output current evaluation function as shown in FIG. 3J a Obtaining 19 evaluation functions with different valuesJ a Output current evaluation functionJ a The formula is:
Figure SMS_20
，
wherein,J a in order to output the current evaluation function,I sator_dRef 、I sator_qRef for the purpose of outputting a current in the target,I d (t+1)、I q (t+1) istOutputting current at +1 moment; each value of the output current evaluation function corresponds to a basic voltage vector, and 19 evaluation functions are obtainedJ a Ordered from small to large and respectively namedJ a1 ~J a19 Selecting the value that minimizes the output current evaluation functionn（nGtoreq.2) output current evaluation functionsJ a1 ~J na 。
Step 6: FIG. 3 is a flowchart showing the algorithm steps provided by the present invention, wherein 19 evaluation functions are shown in FIG. 3J a Ordered from small to large and respectively namedJ a1 ~J a19 The method comprises the steps of carrying out a first treatment on the surface of the The output current evaluation functionJ a1 ~J a19 Corresponds to a basic voltage vector, and is selected so that the output current evaluation function value is minimizedn（nGtoreq.2) output current evaluation functionsJ a1 ~J na The method comprises the steps of carrying out a first treatment on the surface of the Corresponding it tonThe basic voltage vectors are put into a neutral point potential prediction formula to obtainnPersonal (S)tPredicted value of midpoint potential at +1 timeU np (t+1), the midpoint potential prediction formula is:
Figure SMS_21
，
wherein the method comprises the steps ofU np (t+1) istThe midpoint potential at the time +1,T S for the sampling period of the output current,Cis the capacitance value of the direct current bus capacitor,I a (t) Is thattThe current is output by the phase a at the moment,I b (t) Is thattThe current is output by the phase b at the moment,I c (t) Is thattOutputting current at the moment c;
wherein K is a 、K b 、K c As a function of the switch,x=a, b, c represent three-phase legs of a three-level NPC inverter, respectively; p represents a switching tube K x1 And K x2 Conduction, K x3 And K x4 Turning off; o represents a switch tube K x2 And K x3 Conduction, K x1 And K x4 Turning off; n represents a switching tube K x3 And K x4 Conduction, K x1 And K x2 Turning off;x=a, b, c represent three-phase legs, K of a three-level NPC inverter, respectively x1 、K x2 、K x3 、K x4 Sequentially representx4 switching tubes from top to bottom of the phase bridge arm; evaluating the output currentJ a1 ~J na Corresponding tonThe basic voltage vector is brought into a neutral point potential prediction model formula to obtainnPersonal (S)tPredicted value of midpoint potential at +1 timeU np (t+1)。
Step 7: FIG. 3 is a flowchart showing the steps of the algorithm according to the present invention, as shown in FIG. 3nPersonal (S)tPredicted value of midpoint potential at +1 timeU np (t+1) carry-in midpoint potential evaluation functionJ b In (1) to obtainnMidpoint potential evaluation function of different valuesJ b Neutral potential evaluation functionJ b The formula is:
Figure SMS_22
，
wherein,J b as a function of the evaluation of the midpoint potential, U np （t+1) istMidpoint potential at +1; will benPersonal (S)tPredicted value of midpoint potential at +1 timeU np (t+1) carry-in midpoint potential evaluation functionJ b In (1) to obtainnMidpoint potential evaluation function of different valuesJ b 。
Step 8: FIG. 3 is a flowchart showing the steps of the algorithm according to the present invention, as shown in FIG. 3Selecting the minimum midpoint potential evaluation functionJ b The corresponding basic voltage vector acts on the switching device as the final modulated wave vector.
FIG. 4 is a graph showing the experimental results of the algorithm of the present invention, wherein a is the output phase voltage of the present invention, b is the output current of the present invention, and c is the output common mode voltage of the present invention; as shown in FIG. 4, the low-common-mode voltage model predictive control algorithm without the weight coefficient provided by the invention can effectively inhibit common-mode voltage, so that the common-mode voltage amplitude is reduced, and meanwhile, the midpoint potential is kept stable.
FIG. 5 is a graph showing the experimental results of a conventional algorithm, wherein a is the output current, and b is a graph of harmonic analysis of the output current; FIG. 6 is a graph showing the experimental results of the algorithm of the present invention, wherein a is the output current, and b is the harmonic analysis of the output current; comparing fig. 5 and fig. 6, it can be found that the prediction control algorithm of the model with low common-mode voltage and no weight coefficient provided by the invention can effectively reduce current harmonic waves.
Although specific embodiments of the invention have been described in detail with reference to the accompanying drawings, it should not be construed as limiting the scope of protection of the present patent. Various modifications and variations which may be made by those skilled in the art without the creative effort are within the scope of the patent described in the claims.

Claims (1)

1. The model predictive control algorithm for the low common-mode voltage without the weight coefficient is characterized by comprising the following steps of:
s1, according to a three-level diode clamping inverter topological structure, 27 groups of switching states can be obtained, and 27 basic voltage vectors corresponding to the switching states under an alpha beta coordinate system can be obtained;
wherein 27 basic voltage vectors areV ooo 、V ppp 、V nnn 、V poo 、V onn 、V ppo 、V oon 、V opo 、V non 、V opp 、V noo 、V oop 、V nno 、V pop 、V ono 、V pnn 、V pon 、V ppn 、V opn 、V npn 、V npo 、V npp 、V nop 、V nnp 、V onp 、V pnp 、V pno ；
S2, discarding the common-mode voltage amplitude value to beU dc 2 sumU dc 8 basic voltage vectors of/3 and the rest 19 basic voltage vectors form a low common-mode voltage vector control set under an alpha beta coordinate system;
wherein the basic voltage vector with the common-mode voltage amplitude of 0 isV ooo 、V pon 、V opn 、V npo 、V nop 、V onp 、V pno The method comprises the steps of carrying out a first treatment on the surface of the Common-mode voltage amplitude isU dc The basic voltage vector of/6 isV poo 、V oon 、V opo 、V noo 、V oop 、V ono 、V pnn 、V ppn 、V npn 、V npp 、V nnp 、V pnp The method comprises the steps of carrying out a first treatment on the surface of the Common-mode voltage amplitude isU dc The basic voltage vector of/3 isV onn 、V non 、V nno 、V ppo 、V opp 、V pop The method comprises the steps of carrying out a first treatment on the surface of the Common-mode voltage amplitude isU dc The basic voltage vector of/2 isV ppp 、V nnn The method comprises the steps of carrying out a first treatment on the surface of the Reject common mode voltage amplitude ofU dc 2 sumU dc 8 basic voltage vectors of/3V ppp 、V nnn 、V onn 、V non 、V nno 、V ppo 、V opp 、V pop The remaining 19 basic voltage vectors areV ooo 、V pon 、V opn 、V npo 、V nop 、V onp 、V pno 、V poo 、V oon 、V opo 、V noo 、V oop 、V ono 、V pnn 、V ppn 、V npn 、V npp 、V nnp 、V pnp ；
S3, performing Park coordinate transformation on the low common-mode voltage vector set in the alpha beta coordinate system to obtain the low common-mode voltage vector set in the dq coordinate system; the Park coordinate transformation formula is:
Figure QLYQS_1
，
wherein,
Figure QLYQS_2
、/>
Figure QLYQS_3
for the quantity in the alpha beta coordinate system,f d 、f q for the quantity in the dq coordinate system, +.>
Figure QLYQS_4
The electric angle of the rotor of the permanent magnet synchronous motor is set;
s4, establishing an output current prediction model, whereintSubstituting 19 basic voltage vectors in the low common-mode voltage vector set under the dq coordinate system at the moment respectively to calculate 19tPredicted output current at +1;
the output current prediction formula is as follows:
Figure QLYQS_5
，
wherein,I d (t+1)、I q (t+1) are respectivelytThe d-axis and q-axis output currents at +1 moment,I d (t)、I q (t) Respectively istThe current is output by the d axis and the q axis at the moment,U dGroup 、U qGroup respectively low in dq coordinate systemThe fundamental voltage vector in the common mode voltage vector set,Rfor the resistance of the stator,T c the control period of the output current is indicated,L d for d-axis inductance、L q For the q-axis inductance,
Figure QLYQS_6
for rotor angular velocity>
Figure QLYQS_7
Is a permanent magnet flux linkage;
s5, respectively taking 19 predicted output currents into an output current evaluation functionJ a Obtaining 19 different valuesJ a The 19 different values are comparedJ a Ordered from small to large and respectively namedJ a1 ~J a19 ；
Wherein the output current evaluation functionJ a The formula is:
Figure QLYQS_8
，
wherein,J a in order to output the current evaluation function,I sator_dRef 、I sator_qRef for the purpose of outputting a current in the target,I d (t+1)、I q (t+1) istD-axis and q-axis output currents at +1 moment;
s6, selectingnIndividual output current evaluation functionsJ a1 ~J na （nNot less than 2), corresponding to itnSubstituting the values of the switch states of the basic voltage vectors into a neutral point potential prediction formula to obtainnPersonal (S)tPredicted value of midpoint potential at +1 timeU np (t+1)；
The neutral potential prediction model formula is as follows:
Figure QLYQS_9
，
wherein the method comprises the steps ofU np (t+1) istThe midpoint potential at the time +1,T S for the sampling period of the output current,Cis the capacitance value of the direct current bus capacitor,I a (t) Is thattThe current is output by the phase a at the moment,I b (t) Is thattThe current is output by the phase b at the moment,I c (t) Is thattOutputting current at the moment c;K x (t) Is thattTime of dayxThe value of the switching state of the phase bridge armx=a, b, c), whenxPhase switching tube K x1 And K x2 Conduction, K x3 And K x4 K at turn-off x When =1xPhase switching tube K x2 And K x3 Conduction, K x1 And K x4 K at turn-off x When=0, whenxPhase switching tube K x3 And K x4 Conduction, K x1 And K x2 K at turn-off x =-1；K x1 、K x2 、K x3 、K x4 Sequentially representx4 switching tubes from top to bottom of the phase bridge arm;
s7, willnPersonal (S)tPredicted value of midpoint potential at +1 timeU np (t+1) carry-in midpoint potential evaluation functionJ b In (1) to obtainnOf different valuesJ b ；
Wherein the midpoint potential evaluation functionJ b The formula is:
Figure QLYQS_10
，
wherein,J b as a function of the evaluation of the midpoint potential,U np （t+1) istMidpoint potential at +1;
s8, selecting the minimum valueJ b The corresponding basic voltage vector is used as the optimal wave generating vector to act on the switching device.
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