CN115411964A - Marine microgrid inverter, modulation strategy and control method - Google Patents

Abstract

The invention discloses a marine microgrid inverter, a modulation strategy and a control method. The inverter comprises a direct current supporting capacitor, an A-phase bridge type inversion unit, a B-phase bridge type inversion unit, a C-phase bridge type inversion unit, a three-winding transformer and a three-phase alternating current filter, wherein direct current voltage input from the outside is applied to two ends of the direct current supporting capacitor to form a direct current bus, the three bridge type inversion units have the same circuit topology, the input direct current voltage is converted into alternating current PWM voltage, and the alternating current PWM voltage is connected in series and superposed through a transformer winding. The utilization rate of the direct-current voltage of the inverter and the quality of the output voltage are improved through the matching of transformer windings, and the power density is improved through the magnetic integration technology of the transformer; the modulation strategy of the invention restrains common-mode voltage and primary side circulation of the transformer while realizing equivalent frequency multiplication of switching frequency, reduces vibration noise of the inverter and improves power density; the control method can improve the quality of the output voltage of the inverter with the nonlinear load, realize the effective inhibition of the primary side current harmonic of the transformer and reduce the vibration energy level of the inverter.

Description

A marine microgrid inverter, modulation strategy and control method

technical field

The invention belongs to the technical field of inverter power supplies, and in particular relates to a marine microgrid inverter, a modulation strategy and a control method.

Background technique

At present, known microgrid inverters generally adopt a three-phase three-leg inverter topology, which has the advantages of simple circuit topology, small number of components, low cost, and mature control strategies. However, special marine microgrid inverters have higher requirements on power density, DC voltage utilization, output voltage quality, and vibration and noise indicators than general microgrid inverters. The known circulating current suppression measures mainly include methods such as connecting the circulating current limiting inductor in series and changing the carrier phase of the inverter unit. Among them, the series connection of circulating current filter inductors inevitably increases the size and cost of the inverter; although changing the carrier phase of the inverter unit does not increase the hardware cost, the carrier phase determines the multiple harmonic superposition characteristics, while suppressing the circulating current. It may bring new problems such as output voltage waveform quality and electromagnetic compatibility index decline.

At the same time, the three-phase inverter generally adopts the voltage-current double closed-loop control strategy in the synchronous rotating coordinate system, and its regulator generally uses a PI regulator, which can realize the static-free tracking of the fundamental voltage command. However, the PI regulator has limited ability to suppress harmonics, so the quality of the inverter output voltage is significantly reduced when it is loaded with a nonlinear load. In order to improve the harmonic suppression capability of the inverter, the closed-loop control bandwidth can be increased, but the system stability margin may be reduced, resulting in an increase in voltage overshoot or even instability during the load dynamic process. In addition, the multi-superposition topology based on the transformer is adopted, and the harmonics of the primary current of the transformer are relatively abundant, which will inevitably increase the vibration level of the transformer, so it is necessary to take effective suppression measures for the primary current harmonics.

Contents of the invention

Aiming at the above-mentioned technical defects, the present invention proposes a marine microgrid inverter, a modulation strategy and a control method that can improve the inverter's power density, DC voltage utilization rate, and output voltage quality while reducing vibration and noise levels.

In order to achieve the above purpose, the present invention proposes a marine microgrid inverter, including a DC support capacitor 1, an A-phase bridge inverter unit 2, a B-phase bridge inverter unit 3, and a C-phase bridge inverter unit 4 , a three-winding transformer 5 and a three-phase AC filter 6, wherein the external input DC voltage is applied to both ends of the DC support capacitor 1 to form a DC bus; the three bridge inverter units have the same circuit topology, each bridge inverter The variable units are composed of four inverter bridge arms connected in parallel to the DC bus, each inverter bridge arm is composed of two series switching devices, and two sets of inverter bridge arms form an H-bridge inverter unit.

Further, the A-phase bridge inverter unit 2 includes a first switching device 20 connected in series with a second switching device 21 to form a first inverter bridge arm, and a third switching device 22 connected in series with a fourth switching device 23 to form a second inverter bridge arm. Transforming bridge arms, these two inverter bridge arms form an H-bridge inverter unit connected in parallel with the DC bus; the fifth switching device 24 and the sixth switching device 25 are connected in series, and the seventh switching device 26 and the eighth switching device 27 are connected in series Another H-bridge inverter unit formed is connected in parallel with the DC bus; one of the two H-bridge inverter units is connected to the first winding 50 of the primary side of the A phase, and the other is connected to the second winding 51 of the primary side of the A phase; The PWM voltage output by the H-bridge inverter unit is superimposed in series through the transformer winding, and the output of the secondary side winding 52 of the A phase obtains the doubled PWM voltage; the doubled PWM voltage is input to the three-phase AC filter of the inverter The device 6 obtains the power frequency output voltage after filtering.

Also provided is a modulation strategy for the above-mentioned marine microgrid inverter, the modulation strategy is: the triangle carrier wave of the first inverter bridge arm and the triangle carrier wave of the second inverter bridge arm of each phase bridge inverter unit, The phase difference between the triangular carrier wave of the third inverter bridge arm and the triangular carrier wave of the fourth inverter bridge arm is 180°, the triangular carrier wave of the first inverter bridge arm of each phase bridge inverter unit and the triangular carrier wave of the third inverter bridge arm , The phase difference between the triangular carrier wave of the second inverter bridge arm and the triangular carrier wave of the fourth inverter bridge arm is 90°; taking the phase of the triangular carrier wave of the first inverter bridge arm of the A-phase bridge inverter unit as a reference, the B-phase bridge The phase difference between the triangular carriers of each inverter bridge arm of the type inverter unit and the triangular carriers of each inverter bridge arm corresponding to the A-phase bridge inverter unit is α, and the phase difference of each inverter bridge arm triangle carrier of the C-phase bridge inverter unit The phase difference between the carrier wave and the triangular carrier wave of each inverter bridge arm corresponding to the A-phase bridge inverter unit is β.

Further, said α=180°, β=180°.

q轴正序基波电压标幺值

反向同步旋转坐标系下d轴负序基波电压标幺值

及q轴正序基波电压标幺值

与

作为正序电压控制器82的反馈值参与逆变器输出正序电压控制，

与

作为负序电压控制器83的反馈值参与逆变器输出负序电压控制；正序电压控制器82的d轴标幺值电压参考值为1，减去反馈值

后输入至正序电压控制器82的d轴正序电压PI调节器821，正序电压控制器82的q轴标幺值电压参考值为0，减去反馈值

后输入至正序电压控制器82的q轴正序电压PI调节器822；It also provides a control method for a marine microgrid inverter as described above, the sampled three-phase voltages u _oa (t), u _ob (t) and u _oc (t) are passed through the positive sequence and negative sequence bases of DSC Wave voltage decomposition and dq transformation to obtain the per unit value of d-axis positive sequence fundamental wave voltage in the forward synchronous rotating coordinate system

q-axis positive sequence fundamental wave voltage per unit value

d-axis negative-sequence fundamental wave voltage per unit value in the reverse synchronous rotating coordinate system

and q-axis positive sequence fundamental wave voltage per unit value

and

As the feedback value of the positive sequence voltage controller 82, it participates in the inverter output positive sequence voltage control,

and

As the feedback value of the negative-sequence voltage controller 83, it participates in the control of the inverter output negative-sequence voltage; the d-axis per unit voltage reference value of the positive-sequence voltage controller 82 is 1, minus the feedback value

Then input to the d-axis positive-sequence voltage PI regulator 821 of the positive-sequence voltage controller 82, the q-axis per unit voltage reference value of the positive-sequence voltage controller 82 is 0, minus the feedback value

Then input to the q-axis positive-sequence voltage PI regulator 822 of the positive-sequence voltage controller 82;

后输入至负序电压控制器83的d轴负序电压PI调节器831，d轴负序电压PI调节器831的输出与

乘以交叉解耦项ωC_f相减后输出到负序dq到正序dq变换模块86的d轴输入端口；负序电压控制器83的q轴标幺值电压参考值为0，减去反馈值

后输入至负序电压控制器83的q轴负序电压PI调节器832，q轴负序电压PI调节器832的输出与

乘以交叉解耦项ωC_f相减后输出到负序dq到正序dq变换模块85的q轴输入端口；The d-axis per unit voltage reference value of the negative sequence voltage controller 83 is 0, minus the feedback value

Then input to the d-axis negative sequence voltage PI regulator 831 of the negative sequence voltage controller 83, the output of the d-axis negative sequence voltage PI regulator 831 and

Multiplied by the cross-decoupling term ωC _f and subtracted, output to the d-axis input port of the negative-sequence dq to positive-sequence dq transformation module 86; the q-axis per-unit voltage reference value of the negative-sequence voltage controller 83 is 0, minus the feedback value

Then input to the q-axis negative-sequence voltage PI regulator 832 of the negative-sequence voltage controller 83, the output of the q-axis negative-sequence voltage PI regulator 832 and

Multiplied by the cross-decoupling term _ωCf and subtracted, output to the q-axis input port of the negative sequence dq to positive sequence dq transformation module 85;

与q轴电流参考值

Negative sequence dq to positive sequence dq transformation module 85 transforms the dq component of the negative sequence dq coordinate system rotating at -ω _o speed into the positive sequence dq coordinate system rotating at ω _o speed through -2ω _o , and the transformed dq axis components Superimposed with the dq axis component output of the positive sequence voltage controller 82 to obtain the d axis current reference value of the current controller 84

with the q-axis current reference value

加上d轴输出电流

前馈减去d轴电感电流反馈

输入到在PI调节器的基础上增加了谐振控制环节构成的d轴电流PI+R调节器843；q轴电流

交叉解耦项ωL_m以及输出电压

前馈项，再与变压器d轴原边零序电流的三次谐波比例谐振PR控制器841的输出相叠加得到调制波d轴分量u^d；

加上q轴输出电流

前馈减去q轴电感电流反馈

输入到q轴电流PI+R调节器844；d轴电流

交叉解耦项ωL_m以及输出电压

前馈项，再与变压器q轴原边零序电流的三次谐波比例谐振PR控制器842的输出相叠加得到调制波q轴分量u^q；

plus the d-axis output current

Feedforward minus d-axis inductor current feedback

Input to the d-axis current PI+R regulator 843 formed by adding a resonance control link on the basis of the PI regulator; the q-axis current

The cross-decoupling term ωL _m and the output voltage

The feedforward term is superimposed with the output of the third harmonic proportional resonance PR controller 841 of the zero-sequence current of the d-axis primary side of the transformer to obtain the d-axis component u ^d of the modulated wave;

plus the q-axis output current

Feedforward minus q-axis inductor current feedback

Input to q-axis current PI+R regulator 844; d-axis current

The cross-decoupling term ωL _m and the output voltage

The feed-forward term is superimposed with the output of the third harmonic proportional resonance PR controller 842 of the transformer q-axis primary zero-sequence current to obtain the modulated wave q-axis component u ^q ;

The d-axis component u ^d and the q-axis component u ^q of the modulation wave are converted to dq inversely to obtain the ABC three-phase modulation wave voltages u _A , u _B and u _C , and then the driving pulses of the corresponding switching devices are obtained through SPWM modulation.

Compared with the prior art, the beneficial effect of the present invention is that the multiple combined inverter topology realizes the improvement of the DC voltage utilization rate of the inverter and the quality of the output voltage through the cooperation of the transformer windings, and improves the efficiency of the inverter through the magnetic integration technology of the transformer. Power density; the modulation strategy of the present invention, while realizing the equivalent frequency doubling of the switching frequency, suppresses the common-mode voltage and the primary side circulation of the transformer, reduces the vibration noise of the inverter, and improves the power density; the control method of the present invention can improve The quality of the output voltage of the inverter with a nonlinear load, and the effective suppression of the current harmonics of the primary side of the transformer, and the reduction of the vibration level of the inverter.

Description of drawings

Fig. 1 is the topological structure diagram of the marine microgrid inverter of the present invention;

Figure 2 is a waveform diagram of the H-bridge inverter unit power frequency modulation wave and two triangular carrier waves;

Fig. 3 is the H-bridge inverter unit output PWM voltage waveform diagram;

Fig. 4 is a bridge inverter unit power frequency modulation wave and four triangular carrier waves;

Fig. 5 is a PWM voltage waveform diagram output by a bridge inverter unit;

Figure 6 shows the influence of the phase-to-phase carrier shift angle on the output common-mode voltage amplitude of the inverter;

Figure 7 shows the influence of phase-to-phase carrier phase shift angle on the effective value of common-mode voltage output by the inverter;

Fig. 8 is the influence of phase-to-phase carrier shift phase angle on the effective value of A-phase circulating current;

Fig. 9 is the influence of phase-to-phase carrier shift phase angle on the effective value of B-phase circulating current;

Fig. 10 is the influence of phase-to-phase carrier shift phase angle on the effective value of C-phase circulating current;

Fig. 11 is a block diagram of the inverter control system of the present invention.

In the figure 1. DC support capacitor, 2. A-phase bridge inverter unit, 3. B-phase bridge inverter unit, 4. C-phase bridge inverter unit, 5. Multi-winding transformer, 6. Three-phase AC filter 20. A-phase bridge inverter unit first switching device, 21. A-phase bridge inverter unit second switching device, 22. A-phase bridge inverter unit third switching device, 23. A-phase bridge inverter unit The fourth switching device of the inverter unit, 24. the fifth switching device of the A-phase bridge inverter unit, 25. the sixth switching device of the A-phase bridge inverter unit, 26. the seventh switching device of the A-phase bridge inverter unit, 27. The eighth switching device of the A-phase bridge inverter unit, 50. The first winding on the primary side of the A-phase, 51. the second winding on the primary side of the A-phase, 52. the secondary-side winding of the A-phase, 53. the first winding on the primary side of the B-phase First winding, 54.B phase primary side second winding, 55.B phase secondary side winding, 56.C phase primary side first winding, 57.C phase primary side second winding, 58.C phase secondary side winding , 70. Power frequency modulation wave, 71. Triangle carrier wave of the first inverter bridge arm, 72. Triangle carrier wave of the third inverter bridge arm, 73. Triangle carrier wave of the second inverter bridge arm, 74. Triangle carrier wave of the fourth inverter bridge arm Carrier, 75. Transformer primary winding PWM voltage, 76. Transformer secondary winding PWM voltage, 82. Positive sequence voltage controller, 83. Negative sequence voltage controller, 84. Current controller, 85. Negative sequence dq to positive Sequence dq conversion, 821.d-axis positive sequence voltage PI regulator, 822.q-axis positive-sequence voltage PI regulator 822, 831.d-axis negative-sequence voltage PI regulator, 832.q-axis negative-sequence voltage PI regulator, 841 .Third harmonic proportional resonant PR controller of transformer d-axis primary side zero-sequence current, 842. Transformer q-axis primary side zero-sequence current third harmonic proportional resonant PR controller, 843.d-axis current PI+R regulator, 844. Q-axis current PI+R regulator.

Detailed ways

In order to make the object, technical solution and advantages of the present invention clearer, the present invention will be further described in detail below in conjunction with the accompanying drawings.

By analyzing the influence of the three-phase carrier phase shift angle of the inverter unit on the output common-mode voltage and circulating current, the present invention obtains a combination of carrier phase-shifting angles that take into account both common-mode voltage and circulating current suppression, which is easier to implement and can reduce circulating current filter inductance and EMI The cost of the filter will be further described below in conjunction with the accompanying drawings and specific implementation methods.

Fig. 1 is a topological structure diagram of a marine microgrid inverter of the present invention, including a DC support capacitor 1, an A-phase bridge inverter unit 2, a B-phase bridge inverter unit 3, a C-phase bridge inverter unit 4, three A winding transformer 5 and a three-phase AC filter 6, wherein an externally input DC voltage is applied to both ends of the DC support capacitor 1 to form a DC bus. The three bridge inverter units have the same circuit topology. Each bridge inverter unit is composed of four inverter bridge arms connected in parallel to the DC bus. Each inverter bridge arm is composed of two series switching devices. Two sets of inverter bridge arms constitute an H-bridge inverter unit. Taking the A-phase bridge inverter unit 2 as an example: the first switching device 20 and the second switching device 21 are connected in series to form the first inverter bridge arm, and the third switching device 22 and the fourth switching device 23 are connected in series to form the second inverter bridge. The two inverter bridge arms form an H-bridge inverter unit and are connected in parallel with the DC bus; similarly, the fifth switching device 24 and the sixth switching device 25 are connected in series, and the seventh switching device 26 and the eighth switching device 27 are connected in series. Another H-bridge inverter unit formed later is connected in parallel with the DC bus; one of the two H-bridge inverter units is connected to the first winding 50 of the primary side of the A phase, and the other is connected to the second winding 51 of the primary side of the A phase; The PWM voltage output by the two H-bridge inverter units is superimposed in series through the transformer winding, and the output of the secondary side winding 52 of the A phase obtains the doubled PWM voltage; the doubled PWM voltage is input to the three-phase inverter The AC filter 6 obtains the power frequency output voltage after filtering.

In order to convert the input DC voltage into AC PWM voltage, the inverter topology shown in Figure 1 can adopt a carrier phase-shift modulation strategy based on unipolar frequency multiplication modulation: the first inverter bridge of each phase bridge inverter unit The phase difference between the arm triangular carrier wave and the second inverter bridge arm triangular carrier wave, the third inverter bridge arm triangular carrier wave and the fourth inverter bridge arm triangular carrier wave are all 180°, the first inverter of each phase bridge inverter unit The phase difference between the triangular carrier wave of the bridge arm and the triangular carrier wave of the third inverter bridge arm, the triangular carrier wave of the second inverter bridge arm and the triangular carrier wave of the fourth inverter bridge arm is 90°; The phase of the triangular carrier wave of the inverter bridge arm is the reference, and the phase difference between the triangular carrier waves of each inverter bridge arm of the B-phase bridge inverter unit and the corresponding phase difference of each inverter bridge arm triangular carrier wave of the A-phase bridge inverter unit is α, C The phase difference between the triangular carriers of the inverter bridge arms of the phase bridge inverter unit and the triangular carriers of the inverter bridge arms corresponding to the A-phase bridge inverter unit is β, and α=180°, β=180°.

It is precisely because of the existence of the phase shift angle of the triangular carrier wave that the high-frequency harmonics near the frequency of the triangular carrier wave and its multiples are coupled through the windings of the shared transformer core, which will generate high-frequency circulating currents. In order to suppress the high-frequency circulating current between the inverter units, the present invention utilizes the control degree of freedom of the three-phase carrier phase shift angle of the inverter, analyzes the influence of the interphase carrier phase shift angle on the circulating current, and finally takes into account the inverter AC output common mode Voltage suppression, the present invention provides the optimized phase angles of A-phase, B-phase, and C-phase carriers. Using the combination of carrier phase angles can not only suppress the circulation between inverter units, but also reduce the output common-mode voltage and improve electromagnetic compatibility performance .

In this embodiment, it is assumed that the triangular carrier phase of the first inverter bridge arm of the A-phase bridge inverter unit is 0°, then the triangular carrier phase of the second inverter bridge arm is -180°, and the triangular carrier phase of the third inverter bridge arm is -90°, the triangular carrier phase of the fourth inverter bridge arm is -270°, the triangular carrier phase of the first inverter bridge arm of the B-phase bridge inverter unit is α, and the carrier phase of the second bridge arm is α-180°. The carrier phase of the third bridge arm is α-90°, the carrier phase of the fourth bridge arm is α-270°, the carrier phase of the first inverter bridge arm of the C-phase bridge inverter unit is β, and the carrier phase of the second bridge arm is β -180°, the carrier phase of the third bridge arm is β-90°, and the carrier phase of the fourth bridge arm is β-270°;

The angle between α and β can be adjusted according to specific requirements, and it does not affect the output power frequency voltage of the inverter and the equivalent quadruple frequency and five-level effects, but different carrier phases will affect the circulation and AC between three-phase inverter units To output common mode voltage, the specific implementation method of carrier phase selection among three-phase inverter units is given below:

Note that the voltage of the first winding 50 on the primary side of phase A is V ₁ , and the current is I ₁ , the voltage of the second winding 51 on the primary side of phase A is V ₂ , and the current is I ₂ , the voltage of the secondary winding 52 of phase A is V ₃ , and the current is is I ₃ ; the voltage of the first winding 53 on the primary side of the B-phase is V ₄ , and the current is I ₄ , the voltage of the second winding 54 on the primary side of the B-phase is V ₅ , and the current is I ₅ , and the voltage of the secondary winding 55 of the B-phase is V _6. The current is I ₆ ; the voltage of the first winding 56 on the C-phase primary side is V ₇ and the current is I ₇ , the voltage of the second winding 57 on the C-phase primary side is V ₈ , and the current is I ₈ , and the C-phase secondary side winding 58 The voltage is V ₉ , and the current is I ₉ . According to the transformer port voltage and current equation can be obtained:

Where, V＝[V ₁ V ₂ V ₃ V ₄ V ₅ V ₆ V ₇ V ₈ V ₉ ] ^T , I＝[I ₁ I ₂ I ₃ I ₄ I ₅ I ₆ I ₇ I ₈ I ₉ ] ^T ,

The voltage and current coupling relationship at the transformer port is complicated. Among them, the resistance value R _i (i=1, 2, 3, 4, 5, 6, 7, 8, 9) in the resistance matrix R is the i-th winding resistance of the transformer; the inductance matrix L is the transformer model established based on the coupled inductance There are 81 inductance parameters, among which the inductance L _ii (i=1,2,3,4,5,6,7,8,9) on the diagonal of the inductance matrix is the self-inductance of the 9 windings of the transformer, and the inductance L _ij =L _ji (i=1, 2, 3, 4, 5, 6, 7, 8, 9) is the mutual inductance between winding i and winding j, with a total of 72 inductance values. Due to the large number of circuit parameters, it is difficult to intuitively use calculation formulas to express the relationship between the common-mode voltage of the inverter's AC output, the circulating current between the units, and the carrier phase shift angle between the power units.

Define the inverter AC output common-mode voltage V _cm :

V _cm = (V ₃ +V ₆ +V ₉ )/3 (2)

Taking the A-phase carrier as the reference, Fig. 4 shows the change rule of the inverter output common-mode voltage amplitude with the carrier phase shift angle between B-A phase and C-A phase carrier; Fig. 5 shows the inverter output common-mode voltage The change law of the voltage effective value with the carrier phase shift angle between B-A phase and the carrier phase shift angle between C-A phase. According to the principle of volt-second balance, changing the phase of the carrier wave will not affect the change of the amplitude of the output power frequency voltage. Under the combination of (0, -120°, 120°) and (0, 120°, -120°) of the three-phase carrier phase, both the common-mode voltage amplitude and the effective value are the smallest.

Define the circulating current between inverter units of phase A as I _{cir_A} , the circulating current between inverter units of phase B as I _{cir_B} , and the circulating current between inverter units of phase C as I _{cir_C} :

Fig. 6, Fig. 7 and Fig. 8 show the change law of the effective value of the circulating current of the A-phase, B-phase and C-phase inverter units with the phase shift angle of the B-A phase carrier and the phase shift angle of the C-A phase carrier. Also take A-phase carrier phase as the reference. Through the simulation traversal of phase B and C phase carrier phase angles, the effective value of the three-phase circulating current under different carrier phase combinations can be obtained. When the phase shift angles of phase A, phase B and phase C are (0°, 0°, 0°), (0°, 0°, 180°), (0°, 180°, 0°) and (0° , 180°, 180°) the circulation of each phase is the smallest when these four groups are combined.

Therefore, combining the analysis results of the AC output common-mode voltage and the circulating current between the inverter units, the phase-shift angles for the minimum common-mode voltage and the minimum circulating current are different, and a compromise should be made between the two. The priority is to ensure the suppression of circulating current, and select (0, 180°, 180°) as the optimized carrier phase angle, that is, α=180°, β=180°.

Since the three-phase bridge inverter unit of the inverter has three power frequency modulation waves, namely, the power frequency modulation wave of the A-phase bridge inverter unit, the power frequency modulation wave of the B-phase bridge inverter unit, and the C-phase bridge inverter unit. The power frequency modulation wave of the inverter unit, and the difference between the three-phase power frequency modulation waves is 120°; therefore, the power frequency modulation wave of the A-phase bridge inverter unit is used as the four inverters of the A-phase bridge inverter unit For the power frequency modulation wave shared by the bridge arms, the power frequency modulation wave of the B-phase bridge inverter unit is used as the power frequency modulation wave shared by the four inverter bridge arms of the B-phase bridge inverter unit, and the C-phase bridge The power frequency modulation wave of the inverter unit is used as the power frequency modulation wave shared by the four inverter bridge arms of the C-phase bridge inverter unit.

Combined with the modulation strategy of an H-bridge inverter unit shown in Figures 2 and 3, when the power frequency modulation wave 70 is greater than the triangular carrier wave 71 of the first inverter bridge arm, the upper switch tube of the first inverter bridge arm is driven to conduct and the lower switch The tube is turned off; the second inverter bridge arm triangle carrier wave 73 can be obtained by inverting the phase of the first inverter bridge arm triangular carrier wave 71, and the same power frequency modulation wave 70 is used for comparison. When the power frequency modulation wave 70 is greater than the second inverter When the bridge arm triangular carrier wave 73, the upper switching tube of the second inverter bridge arm is driven to be turned on and the lower switching tube is turned off; because the phase difference between the first inverter bridge arm triangular carrier wave 71 and the second inverter bridge arm triangular carrier wave 73 is 180 °, after adopting this modulation strategy, the polarity of the transformer primary winding PWM voltage 75 is the same as that of the power frequency modulation wave, and its equivalent output harmonic frequency is equal to twice the carrier frequency.

Combining the carrier signals corresponding to the four inverter bridge arms in a bridge inverter unit shown in Figures 4 and 5, the triangular carrier wave 72 of the third inverter bridge arm and the triangular carrier wave 74 of the fourth inverter bridge arm lag behind the first inverter bridge arm respectively. The variable bridge arm triangular carrier 71 and the second inverter bridge arm triangular carrier 73 are each 90°. The PWM output voltages of the two H-bridge inverter units are superimposed through the transformer winding, and the winding voltage 76 on the secondary side of the transformer presents a five-level PWM characteristic of four times the carrier frequency. Therefore, the waveform of the PWM voltage 76 of the secondary winding of the transformer superimposed by the transformer presents four times the carrier frequency and five-level characteristics, which improves the power density of the inverter and reduces the vibration level within the test range of the inverter.

The present invention explains its ability to improve the output voltage waveform quality and suppress harmonics by analyzing the structure of the inverter control system. Further description will be made below in conjunction with the accompanying drawings and specific implementation methods.

q轴正序基波电压标幺值

反向同步旋转坐标系下d轴负序基波电压标幺值

及q轴正序基波电压标幺值

与

作为正序电压控制器82的反馈值参与逆变器输出正序电压控制，

与

作为负序电压控制器83的反馈值参与逆变器输出负序电压控制。Fig. 9 shows the control algorithm block diagram of the present invention, the three-phase voltages u _oa (t), u _ob (t) and u _oc (t) obtained by sampling are decomposed by DSC positive sequence, negative sequence fundamental wave voltage and dq transformation to obtain positive d-axis positive sequence fundamental wave voltage per unit value in the synchronous rotating coordinate system

q-axis positive sequence fundamental wave voltage per unit value

d-axis negative-sequence fundamental wave voltage per unit value in the reverse synchronous rotating coordinate system

and q-axis positive sequence fundamental wave voltage per unit value

and

As the feedback value of the positive sequence voltage controller 82, it participates in the inverter output positive sequence voltage control,

and

As the feedback value of the negative-sequence voltage controller 83, it participates in the control of the inverter output negative-sequence voltage.

后输入至正序电压控制器82的d轴正序电压PI调节器821，正序电压控制器82的q轴标幺值电压参考值为0，减去反馈值

后输入至正序电压控制器82的q轴正序电压PI调节器822；The d-axis per unit voltage reference value of the positive sequence voltage controller 82 is 1, minus the feedback value

Then input to the d-axis positive-sequence voltage PI regulator 821 of the positive-sequence voltage controller 82, the q-axis per unit voltage reference value of the positive-sequence voltage controller 82 is 0, minus the feedback value

Then input to the q-axis positive-sequence voltage PI regulator 822 of the positive-sequence voltage controller 82;

后输入至负序电压控制器83的d轴负序电压PI调节器831，d轴负序电压PI调节器831的输出与

乘以交叉解耦项ωC_f相减后输出到负序dq到正序dq变换模块86的d轴输入端口；负序电压控制器83的q轴标幺值电压参考值为0，减去反馈值

后输入至负序电压控制器83的q轴负序电压PI调节器832，q轴负序电压PI调节器832的输出与

乘以交叉解耦项ωC_f相减后输出到负序dq到正序dq变换模块85的q轴输入端口；The d-axis per unit voltage reference value of the negative sequence voltage controller 83 is 0, minus the feedback value

Then input to the d-axis negative sequence voltage PI regulator 831 of the negative sequence voltage controller 83, the output of the d-axis negative sequence voltage PI regulator 831 and

Multiplied by the cross-decoupling term ωC _f and subtracted, output to the d-axis input port of the negative-sequence dq to positive-sequence dq transformation module 86; the q-axis per-unit voltage reference value of the negative-sequence voltage controller 83 is 0, minus the feedback value

Then input to the q-axis negative-sequence voltage PI regulator 832 of the negative-sequence voltage controller 83, the output of the q-axis negative-sequence voltage PI regulator 832 and

Multiplied by the cross-decoupling term _ωCf and subtracted, output to the q-axis input port of the negative sequence dq to positive sequence dq transformation module 85;

与q轴电流参考值

Negative sequence dq to positive sequence dq transformation module 85 transforms the dq component of the negative sequence dq coordinate system rotating at -ω _o speed into the positive sequence dq coordinate system rotating at ω _o speed through -2ω _o , and the transformed dq axis components Superimposed with the dq axis component output of the positive sequence voltage controller 82 to obtain the d axis current reference value of the current controller 84

with the q-axis current reference value

加上d轴输出电流

前馈减去d轴电感电流反馈

输入到在PI调节器的基础上增加了谐振控制环节构成的d轴电流PI+R调节器843，可有效抑制输出电压波形中的5、7次谐波畸变；q轴电流

交叉解耦项ωL_m以及输出电压

前馈项，再与变压器d轴原边零序电流的三次谐波比例谐振PR控制器841的输出相叠加得到调制波d轴分量u^d。

加上q轴输出电流

前馈减去q轴电感电流反馈

输入到q轴电流PI+R调节器844；d轴电流

交叉解耦项ωL_m以及输出电压

前馈项，再与变压器q轴原边零序电流的三次谐波比例谐振PR控制器842的输出相叠加得到调制波q轴分量u^q。

plus the d-axis output current

Feedforward minus d-axis inductor current feedback

Input to the d-axis current PI+R regulator 843 composed of a resonance control link added on the basis of the PI regulator, which can effectively suppress the 5th and 7th harmonic distortion in the output voltage waveform; the q-axis current

The cross-decoupling term ωL _m and the output voltage

The feedforward term is superimposed with the output of the third harmonic proportional resonance PR controller 841 of the zero-sequence current of the d-axis primary side of the transformer to obtain the d-axis component ^ud of the modulated wave.

plus the q-axis output current

Feedforward minus q-axis inductor current feedback

Input to q-axis current PI+R regulator 844; d-axis current

The cross-decoupling term ωL _m and the output voltage

The feed-forward term is superimposed with the output of the third harmonic proportional resonance PR controller 842 of the transformer q-axis primary zero-sequence current to obtain the modulated wave q-axis component u ^q .

The d-axis component u ^d and the q-axis component u ^q of the modulated wave are converted to dq inversely to obtain the ABC three-phase modulated wave voltages u _A , u _B and u _C . Then the driving pulse corresponding to the switching device is obtained through SPWM modulation. Using this control algorithm can suppress the negative sequence component in the fundamental voltage and improve the unbalanced load capacity of the inverter; it can also suppress the harmonics brought by the nonlinear load and the primary current harmonics of the transformer, and improve the quality of the output voltage waveform. Reduce the vibration level of the inverter. Based on the proposed topology, the invention discloses an inverter output voltage harmonic suppression strategy. The control strategy is based on the voltage-current double closed-loop control architecture under the synchronous rotating coordinate system, and the PI regulator is used to realize the non-static tracking of the fundamental voltage command. Further, in order to suppress the harmonics brought by nonlinear loads, the present invention adds a resonance link on the basis of the PI regulator, thereby improving the control system’s ability to suppress harmonics, improving the quality of the output voltage waveform, and reducing the The vibration energy level of the transformer.

The special marine micro-grid inverter disclosed in the present invention adopts a multiple combined inverter topology. The combined three-phase inverter is composed of three single-phase bridge inverter units, and each phase inverter unit is relatively independent. , its design and control method are similar to those of single-phase inverters. Compared with the three-phase half-bridge inverter, the combined three-phase inverter has the advantages of high utilization rate of DC voltage, flexible control, and strong load capacity with unbalanced load.

In the present invention, the output phases of two or more inverter units are staggered and superimposed together through transformers or inductors, so that the level number of the output voltage is increased. Although the switching frequency of each unit remains unchanged, the overall efficiency of the inverter is doubled. output equivalent switching frequency. Therefore, the present invention can reduce the output voltage harmonics, reduce the volume and weight of the output filter, reduce the heat dissipation pressure of the power semiconductor device, and do not need to improve the power level of the inverter through the series-parallel design of the complicated power devices; The transformer coupling is used on the secondary side to achieve multiplexing and electrical isolation on the secondary side.

The multi-combined inverter topology of the present invention realizes the improvement of the DC voltage utilization rate of the inverter and the quality of the output voltage through the cooperation of the transformer windings, and improves the power density through the magnetic integration technology of the transformer; the modulation strategy of the present invention, in realizing While the switching frequency is multiplied equivalently, the common mode voltage and the transformer primary side circulation are suppressed, the vibration noise of the inverter is reduced, and the power density is improved; the control method of the present invention can improve the output of the inverter with a nonlinear load Voltage quality, and effective suppression of transformer primary current harmonics, reducing the vibration level of the inverter.

Claims (5)

1. A marine microgrid inverter, characterized in that: it comprises a DC support capacitor (1), an A-phase bridge inverter unit (2), a B-phase bridge inverter unit (3), a C-phase bridge inverter Transformer unit (4), three-winding transformer (5) and three-phase AC filter (6), wherein the DC voltage input from the outside is applied to both ends of the DC support capacitor (1) to form a DC bus; three bridge inverter units With the same circuit topology, each bridge inverter unit is composed of four inverter bridge arms connected in parallel to the DC bus, each inverter bridge arm is composed of two series switching devices, and two sets of inverter bridge arms are composed An H-bridge inverter unit. 2. The marine microgrid inverter according to claim 1, characterized in that: the A-phase bridge inverter unit (2) includes a first switching device (20) connected in series with a second switching device (21) to form a second An inverter bridge arm, the third switching device (22) and the fourth switching device (23) are connected in series to form a second inverter bridge arm, and these two inverter bridge arms form an H-bridge inverter unit connected in parallel with the DC bus; Another H-bridge inverter unit formed after the fifth switching device (24) and the sixth switching device (25) are connected in series, and the seventh switching device (26) and the eighth switching device (27) are connected in parallel with the DC bus; Two H-bridge inverter units, one of which is connected to the first winding (50) of the A-phase primary side, and the other is connected to the second winding (51) of the A-phase primary side; the PWM voltage output by the two H-bridge inverter units is passed through the transformer winding Superposed in series, the output of the A-phase secondary side winding (52) obtains the doubled PWM voltage; the doubled PWM voltage is input to the three-phase AC filter (6) of the inverter, and the working voltage is obtained after filtering. frequency output voltage. 3. A modulation strategy of the marine microgrid inverter as claimed in claim 1, wherein the modulation strategy is: the triangular carrier wave of the first inverter bridge arm of each phase bridge inverter unit and the second inverter The phase difference between the variable bridge arm triangular carrier wave, the third inverter bridge arm triangular carrier wave and the fourth inverter bridge arm triangular carrier wave is 180°, the first inverter bridge arm triangular carrier wave of each phase bridge inverter unit is different from the third The phase difference between the triangular carrier wave of the inverter bridge arm, the triangular carrier wave of the second inverter bridge arm and the triangular carrier wave of the fourth inverter bridge arm is 90°; As a reference, the phase difference between the triangular carrier wave of each inverter bridge arm of the B-phase bridge inverter unit and the corresponding phase difference of each inverter bridge arm triangular carrier wave of the A-phase bridge inverter unit is α, and the phase difference of each inverter bridge arm triangular carrier wave of the C-phase bridge inverter unit The phase difference between each inverter bridge arm triangular carrier wave and each inverter bridge arm triangular carrier wave corresponding to the A-phase bridge inverter unit is β. 4. The modulation strategy of the marine microgrid inverter according to claim 3, characterized in that: α=180°, β=180°. 5. A control method for a marine microgrid inverter as claimed in claim 1, characterized in that: the three-phase voltage u _oa (t), u _ob (t) and u _oc (t) obtained by sampling are passed through the DSC Positive-sequence and negative-sequence fundamental wave voltage decomposition and dq transformation to obtain the d-axis positive-sequence fundamental wave voltage per unit value in the positive synchronous rotating coordinate system

q-axis positive sequence fundamental wave voltage per unit value

d-axis negative-sequence fundamental wave voltage per unit value in the reverse synchronous rotating coordinate system

and q-axis positive sequence fundamental wave voltage per unit value

and

As the feedback value of the positive sequence voltage controller (82), it participates in the inverter output positive sequence voltage control,

and

As the feedback value of the negative-sequence voltage controller (83), it participates in the inverter output negative-sequence voltage control; the d-axis per unit voltage reference value of the positive-sequence voltage controller (82) is 1, minus the feedback value

Then input to the d-axis positive-sequence voltage PI regulator (821) of the positive-sequence voltage controller (82), the q-axis per unit voltage reference value of the positive-sequence voltage controller (82) is 0, minus the feedback value

Then input to the q-axis positive sequence voltage PI regulator (822) of the positive sequence voltage controller (82); The d-axis per unit voltage reference value of the negative sequence voltage controller (83) is 0, minus the feedback value

The d-axis negative-sequence voltage PI regulator (831) input to the negative-sequence voltage controller (83), the output of the d-axis negative-sequence voltage PI regulator (831) and

Multiplied by the cross-decoupling term ωC _f and subtracted, output to the d-axis input port of the negative-sequence dq to positive-sequence dq transformation module (86); the q-axis per-unit voltage reference value of the negative-sequence voltage controller (83) is 0 , minus the feedback value

The q-axis negative-sequence voltage PI regulator (832) that is input to the negative-sequence voltage controller (83) afterward, the output of the q-axis negative-sequence voltage PI regulator (832) and

After being multiplied by the cross-decoupling term ωC _f subtracted, it is output to the q-axis input port of the negative sequence dq to the positive sequence dq transformation module (85); Negative sequence dq to positive sequence dq transformation module (85) transforms the dq component of the negative sequence dq coordinate system rotated at -ω _o speed into the positive sequence dq coordinate system rotated at ω _o speed through -2ω _o , the transformed dq The axis component and the dq axis component output of the positive sequence voltage controller (82) are superimposed to obtain the d axis current reference value of the current controller (84)

with the q-axis current reference value

plus the d-axis output current

Feedforward minus d-axis inductor current feedback

Input to the d-axis current PI+R regulator (843) that has increased the resonance control link on the basis of the PI regulator; the q-axis current

The cross-decoupling term ωL _m and the output voltage

The feedforward item is superimposed with the output of the third harmonic proportional resonance PR controller (841) of the zero-sequence current of the primary side of the d-axis of the transformer to obtain the d-axis component u ^d of the modulated wave;

plus the q-axis output current

Feedforward minus q-axis inductor current feedback

Input to q-axis current PI+R regulator (844); d-axis current

The cross-decoupling term ωL _m and the output voltage

The feed-forward term is superimposed with the output of the third harmonic proportional resonance PR controller (842) of the transformer q-axis primary zero-sequence current to obtain the modulated wave q-axis component u ^q ; The d-axis component u ^d and the q-axis component u ^q of the modulation wave are converted to dq inversely to obtain the ABC three-phase modulation wave voltages u _A , u _B and u _C , and then the driving pulses of the corresponding switching devices are obtained through SPWM modulation.

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