CN115411964A - Marine microgrid inverter, modulation strategy and control method - Google Patents

Marine microgrid inverter, modulation strategy and control method Download PDF

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Publication number
CN115411964A
CN115411964A CN202211006428.8A CN202211006428A CN115411964A CN 115411964 A CN115411964 A CN 115411964A CN 202211006428 A CN202211006428 A CN 202211006428A CN 115411964 A CN115411964 A CN 115411964A
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inverter
phase
voltage
axis
current
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揭贵生
肖飞
高山
季圣贤
王瑞田
范学鑫
张磊
王恒利
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Naval University of Engineering PLA
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H02M1/123—Suppression of common mode voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02E—REDUCTION OF GREENHOUSE GAS [GHG] EMISSIONS, RELATED TO ENERGY GENERATION, TRANSMISSION OR DISTRIBUTION
    • Y02E40/00—Technologies for an efficient electrical power generation, transmission or distribution
    • Y02E40/40—Arrangements for reducing harmonics

Abstract

The invention discloses a marine microgrid inverter, a modulation strategy and a control method. The inverter comprises a direct current supporting capacitor, an A-phase bridge type inversion unit, a B-phase bridge type inversion unit, a C-phase bridge type inversion unit, a three-winding transformer and a three-phase alternating current filter, wherein direct current voltage input from the outside is applied to two ends of the direct current supporting capacitor to form a direct current bus, the three bridge type inversion units have the same circuit topology, the input direct current voltage is converted into alternating current PWM voltage, and the alternating current PWM voltage is connected in series and superposed through a transformer winding. The utilization rate of the direct-current voltage of the inverter and the quality of the output voltage are improved through the matching of transformer windings, and the power density is improved through the magnetic integration technology of the transformer; the modulation strategy of the invention restrains common-mode voltage and primary side circulation of the transformer while realizing equivalent frequency multiplication of switching frequency, reduces vibration noise of the inverter and improves power density; the control method can improve the quality of the output voltage of the inverter with the nonlinear load, realize the effective inhibition of the primary side current harmonic of the transformer and reduce the vibration energy level of the inverter.

Description

Marine microgrid inverter, modulation strategy and control method
Technical Field
The invention belongs to the technical field of inverter power supplies, and particularly relates to a marine micro-grid inverter, a modulation strategy and a control method.
Background
At present, a known microgrid inverter generally adopts a three-phase three-leg inverter topology, and has the advantages of simple circuit topology, small number of devices, low cost, mature control strategy and the like. But the special marine micro-grid inverter has higher requirements on power density, direct-current voltage utilization rate, output voltage quality and vibration noise indexes than a common micro-grid inverter. The known circulating current suppression measures mainly comprise methods of connecting circulating current limiting inductors in series, changing the carrier phase of an inversion unit and the like. The size and the cost of the inverter are inevitably increased by connecting the loop current filter inductors in series; although the hardware cost is not increased by changing the carrier phase of the inversion unit, the carrier phase determines the multiple harmonic superposition characteristics, and the new problems of output voltage waveform quality, electromagnetic compatibility index reduction and the like can be brought while the circulating current is restrained.
Meanwhile, a three-phase inverter generally adopts a voltage-current double closed-loop control strategy under a synchronous rotating coordinate system, and a regulator of the three-phase inverter generally adopts a PI (proportional integral) regulator, so that the static-error-free tracking of fundamental voltage instructions can be realized. However, since the PI regulator has a limited ability to suppress harmonics, the quality of the inverter output voltage is significantly reduced when a nonlinear load is applied. In order to improve the harmonic suppression capability of the inverter, the closed-loop control bandwidth can be increased, but the system stability margin can be reduced, so that the voltage overshoot is increased and even the voltage is unstable in the dynamic process of the load. In addition, due to the adoption of the multiple superposition topology based on the transformer, the primary current harmonics of the transformer are abundant, and the vibration level of the transformer is inevitably increased, so that effective primary current harmonic suppression measures are required.
Disclosure of Invention
Aiming at the technical defects, the invention provides the marine micro-grid inverter, a modulation strategy and a control method which can improve the power density, the direct-current voltage utilization rate and the output voltage quality of the inverter and simultaneously reduce the vibration noise energy level.
In order to achieve the purpose, the invention provides a micro-grid inverter for a ship, which comprises a direct-current supporting capacitor 1, an A-phase bridge inverter unit 2, a B-phase bridge inverter unit 3, a C-phase bridge inverter unit 4, a three-winding transformer 5 and a three-phase alternating-current filter 6, wherein direct-current voltage input from the outside is applied to two ends of the direct-current supporting capacitor 1 to form a direct-current bus; the three bridge-type inverter units have the same circuit topology, each bridge-type inverter unit consists of four inverter bridge arms connected in parallel with a direct-current bus, each inverter bridge arm consists of two series-connected switch devices, and two groups of inverter bridge arms form an H-bridge inverter unit.
Further, the a-phase bridge inverter unit 2 includes a first switching device 20 and a second switching device 21 connected in series to form a first inverter bridge arm, a third switching device 22 and a fourth switching device 23 connected in series to form a second inverter bridge arm, and the two inverter bridge arms form an H-bridge inverter unit connected in parallel with the dc bus; another H-bridge inverter unit formed by connecting the fifth switching device 24 and the sixth switching device 25 in series and connecting the seventh switching device 26 and the eighth switching device 27 in series is connected in parallel with the dc bus; one of the two H-bridge inverter units is connected to the A-phase primary side first winding 50, and the other is connected to the A-phase primary side second winding 51; the PWM voltages output by the two H-bridge inverter units are superposed in series through a transformer winding, and a doubled PWM voltage is output from an A-phase secondary side winding 52; the doubled PWM voltage is input to a three-phase ac filter 6 of the inverter, and filtered to obtain a power frequency output voltage.
The modulation strategy of the marine microgrid inverter is further provided, and is as follows: the phase difference between the first inverter bridge arm triangular carrier wave and the second inverter bridge arm triangular carrier wave and between the third inverter bridge arm triangular carrier wave and the fourth inverter bridge arm triangular carrier wave of each phase bridge type inverter unit is 180 degrees, and the phase difference between the first inverter bridge arm triangular carrier wave and the third inverter bridge arm triangular carrier wave and between the second inverter bridge arm triangular carrier wave and the fourth inverter bridge arm triangular carrier wave of each phase bridge type inverter unit is 90 degrees; and taking the phase of the first inverter bridge arm triangular carrier of the A-phase bridge inverter unit as a reference, wherein the phase difference between each inverter bridge arm triangular carrier of the B-phase bridge inverter unit and each inverter bridge arm triangular carrier corresponding to the A-phase bridge inverter unit is alpha, and the phase difference between each inverter bridge arm triangular carrier of the C-phase bridge inverter unit and each inverter bridge arm triangular carrier corresponding to the A-phase bridge inverter unit is beta.
Further, α =180 °, β =180 °.
The control method of the marine micro-grid inverter is also provided, and the three-phase voltage u obtained by sampling is used oa (t)、u ob (t) and u oc (t) obtaining d-axis positive sequence fundamental wave voltage per unit value under the positive synchronous rotation coordinate system through DSC positive sequence, negative sequence fundamental wave voltage decomposition and dq conversion
Figure BDA0003808791180000021
q-axis positive sequence fundamental wave voltage per unit value
Figure BDA0003808791180000022
D-axis negative sequence fundamental wave voltage per unit value under reverse synchronous rotation coordinate system
Figure BDA0003808791180000023
And q-axis positive sequence fundamental wave voltage per unit value
Figure BDA0003808791180000031
And
Figure BDA0003808791180000032
participate in the inverter output positive sequence voltage control as a feedback value to the positive sequence voltage controller 82,
Figure BDA0003808791180000033
and
Figure BDA0003808791180000034
participate in the inverter output negative sequence voltage control as a feedback value of the negative sequence voltage controller 83; the d-axis per-unit voltage reference value of the positive sequence voltage controller 82 is 1, minus the feedback value
Figure BDA0003808791180000035
Then, the q-axis voltage is input to the d-axis positive sequence voltage PI regulator 821 of the positive sequence voltage controller 82, the q-axis per-unit voltage reference value of the positive sequence voltage controller 82 is 0, and the feedback value is subtracted
Figure BDA0003808791180000036
A q-axis positive sequence voltage PI regulator 822 that inputs the q-axis positive sequence voltage PI to the positive sequence voltage controller 82;
the d-axis per-unit voltage reference value of the negative sequence voltage controller 83 is 0, minus the feedback value
Figure BDA0003808791180000037
The d-axis negative sequence voltage PI regulator 831 subsequently inputted to the negative sequence voltage controller 83, the output of the d-axis negative sequence voltage PI regulator 831, and
Figure BDA0003808791180000038
multiplication by a cross-decoupling term ω C f The subtracted signals are output to a d-axis input port of a negative-sequence dq to positive-sequence dq conversion module 86; the q-axis per-unit voltage reference value of the negative sequence voltage controller 83 is 0, minus the feedback value
Figure BDA0003808791180000039
Then input to the q-axis negative sequence voltage PI regulator 832 of the negative sequence voltage controller 83, the output of the q-axis negative sequence voltage PI regulator 832Go out and
Figure BDA00038087911800000310
multiplication by a cross-decoupling term ω C f The subtracted signals are output to a q-axis input port of a negative sequence dq to positive sequence dq conversion module 85;
negative-sequence dq to positive-sequence dq conversion module 85 converts- ω o The dq component of the negative sequence dq coordinate system of the speed rotation passes through-2 ω o Conversion to ω o In a positive sequence dq coordinate system of speed rotation, the converted dq axis component is superposed with the dq axis component output of the positive sequence voltage controller 82 to obtain a d-axis current reference value of the current controller 84
Figure BDA00038087911800000311
And q-axis current reference value
Figure BDA00038087911800000312
Figure BDA00038087911800000313
Plus d-axis output current
Figure BDA00038087911800000314
Feed forward minus d-axis inductor current feedback
Figure BDA00038087911800000315
The current is input to a d-axis current PI + R regulator 843 formed by adding a resonance control link on the basis of a PI regulator; q axis current
Figure BDA00038087911800000316
Cross decoupling term ω L m And an output voltage
Figure BDA00038087911800000317
The feedforward term is superposed with the output of a third harmonic proportional resonant PR controller 841 of the zero-sequence current of the primary side of the d-axis of the transformer to obtain a modulated wave d-axis component u d ；
Figure BDA00038087911800000318
Plus q-axis output current
Figure BDA00038087911800000319
Feed forward minus q-axis inductor current feedback
Figure BDA00038087911800000320
Input to q-axis current PI + R regulator 844; d axis current
Figure BDA00038087911800000321
Cross decoupling term ω L m And an output voltage
Figure BDA00038087911800000322
The feedforward term is superposed with the output phase of the third harmonic proportional resonant PR controller 842 of the zero-sequence current of the primary side of the q axis of the transformer to obtain the q axis component u of the modulation wave q ；
For d-axis component u of modulated wave d And q-axis component u q Carrying out inverse transformation to dq to obtain ABC three-phase modulation wave voltage u A 、u B And u C And then the driving pulse of the corresponding switching device is obtained through SPWM modulation.
Compared with the prior art, the invention has the beneficial effects that: the multiple combined inversion topology realizes the improvement of the utilization rate of the direct-current voltage and the quality of the output voltage of the inverter through the matching of transformer windings, and improves the power density through the magnetic integration technology of the transformer; the modulation strategy of the invention restrains common mode voltage and transformer primary side circulation while realizing equivalent frequency multiplication of switching frequency, reduces vibration noise of the inverter and improves power density; the control method can improve the quality of the output voltage of the inverter with the nonlinear load, realize the effective inhibition of the primary current harmonic of the transformer and reduce the vibration energy level of the inverter.
Drawings
Fig. 1 is a topological structure diagram of a marine microgrid inverter according to the present invention;
FIG. 2 is a waveform diagram of a power frequency modulation wave and two triangular carrier waves of an H-bridge inverter unit;
FIG. 3 is a graph of the output PWM voltage waveform of the H-bridge inverter unit;
FIG. 4 is a waveform diagram of a power frequency modulation wave and four triangular carrier waves of a bridge inverter unit;
FIG. 5 is a graph of the PWM voltage waveform output by a bridge inverter unit;
FIG. 6 is a graph of the effect of phase-to-phase carrier phase shift angle on the amplitude of the common-mode voltage output by the inverter;
FIG. 7 is a graph showing the effect of the phase shift angle of the phase-to-phase carrier on the effective value of the common mode voltage output by the inverter;
FIG. 8 is a diagram illustrating the effect of the phase shift angle of the phase-to-phase carrier on the effective value of the phase A circulating current;
FIG. 9 is a diagram illustrating the effect of the phase shift angle of the inter-phase carrier on the effective value of the B-phase circulating current;
FIG. 10 is a graph showing the effect of the phase shift angle of the inter-phase carrier on the effective value of the C-phase circulating current;
fig. 11 is a block diagram of an inverter control system of the present invention.
In the figure, 1, a direct current support capacitor, 2.A phase bridge inversion unit, 3.B phase bridge inversion unit, 4.C phase bridge inversion unit, 5, a multi-winding transformer, 6, a three-phase alternating current filter, 20.A phase bridge inversion unit first switch device, 21.A phase bridge inversion unit second switch device, 22.A phase bridge inversion unit third switch device, 23.A phase bridge inversion unit fourth switch device, 24.A phase bridge inversion unit fifth switch device, 25.A phase bridge inversion unit sixth switch device, 26.a phase bridge inversion unit seventh switch device, 27.A phase bridge inversion unit eighth switch device, 50.a phase primary side first winding, 51.a phase primary side second winding, 52.a phase primary side secondary winding, 53.b phase first winding, 54.b phase primary side second winding, 55.b phase secondary side winding, 56.c phase first winding, 57.c phase second winding, 58.C-phase secondary side winding, 70 power frequency modulation wave, 71 first inverter bridge arm triangular carrier wave, 72 third inverter bridge arm triangular carrier wave, 73 second inverter bridge arm triangular carrier wave, 74 fourth inverter bridge arm triangular carrier wave, 75 transformer primary side winding PWM voltage, 76 transformer secondary side winding PWM voltage, 82 positive sequence voltage controller, 83 negative sequence voltage controller, 84 current controller, 85 negative sequence dq to positive sequence dq conversion, 821.D axis positive sequence voltage PI regulator, 822.Q axis positive sequence voltage PI regulator, 831.d axis negative sequence voltage PI regulator, 832.Q axis negative sequence voltage PI regulator, 841 transformer d axis primary side zero sequence current third harmonic proportional resonant PR controller, 842 transformer q axis primary side zero sequence current third harmonic proportional resonant PR controller, 843.D axis current PI + R regulator, 844.Q axis current PI + R regulator.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention is further described in detail below with reference to the accompanying drawings.
The invention obtains the carrier phase shift angle combination giving consideration to common mode voltage and circulation current inhibition by analyzing the influence of three inter-phase carrier phase shift angles of the inversion unit on the output common mode voltage and circulation current, is easy to realize, can reduce the cost of circulation current filter inductance and an EMI filter, and is further explained by combining the attached drawings and the specific implementation mode.
Fig. 1 is a topological structure diagram of the marine micro-grid inverter, and the marine micro-grid inverter includes a dc support capacitor 1, an a-phase bridge inverter unit 2, a B-phase bridge inverter unit 3, a C-phase bridge inverter unit 4, a three-winding transformer 5, and a three-phase ac filter 6, wherein an externally input dc voltage is applied to two ends of the dc support capacitor 1 to form a dc bus. The three bridge type inversion units have the same circuit topology, each bridge type inversion unit is composed of four inversion bridge arms connected into a direct current bus in parallel, each inversion bridge arm is composed of two series-connected switch devices, and two groups of inversion bridge arms form an H-bridge inversion unit. Taking the a-phase bridge inverter unit 2 as an example: the first switching device 20 and the second switching device 21 are connected in series to form a first inverter bridge arm, the third switching device 22 and the fourth switching device 23 are connected in series to form a second inverter bridge arm, and the two inverter bridge arms form an H-bridge inverter unit which is connected with a direct-current bus in parallel; similarly, another H-bridge inverter unit formed by connecting the fifth switching device 24 and the sixth switching device 25 in series and connecting the seventh switching device 26 and the eighth switching device 27 in series is connected in parallel with the dc bus; one of the two H-bridge inverter units is connected into the A-phase primary side first winding 50, and the other one is connected into the A-phase primary side second winding 51; the PWM voltages output by the two H-bridge inverter units are superposed in series through a transformer winding, and a doubled PWM voltage is output from an A-phase secondary side winding 52; the doubled PWM voltage is input to a three-phase ac filter 6 of the inverter, and filtered to obtain a power frequency output voltage.
To convert the input dc voltage into ac PWM voltage, the inverter topology shown in fig. 1 may employ a carrier phase shift modulation strategy based on unipolar frequency multiplication modulation as follows: the phase difference between the first inverter bridge arm triangular carrier wave and the second inverter bridge arm triangular carrier wave and between the third inverter bridge arm triangular carrier wave and the fourth inverter bridge arm triangular carrier wave of each phase bridge inverter unit is 180 degrees, and the phase difference between the first inverter bridge arm triangular carrier wave and the third inverter bridge arm triangular carrier wave and between the second inverter bridge arm triangular carrier wave and the fourth inverter bridge arm triangular carrier wave of each phase bridge inverter unit is 90 degrees; the phase difference between each inverter bridge arm triangular carrier of the B-phase bridge inverter unit and each inverter bridge arm triangular carrier corresponding to the A-phase bridge inverter unit is alpha, the phase difference between each inverter bridge arm triangular carrier of the C-phase bridge inverter unit and each inverter bridge arm triangular carrier corresponding to the A-phase bridge inverter unit is beta, and the phase difference is alpha =180 degrees and beta =180 degrees.
Due to the existence of the phase shift angle of the triangular carrier, the triangular carrier and the high-frequency harmonic waves near the multiple frequency of the triangular carrier are coupled through the windings of the shared transformer core, and high-frequency circulating current can be generated. In order to restrain high-frequency circulation among inversion units, the influence of phase-to-phase carrier phase shift angles on circulation is analyzed by utilizing the control freedom degree of the three phase-to-phase carrier phase shift angles of the inverter, and finally, the AC output common mode voltage restraint of the inverter is considered.
In this embodiment, it is assumed that the phase of the triangular carrier of the first inverter bridge arm of the a-phase bridge inverter unit is 0 °, the phase of the triangular carrier of the second inverter bridge arm is-180 °, the phase of the triangular carrier of the third inverter bridge arm is-90 °, the phase of the triangular carrier of the fourth inverter bridge arm is-270 °, the phase of the triangular carrier of the first inverter bridge arm of the B-phase bridge inverter unit is α, the phase of the carrier of the second bridge arm is α -180 °, the phase of the carrier of the third bridge arm is α -90 °, the phase of the carrier of the fourth bridge arm is α -270 °, the phase of the triangular carrier of the first inverter bridge arm of the C-phase bridge inverter unit is β, the phase of the carrier of the second bridge arm is β -180 °, the phase of the carrier of the third bridge arm is β -90 °, and the phase of the carrier of the fourth bridge arm is β -270 °;
the angle between alpha and beta can be adjusted according to specific requirements, the output power frequency voltage of the inverter and equivalent quadruple frequency and five-level effects are not influenced, but different carrier phases can influence the circulating current and alternating current output common mode voltage among the three-phase inversion units, and a specific implementation method for selecting the carrier phases among the three-phase inversion units is given as follows:
note that the voltage of the primary side first winding 50 of the a phase is V 1 Current is I 1 The voltage of the A-phase primary side secondary winding 51 is V 2 Current is I 2 The voltage of the A-phase secondary winding 52 is V 3 Current is I 3 (ii) a The voltage of the B-phase primary side first winding 53 is V 4 Current is I 4 The voltage of the B-phase primary side secondary winding 54 is V 5 Current is I 5 The voltage of the B-phase secondary winding 55 is V 6 Current is I 6 (ii) a The voltage of the first winding 56 on the primary side of the C phase is V 7 Current is I 7 And the voltage of the C-phase primary side secondary winding 57 is V 8 Current is I 8 The voltage of the C-phase secondary side winding 58 is V 9 Current is I 9 . According to the voltage and current equation of the transformer port, the following can be obtained:
Figure BDA0003808791180000071
wherein, V = [ V ] 1 V 2 V 3 V 4 V 5 V 6 V 7 V 8 V 9 ] T ，I＝[I 1 I 2 I 3 I 4 I 5 I 6 I 7 I 8 I 9 ] T ，
Figure BDA0003808791180000072
Figure BDA0003808791180000073
The voltage and current coupling relation of the transformer port is complex. Wherein the resistance value R in the resistance matrix R i (i =1,2,3,4,5,6,7,8, 9) is the transformer ith winding resistance; the inductance matrix L has 81 inductance parameters for the transformer model established based on the coupling inductance, wherein the inductance L on the diagonal line of the inductance matrix ii (i =1,2,3,4,5,6,7,8,9) is the self-inductance of the 9 windings of the transformer, the inductance L ij ＝L ji (i =1,2,3,4,5,6,7,8, 9) is the mutual inductance of the i and j windings, for a total inductance of 72. Because of numerous circuit parameters, the relation between the AC output common-mode voltage of the inverter, the circulating current among the units and the phase shift angle of the carrier among the power units is difficult to express intuitively by a calculation formula.
Defining inverter AC output common mode voltage V cm ：
V cm ＝(V 3 +V 6 +V 9 )/3 (2)
With the phase-A carrier as ase:Sub>A reference, FIG. 4 shows the variation law of the amplitude of the common-mode voltage output by the inverter along with the phase shift angle of the phase-B-A phase carrier and the phase shift angle of the phase-C-A phase carrier; fig. 5 shows the change rule of the effective value of the common-mode voltage output by the inverter along with the phase shift angle of the carrier between the phases B and A and the phase shift angle of the carrier between the phases C and A. According to the volt-second balance principle, the change of the phase of the carrier does not affect the change of the amplitude of the output power frequency voltage. The three-phase carrier phase is at the minimum of the common-mode voltage amplitude and the effective value under the two combinations of (0, -120 degrees, 120 degrees) and (0, 120 degrees).
Defining the circulation current between A phase inversion units as I cir_A And the circulation current between B phase inversion units is I cir_B The circulation current between the C phase inversion units is I cir_C ：
Figure BDA0003808791180000081
Fig. 6, fig. 7 and fig. 8 show the change law of circulation effective values of the phase inversion units of phase ase:Sub>A, phase B and phase C with the carrier phase shift angle between phase B-ase:Sub>A and the carrier phase shift angle between phase C-ase:Sub>A. Again with reference to the a-phase carrier phase. And (4) simulating and traversing the carrier phase angles of the phase B and the phase C to obtain three-phase circulating current effective values under different carrier phase combinations. The circulation of each phase is minimized when the phase shift angles of the a, B and C phases are four combinations of (0 °,0 °,0 °), (0 °,0 °,180 °), (0 °,180 °,0 °), and (0 °,180 °,180 °).
Therefore, the analysis result of the circulation current between the alternating current output common mode voltage and the inversion unit is integrated, so that the phase shift angle of the minimum common mode voltage and the minimum circulation current is different, and if the phase shift angle is in compromise between the common mode voltage and the minimum circulation current. Preferably, the circulating current is restrained, and (0, 180 degrees and 180 degrees) are selected as the optimized carrier phase angles, namely, alpha =180 degrees and beta =180 degrees.
Figure BDA0003808791180000082
Three power frequency modulation waves are shared by the three-phase bridge type inversion units of the inverter, namely, the power frequency modulation wave of the A-phase bridge type inversion unit, the power frequency modulation wave of the B-phase bridge type inversion unit and the power frequency modulation wave of the C-phase bridge type inversion unit, and the three-phase power frequency modulation waves are different by 120 degrees; therefore, the power frequency modulation wave of the A-phase bridge type inverter unit is used as a power frequency modulation wave shared by four inverter bridge arms of the A-phase bridge type inverter unit, the power frequency modulation wave of the B-phase bridge type inverter unit is used as a power frequency modulation wave shared by four inverter bridge arms of the B-phase bridge type inverter unit, and the power frequency modulation wave of the C-phase bridge type inverter unit is used as a power frequency modulation wave shared by four inverter bridge arms of the C-phase bridge type inverter unit.
With reference to the modulation strategy of an H-bridge inverter unit shown in fig. 2 and 3, when the power frequency modulation wave 70 is greater than the triangular carrier wave 71 of the first inverter bridge arm, the upper switch tube of the first inverter bridge arm is driven to be turned on, and the lower switch tube is driven to be turned off; inverting the phase of the first inverter bridge arm triangular carrier wave 71 to obtain a second inverter bridge arm triangular carrier wave 73, comparing by using the same power frequency modulation wave 70, and driving the upper switch tube of the second inverter bridge arm to be switched on and the lower switch tube to be switched off when the power frequency modulation wave 70 is greater than the second inverter bridge arm triangular carrier wave 73; because the phase difference between the first inverter bridge arm triangular carrier wave 71 and the second inverter bridge arm triangular carrier wave 73 is 180 degrees, after the modulation strategy is adopted, the polarity of the primary side winding PWM voltage 75 of the transformer is the same as that of a power frequency modulation wave, and the equivalent output harmonic frequency of the transformer is equal to twice of the carrier frequency.
With reference to the carrier signals corresponding to the 4 inverter legs in one bridge inverter unit shown in fig. 4 and 5, the third inverter leg triangular carrier 72 and the fourth inverter leg triangular carrier 74 lag behind the first inverter leg triangular carrier 71 and the second inverter leg triangular carrier 73 by 90 °, respectively. The PWM output voltages of the two H-bridge inverter units are superposed by the transformer windings, and a five-level PWM characteristic of quadruple carrier frequency is presented in the transformer secondary winding voltage 76. Therefore, the waveform of the transformer secondary side winding PWM voltage 76 after the superposition of the transformer presents quadruple carrier frequency and five-level characteristics, the power density of the inverter is improved, and the vibration energy level in the inverter testing range is reduced.
The capability of improving the quality of the output voltage waveform and suppressing the harmonic wave of the inverter control system is illustrated by analyzing the architecture of the inverter control system, and the following description is further provided by combining the attached drawings and the specific implementation mode.
FIG. 9 shows a block diagram of a control algorithm of the present invention, sampled three-phase voltages u oa (t)、u ob (t) and u oc (t) obtaining d-axis positive sequence fundamental wave voltage per unit value under the positive synchronous rotation coordinate system through DSC positive sequence, negative sequence fundamental wave voltage decomposition and dq conversion
Figure BDA0003808791180000091
q-axis positive sequence fundamental wave voltage per unit value
Figure BDA0003808791180000092
D-axis negative sequence fundamental wave voltage per unit value under reverse synchronous rotation coordinate system
Figure BDA0003808791180000093
And q-axis positive sequence fundamental wave voltage per unit value
Figure BDA0003808791180000094
And with
Figure BDA0003808791180000095
Participate in the inverter output positive sequence voltage control as a feedback value to the positive sequence voltage controller 82,
Figure BDA0003808791180000096
and
Figure BDA0003808791180000097
and participates in the inverter output negative sequence voltage control as a feedback value of the negative sequence voltage controller 83.
The d-axis per unit voltage reference value of the positive sequence voltage controller 82 is 1, minus the feedback value
Figure BDA0003808791180000098
Then, the q-axis voltage is input to a d-axis positive sequence voltage PI regulator 821 of the positive sequence voltage controller 82, the q-axis per-unit voltage reference value of the positive sequence voltage controller 82 is 0, and the feedback value is subtracted
Figure BDA0003808791180000099
A q-axis positive sequence voltage PI regulator 822 that inputs the q-axis positive sequence voltage PI to the positive sequence voltage controller 82;
the d-axis per-unit voltage reference value of the negative sequence voltage controller 83 is 0, minus the feedback value
Figure BDA00038087911800000910
The d-axis negative sequence voltage PI regulator 831 subsequently inputted to the negative sequence voltage controller 83, the output of the d-axis negative sequence voltage PI regulator 831, and
Figure BDA0003808791180000101
multiplication by a cross-decoupling term ω C f The subtracted signals are output to a d-axis input port of a negative-sequence dq to positive-sequence dq conversion module 86; the q-axis per-unit voltage reference value of the negative sequence voltage controller 83 is 0, minus the feedback value
Figure BDA0003808791180000102
Is then input toQ-axis negative sequence voltage PI regulator 832 of negative sequence voltage controller 83, output of q-axis negative sequence voltage PI regulator 832 and
Figure BDA0003808791180000103
multiplication by a cross-decoupling term ω C f The subtracted signals are output to a q-axis input port of a negative sequence dq to positive sequence dq conversion module 85;
negative-sequence dq to positive-sequence dq conversion module 85 converts- ω o The dq component of the negative sequence dq coordinate system of the speed rotation passes through-2 ω o Conversion to ω o In a positive sequence dq coordinate system of speed rotation, the converted dq axis component is superposed with the dq axis component output of the positive sequence voltage controller 82 to obtain a d-axis current reference value of the current controller 84
Figure BDA0003808791180000104
And q-axis current reference value
Figure BDA0003808791180000105
Figure BDA0003808791180000106
Plus d-axis output current
Figure BDA0003808791180000107
Feed forward minus d-axis inductor current feedback
Figure BDA0003808791180000108
The output voltage is input to a d-axis current PI + R regulator 843 formed by adding a resonance control link on the basis of a PI regulator, and 5 and 7 harmonic distortion in the output voltage waveform can be effectively inhibited; current of q axis
Figure BDA0003808791180000109
Cross decoupling term ω L m And an output voltage
Figure BDA00038087911800001010
The feedforward term is superposed with the output of the third harmonic proportional resonant PR controller 841 of the zero sequence current of the primary side of the d-axis of the transformerTo d-axis component u of modulated wave d 。
Figure BDA00038087911800001011
Plus q-axis output current
Figure BDA00038087911800001012
Feed forward minus q-axis inductor current feedback
Figure BDA00038087911800001013
An input to q-axis current PI + R regulator 844; d axis current
Figure BDA00038087911800001014
Cross decoupling term ω L m And an output voltage
Figure BDA00038087911800001015
The feedforward term is superposed with the output phase of the third harmonic proportional resonant PR controller 842 of the zero-sequence current of the primary side of the q axis of the transformer to obtain the q axis component u of the modulation wave q 。
For d-axis component u of modulated wave d And q-axis component u q Carrying out inverse transformation to dq to obtain ABC three-phase modulation wave voltage u A 、u B And u C . And then the driving pulse of the corresponding switching device is obtained through SPWM modulation. The control algorithm can inhibit the negative sequence component in the fundamental voltage and improve the unbalanced load capacity of the inverter; and harmonic waves caused by nonlinear loads and primary side current harmonic waves of the transformer can be restrained, the quality of output voltage waveform is improved, and the vibration energy level of the inverter is reduced. The invention discloses an inverter output voltage harmonic suppression strategy based on the proposed topology. The control strategy is based on a voltage-current double closed-loop control framework under a synchronous rotating coordinate system, and static error-free tracking of a fundamental voltage instruction is realized by adopting a PI (proportional-integral) regulator. Furthermore, in order to suppress harmonic waves caused by nonlinear loads, the invention adds a resonance link on the basis of the PI regulator, thereby improving the suppression capability of a control system on the harmonic waves, improving the quality of the output voltage waveform and reducing the vibration energy level of the transformer.
The invention discloses a special marine micro-grid inverter, which adopts a multiple combined inversion topology, wherein a combined three-phase inverter is formed by combining three single-phase bridge inversion units, each phase inversion unit is relatively independent, and the design and control method of the inverter are similar to those of a single-phase inverter. Compared with a three-phase half-bridge inverter, the combined three-phase inverter has the advantages of high direct-current voltage utilization rate, flexible control, strong unbalanced load capacity and the like.
The invention staggers the output phases of two or more inversion units and then superposes the output phases together through a transformer or an inductor, so that the level number of the output voltage is increased, and the integral output equivalent switching frequency of the inverter is multiplied although the switching frequency of each unit is unchanged. Therefore, the invention can reduce the output voltage harmonic wave, reduce the volume and the weight of the output filter, reduce the heat dissipation pressure of the power semiconductor device, and improve the power grade of the inverter without the series-parallel design of complex power devices; by using transformer coupling at the output, multiplexing and electrical isolation can be achieved on the secondary side.
According to the multiple combined inversion topology, the utilization rate of direct-current voltage and the quality of output voltage of the inverter are improved through the matching of transformer windings, and the power density is improved through the magnetic integration technology of the transformer; the modulation strategy of the invention restrains common-mode voltage and primary side circulation of the transformer while realizing equivalent frequency multiplication of switching frequency, reduces vibration noise of the inverter and improves power density; the control method can improve the quality of the output voltage of the inverter with the nonlinear load, realize the effective inhibition of the primary side current harmonic of the transformer and reduce the vibration energy level of the inverter.

Claims (5)

1. The utility model provides a marine microgrid inverter which characterized in that: the three-phase inverter comprises a direct-current supporting capacitor (1), an A-phase bridge inverter unit (2), a B-phase bridge inverter unit (3), a C-phase bridge inverter unit (4), a three-winding transformer (5) and a three-phase alternating-current filter (6), wherein direct-current voltage input from the outside is applied to two ends of the direct-current supporting capacitor (1) to form a direct-current bus; the three bridge-type inverter units have the same circuit topology, each bridge-type inverter unit consists of four inverter bridge arms connected in parallel with a direct-current bus, each inverter bridge arm consists of two series-connected switch devices, and two groups of inverter bridge arms form an H-bridge inverter unit.
2. The marine microgrid inverter of claim 1, wherein: the A-phase bridge type inversion unit (2) comprises a first switch device (20) and a second switch device (21) which are connected in series to form a first inversion bridge arm, a third switch device (22) and a fourth switch device (23) which are connected in series to form a second inversion bridge arm, and the two inversion bridge arms form an H-bridge inversion unit which is connected with a direct-current bus in parallel; the other H-bridge inverter unit formed by connecting a fifth switching device (24) and a sixth switching device (25) in series and connecting a seventh switching device (26) and an eighth switching device (27) in series is connected with the direct-current bus in parallel; one of the two H-bridge inverter units is connected into the A-phase primary side first winding (50), and the other one is connected into the A-phase primary side second winding (51); PWM voltages output by the two H-bridge inverter units are superposed in series through a transformer winding, and a doubled PWM voltage is obtained through output of an A-phase secondary side winding (52); the doubled PWM voltage is input to a three-phase alternating current filter (6) of the inverter, and power frequency output voltage is obtained after filtering.
3.A modulation strategy for the marine microgrid inverter of claim 1, characterized in that: the modulation strategy is as follows: the phase difference between the first inverter bridge arm triangular carrier wave and the second inverter bridge arm triangular carrier wave and between the third inverter bridge arm triangular carrier wave and the fourth inverter bridge arm triangular carrier wave of each phase bridge inverter unit is 180 degrees, and the phase difference between the first inverter bridge arm triangular carrier wave and the third inverter bridge arm triangular carrier wave and between the second inverter bridge arm triangular carrier wave and the fourth inverter bridge arm triangular carrier wave of each phase bridge inverter unit is 90 degrees; and taking the phase of the first inverter bridge arm triangular carrier of the A-phase bridge inverter unit as a reference, wherein the phase difference between each inverter bridge arm triangular carrier of the B-phase bridge inverter unit and each inverter bridge arm triangular carrier corresponding to the A-phase bridge inverter unit is alpha, and the phase difference between each inverter bridge arm triangular carrier of the C-phase bridge inverter unit and each inverter bridge arm triangular carrier corresponding to the A-phase bridge inverter unit is beta.
4. The modulation strategy of the marine microgrid inverter of claim 3, characterized in that: α =180 °, β =180 °.
5.A method for controlling the marine microgrid inverter according to claim 1, characterized in that: three-phase voltage u obtained by sampling oa (t)、u ob (t) and u oc (t) obtaining d-axis positive sequence fundamental wave voltage per unit value under the positive synchronous rotation coordinate system through DSC positive sequence, negative sequence fundamental wave voltage decomposition and dq conversion
Figure FDA0003808791170000021
q-axis positive sequence fundamental wave voltage per unit value
Figure FDA0003808791170000022
D-axis negative sequence fundamental wave voltage per unit value under reverse synchronous rotation coordinate system
Figure FDA0003808791170000023
And q-axis positive sequence fundamental wave voltage per unit value
Figure FDA0003808791170000024
And with
Figure FDA0003808791170000025
Participate in the inverter output positive sequence voltage control as a feedback value of a positive sequence voltage controller (82),
Figure FDA0003808791170000026
and
Figure FDA0003808791170000027
the feedback value of the negative sequence voltage controller (83) is used for participating in the output negative sequence voltage control of the inverter; the d-axis per unit value voltage reference value of the positive sequence voltage controller (82) is 1, and the feedback value is subtracted
Figure FDA0003808791170000028
A d-axis positive sequence voltage PI regulator (821) which is input to the positive sequence voltage controller (82), wherein the q-axis per-unit value voltage reference value of the positive sequence voltage controller (82) is 0, and the feedback value is subtracted
Figure FDA0003808791170000029
A q-axis positive sequence voltage PI regulator (822) which is input to the positive sequence voltage controller (82) later;
the d-axis per-unit value voltage reference value of the negative sequence voltage controller (83) is 0, and the feedback value is subtracted
Figure FDA00038087911700000210
A d-axis negative sequence voltage PI regulator (831) input to the negative sequence voltage controller (83), and the output of the d-axis negative sequence voltage PI regulator (831)
Figure FDA00038087911700000211
Multiplication by a cross-decoupling term ω C f After subtraction, the output is sent to a d-axis input port of a negative sequence dq to positive sequence dq conversion module (86); the q-axis per unit voltage reference value of the negative sequence voltage controller (83) is 0, and the feedback value is subtracted
Figure FDA00038087911700000212
A q-axis negative sequence voltage PI regulator (832) which is input to the negative sequence voltage controller (83), and the output of the q-axis negative sequence voltage PI regulator (832)
Figure FDA00038087911700000213
Multiplication by a cross-decoupling term ω C f After subtraction, the output is input to a q-axis input port of a negative sequence dq to positive sequence dq conversion module (85);
a negative-sequence dq to positive-sequence dq conversion module (85) converts-omega o The dq component of the negative sequence dq coordinate system of speed rotation passes through-2 omega o Conversion to ω o In a positive sequence dq coordinate system of speed rotation, the converted dq axis component is superposed with the dq axis component output of a positive sequence voltage controller (82) to obtain a d-axis current reference value of a current controller (84)
Figure FDA00038087911700000214
And q-axis current reference value
Figure FDA00038087911700000215
Figure FDA00038087911700000216
Plus d-axis output current
Figure FDA00038087911700000217
Feed forward minus d-axis inductor current feedback
Figure FDA00038087911700000218
The current is input to a d-axis current PI + R regulator (843) which is formed by adding a resonance control link on the basis of a PI regulator; q axis current
Figure FDA00038087911700000219
Cross decoupling term ω L m And an output voltage
Figure FDA00038087911700000220
The feedforward term is superposed with the output of a third harmonic proportional resonant PR controller (841) of the zero-sequence current of the primary side of the d-axis of the transformer to obtain a modulated wave d-axis component u d ；
Figure FDA00038087911700000221
Plus q-axis output current
Figure FDA00038087911700000222
Feed forward minus q-axis inductor current feedback
Figure FDA00038087911700000223
An input to a q-axis current PI + R regulator (844); d axis current
Figure FDA00038087911700000224
Cross decoupling term ω L m And an output voltage
Figure FDA00038087911700000225
A feedforward term is superposed with the output phase of a third harmonic proportional resonant PR controller (842) of the transformer q-axis primary zero-sequence current to obtain a modulation wave q-axis component u q ；
For modulated wave d-axis component u d And q-axis component u q Carrying out inverse transformation to dq to obtain ABC three-phase modulation wave voltage u A 、u B And u C And then the driving pulse of the corresponding switching device is obtained through SPWM modulation.
CN202211006428.8A 2022-08-22 2022-08-22 Marine microgrid inverter, modulation strategy and control method Pending CN115411964A (en)

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN116418219A (en) * 2023-01-13 2023-07-11 哈尔滨工业大学（深圳） Isolation switch power supply design of proportion balance structure with low common mode electromagnetic noise

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN116418219A (en) * 2023-01-13 2023-07-11 哈尔滨工业大学（深圳） Isolation switch power supply design of proportion balance structure with low common mode electromagnetic noise

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