CN114268329B - Dual-frequency high-linearity demodulator - Google Patents
Dual-frequency high-linearity demodulator Download PDFInfo
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- CN114268329B CN114268329B CN202111523522.6A CN202111523522A CN114268329B CN 114268329 B CN114268329 B CN 114268329B CN 202111523522 A CN202111523522 A CN 202111523522A CN 114268329 B CN114268329 B CN 114268329B
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Abstract
The invention discloses a dual-frequency high-linearity demodulator, which comprises two mixing cores, a local oscillator quadrature generation network and a 90-degree mixing network, wherein the mixing cores are connected with the local oscillator quadrature generation network; the two frequency mixing cores are respectively used for mixing one half of received radio frequency input signals with local oscillator quadrature signals input by a local oscillator quadrature generation network to form two paths of intermediate frequency signals and outputting the two paths of intermediate frequency signals to the same 90-degree hybrid network; the local oscillator quadrature generation network is connected with the two frequency mixing cores and is used for converting a local oscillator single-ended input signal into four paths of local oscillator quadrature signals and then inputting the quadrature signals into the frequency mixing cores; and the 90-degree hybrid network is connected with the two frequency mixing cores and is used for synthesizing and outputting the two paths of demodulated intermediate frequency signals. The invention has scientific structural design, can work in two working frequency bands of 24-30 GHz and 36-42 GHz simultaneously, has better linearity and mirror image inhibition degree in the whole frequency band, and meets the use requirement of a multi-standard receiver.
Description
Technical Field
The invention relates to the technical field of integrated circuits, in particular to a dual-frequency high-linearity demodulator.
Background
In recent years, application of the millimeter wave band to 5G communication has gradually become a hotspot. Compared with the current Sub-6G system, the communication is carried out in the millimeter wave frequency band, so that larger bandwidth and larger communication speed can be realized, various communication applications can be supported, and for communication, the high-frequency communication can effectively reduce the size requirement on an antenna, and the miniaturization of equipment is easier to realize.
In order to better realize the reception of the high-frequency signal, a mixer is needed to perform frequency conversion, and in particular, in a variable frequency receiver, a demodulator is needed to shift the high-frequency signal to a low frequency so as to realize the transmission of the signal.
In the conventional receiver, the Hartley receiver architecture and the Weffy receiver architecture are widely applied to an image rejection system, so that image signals can be effectively suppressed, and efficient transmission of the signals is realized. However, with the increase of the operating frequency, it has become increasingly difficult to meet the use requirements of discrete components and module circuits used in the conventional architecture, and with the current miniaturization and integration requirements, it is very important to design a fully integrated multi-mode broadband demodulator for 5G millimeter wave communication applications.
Disclosure of Invention
The invention aims at overcoming the technical defects in the prior art and provides a dual-frequency high-linearity demodulator.
To this end, the invention provides a dual-frequency high linearity demodulator comprising two mixing cores, a local oscillator quadrature generation network and a 90 ° hybrid network, wherein:
the two frequency mixing cores are respectively used for mixing one half of received radio frequency input signals with local oscillator quadrature signals input by a local oscillator quadrature generation network to form two paths of intermediate frequency signals and outputting the two paths of intermediate frequency signals to the same 90-degree hybrid network;
the local oscillator quadrature generation network is connected with the two frequency mixing cores and is used for converting a local oscillator single-ended input signal into four paths of local oscillator quadrature signals, and then respectively outputting the two paths of quadrature signals to a frequency mixer in one frequency mixing core, so that the quadrature signals and the radio frequency input signal are mixed;
the 90-degree hybrid network is connected with the two frequency mixing cores and is used for receiving four paths of intermediate frequency signals input by the two frequency mixing cores, respectively combining the two paths of intermediate frequency signals of each frequency mixing core to form two paths of combined intermediate frequency signals, and outputting the two paths of intermediate frequency signals outwards.
Compared with the prior art, the dual-frequency high-linearity demodulator provided by the invention has scientific structural design, can work at two working frequency bands of 24-30 GHz and 36-42 GHz at the same time, has better linearity and mirror image suppression degree in the whole frequency band, has better balance among power consumption, gain, isolation and design cost, meets the use requirement of a multi-standard receiver, and has great practical significance.
Drawings
Fig. 1 is a block diagram of a dual-band high linearity demodulator according to the present invention;
FIG. 2 is a schematic circuit diagram of a dual-band high linearity demodulator according to the present invention;
FIG. 3 is a diagram showing simulation results of conversion gain with frequency variation of a RF signal in a specific embodiment of a dual-band high linearity demodulator according to the present invention;
FIG. 4 is a diagram showing simulation results of the image suppression degree according to the frequency variation of the RF signal in the embodiment of the dual-band high-linearity demodulator according to the present invention;
fig. 5 is a diagram of simulation results of IP1dB versus frequency of a radio frequency signal in a specific embodiment of the dual-band high linearity demodulator provided by the present invention;
fig. 6a is a diagram of a spectrum simulation result at a fixed frequency point in a specific embodiment of a dual-band high linearity demodulator according to the present invention;
fig. 6b is a second diagram of a spectrum simulation result at a fixed frequency point in a specific embodiment of the dual-band high linearity demodulator provided by the present invention.
Detailed Description
The technical solutions of the present invention will be clearly and completely described in conjunction with the embodiments of the present invention, and it is apparent that the described embodiments are only some embodiments of the present invention, not all embodiments. All other embodiments, which can be made by those skilled in the art based on the embodiments of the invention without making any inventive effort, are intended to be within the scope of the invention.
In the description of the present invention, it should be understood that the terms "center", "longitudinal", "lateral", "upper", "lower", "front", "rear", "left", "right", "vertical", "horizontal", "top", "bottom", "inner", "outer", etc. indicate orientations or positional relationships based on the orientations or positional relationships shown in the drawings, are merely for convenience in describing the present invention and simplifying the description, and do not indicate or imply that the devices or elements referred to must have a specific orientation, be configured and operated in a specific orientation, and thus should not be construed as limiting the present invention. Furthermore, the terms "first," "second," and the like, are used for descriptive purposes only and are not to be construed as indicating or implying a relative importance or implicitly indicating the number of technical features indicated. Thus, a feature defining "a first", "a second", etc. may explicitly or implicitly include one or more such feature. In the description of the present invention, unless otherwise indicated, the meaning of "a plurality" is two or more.
In the description of the present invention, it should be noted that, unless explicitly specified and limited otherwise, the terms "mounted," "connected," and "connected" are to be construed broadly, and may be either fixedly connected, detachably connected, or integrally connected, for example; can be mechanically or electrically connected; can be directly connected or indirectly connected through an intermediate medium, and can be communication between two elements. The specific meaning of the above terms in the present invention can be understood by those of ordinary skill in the art in a specific case.
The invention will be described in detail below with reference to the drawings in connection with embodiments.
Referring to fig. 1 to 6b, the present invention provides a dual-band high linearity demodulator comprising two mixing cores 100, an In-phase component/Quadrature component (I/Q) generating network 200 and a 90 ° hybrid network 300, wherein:
the two frequency mixing cores 100 are respectively configured to mix one half of the received radio frequency input signals with local oscillator quadrature (I/Q) signals input by the local oscillator quadrature (I/Q) generation network 200, and form two paths of intermediate frequency signals to output to the same 90 ° hybrid network;
a local oscillator quadrature (I/Q) generation network 200 connected to the two frequency mixing cores 100, for converting a local oscillator single-ended input signal into four local oscillator quadrature signals, and then outputting the two local oscillator quadrature signals to the frequency mixer 101 in one frequency mixing core 100, respectively, so as to mix the quadrature signals with the radio frequency input signal;
the local oscillator single-ended input signal may be provided by an external signal source connected to a signal receiving end of the local oscillator quadrature (I/Q) generation network 200.
The 90 ° hybrid network is connected to the two mixing cores 100, and is configured to receive four paths of intermediate frequency signals input by the two mixing cores 100 (each mixing core 100 inputs two paths of intermediate frequency signals), and perform combination processing (specifically, performing in-phase signal superposition and image signal suppression processing) on the two paths of intermediate frequency signals of each mixing core 100, so as to form two paths of combined intermediate frequency signals, and then output the two paths of intermediate frequency signals, where the output intermediate frequency signals can be processed by performing secondary mixing to baseband through an external device or directly connected to baseband for processing.
It should be noted that, for the present invention, the single-ended signal (i.e. the rf input signal) input by the rf is directly split into two parts, and is input to two mixing cores respectively, the local oscillator single-ended input signal is converted into four-way quadrature signal via the I/Q generating network, and mixed with the rf input signal, the upper and lower intermediate frequency signals output by the two mixing cores 100 are synthesized by the 90 ° hybrid network, so as to realize in-phase signal superposition, image signal suppression, and finally the intermediate frequency signals demodulated by the upper and lower sidebands of the rf are output via the "intermediate frequency output 1" port and the "intermediate frequency output 2" port respectively.
In the present invention, each mixing core 100 includes a mixer 101, a single slip transformer network 102 (i.e. the rf balun in fig. 1), and an intermediate frequency balun (i.e. the Ma Xiang Marchand balun) 103;
a single slip transformer network 102 for converting one half of the rf input signal into a differential rf input signal and outputting the differential rf input signal to the mixer 101;
a mixer 101 for receiving differential RF input signals, mixing with local oscillator quadrature signals input from a local oscillator quadrature (I/Q) generation network 200 to form two paths of differential intermediate frequency signals, and outputting the two paths of differential intermediate frequency signals to
Intermediate frequency balun (Ma Xiang Marchand balun) 103 is used for converting two paths of differential intermediate frequency signals into two paths of single-ended intermediate frequency signals, and outputting the two paths of single-ended intermediate frequency signals to the 90-degree hybrid network.
In a specific implementation, the mixer 101 comprises a radio frequency transconductance stage and a local oscillation switching stage;
the radio frequency transconductance stage of mixer 101 comprises transistor M 1 ～M 4 Capacitance C 1 ～C 4 Inductance L 1 ～L 4 ；
Wherein the inductance L 1 And L 3 The coupling forms a transformer;
inductance L 2 And L 4 The coupling becomes a transformer;
one end of the capacitor C1 and one end of the capacitor C3 are connected with the differential radio frequency signal end RF+;
the other end of the capacitor C1 is connected with one end of the resistor R1 and the transistor M respectively 3 Is connected with the grid electrode of the power supply;
the other end of the capacitor C3 is connected with one end of the resistor R3 and the transistor M respectively 1 Is connected with the grid electrode of the power supply;
the other end of the resistor R1 is connected with an auxiliary voltage end Va, and the auxiliary voltage end Va is used for supplying power to the transconductance stage.
The other end of the resistor R3 is connected to a main voltage terminal Vm, which is used to supply power to the transconductance stage.
Transistor M 3 And the drain electrode of transistor M 1 The drain electrode of the transistor is connected;
transistor M 1 Through inductance L 1 Grounding;
transistor M 3 Is grounded;
transistor M 1 Drain of (d) and transistor M 3 After the drain electrodes of the capacitors are converged, the inductor L is used for 3 Transistor M in local oscillator switching stage with mixer 101 5 And M 6 The source electrodes of the electrodes are connected;
in the concrete implementation, one end of the capacitor C2 and one end of the capacitor C4 are connected with a differential radio frequency signal end RF-;
the other end of the capacitor C2 is connected with one end of the resistor R2 and the transistor M respectively 4 Is connected with the grid electrode of the power supply;
the other end of the capacitor C4 is connected with one end of the resistor R4 and the transistor M respectively 2 Is connected with the grid electrode of the power supply;
the other end of the resistor R2 is connected with an auxiliary voltage end Va, and the auxiliary voltage end Va is used for supplying power to the transconductance stage.
The other end of the resistor R4 is connected to a main voltage terminal Vm, which is used to supply the transconductance stage.
Transistor M 4 And the drain electrode of transistor M 2 The drain electrode of the transistor is connected;
transistor M 2 Through inductance L 2 Grounding;
transistor M 4 Is grounded;
transistor M 2 Drain of (d) and transistor M 4 After the drain electrodes of the capacitors are converged, the inductor L is used for 4 Transistor M in local oscillator switching stage with mixer 101 7 And M 8 The source electrodes of the electrodes are connected;
in particular implementation, the local oscillator switching stage of the mixer 101 comprises a transistor M 5 ～M 8 Inductance L 5 ～L 6 ；
Transistor M 5 And M 6 Drain electrodes of (a) are respectively connected with the inductor L 5 And L 6 One end of the first part is connected;
inductance L 5 And L 6 The other end of the power supply circuit is connected with the total power supply voltage VDD;
transistor M 5 The grid electrode of the oscillator is connected with one local oscillator quadrature signal LO+ and is used for realizing the input of local oscillator signals;
transistor M 7 Drain electrodes of (a) are respectively connected with the inductor L 5 Is connected with the intermediate frequency output IF+;
transistor M 8 Drain electrodes of (a) are respectively connected with the inductor L 6 Is connected with the intermediate frequency output IF-;
transistor M 7 Gate of (c) and transistor M 6 Is connected with the grid electrode of the local oscillator quadrature signal LO-at the same time;
transistor M 8 The grid electrode of the oscillator is connected with one local oscillator quadrature signal LO+ and is used for realizing the input of local oscillator signals;
it should be noted that, the rf input signal is converted into a differential signal through the single slip transformer network 102, and then passes through the transistor M in the mixer 101 1 ～M 4 Gate access to (c); capacitor C 1 ～C 4 The method is mainly used for isolating straight-through traffic; inductance L 1 ～L 4 On the one hand, is used for reducing the transistor M 1 ～M 4 On the other hand, to improve the linearity of the transconductance stage. The input radio frequency signal RF (i.e. differential radio frequency signal) passes through the transistor M 1 ～M 4 The drain electrode output of (1) is a current signal, and enters a local oscillation switching stage of the mixer 101 to be mixed;
it should be noted that, the radio frequency signal RF (i.e. the differential radio frequency signal output from the radio frequency transconductance stage) passes through the switching tube M 5 ～M 8 The source electrode of the (a) enters a local oscillation switching stage, and a local oscillation signal LO passes through a switching tube M 5 ～M 8 The grid electrode of the (B) is fed into a local oscillation switching stage, and the intermediate frequency signal after mixing passes through a switching tube M 5 ～M 8 Is output to the medium frequency load inductance L 5 ～L 6 ；
It should be noted that, for the present invention, the mixer core 100 is designed based on the gilbert cell, the radio frequency transconductance stage of the mixer core 100 is designed based on the multi-transistor parallel technology, and includes a main circuit and an auxiliary circuit, an input signal enters from the gate of the transistor, is converted into a current signal and then output from the drain, and further enters the local oscillation switching stage of the mixer core 100 to perform mixing, and through the structural design of the main circuit and the auxiliary circuit, the third-order intermodulation of the mixer can be effectively improved. The source electrode of the main circuit transistor is connected with a source electrode degeneration inductance L 1 And drain inductance L 3 A weakly coupled transformer is formed to effectively increase the input 1dB compression point of the mixer.
In the invention, the single slip transformer network is realizedA network 102 comprising an inductor L 7 And inductance L 8 Formed transformer, matching capacitor C 5 ～C 6 Matching inductance L 9 ～L 10 ；
Capacitor C 5 And inductance L 7 And a single-ended RF signal input S in Is connected with each other;
it should be noted that the network is a balun, and the analysis of the balun is generally an ensemble for single-slipping the signal;
capacitor C 5 And inductance L 7 The other ends of the two are grounded;
capacitor C 6 And inductance L 8 And a differential signal terminal S out+ Is connected with each other;
capacitor C 6 And inductance L 8 And the other end of the differential signal terminal S out -a phase connection;
differential signal terminal S out+ And differential signal terminal S out Respectively coupled to differential radio frequency signal terminals RF + and RF-of mixer 101.
It should be noted that, the single-slip transformer network 102 is mainly used for converting an input single-ended signal (i.e. a single-ended rf signal) into a differential signal, where a single-ended signal end of the single-slip transformer network 102 is connected to one end of an rf input by two, and a differential signal end (S out+ S out- ) Rf+ and RF-connected to the mixer 101;
in the present invention, the intermediate frequency balun 103 comprises four quarter-wavelength transmission lines;
differential input terminal IF in+ And IF (IF) in- Connecting the quarter wavelength transmission lines (1) and (2) to ground, respectively;
single ended output IF out Two sections of serially connected quarter-wavelength transmission lines (3) and (4) are connected, and the opposite transmission lines are coupled, so that the conversion from a medium-frequency differential signal to a single-end signal can be realized.
The differential input terminal IF of the intermediate frequency balun 103 in+ And IF (IF) in- And intermediate frequency outputs if+ and IF-phases of mixer 101Single-ended output IF of intermediate frequency balun 103 out Connected to inputs IF1 and IF2 of 90 hybrid network 300.
It should be noted that, for the present invention, the intermediate frequency balun is designed by using Marchand (Ma Xiang) balun, so that the conversion from the differential signal with the lower intermediate frequency to the single-ended signal can be realized. The local oscillator I/Q (quadrature) generation network 200 is designed based on a transformer for converting differential signals into four-way quadrature signals. The 90 ° hybrid network 300 is based on a design mechanism of transmission line coupling, and by increasing inductance, the use line length is effectively reduced, while the amplitude phase error is reduced.
In the present invention, in a specific implementation, the local oscillator quadrature (I/Q) generating network 200 is designed based on a transformer, and is a passive network structure, so that conversion from a single-ended local oscillator signal to a four-way quadrature signal can be realized.
For local oscillator quadrature (I/Q) generation network 200, it includes inductance L 11 ～L 14 ；
Inductance L 11 One end is connected with the input signal IN+ and the other end is connected with the output I+;
inductance L 14 One end is connected with the input signal IN-and the other end is connected with the output I-;
inductance L 12 One end is connected with the output signal Q-and the other end is connected with the resistor R 5 Is connected to the ground;
inductance L 13 One end is connected with the output signal Q+ and the other end is connected with the ground through a resistor R6;
the input signals IN+ and IN-are IN constant amplitude inversion, the output signals I+, I-, Q-are IN constant amplitude, the phases are different by 90 degrees IN sequence, and the inductance L 11 And inductance L 12 Is mutually coupled with inductance L 13 And inductance L 14 Mutual coupling, thereby realizing the generation from a differential input signal to four-way orthogonal single-ended signals;
note that, since the local oscillator input is a single-ended signal, it is necessary to implement differential local oscillator signal generation through the same network as the single-slip transformer network 102, and then implement four-way quadrature single-ended signal generation through the I/Q generation network, and the input of the local oscillator quadrature generation network 200 may be provided by an external signal source, and outputs lo+ and LO-connected to the mixer 101.
In the present invention, in a specific implementation, the 90 ° hybrid network 300 is designed based on the principle of transmission line coupling, and is a four-port passive network, so that 90 ° phase shift of a target signal can be realized.
For a 90 hybrid network 300, it includes an inductance L 15 ～L 16 ；
Inductance L 15 One end of the capacitor C and the input signal IF1 7 Connected at the other end to the output OUT1 and the capacitor C 8 Are connected;
inductance L 16 One end of the capacitor C and the input signal IF2 8 Connected at the other end to the output OUT2 and the capacitor C 7 Are connected;
the IF1 and IF2 outputted from the mixer core 100 achieve sideband suppression through the 90 ° hybrid network 300, and inductance L 15 And inductance L 16 Mutual coupling to realize the change of the phase of the input signal, and a capacitor C 7 And C 8 OUT1 and OUT2 output from the 90 ° quadrature generation network 300 correspond to the intermediate frequency output 1 and the intermediate frequency output 2, respectively, using inductance values for reducing inductance.
In the invention, the whole topology is designed based on a Hartley architecture, and is improved on the basis, so that double-frequency work and image suppression are realized.
In the invention, aiming at broadband matching design, the local oscillator and the radio frequency port both adopt a transformer-based multi-order matching network, and better matching and gain flattening are realized by adjusting parameters of each element in the matching network.
In the invention, for dual-band operation, a local oscillator operating band is selected to be between radio frequency dual-bands, and the upper and lower sidebands are effectively utilized to realize simultaneous demodulation of the upper and lower sidebands.
In the invention, the implementation can be realized under a CMOS process, and the transistor is a field effect transistor.
In the invention, as shown in fig. 3, the simulation result diagram of the conversion gain along with the frequency change of the radio frequency signal in the embodiment of the invention is shown in the specification, the simulation result diagram shows that the intermediate frequency is 6GHz, the signal with the frequency of 24-30 GHz is output through an intermediate frequency output 1 port in a 90-degree hybrid network 300 after being demodulated, the signal with the frequency of 36-42 GHz is output through an intermediate frequency output 2 port in the 90-degree hybrid network 300 after being demodulated, the local oscillation frequency is 30-36 GHz, the lower sideband frequency conversion gain is-9.0 to-7.7 dB, and the upper sideband frequency conversion gain is-10.6 to-10.3 dB.
In the invention, as shown in fig. 4, the simulation result diagram of the image rejection degree along with the frequency change of the radio frequency signal in the embodiment of the invention is shown in the detailed implementation, the simulation result diagram of the image rejection degree along with the frequency change of the radio frequency signal in the embodiment of the invention is shown in the specification, the intermediate frequency is 6GHz, the signal with the radio frequency of 24-30 GHz is output through an intermediate frequency output 1 port in the 90-degree hybrid network 300 after being demodulated, the signal with the radio frequency of 36-42 GHz is output through an intermediate frequency output 2 port in the 90-degree hybrid network 300 after being demodulated, the local oscillation frequency is 30-36 GHz, the lower sideband image rejection ratio is more than 23dB, the optimal value is 31dB, the upper sideband image rejection ratio is more than 15dB, and the optimal value is 18dB.
In the present invention, as shown in fig. 5, the embodiment of the present invention is IP 1dB According to the simulation result graph of the frequency change of the radio frequency signals, the intermediate frequency is 6GHz, the signals with the radio frequency between 24 and 30GHz are output through an intermediate frequency output 1 port in the 90-degree hybrid network 300 after being demodulated, the signals with the radio frequency between 36 and 42GHz are output through an intermediate frequency output 2 port in the 90-degree hybrid network 300 after being demodulated, the local oscillation frequency is between 30 and 36GHz, and the in-band linearity is integrally larger than 6.3dBm, and can be optimally up to 9dBm.
In the invention, in particular implementation, as shown in fig. 6, a spectrum simulation result diagram under a fixed frequency point in the embodiment of the invention is shown, wherein the intermediate frequency of fig. 6a is 6GHz, the local oscillation frequency is 33GHz, the radio frequency is 27GHz, and the radio frequency input power is 0dBm; the intermediate frequency of FIG. 6b is 6GHz, the local oscillation frequency is 33GHz, the radio frequency is 39GHz, and the radio frequency input signal is 0dBm.
Compared with the prior art, the dual-frequency high-linearity demodulator provided by the invention has the following beneficial effects:
1. the dual-frequency high-linearity demodulator provided by the invention can realize signal demodulation in dual frequency bands by adopting a method of combining an active frequency mixing core with a passive broadband network.
2. The dual-frequency high-linearity demodulator provided by the invention integrally adopts an image rejection architecture, and can realize a better image rejection effect in dual frequency bands.
3. According to the dual-frequency high-linearity demodulator provided by the invention, good linearity can be realized in dual frequency bands by adopting the transconductance stage based on the source-drain weak coupling transformer.
In summary, compared with the prior art, the dual-frequency high-linearity demodulator provided by the invention has scientific structural design, can work in two working frequency bands of 24-30 GHz and 36-42 GHz at the same time, has better linearity and image suppression degree in the whole frequency band, has better balance among power consumption, gain, isolation and design cost, meets the use requirement of a multi-standard receiver, and has great practical significance.
The foregoing is merely a preferred embodiment of the present invention and it should be noted that modifications and adaptations to those skilled in the art may be made without departing from the principles of the present invention, which are intended to be comprehended within the scope of the present invention.
Claims (8)
1. A dual-band high linearity demodulator comprising two mixing cores (100), a local oscillator quadrature generation network (200) and a 90 ° hybrid network (300), wherein:
the two frequency mixing cores (100) are respectively used for mixing one half of received radio frequency input signals with local oscillator quadrature signals input by the local oscillator quadrature generation network (200) to form two paths of intermediate frequency signals and outputting the two paths of intermediate frequency signals to the same 90-degree hybrid network;
the local oscillator quadrature generation network (200) is connected with the two frequency mixing cores (100) and is used for converting a local oscillator single-ended input signal into four paths of local oscillator quadrature signals, and then outputting the two paths of quadrature signals to a frequency mixer (101) in one frequency mixing core (100) respectively, so that the quadrature signals and the radio frequency input signal are mixed;
the 90-degree hybrid network is connected with the two frequency mixing cores (100) and is used for receiving four paths of intermediate frequency signals input by the two frequency mixing cores (100), respectively combining the two paths of intermediate frequency signals of each frequency mixing core (100) to form two paths of combined intermediate frequency signals, and outputting the two paths of intermediate frequency signals outwards;
each mixing core (100) comprises a mixer (101), a single slip transformer network (102) and an intermediate frequency balun (103);
a single slip transformer network (102) for converting one half of the radio frequency input signal into a differential radio frequency input signal and then outputting to the mixer (101);
a mixer (101) for receiving differential RF input signals, mixing with local oscillator quadrature signals input by a local oscillator quadrature generation network (200), forming two paths of differential intermediate frequency signals, and outputting the signals to
And the intermediate frequency balun (103) is used for converting the two paths of differential intermediate frequency signals into two paths of single-ended intermediate frequency signals and outputting the two paths of single-ended intermediate frequency signals to the 90-degree hybrid network.
2. The dual-frequency high linearity demodulator as claimed in claim 1 wherein the mixer (101) includes a radio frequency transconductance stage and a local oscillator switching stage;
radio frequency transconductance stage of mixer (101) comprising transistor M 1 ～M 4 Capacitance C 1 ～C 4 Inductance L 1 ～L 4 ；
Wherein the inductance L 1 And L 3 The coupling forms a transformer;
inductance L 2 And L 4 The coupling becomes a transformer;
one end of the capacitor C1 and one end of the capacitor C3 are connected with the differential radio frequency signal end RF+;
the other end of the capacitor C1 is connected with one end of the resistor R1 and the transistor M respectively 3 Is connected with the grid electrode of the power supply;
the other end of the capacitor C3 is connected with one end of the resistor R3 and the transistor M respectively 1 Is connected with the grid electrode of the power supply;
the other end of the resistor R1 is connected with an auxiliary voltage end Va;
the other end of the resistor R3 is connected with a main voltage end Vm;
transistor M 3 And the drain electrode of transistor M 1 The drain electrode of the transistor is connected;
transistor M 1 Through inductance L 1 Grounding;
transistor M 3 Is grounded;
transistor M 1 Drain of (d) and transistor M 3 After the drain electrodes of the capacitors are converged, the inductor L is used for 3 Transistor M in local oscillator switching stage with mixer (101) 5 And M 6 The source electrodes of the electrodes are connected;
one end of the capacitors C2 and C4 is connected with the differential radio frequency signal end RF-;
the other end of the capacitor C2 is connected with one end of the resistor R2 and the transistor M respectively 4 Is connected with the grid electrode of the power supply;
the other end of the capacitor C4 is connected with one end of the resistor R4 and the transistor M respectively 2 Is connected with the grid electrode of the power supply;
the other end of the resistor R2 is connected with an auxiliary voltage end Va;
the other end of the resistor R4 is connected with a main voltage end Vm;
transistor M 4 And the drain electrode of transistor M 2 The drain electrode of the transistor is connected;
transistor M 2 Through inductance L 2 Grounding;
transistor M 4 Is grounded;
transistor M 2 Drain of (d) and transistor M 4 After the drain electrodes of the capacitors are converged, the inductor L is used for 4 Transistor M in local oscillator switching stage with mixer (101) 7 And M 8 Is connected with the source electrode of the transistor.
3. A dual frequency high linearity demodulator as claimed in claim 2 wherein the local oscillator switching stage of the mixer (101) includes a transistor M 5 ～M 8 Inductance L 5 ～L 6 ；
Transistor M 5 And M 6 Drain electrodes of (a) are respectively connected with the inductor L 5 And L 6 One end of the first part is connected;
inductance L 5 And L 6 The other end of the power supply circuit is connected with the total power supply voltage VDD;
transistor M 5 The grid electrode of the oscillator is connected with one local oscillator quadrature signal LO+;
transistor M 7 Drain electrodes of (a) are respectively connected with the inductor L 5 Is connected with the intermediate frequency output IF+;
transistor M 8 Drain electrodes of (a) are respectively connected with the inductor L 6 Is connected with the intermediate frequency output IF-;
transistor M 7 Gate of (c) and transistor M 6 Is connected with the grid electrode of the local oscillator quadrature signal LO-at the same time;
transistor M 8 Is connected with a local oscillator quadrature signal LO+.
4. The dual frequency high linearity demodulator as recited in claim 1 wherein the single slip transformer network (102) includes a capacitor consisting of an inductance L 7 And inductance L 8 Formed transformer, matching capacitor C 5 ～C 6 Matching inductance L 9 ～L 10 ；
Capacitor C 5 And inductance L 7 And a single-ended RF signal input S in Is connected with each other;
capacitor C 5 And inductance L 7 The other ends of the two are grounded;
capacitor C 6 And inductance L 8 And a differential signal terminal S out+ Is connected with each other;
capacitor C 6 And inductance L 8 And the other end of the differential signal terminal S out- Is connected with each other;
differential signal terminal S out+ And differential signal terminal S out Are connected to differential radio frequency signal terminals rf+ and RF-of the mixer (101), respectively.
5. The dual-frequency high linearity demodulator as claimed in claim 1 wherein the intermediate frequency balun (103) comprises four quarter wavelength transmission lines;
differential input terminal IF in+ And IF (IF) in- Connecting the quarter wavelength transmission lines (1) and (2) to ground, respectively;
single endOutput terminal IF out Two sections of serially connected quarter-wavelength transmission lines (3) and (4) are connected, and the opposite transmission lines are coupled.
6. The dual-frequency high linearity demodulator as claimed in claim 5 wherein the differential input IF of the intermediate frequency balun (103) in+ And IF (IF) in- Connected to the intermediate frequency outputs IF+ and IF-of the mixer (101), a single-ended output IF of the intermediate frequency balun (103) out Is connected to inputs IF1 and IF2 of the 90 hybrid network (300).
7. The dual frequency high linearity demodulator as recited in claim 1 wherein for a local oscillator quadrature generation network (200) including an inductance L 11 ～L 14 ；
Inductance L 11 One end is connected with the input signal IN+ and the other end is connected with the output I+;
inductance L 14 One end is connected with the input signal IN-and the other end is connected with the output I-;
inductance L 12 One end is connected with the output signal Q-and the other end is connected with the resistor R 5 Is connected to the ground;
inductance L 13 One end is connected to the output signal q+ and the other end is connected to ground with a resistor R6.
8. The dual-frequency high linearity demodulator as claimed in any of the claims 1 to 7, wherein for a 90 ° hybrid network (300), it comprises an inductance L 15 ～L 16 ；
Inductance L 15 One end of the capacitor C and the input signal IF1 7 Connected at the other end to the output OUT1 and the capacitor C 8 Are connected;
inductance L 16 One end of the capacitor C and the input signal IF2 8 Connected at the other end to the output OUT2 and the capacitor C 7 Are connected.
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