CN112532049B

CN112532049B - Power supply conversion system

Info

Publication number: CN112532049B
Application number: CN202010772969.6A
Authority: CN
Inventors: 金达; 熊雅红; 宿清华
Original assignee: Delta Electronics Inc
Current assignee: Delta Electronics Inc
Priority date: 2019-09-18
Filing date: 2020-08-04
Publication date: 2022-04-05
Anticipated expiration: 2040-08-04
Also published as: CN112532049A; CN112532051A; CN112532052B; CN112532050B; CN112532050A; CN112532051B; CN112532052A

Abstract

The present disclosure provides a power conversion system, which includes a power conversion circuit, a bootstrap power supply circuit, and at least N driving circuits. The power conversion circuit comprises an input end, an output end, N switching power conversion units and N nodes. N is an integer greater than 1. The switching power conversion unit includes a first switch and a second switch. The bootstrap power supply circuit comprises N bootstrap capacitors and N bootstrap switches, wherein the N bootstrap switches are sequentially connected in series, two ends of each bootstrap capacitor are respectively connected to the corresponding node and the second end of the corresponding bootstrap switch, and the first end of the Nth bootstrap switch receives a power supply voltage. The driving circuit is connected to the corresponding bootstrap capacitor and outputs two driving signals according to the voltage of the anode terminal of the bootstrap capacitor so as to control the switch of the corresponding switching power conversion unit.

Description

Power supply conversion system

Technical Field

The present disclosure relates to power conversion systems, and more particularly to a power conversion system having a bootstrap power supply circuit.

Background

In the existing non-isolated high-current step-down application, a 2-phase step-down circuit topology connected in parallel is mostly adopted, as shown in fig. 1, the current stress on each switching tube is reduced by parallel connection, and each phase of step-down circuit is driven by a switch with a duty ratio staggered by 180 degrees, so that the current ripple is reduced. However, the voltage transfer ratio between the output voltage and the input voltage of the buck circuit topology is the switching duty ratio, i.e., Vo is Vin × D. In the application occasion that the input voltage is larger than the output voltage, the switching duty ratio is correspondingly reduced, so that the voltage reduction circuit topology cannot work in an optimal state.

Therefore, how to develop a power conversion system that can improve the above-mentioned prior art is a urgent need.

Disclosure of Invention

The present disclosure is directed to a power conversion system, in which a power conversion circuit of the power conversion system is a multi-phase buck converter with an expandable duty ratio, and compared to the existing buck converter, the power conversion system can effectively increase the duty ratio under the same input/output condition, and correspondingly reduce the corresponding jump voltage variation of a switch during conduction or off-state, thereby reducing the switching loss and increasing the efficiency. In addition, the bootstrap power supply function is realized and the switch is correspondingly controlled by arranging the bootstrap power supply circuit and the driver, so that the applicability of the power conversion circuit is greatly improved, the miniaturization of a power conversion system product is facilitated, and the bootstrap power supply circuit has a simple structure and is low in cost.

To achieve the above objective, the present disclosure provides a power conversion system, which includes X power conversion circuits, X bootstrap power supply circuits, and at least N driving circuits, where X is an integer greater than or equal to 1. The power conversion circuit comprises an input end, an output end, N cascaded switch power conversion units and N nodes. The input end is used for receiving an input voltage. The output end is used for outputting output voltage. Each switching power conversion unit comprises a first switch and a second switch, the second switch is connected with the first switch in series and is grounded, the first switch of the 1 st switching power conversion unit is coupled with the input end, the first switches of the other switching power conversion units are sequentially connected with the first switch of the previous switching power conversion unit in series and are coupled, and N is a positive integer greater than or equal to 2. The nth node is located between the first switch of the nth switching power conversion unit and the first switch of the (N +1) th switching power conversion unit, the nth node is located between the first switch and the second switch of the nth switching power conversion unit, and N is a positive integer greater than or equal to 1 and less than N. The bootstrap power supply circuit comprises N bootstrap capacitors and N bootstrap switches, wherein the N bootstrap switches are sequentially coupled in series, a negative electrode end of the nth bootstrap capacitor is electrically connected to the nth node, a positive electrode end of the nth bootstrap capacitor is electrically connected to a second end of the nth bootstrap switch, a negative electrode end of the nth bootstrap capacitor is electrically connected to the nth node, a positive electrode end of the nth bootstrap capacitor is electrically connected to a second end of the nth bootstrap switch, and a first end of the nth bootstrap switch receives a power supply voltage. Each bootstrap power supply circuit is electrically connected to the corresponding N driving circuits, and in any bootstrap power supply circuit and the corresponding N driving circuits thereof, the driving circuit is connected to the corresponding bootstrap capacitor, and respectively supplies power to the driving of the corresponding first switch and second switch according to the positive terminal voltage and the power supply voltage of the bootstrap capacitor, and outputs a first driving signal and a second driving signal to control the corresponding first switch and second switch.

To achieve the above objective, the present disclosure provides a power conversion system, which includes a power conversion circuit, a bootstrap power supply circuit, and a driving circuit unit. The power conversion circuit comprises an input end, an output end and at least one switch bridge arm. The input end is used for receiving an input voltage. The output end is used for outputting output voltage. Each switch bridge arm comprises an upper switch, a middle switch, a lower switch and two nodes, wherein the upper switch, the middle switch and the lower switch are sequentially connected in series between the positive end and the negative end of the input end, the 1 st node is positioned between the upper switch and the middle switch, and the 2 nd node is positioned between the middle switch and the lower switch. The bootstrap power supply circuit comprises two bootstrap capacitors and two bootstrap switches, wherein the first end of the 2 nd bootstrap switch receives a power supply voltage, the second end of the 2 nd bootstrap switch is electrically connected to the first end of the 1 st bootstrap switch, the second end of the 1 st bootstrap switch is electrically connected to the positive terminal of the 1 st bootstrap capacitor, the negative terminal of the 1 st bootstrap capacitor is electrically connected to the 1 st node, the negative terminal of the 2 nd bootstrap capacitor is electrically connected to the 2 nd node, and the positive terminal of the 2 nd bootstrap capacitor is electrically connected to the second end of the 2 nd bootstrap switch. The driving circuit unit is electrically connected to the bootstrap power supply circuit and is configured to output a driving signal to control the switch in the switch bridge arm according to the voltage of the 2 nodes, the power supply voltage, and the at least one control signal.

Drawings

Fig. 1 is a schematic circuit diagram of a conventional 2-phase step-down circuit.

Fig. 2A is a schematic circuit structure diagram of a first power conversion circuit according to a preferred embodiment of the disclosure.

Fig. 2B is a schematic circuit structure diagram of a second power conversion circuit according to a preferred embodiment of the disclosure.

Fig. 3A and 3B are waveform diagrams of driving signals of switches in the first power conversion circuit at different duty ratios.

Fig. 3C is a waveform diagram of driving signals of switches in the second power conversion circuit at different duty ratios.

Fig. 4A is a schematic circuit structure diagram of a first power conversion system according to a first embodiment of the disclosure.

Fig. 4B is a waveform diagram showing the switch driving signals and the corresponding voltage variations in fig. 4A.

Fig. 5A is a schematic circuit structure diagram of a first power conversion system according to a second embodiment of the disclosure.

Fig. 5B is a waveform diagram showing the switch driving signals and the corresponding voltage variations in fig. 5A.

Fig. 5C is a schematic circuit structure diagram of a second power conversion system according to a third embodiment of the disclosure.

Fig. 5D is a schematic circuit structure diagram of a second power conversion system according to a fourth embodiment of the disclosure.

Fig. 6A is a schematic circuit structure diagram of a first power conversion circuit according to a fifth embodiment.

Fig. 6B is a schematic circuit structure diagram of a power circuit and a precharge circuit applied to the first power conversion circuit of fig. 6A according to a fifth embodiment of the disclosure.

Fig. 6C is a schematic circuit structure diagram of a second power conversion circuit according to a sixth embodiment.

Fig. 6D is a control sequence corresponding to fig. 6C.

Fig. 7A is a circuit structure diagram of a first power conversion circuit and a clamping circuit according to a seventh embodiment of the disclosure.

Fig. 7B is a circuit structure diagram of a first power conversion circuit and a clamping circuit according to an eighth embodiment of the disclosure.

Fig. 7C is a schematic circuit structure diagram of a first power conversion circuit and a clamping circuit according to a ninth embodiment of the disclosure.

Fig. 7D is a circuit structure diagram of a second power conversion circuit and a clamp circuit thereof according to a tenth embodiment of the disclosure.

Fig. 7E is a circuit structure diagram of a second power conversion circuit and a clamping circuit thereof according to an eleventh embodiment of the disclosure.

Fig. 7F is a schematic circuit diagram of a second power conversion circuit and a clamping circuit thereof according to a twelfth embodiment of the disclosure.

Fig. 8A is a schematic circuit structure diagram of a first power conversion circuit, a bootstrap power circuit, and a driving circuit according to a thirteenth embodiment of the disclosure.

Fig. 8B is a schematic circuit structure diagram of a first power conversion circuit, a bootstrap power circuit, and a driving circuit according to a fourteenth embodiment of the disclosure.

Fig. 8C is a schematic circuit structure diagram of a second power conversion circuit, a bootstrap power circuit, and a driving circuit according to a fifteenth embodiment of the disclosure.

Fig. 9A is a schematic view of a portion of a magnetic core assembly according to a preferred embodiment of the present disclosure.

Fig. 9B and 9C are schematic diagrams of ac magnetic flux directions of the magnetic core assembly of fig. 9A at different time periods within one switching cycle.

Fig. 9D is a schematic view of a portion of a magnetic core assembly according to another preferred embodiment of the present disclosure.

Fig. 9E is a schematic view of the ac flux direction of the core assembly of fig. 9D.

Fig. 9F is a schematic view of a portion of a magnetic core assembly according to another preferred embodiment of the present disclosure.

Fig. 10A is a schematic view of a portion of a magnetic core assembly according to another preferred embodiment of the present disclosure.

FIG. 10B is a schematic view of the AC flux direction of the magnetic core assembly of FIG. 10A at different time periods within a switching cycle.

Fig. 10C is a waveform diagram of an amount of change in ac magnetic flux in the core assembly of fig. 10A over one switching cycle.

Fig. 10D, 10G, 10H, and 10I are partial schematic structural views of various modifications of the core assembly of fig. 10A.

Fig. 10E and 10F are top views of fig. 10D, illustrating two winding methods of inductors.

Fig. 11A and 11B are schematic diagrams illustrating a circuit configuration in which a plurality of first power conversion circuits are alternately connected in parallel.

Fig. 11C is a schematic diagram illustrating a circuit structure in which a plurality of second power conversion circuits are alternately connected in parallel.

Fig. 12 is a schematic circuit diagram of a power conversion system including two first power conversion circuits.

Fig. 13A and 13B are schematic circuit diagrams of a precharge circuit of the power conversion system of fig. 12.

Fig. 14 is a schematic circuit diagram of a bootstrap power circuit of the power conversion system of fig. 12.

Fig. 15 is a schematic circuit diagram of a power conversion system including two second power conversion circuits.

Fig. 16 is a schematic circuit diagram of a precharge circuit of the power conversion system of fig. 15.

Description of reference numerals:

10. 100, and (2) a step of: controller

11. 12, 13, a1, a2, AX: power supply conversion circuit

21. 22: power supply circuit

31. 32, 33, 34, 35, 36, 37: pre-charging circuit

Cb. Cb1, Cb2, Cb10, Cb11, Cb12, Cvcc: capacitor with a capacitor element

SWA, SWC, SWAa, SWAb, SWCa, SWE, SWG, SWI, SWec, SWGc, SWEd, SWGd: first node

SWB, SWD, SWBa, SWBb, SWCa, SWF, SWH, SWJ, SWFc, SWHc, SWFd, SWHd: second node

M11, M12, M13, M11a, M12a, M11b, M12b, S11, S12, S13, S11c, S12c, S11d, S12 d: first switch

M21, M22, M23, M21a, M22a, M21b, M22b, S21, S22, S23, S21c, S22c, S21d, S22 d: second switch

SR1, SR2, SR3, SR1c, SR2c, SR1d, SR2 d: third switch

S1: the fourth switch

S2: fifth switch

S3: sixth switch

L1, L2, L1a, L2a, L1b, L2 b: inductance

Ts: period of switching

D. And (3) Daux: duty cycle

T1: magnetic element

T4, T2, T3, T31, T32, T3a, T3b, T3c, T3 d: winding wire

T2 a: first end

T2 b: second end

Vcc: supply voltage

D1: first diode

D2: second diode

D3, D3a, D3b, D31, D32, D3c, D3D: third diode

D4: fourth diode

D41, D42, MD 1', D43, D44: absorption diode

D51, D52, D53, D54: discharge diode

D61, D62, D63, D11, D12, D21, D22: bootstrap diode

D71, D72, D73, D74, D71c, D72c, D73c, D74c, D71D, D72D, D73D, D74D: isolation diode

MD 1: parasitic diode

C1: first capacitor

C2: second capacitor

C3, C31, C32, C3a, C3b, C3C, C3 d: third capacitor

C41, C42, C43, C44, Cin ', Cb', Cin1 ', Cb 1', Cb2 ', Cin 2': absorption capacitor

C51, C52, C53, C14, C15, C24, C25: bootstrap capacitor

Cin: input capacitance

R1: a first resistor

R2: second resistance

R3, R31, R32, R3a, R3b, R3c, R3 d: third resistance

R4, R41, R42, R4a, R4b, R4c, R4 d: fourth resistor

IC1, IC2, IC3, IC11, IC12, IC21, IC 22: driving circuit

P1, P2, P3: node point

VT1, VT2, VC1, VC2, VC3, VC31, VC32, Vcc1, Vcc2, Vcc3, VC3a, VC3b, Vcc1a, Vcc2a, Vcc1b, Vcc2b, Vcc11, Vcc12, Vcc21, Vcc 22: voltage of

Vin: input voltage

Vo: output voltage

Dri-M11, Dri-M12, Dri-M13, Dri-M21, Dri-M22, Dri-M23, Dri-S11, Dri-S12, Dri-S21, Dri-S22, Dri-SR1, Dri-SR 2: drive signal

PWM1, PWM2, PWM11, PWM12, PWM21, PWMX1, PWMX2, PWMY1, PWMY 2: control signal

4. 5: lower magnetic core

41. 42, 61, 62, 63, 64, 71, 72, 73, 74, 81, 82, 83, 84, 91, 92, 93, 94: wrapping post

51: first wrapping post

52: second wrapping post

53: third wrapping post

54: fourth wrapping post

43. 44, 95, 96: side column

55: first center pillar

56: second center pillar

65. 75, 85: center post

45. 57, 67, 77, 87, 97: substrate

Φ ac1, Φ ac2, Φ ac3, Φ ac 4: AC magnetic flux

to, t1, t2, t3, t 4: time of day

Detailed Description

Some exemplary embodiments that incorporate the features and advantages of the present disclosure will be described in detail in the specification which follows. It is to be understood that the disclosure is capable of various modifications in various embodiments without departing from the scope of the disclosure, and that the description and drawings are to be regarded as illustrative in nature, and not as restrictive.

Fig. 2A is a circuit structure diagram of a first power conversion circuit according to a preferred embodiment of the present disclosure, fig. 3A and 3B are waveform diagrams of driving signals of switches in the first power conversion circuit at different duty ratios, fig. 4A is a circuit structure diagram of a first power conversion system according to the first embodiment of the present disclosure, fig. 4B is a waveform diagram showing a switch driving signal and a corresponding voltage change in fig. 4A, fig. 5A is a circuit structure diagram of a first power conversion system according to a second embodiment of the present disclosure, and fig. 5B is a waveform diagram showing a switch driving signal and a corresponding voltage change in fig. 5A. As shown in fig. 4A and 5A, the first power conversion system of the present disclosure includes the first power conversion circuit, the power circuit and at least one pre-charge circuit as shown in fig. 2A.

The first power conversion circuit comprises an input end, an output end, N switching power conversion units and N-1 energy storage devices, wherein N is a positive integer greater than or equal to 2. The input terminal and the output terminal are respectively used for receiving an input voltage Vin and generating an output voltage Vo. Each switching power conversion unit comprises a first switch and a second switch which are connected in series, wherein the first switch and the second switch have a switching period and are periodically actuated along with the switching period, and the switching period has a duty ratio. The first switches of the 1 st switching power conversion unit are connected to the input end, and the first switches of the other switching power conversion units are sequentially connected in series to the first switch of the previous switching power conversion unit. N-1 energy storage devices are connected in series between the input end and the output end, two ends of each energy storage device are respectively provided with a first node SWA and a second node SWB, and N first switches are arranged between the nth energy storage device and the input end, wherein N is a positive integer which is more than or equal to 1 and less than or equal to N-1. The energy storage device may be, for example, but not limited to, a capacitor. The energy storage device is used for dividing the input voltage, stores energy and transmits the energy to the output end in a switching period, and the energy stored by the energy storage device is in direct proportion to the duty ratio. In some embodiments, the power conversion circuit further includes a controller 10, and the controller 10 is configured to output at least two control signals PWM1 and PWM2 to control the switches of the switching power conversion units in the first power conversion circuit, respectively. Therefore, the first power conversion circuit disclosed by the disclosure is a multi-phase buck converter with an expandable duty ratio, and compared with the existing buck circuit, the multi-phase buck converter can effectively improve the duty ratio under the condition of the same input and output, correspondingly reduce the jump voltage variation corresponding to the switch in the on or off state, reduce the switching loss, and improve the efficiency.

The first power conversion system further comprises a power circuit. The power circuit is electrically connected to the first power conversion circuit for receiving the input voltage Vin, and the power circuit includes a magnetic element T1, the magnetic element T1 may be, for example, but not limited to, an inductor or a transformer. In an embodiment, the power circuit may further output a power supply voltage Vcc to the power conversion circuit, so as to supply power to the control chip and the driving chip in the power conversion circuit.

The first power conversion system further includes N-1 precharge circuits. The N-1 pre-charging circuits are in one-to-one correspondence with the N-1 energy storage devices so as to respectively charge the corresponding energy storage devices. Each pre-charging circuit comprises a winding and a rectifying and filtering circuit which are connected. The winding is coupled to the magnetic element T1 to receive a switched voltage. The output of the rectifying and filtering circuit is electrically connected with a first node and a second node at two ends of the corresponding energy storage device so as to receive the converted voltage and output a charging voltage to the corresponding energy storage device, wherein the turn ratio of a winding of a pre-charging circuit for charging the nth energy storage device to a magnetic element is (N-N): and N is added.

Therefore, before the first power conversion circuit performs voltage conversion, the voltage on the magnetic element T1 in the first power circuit and the pre-charging circuit can be utilized to pre-charge the energy storage device, then the first power conversion circuit enters a soft start state of output voltage, and after the first switch is conducted, the terminal voltage stress of the second switch is lower, so that the second switch can meet the requirement by selecting a low-voltage-resistant switching device, the cost can be effectively reduced, the low-voltage-resistant switching device has low conduction internal resistance, the power conversion efficiency can be improved, and the loss can be reduced.

In addition, the specific embodiments of the power supply circuit and the precharge circuit vary depending on the type of the power supply circuit.

In an embodiment, the power circuit may be a voltage-reducing circuit, as shown in fig. 4A, the power circuit 21 includes a magnetic element T1, a fourth switch S1 and a fifth switch S2, one end of the fourth switch S1 is electrically connected to the input end, the other end of the fourth switch S1 is connected to one end of the fifth switch S2 and one end of the magnetic element T1, the other end of the fifth switch S2 is grounded, the other end of the magnetic element T1 outputs a positive end of a power supply voltage Vcc, wherein the fourth switch S1 and the fifth switch S2 are complementarily turned on. In this case, in each pre-charge circuit 31, the winding T2 has a first end and a second end, the rectifier filter circuit includes a first diode D1, a first capacitor C1, a second diode D2 and a second capacitor C2, the first end T2a of the winding T2 is connected to the anode of the first diode D1 and the cathode of the second diode D2, the second end T2b of the winding T2 is connected to the cathode of the first capacitor C1 and the anode of the second capacitor C2, the cathode of the first diode D1 and the anode of the first capacitor C1 are electrically connected to the corresponding first node SWA, and the anode of the second diode D2 and the cathode of the second capacitor C2 are electrically connected to the corresponding second node SWB, so as to pre-charge the energy storage device Cb. In practical applications, the capacitance values of the first capacitor C1 and the second capacitor C2 are adjusted to limit the charging current output from the rectifying-filtering circuit 31 to the energy storage device Cb, so as to limit the current. In some embodiments, the rectifier filter circuit 31 further includes a first resistor R1 and a second resistor R2, one end of the first resistor R1 is connected to the cathode of the first diode D1 and the anode of the first capacitor C1, the other end of the first resistor R1 is connected to the corresponding first node SWA, one end of the second resistor R2 is connected to the anode of the second diode D2 and the cathode of the second capacitor C2, and the other end of the second resistor R2 is connected to the corresponding second node SWB, so that the first resistor R1 and the second resistor R2 can serve as current limiting resistors to limit the charging current output to the energy storage device Cb. In addition, in some embodiments, the power circuit 21 further includes a capacitor Cvcc, one end of the capacitor Cvcc is connected to the other end of the magnetic element T1, i.e., the positive end of the power supply voltage Vcc, and the other end of the capacitor Cvcc is grounded.

In another embodiment, the power circuit may be a flyback circuit, as shown in fig. 5A, the power circuit 22 includes a magnetic element T1 and a sixth switch S3 connected in series, in each of the pre-charge circuits 32, the winding T3 is coupled to the magnetic element T1 in the forward direction, a first end of the winding T3 is connected to the anode of a third diode D3, the cathode of the third diode D3 is connected to the anode of a third capacitor C3 and electrically connected to the corresponding first node SWA, and a second end of the winding T3 is connected to the cathode of a third capacitor C3 and electrically connected to the corresponding second node SWB, so as to pre-charge the energy storage device Cb. In some embodiments, the rectifying-filtering circuit 32 further includes a third resistor R3 and a fourth resistor R4. One end of the third resistor R3 is connected to the cathode of the third diode D3 and the anode of the third capacitor C3, the other end of the third resistor R3 is connected to the corresponding first node SWA, one end of the fourth resistor R4 is connected to the second end of the winding T3 and the cathode of the third capacitor C3, and the other end of the fourth resistor R4 is connected to the corresponding second node SWB, so that the third resistor R3 and the fourth resistor R4 can be used as current-limiting resistors to limit the charging current output to the energy storage device Cb. In some embodiments, the power circuit 22 further includes a winding T4, a fourth diode D4, and a capacitor Cvcc, one end of the winding T4 is connected to the anode of the fourth diode D4, the cathode of the fourth diode D4 is connected to one end of the capacitor Cvcc and the power conversion circuit, and the other end of the winding T4 is connected to the other end of the capacitor Cvcc and grounded, so as to be reversely coupled to the magnetic element T1 through the winding T4 to generate the power supply voltage Vcc to power the power conversion circuit.

The following describes an example of an actual implementation of the first power conversion circuit when N is equal to 2, with reference to the embodiments shown in fig. 4A and 5A.

In the first embodiment of the disclosure shown in fig. 4A, N is equal to 2, the power conversion circuit 11 includes two switching power conversion units and an energy storage device, wherein the energy storage device is a capacitor Cb, two ends of the capacitor Cb have a first node SWA and a second node SWB, the 1 st switching power conversion unit includes a first switch M11, a second switch M21 and an inductor L1, and the 2 nd switching power conversion unit includes a first switch M12, a second switch M22 and an inductor L2. Referring to fig. 3A and 3B, Ts is a switching period, D is a duty cycle of the switching period, and fig. 3A and 3B respectively show the switching driving signals of the power conversion circuit 11 when the duty cycle D is less than 50% and the duty cycle D is greater than 50%, wherein the control timings of the first switch M11 and the M12 are staggered by 180 degrees, the driving signals of the first switch M11 and the second switch M21 are complementary, and the driving signals of the first switch M12 and the second switch M22 are complementary.

The winding T2 of the pre-charge circuit 31 has a first end T2a and a second end T2b, and the turns ratio of the magnetic element T1 to the winding T2 of the power circuit 21 is 2: 1. referring to the waveform diagram shown in fig. 4B, Daux is the duty cycle of the auxiliary power supply. When the fourth switch S1 is turned on and the fifth switch S2 is turned off, the voltage across the magnetic element T1 is Vin-Vcc, the voltage VT2 across the winding T2 is (Vin-Vcc)/2, the voltage at the first terminal T2a is positive, the voltage at the second terminal T2b is negative, the first diode D1 is turned on, and the voltage VC1 across the first capacitor C1 is (Vin-Vcc)/2. When the fourth switch S1 is turned off and the fifth switch S2 is turned on, the voltage VT1 across the magnetic device T1 is-Vcc, the voltage potential at the first terminal T2a is negative, the voltage potential at the second terminal T2b is positive, and the second diode D2 is turned on, so the voltage VC2 across the second capacitor C2 is Vcc/2. As can be seen from the above description, the superimposed voltage (Vin/2) between the first capacitor C1 and the second capacitor C2 charges the capacitor Cb through the first resistor R1 and the second resistor R2, so that the voltage between the two terminals of the capacitor Cb is Vin/2. When the first switch M11 is turned on, the voltage stress on the second switches M21 and M22 is the difference between the input voltage Vin and the voltage across the capacitor Cb (i.e., Vin-Vin/2 is Vin/2), and the second switches M21 and M22 can be switches with low breakdown voltage due to the low voltage stress.

In the second embodiment of the disclosure shown in fig. 5A, N is equal to 2, and the power conversion circuit 11 is the same as the power conversion circuit 11 shown in fig. 2A and fig. 4A, and therefore, the description thereof is omitted. The turns ratio of the magnetic element T1 of the power circuit 22 to the winding T3 of the pre-charge circuit 32 is 2:1, and winding T3 is positively coupled to the magnetic element T1. Referring to the waveform diagram shown in fig. 5B, when the sixth switch S3 is turned on, the voltage VT1 across the magnetic element T1 is Vin, the voltage across the winding T3 is Vin/2, and the third diode D3 is turned on, so the voltage VC3 across the third capacitor C3 is Vin/2. When the sixth switch S3 is turned off, the supply voltage Vcc is generated by the reverse coupling of the winding T4 and the magnetic element T1, and at the same time, the voltage VC3 across the third capacitor C3 charges the capacitor Cb through the third resistor R3 and the fourth resistor R4, so that the voltage across the capacitor Cb is Vin/2. When the first switch M11 is turned on, the voltage stress on the second switches M21 and M22 is the difference between the input voltage Vin and the voltage across the capacitor Cb1 (i.e., Vin-Vin/2 is Vin/2), and the second switches M21 and M22 can be switches with low breakdown voltage due to low voltage stress.

Fig. 2B is a schematic circuit structure diagram of a second power conversion circuit according to a preferred embodiment of the disclosure, fig. 3C is a waveform diagram of driving signals of switches in the second power conversion circuit at different duty ratios, fig. 5C is a schematic circuit structure diagram of a second power conversion system according to a third embodiment of the disclosure, and fig. 5D is a schematic circuit structure diagram of a second power conversion system according to a fourth embodiment of the disclosure. As shown in fig. 5C and 5D, the second power conversion system of the present disclosure includes the second power conversion circuit, the power circuit and at least one pre-charge circuit as shown in fig. 2B.

The second power conversion circuit comprises an input end, an output end, N switching power conversion units and N energy storage devices, wherein N is a positive integer greater than or equal to 2. The input terminal and the output terminal are respectively used for receiving an input voltage Vin and generating an output voltage Vo. Each switching power conversion unit comprises a first switch, a second switch and a third switch, and the first switch, the second switch and the third switch are periodically operated along a switching period, wherein the switching period has a duty ratio. The first end of the first switch is electrically connected to the input end, the first end of the energy storage device is electrically connected to the second end of the first switch, the second end of the energy storage device is electrically connected to the second end of the second switch and one end of the third switch, and the other end of the third switch is grounded. The first end of the second switch is electrically connected to the second end of the first switch of another switching power conversion unit.

The second power conversion system further comprises a power circuit. The power circuit is electrically connected to the second power conversion circuit for receiving the input voltage Vin, and the power circuit includes a magnetic element T1, the magnetic element T1 may be, for example, but not limited to, an inductor or a transformer. In an embodiment, the power circuit may further output a power supply voltage Vcc to the power conversion circuit, so as to supply power to the control chip and the driving chip in the power conversion circuit. As shown in fig. 5C and 5D, the power circuit can be a flyback circuit as shown in fig. 5A, and will not be described in detail here.

The second power conversion system may further include a plurality of pre-charge circuits, wherein the number of the pre-charge circuits is greater than or equal to 1 and less than or equal to N. Each pre-charging circuit comprises a winding and a rectifying and filtering circuit which are connected. The winding is coupled to the magnetic element T1 to receive a switched voltage. The output of the rectification filter circuit is electrically connected with a first node and a second node at two ends of the corresponding energy storage device so as to receive the conversion voltage and output the charging voltage to the corresponding energy storage device, wherein the turn ratio of a winding of a pre-charging circuit for charging each energy storage device to a magnetic element is 1: 2.

in one embodiment, the second power conversion system may include 1 pre-charge circuit outputting a charging voltage to the N energy storage devices. In another embodiment, the second power conversion system may include N pre-charge circuits, where the N pre-charge circuits are in one-to-one correspondence with the energy storage devices to charge the energy storage devices respectively. In another embodiment, a part of the N energy storage devices share a pre-charge circuit. The precharge circuit is described below by taking as an example the second power conversion circuit including 2 switching power conversion units, wherein the second power conversion circuit including 2 switching power conversion units includes two energy storage devices, such as energy storage capacitors Cb10 and Cb 11.

In one embodiment, as shown in fig. 5C, the second power conversion system of the present disclosure includes two

precharge circuits

36, 37. In the two

pre-charge circuits

36 and 37, the two windings T31 and T32 are both coupled to the magnetic element T1 in the forward direction, the two windings T31 and T32 have a first end and a second end, and the two rectifying and filtering circuits respectively include a third diode D31 and a third capacitor C31 and a third diode D32 and a third capacitor C32. A first end of the winding T31 is connected to the anode of the third diode D31, the cathode of the third diode D31 is connected to the anode of the third capacitor C31 and is electrically connected to the first node SWE of the energy storage device Cb10, and a second end of the winding T31 is electrically connected to the second node SWF of the energy storage device Cb10, so as to pre-charge the energy storage device Cb 10; a first end of the winding T32 is connected to the anode of the third diode D32, the cathode of the third diode D32 is connected to the anode of the third capacitor C32 and is electrically connected to the first node SWG of the energy storage device Cb11, and a second end of the winding T32 is electrically connected to the second node SWH of the energy storage device Cb11, so as to pre-charge the energy storage device Cb 11.

In another embodiment, the second power conversion system of the present disclosure includes a pre-charge circuit 35, in the pre-charge circuit 35, the winding T3 is coupled to the magnetic element T1 in a forward direction, the winding T3 has a first terminal and a second terminal respectively, and the rectifier-filter circuit includes a third diode D3 and a third capacitor C3. The first end of the winding T3 is connected to the anode of the third diode D3, the cathode of the third diode D3 is connected to the anode of the third capacitor C3, and two ends of the third capacitor C3 are electrically connected to two sets of isolation diodes (D73, D74) and (D71, D72), and the two sets of isolation diodes are used to respectively precharge the energy storage devices Cb10 and Cb 11. Specifically, the anode of the third capacitor C3 is electrically connected to the anodes of the two isolation diodes D71 and D73, the cathodes of the isolation diodes D71 and D73 are electrically connected to the corresponding first nodes SWE and SWG, respectively, the second end of the winding T3 is electrically connected to the cathodes of the two isolation diodes D72 and D74, and the anodes of the two isolation diodes D72 and D74 are electrically connected to the corresponding second nodes SWF and SWH, respectively, for pre-charging the energy storage devices Cb10 and Cb 11.

The following is an example of an actual implementation of the second power conversion circuit when N is equal to 2 in the embodiments shown in fig. 5C and 5D.

In a third embodiment of the present disclosure shown in fig. 5C, N is equal to 2, the second power conversion circuit 13 includes two switching power conversion units and two energy storage devices, wherein the 1 st switching power conversion unit includes a first switch S11, a second switch S22, a third switch SR1 and an inductor L1, the 2 nd switching power conversion unit includes a first switch S12, a second switch S21, a third switch SR2 and an inductor L2, the energy storage devices are capacitors Cb10 and Cb11, two ends of the capacitor Cb10 have a first node SWE and a second node SWF, and two ends of the capacitor Cb11 have a first node SWG and a second node SWH. Referring to fig. 3C, Ts is a switching period, D is a duty ratio of the switching period, wherein the first switch S11 and the second switch S22 are turned on and off synchronously, the first switch S12 and the second switch S21 are turned on and off synchronously, the control timings of the first switch S11 and the first switch S12 are staggered by 180 degrees, the driving signals of the first switch S11 and the third switch SR1 are complementary, and the driving signals of the first switch S12 and the third switch SR2 are complementary.

The turns ratios of the magnetic element T1 of the power circuit 22 to the windings T31 or T32 of the

pre-charge circuits

36 and 37 are both 2:1, and winding T31 is positively coupled to the magnetic element T1, and winding T32 is positively coupled to the magnetic element T1. Referring to the waveform diagram shown in fig. 5B, when the sixth switch S3 is turned on, the voltage VT1 across the magnetic element T1 is Vin, the voltages across the windings T31 and T32 are both Vin/2, and the third diodes D31 and D32 are turned on, so the voltages VC31 and VC32 across the third capacitors C31 and C32 are Vin/2. When the sixth switch S3 is turned off, the supply voltage Vcc is generated by the reverse coupling of the winding T4 and the magnetic element T1, and at the same time, the voltages across the third capacitors C31 and C32 charge the capacitors Cb10 and Cb11 through the third resistors R31/R32 and the fourth resistors R41/R42, so that the voltages across the capacitors Cb10 and Cb11 are Vin/2. When the first switch S11 and the second switch S22 are turned on, the voltage stress on the third switch SR1 is the difference between the input voltage Vin and the voltage across the capacitor Cb10 (i.e., Vin-Vin/2 is Vin/2), and when the first switch S12 and the second switch S21 are turned on, the voltage stress on the third switch SR2 is the difference between the input voltage Vin and the voltage across the capacitor Cb11 (i.e., Vin-Vin/2 is Vin/2), and the voltage stress is low, so the third switches SR1 and SR2 can be switches with low withstand voltage.

In the fourth embodiment of the present disclosure shown in fig. 5D, N is equal to 2, and the power conversion circuit 13 is the same as the power conversion circuit 13 shown in fig. 2B and fig. 5C, and therefore, the description thereof is omitted. The turns ratio of the magnetic element T1 of the power circuit 22 to the winding T3 of the pre-charge circuit 35 is 2:1, and winding T3 is positively coupled to the magnetic element T1. Referring to the waveform diagram shown in fig. 5B, the operating principle is the same, the voltage VC3 across the third capacitor C3 is Vin/2, the capacitor voltage VC3 charges the capacitor Cb10 through the third resistor R3, the fourth resistor R4, the isolation diodes D71 and D72, so that the voltage across the capacitor Cb10 is Vin/2, and the capacitor voltage VC3 charges the capacitor Cb11 through the third resistor R3, the fourth resistor R4, the isolation diodes D73 and D74, so that the voltage across the capacitor Cb11 is Vin/2. Other working principles are the same as those illustrated in fig. 5C and are not described herein again.

Of course, in the above embodiments, N may be a positive integer greater than or equal to 2, and is not limited to 2, that is, the power conversion circuit may be a 2-phase circuit and a 2-phase or more circuit. To facilitate the description of the variation of the power conversion system of the present disclosure with an increase in N, the power conversion system when N is equal to 3 is exemplified below.

Fig. 6A is a schematic circuit structure diagram of a first power conversion circuit according to a fifth embodiment, and fig. 6B is a schematic circuit structure diagram of a power circuit and a precharge circuit applied to the first power conversion circuit of fig. 6A according to a fifth embodiment of the present disclosure. As shown in fig. 6A and 6B, N is equal to 3, the first power conversion circuit 12 includes three switching power conversion units and two energy storage devices, where the energy storage devices are capacitors Cb1 and Cb2, two ends of the capacitor Cb1 have a first node SWA and a second node SWB, two ends of the capacitor Cb2 have a first node SWC and a second node SWD, the 1 st switching power conversion unit includes a first switch M11, a second switch M21 and an inductor L1, the 2 nd switching power conversion unit includes a first switch M12, a second switch M22 and an inductor L2, and the 3 rd switching power conversion unit includes a first switch M13, a second switch M23 and an inductor L3. The controller 10 generates two control signals PWM1 and PWM2 that are 180 degrees out of phase. The control signal PWM1 is used to control the on of the first switches M11 and M13, the control signal PWM2 is used to control the on of the first switch M12, the control signals of the second switches M21 and M23 are complementary to the control signal PWM1, and the control signal of the second switch M22 is complementary to the control signal PWM 2. The control method can also be extended to a power conversion circuit comprising N cascaded switching power conversion units and N-1 cascaded energy storage devices, wherein in the odd switching power conversion units, the first switch is controlled by the control signal PWM1, the control signal of the second switch is complementary to the control signal PWM1, and in the even switching power conversion units, the first switch is controlled by the control signal PWM2, and the control signal of the second switch is complementary to the control signal PWM 2.

Here, the circuit configurations of the power supply circuit 22 and the precharge circuit 33 are exemplified in the case where the power supply circuit is a flyback circuit, but not limited to this, and a multi-path precharge circuit when the power supply circuit is a step-down circuit may be constructed with reference to the structure shown in fig. 4A. Referring to FIG. 6B, the turn ratio of the magnetic element T1 to the winding T31 of the 1 st precharge circuit 33 is 3: 2, and the winding T31 is positively coupled to the magnetic element T1, the turns ratio of the magnetic element T1 to the winding T32 of the 2 nd precharge circuit 34 is 3: 1, and winding T32 is positively coupled to the magnetic element T1. The output of the 1 st precharge circuit 33 is electrically connected to the first node SWA and the second node SWB at both ends of the capacitor Cb1, and the output of the 2 nd precharge circuit 34 is electrically connected to the first node SWC and the second node SWD at both ends of the capacitor Cb 2. Therefore, the third capacitor C31 of the pre-charge circuit 33 charges the capacitor Cb1 to 2Vin/3 through the third resistor R31 and the fourth resistor R41, and the third capacitor C32 of the 2 nd pre-charge circuit charges the capacitor Cb2 to Vin/3 through the third resistor R32 and the fourth resistor R42. When the first switches M11 and M13 are turned on or when the first switch M12 is turned on, the voltage stress on the second switches M21, M22 and M23 is the difference between the input voltage Vin and the voltage across the capacitor Cb1 (i.e., Vin-2Vin/3 is Vin/3), and the second switches M21, M22 and M23 can be switches with low withstand voltage due to the low voltage stress.

Fig. 6C is a schematic circuit structure diagram of a second power conversion circuit according to a sixth embodiment, and fig. 6D is a control timing sequence corresponding to fig. 6C. As shown in fig. 6C, N is 3, the power conversion circuit 14 includes three switching power conversion units and three energy storage devices, where the energy storage devices are capacitors Cb10, Cb11, and Cb12, two ends of the capacitor Cb10 have a first node SWE and a second node SWF, two ends of the capacitor Cb11 have a first node SWG and a second node SWH, two ends of the capacitor Cb12 have a first node SWI and a second node SWJ, the 1 st switching power conversion unit includes a first switch S11, a second switch S22, a third switch SR1, and an inductor L1, the 2 nd switching power conversion unit includes a first switch S12, a second switch S23, a third switch SR2, and an inductor L2, and the 3 rd switching power conversion unit includes a first switch S13, a second switch S21, a third switch SR3, and an inductor L3. Referring to fig. 6D, Ts is a switching period, D is a duty ratio of the switching period, and the power conversion circuit 14 shown in fig. 6C is configured to drive the switches with the duty ratio D less than 50%, wherein the control signals of the first switch and the second switch in each switching power conversion unit are the same, the control signal of the third switch is complementary to the control signal of the second switch, and the first switch in each switching power conversion unit is out of phase by 120 degrees, that is, the three control signals PWM1, PWM2 and PWM3 generated by the controller 10 are out of phase by 120 degrees in sequence. The control method can also be expanded to include N switch power conversion units, N first switches of the N switch power conversion units are respectively controlled by N control signals, the N control signals are sequentially staggered by 360/N degrees, in any switch power conversion unit, the control signal of a second switch is the same as that of the first switch, and the control signal of a third switch is complementary with that of the first switch.

To achieve the pre-charging of the energy storage device in the second power conversion circuit in fig. 6C, in an embodiment, the second power conversion system may include 3 pre-charging circuits, and each pre-charging circuit may employ the pre-charging circuit shown in fig. 5C, that is, three pre-charging circuits are employed to respectively charge the energy storage device, such as the energy storage capacitors Cb10, Cb11 and Cb 12. Each of the pre-charge circuits includes a winding that is positively coupled to a magnetic element T1 in the power circuit 22, and the turns ratio of the magnetic element to each winding is 2: 1. In another embodiment, the second power conversion system may include 1 pre-charge circuit, and the pre-charge circuit may adopt a structure similar to that shown in fig. 5D, except that: three groups of isolation diodes are adopted, and form three groups of charging output ends to respectively charge the energy storage devices Cb10, Cb11 and Cb 12. Wherein the winding of the pre-charge circuit is positively coupled to the magnetic element T1 in the power circuit 22, and the turn ratio of the magnetic element to the one winding is 2: 1. In yet another embodiment, the second power conversion system may comprise 2 pre-charge circuits, one of which may employ the pre-charge circuit shown in fig. 5C, and the other one of which may employ the pre-charge circuit shown in fig. 5D to charge the energy storage devices Cb10, Cb11 and Cb12, respectively. In the above three embodiments, the operation principle is the same as that in fig. 5C and 5D, and the details are not repeated here.

The energy of the pre-charging circuit in the power conversion system is all from a magnetic element in the power circuit, and a voltage exists on the magnetic element and is coupled with a winding in the pre-charging circuit to provide energy for the pre-charging circuit. Moreover, the diodes in all of the precharge circuits in fig. 4A, 5C, 5D, and 6B, such as D1 and D2 in fig. 4A, D3 in fig. 5A, D31 and D32 in fig. 5C, D3 in fig. 5D, and D31 and D32 in fig. 6B, may be replaced by controllable switches.

In both the first power conversion circuit and the second power conversion circuit, the circuit wiring is complicated due to the existence of the energy storage device, the loop formed by the energy storage device and the switching element is relatively large, and accordingly, the peak voltage generated during the on or off process of the switch is also large, so that a clamping circuit is required to protect the grounded switch. In order to avoid the damage of the peak voltage generated in the process of switching on or off the switch to the grounding switch of the switching power conversion unit, clamping circuits are added at two ends of the grounding switch, each clamping circuit comprises an absorption circuit and a discharge circuit, the absorption circuit is used for absorbing the peak voltage at two ends of the grounding switch so as to protect the grounding switch, and the discharge circuit can feed back the energy generated at the moment of switching on and absorbed by the absorption circuit to an energy storage device in the circuit so as to reduce the energy loss. Specifically, in some embodiments, the power conversion system further includes N clamping circuits, each of which is connected to the ground switch of the corresponding switching power conversion unit to clamp the voltage across the corresponding ground switch.

Fig. 7A is a schematic circuit diagram of a first power conversion circuit and a clamping circuit according to a seventh embodiment of the disclosure, and fig. 7B is a schematic circuit diagram of a first power conversion circuit and a clamping circuit according to an eighth embodiment of the disclosure. As shown in fig. 7A and 7B, any one of the N switching power conversion cells of the first power conversion circuit corresponds to a clamping circuit, for example, for clamping the voltages across the second switches M21 and M22. When the duty ratio of the switching period of the first switch is less than or equal to 50%, as shown in fig. 7A, each clamping circuit includes an absorption circuit and a discharge circuit, the absorption circuit includes an absorption diode and an absorption capacitor, the discharge circuit includes a discharge diode, wherein the absorption diode is connected in series with the absorption capacitor, the anode of the absorption diode is connected to one end of the corresponding second switch, one end of the absorption capacitor is connected to the cathode of the absorption diode, and the other end of the absorption capacitor is connected to the other end of the corresponding second switch; one end of the discharge circuit is connected to the cathode of the absorption diode, and the other end of the discharge circuit is connected to the corresponding first node. When the duty ratio is greater than 50%, as shown in fig. 7B, the connection relationship of the cathodes of the discharge diodes of some of the clamps is different from that of the clamps when the duty ratio is less than or equal to 50%. Under the condition that the duty ratio is larger than 50%, the negative electrode of the discharge diode of the 1 st clamping circuit is connected to the corresponding first node, and the negative electrodes of the discharge diodes of the rest clamping circuits are connected to the positive input end of the power supply conversion circuit.

In the seventh embodiment of the present disclosure shown in fig. 7A, N is equal to 2, the duty ratio of the switching period of the first switch is less than or equal to 50%, and the power conversion circuit is the same as the power conversion circuit shown in fig. 2A. For the 1 st clamping circuit, a snubber circuit and a discharge circuit are included, the snubber circuit includes a snubber diode D41 and a snubber capacitor C41 coupled in series, the cathode of the snubber diode D41 is connected to the snubber capacitor C41, the snubber circuit is connected in parallel to both ends of the second switch M21 and is placed close to the second switch M21 on the layout of the printed circuit board to realize the shortest routing path with the switch M21; the discharge circuit includes a discharge diode D51, the anode of the discharge diode D51 is connected to the cathode of the sinking diode D41, and the cathode of the discharge diode D51 is connected to the first node SWA. When the first switch M11 is turned on, the voltage drop Vin/2 is instantaneously borne by the two ends of the second switch M21, and a spike voltage at the turn-on instant exists on the voltage drop, at this time, the absorption diode D41 is turned on, and the absorption capacitor C41 absorbs the spike voltage at the turn-on instant of the first switch M11; when the first switch M11 is turned off and the second switch M21 is turned on, the withstand voltage Vds across the second switch M21 drops, the snubber diode D41 is turned off in the reverse direction, the discharge diode D51 in the discharge circuit is turned on, and the energy in the snubber capacitor C41 is discharged through the discharge diode D51 and fed back to the energy storage device Cb in the power conversion circuit.

For the 2 nd clamp circuit, also include a snubber circuit and a discharge circuit, the snubber circuit includes a snubber diode D42 and a snubber capacitor C42 coupled in series, the negative pole of the snubber diode D42 is connected to the snubber capacitor C42, the snubber circuit is connected in parallel to both ends of the second switch M22, and is placed close to the second switch M22 on the printed circuit board layout, so as to realize the shortest routing path with the switch M22; the discharge circuit includes a discharge diode D52, the anode of which is connected to the cathode of the sinking diode D42, and the cathode of which is connected to the first node SWA. When the first switch M12 is turned on, the voltage drop Vin/2 is instantaneously borne by the two ends of the second switch M22, and a spike voltage at the turn-on instant exists on the voltage drop, at this time, the absorption diode D42 is turned on, and the absorption capacitor C42 absorbs the spike voltage at the turn-on instant of the first switch M12; when the voltage across the snubber capacitor C42 is higher than Vin/2, the discharge diode D52 is turned on, so that the energy on the snubber capacitor C42 is discharged through the discharge diode D52 and fed back to the energy storage device Cb in the switching converter. Similarly, the energy of the absorption capacitor is fed back to the energy storage device Cb in the power conversion circuit. Thereby, protection of the second switches M21 and M22 may be achieved.

In the prior art, in a conventional Buck circuit, the energy of an absorption capacitor is generally fed back to an input capacitor in the converter circuit, and the voltage clamp at two ends of a switch is Vin; in this embodiment, the energy of the absorption capacitor is fed back to the energy storage device Cb in the power conversion circuit, so that the loss of the peak energy can be reduced, and the efficiency of the power conversion circuit can be improved; and the voltage drop at the two ends of the energy storage device Cb is Vin/2 when the energy storage device works in a steady state, and the voltage drop at the two ends of the second switches M21 and M22 is clamped to be Vin/2, so that the second switch can meet the system requirement by selecting a switch with low voltage withstanding level, and the cost is reduced.

In the eighth embodiment of the present disclosure shown in fig. 7B, N is equal to 2, the duty ratio of the switching period of the first switch is greater than 50%, and the power conversion circuit is the same as the power conversion circuit shown in fig. 2A. The working principle of the 1 st clamping circuit for clamping and protecting the second switch M21 is similar to that of the clamping circuit shown in fig. 7A, and therefore, the description thereof is omitted. For the 2 nd clamp circuit shown in fig. 7B, in a state where the first switch M12 and the second switch M21 are both turned on, at the instant when the first switch M11 is turned on, the voltage drop Vin is instantaneously borne by both ends of the second switch M22, and a peak voltage at the turn-on instant exists on the voltage drop, at this time, the snubber diode D42 is turned on, and the snubber capacitor C42 snugs the peak voltage at the turn-on instant of the first switch M11; when the first switch M12 is turned off and the second switch M22 is turned on, the withstand voltage Vds across the second switch M22 drops, the absorption diode D42 is turned off in the reverse direction, the discharge diode D52 is turned on, and the energy on the absorption capacitor C42 is discharged through the discharge diode D52 and fed back to the input capacitor at the input end of the power conversion circuit.

In the embodiment shown in fig. 7A and 7B, the clamp circuit belongs to a DCD clamp circuit, for example, the absorption capacitor C41 and the absorption diode D41 are connected in parallel across the second switch M21, and the absorption loop has a shorter path and a better absorption effect. However, in this embodiment, a voltage ripple is superimposed on the dc voltage of Vin/2 at the first node SWA, and the voltage ripple causes a certain energy loss when the absorption capacitor C41 discharges to the energy storage device Cb through the discharge diode D51. In another embodiment, to avoid the energy loss, the present invention provides a CDC clamp circuit applied to the first power conversion circuit shown in fig. 2A. The CDC clamp may replace the DCD clamp placed across the second switch M21 of the first switching power conversion unit, while the DCD clamp is still used across the second switches (e.g., M22) of the other switching power conversion units.

Fig. 7C is a schematic circuit structure diagram of a first power conversion circuit and a clamping circuit according to a ninth embodiment of the disclosure. In the embodiment shown in FIG. 7C, N is equal to 2, and the power conversion circuit is the same as the power conversion circuit shown in FIG. 2A. The working principle of the 2 nd clamp circuit for clamping and protecting the second switch M22 is similar to that of the clamp circuit shown in fig. 7A and 7B, and therefore, the description thereof is omitted. As shown in fig. 7C, at the instant when the first switch M11 turns on, the voltage across the second switch M21 (i.e., the voltage at node SWB) abruptly changes to Vin/2, and there is a spike voltage at the instant of the turn on. In this embodiment, a CDC clamp circuit is connected in parallel to the second switch M21, and the CDC clamp circuit forms a mirror symmetry with the energy storage device Cb, the input capacitor Cin, and the parasitic diode MD1 at the two ends of the first switch M11. The CDC clamp circuit includes a snubber circuit including a snubber diode MD1 'and snubber capacitors Cb' and Cin ', one end of the snubber capacitor Cb' is connected in series to the positive terminal of the snubber diode MD1 ', one end of the snubber capacitor Cin' is connected in series to the negative terminal of the snubber diode, and the snubber circuit is connected in parallel across the second switch M21, and on a printed circuit board layout, the shortest path is realized between each element of the snubber circuit and between the snubber circuit and the second switch M21. In this embodiment, a discharge circuit is not additionally arranged, and the anode and the cathode of the absorption diode MD 1' are respectively connected to the first node and the input positive terminal, so that the functions of energy absorption and remote discharge can be realized by using the switch and the capacitor of the power conversion circuit.

The working principle is that when the first switch M11 is turned on, the spike voltage across the second switch M21 is absorbed by the absorption capacitors Cb 'and Cin' in the CDC clamp circuit, and at the same time, the absorption capacitors Cb 'and Cin' discharge to the energy storage device Cb and the input capacitor Cin, so as to achieve the clamp protection of the second switch M21, and in this embodiment, the voltage drop across the absorption capacitor Cin 'is clamped to Vin, and the voltage drop across the absorption capacitor Cb' is clamped to-Vin/2, so in this embodiment, the voltage drop across the second switch M21 is approximately clamped to Vin/2, and a second switch with a low withstand voltage level may also be used. In addition, in the embodiment, the capacitance values of the two capacitors Cb 'and Cin' have no special requirements except for the absorption requirement, and the design is simple and the loss is small.

Fig. 7D is a schematic circuit structure diagram of a second power conversion circuit and a clamp circuit thereof according to a tenth embodiment of the disclosure, fig. 7E is a schematic circuit structure diagram of a second power conversion circuit and a clamp circuit thereof according to an eleventh embodiment of the disclosure, and fig. 7F is a schematic circuit structure diagram of a second power conversion circuit and a clamp circuit thereof according to a twelfth embodiment of the disclosure. The clamping circuits of fig. 7D-7F may be applied to the second power conversion circuit shown in fig. 2B for clamping the voltages across the third switches SR1 and SR 2. As shown in fig. 7D and 7E, each of the N switching power conversion units of the second power conversion circuit corresponds to a clamp circuit, and each clamp circuit includes an absorption circuit and a discharge circuit, the absorption circuit includes an absorption diode and an absorption capacitor, the discharge circuit includes a discharge diode, wherein the absorption diode is connected in series with the absorption capacitor, the anode of the absorption diode is connected to the other end of the corresponding third switch, one end of the absorption capacitor is connected to the cathode of the absorption diode, and the other end of the absorption capacitor is connected to one end, i.e., the ground end, of the corresponding third switch; one end of the discharge circuit is connected to the cathode of the absorption diode, and the other end of the discharge circuit is connected to the first node of the corresponding switching power unit or the first node of the other switching power conversion unit.

In the embodiment of the disclosure shown in fig. 7D, N is equal to 2, the clamp circuit corresponding to the first switching power conversion unit includes a serially coupled absorption diode D43 and an absorption capacitor C43, the cathode of the absorption diode D43 is connected to the absorption capacitor C43, the absorption circuit is connected in parallel to two ends of the third switch SR1 and is disposed close to the third switch SR1 on the printed circuit board layout, so as to achieve the shortest routing path with the third switch SR 1; the discharge circuit includes a discharge diode D53, the anode of the discharge diode D53 is connected to the cathode of the absorption diode D43, and the cathode of the discharge diode D53 is connected to the first node SWG of the second switching power conversion unit. When the first switch S11 and the second switch S22 are turned on, the voltage drop Vin/2 is instantaneously borne across the third switch SR1, and a spike voltage at the turn-on instant exists on the voltage drop, at this time, the absorption diode D43 is turned on, and the absorption capacitor C43 absorbs the spike voltage at the turn-on instant of the first switch S11 and the second switch S22; when the voltage across the absorption capacitor C43 is higher than Vin/2, the discharge diode D53 is turned on, so that the energy on the absorption capacitor C43 is discharged through the discharge diode D53 and fed back to the energy storage device Cb11 of the second switching power conversion unit. Therefore, by utilizing the characteristic that the voltage drop at two ends of the energy storage device Cb11 is Vin/2 in steady-state operation, the voltage drop at two ends of the third switch SR1 is clamped to be Vin/2, and the system requirement can be met by selecting a switch with low voltage withstanding level.

The absorption circuit comprises an absorption diode D44 and an absorption capacitor C44 which are coupled in series, the cathode of the absorption diode D54 is connected to the absorption capacitor C54, the absorption circuit is connected to two ends of the third switch SR2 in parallel and is placed close to the third switch SR2 on the printed circuit board layout, so that the shortest routing path between the absorption circuit and the switch SR2 is realized; the discharge circuit includes a discharge diode D54, the anode of the discharge diode D54 is connected to the cathode of the sinking diode D44, and the cathode of the discharge diode D54 is connected to the first node SWE of the first switching power converting unit. When the first switch S12 and the second switch S21 are turned on, the voltage drop Vin/2 is instantaneously borne across the third switch SR2, and a spike voltage at the turn-on instant exists on the voltage drop, at this time, the absorption diode D44 is turned on, and the absorption capacitor C44 absorbs the spike voltage at the turn-on instant of the first switch S12 and the second switch S21; when the voltage across the absorption capacitor C44 is higher than Vin/2, the discharge diode D54 is turned on, so that the energy on the absorption capacitor C44 is discharged through the discharge diode D54 and fed back to the energy storage device Cb10 of the first switching power conversion unit. Therefore, by utilizing the characteristic that the voltage drop at two ends of the energy storage device Cb10 is Vin/2 in steady-state operation, the voltage drop at two ends of the third switch SR2 is clamped to be Vin/2, and the system requirement can be met by selecting a switch with low voltage withstanding level. Therefore, the third switches SR1 and SR2 can be protected, and absorbed peak energy is fed back to the energy storage devices Cb11 and Cb10 by using the absorption capacitors C43 and C44, so that the loss of the peak energy can be reduced, and the efficiency of the power conversion circuit is improved.

In the embodiment of the present disclosure shown in fig. 7E, N is equal to 2, and the clamp circuits corresponding to the first/second switching power conversion units are the same as the embodiment shown in fig. 7D, except that the connection point of the cathodes of the discharge diodes of the clamp circuits is different, that is, the feedback point at which the discharge circuits feed back the absorbed peak energy to the energy storage device is different. Taking the clamp circuit corresponding to the first switching power conversion unit as an example, the cathode of the discharge diode D53 is connected to the first node SWE. When the first switch S11 and the second switch S22 are turned on, a spike voltage at the moment of turning on the third switch SR1 turns on the absorption diode D43, and the absorption capacitor C43 absorbs the spike voltage; when the first switch S11 and the second switch S22 are turned off and the first switch S12 and the second switch S21 are turned on, the sinking diode D43 is turned off in the reverse direction, and the discharging diode D53 in the discharging circuit is turned on, so that the energy in the sinking capacitor C43 is discharged through the discharging diode D53 and fed back to the energy storage device Cb 10. Therefore, by utilizing the characteristic that the voltage drop at two ends of the energy storage device Cb10 is Vin/2 in steady-state operation, the voltage drop at two ends of the third switch SR1 is clamped to be Vin/2, and the system requirement can be met by selecting a switch with low voltage withstanding level. Similarly, the negative electrode of the clamp circuit corresponding to the second switching power conversion unit is connected to the first node SWG, and the principle is basically the same, and will not be described in detail herein.

In practical applications, the cathode of the discharge diode in each clamping circuit may be set at the first node of the corresponding switching power conversion unit or the first node of another switching power conversion unit according to practical situations, and is not limited herein.

In the embodiments shown in fig. 7D and 7E, a voltage ripple is superimposed on the dc voltage level of Vin/2 at the first nodes SWE and SWG, and the voltage ripple causes a certain energy loss when the absorption capacitors C43 and C44 discharge to the energy storage device Cb11 or Cb10 through the discharge diodes D53 and D54. In the embodiment shown in FIG. 7F, the CDC clamp shown in FIG. 7C may be used as the clamp. The CDC clamp circuit is arranged between the first node and the grounding end.

In the embodiment shown in FIG. 7F, N is equal to 2, and the power conversion circuit is the same as the power conversion circuit shown in FIG. 2B. The CDC clamp circuits corresponding to the first switching power conversion unit and the second switching power conversion unit are basically the same, and only the CDC clamp circuit corresponding to the first switching power conversion unit is taken as an example for description. The CDC clamp circuit includes a snubber circuit including a snubber diode SD1 ' and snubber capacitors Cb1 ' and Cin1 ', one end of the snubber capacitor Cb1 ' is connected in series to the positive terminal of the snubber diode, one end of the snubber capacitor Cin1 ' is connected in series to the negative terminal of the snubber diode, and the snubber circuit is connected in parallel across the third switch SR1, and is implemented with the shortest path between each element of the snubber circuit and the third switch SR1 on a printed circuit board layout. In this embodiment, a discharge circuit is not required to be additionally arranged, and the anode and the cathode of the absorption diode SD 1' are respectively connected to the first node SWE and the input positive terminal Vin + of the first switching power conversion unit, so that the functions of energy absorption and remote discharge by using the switch and the capacitor of the power conversion circuit can be realized.

Similar to the operation principle shown in fig. 7C, when the first switch S11 and the second switch S22 turn on the spike voltage across the third switch SR1 at the moment, the spike voltage is absorbed by the absorption capacitors Cb1 'and Cin 1' in the corresponding CDC clamping circuit, and at the same time, the absorption capacitors Cb1 'and Cin 1' discharge to the energy storage device Cb10 and the input capacitor Cin, so as to implement the clamping protection of the third switch SR 1. The CDC clamping circuit of the second switching power conversion unit also absorbs the voltage spike through the absorption capacitors Cb2 'and Cin 2', and at the same time, the absorption capacitors Cb2 'and Cin 2' discharge to the energy storage device Cb11 and the input capacitor Cin, so as to realize the clamping protection of the third switch SR 2.

In this embodiment, the voltage drop across the snubber capacitors Cin1 'and Cin 2' is clamped to Vin and the voltage drop across the snubber capacitors Cb1 'and Cb 2' is-Vin/2, so that in this embodiment, the voltage drop across the third switches SR1 and SR2 is approximately clamped to Vin/2, and a third switch with a low voltage withstanding level may be used as well. In addition, in the embodiment, the capacitance values of the capacitors Cb1 ' and Cin1 ' and Cb2 ' and Cin2 have no special requirements except for the absorption requirement, and the capacitor has simple design and small loss.

In addition, the diodes in the embodiments of the clamping circuit described above can be replaced by controllable switches, such as D41, D51, D42 and D52 in fig. 7A and 7B, MD1 ', D42 and D52 in fig. 7C, D43 and D44 in fig. 7D and 7E, and SD1 ' and SD2 ' in fig. 7E.

In addition, as shown in the first power conversion circuit of fig. 2A, the sources of the second switches M21 and M22 are grounded, and the sources of the first switches M11 and M12 are connected to the capacitor Cb and the drain of the second switch M22, respectively, so that in practical applications, the driving of the first switches M11 and M12 requires bootstrap power supply. In the second power conversion circuit shown in fig. 2B, the sources of the third switches SR1 and SR2 are grounded, the sources of the second switches S21 and S22 are respectively connected to the drains of the third switches SR2 and SR1, and the sources of the first switches S11 and S12 are respectively connected to the drains of the second switches S21 and S22, so that in practical applications, bootstrap power supply is required for driving the first switches S11 and S12 and the second switches S21 and S22. In some embodiments, the power conversion system further includes a bootstrap power circuit and a plurality of driving circuits to control and drive the first switch or the second switch in each power conversion circuit to realize bootstrap power supply.

Fig. 8A is a schematic circuit structure diagram of a first power conversion circuit, a bootstrap power circuit, and a driving circuit according to a thirteenth embodiment of the disclosure, and fig. 8B is a schematic circuit structure diagram of a first power conversion circuit, a bootstrap power circuit, and a driving circuit according to a fourteenth embodiment of the disclosure. The bootstrap power supply circuit and the driving circuit shown in fig. 8A and 8B can be applied to the first power conversion circuit. As shown in fig. 8A and 8B, the first power conversion circuit further includes N nodes, where the nth node Pn is located between the first switch of the nth switching power conversion unit and the first switch of the (N +1) th switching power conversion unit, and the nth node Pn is located between the first switch and the second switch of the nth switching power conversion unit. The bootstrap power supply circuit includes N bootstrap capacitors and N bootstrap diodes, wherein the N bootstrap diodes are sequentially coupled in series, a negative terminal of an nth bootstrap capacitor is electrically connected to an nth node Pn, a positive terminal of the nth bootstrap capacitor is electrically connected to a negative terminal of the nth bootstrap diode, a negative terminal of the nth bootstrap capacitor is electrically connected to the nth node Pn, a positive terminal of the nth bootstrap capacitor is electrically connected to a negative terminal of the nth bootstrap diode, and a positive terminal of the nth bootstrap diode receives a supply voltage Vcc. The N driving circuits are respectively connected to the corresponding bootstrap capacitors, respectively supply power to the drives corresponding to the first switch and the second switch according to the positive terminal voltage and the power supply voltage Vcc of the bootstrap capacitors, and output a first driving signal and a second driving signal to control the first switch and the second switch of the corresponding power conversion circuit. Therefore, the bootstrap power supply function can be realized and the switch is correspondingly controlled through the bootstrap power supply circuit and the driving circuit, the applicability of the power supply conversion circuit is greatly improved, the miniaturization of a power supply conversion system product is also facilitated, and the bootstrap power supply circuit is simple in structure and low in cost.

In some embodiments, an input terminal of the nth driving circuit is electrically connected to a cathode of the nth bootstrap diode and an nth node, an output terminal of the nth driving circuit outputs a first driving signal to correspondingly control the first switch of the nth switching power conversion unit, and a high level and a low level of the first driving signal output by the nth driving circuit are a voltage at an anode terminal of the nth bootstrap capacitor and a voltage at the nth node, respectively. The input end of the nth driving circuit is electrically connected to the cathode of the nth bootstrap diode and the nth node, the output end of the nth driving circuit outputs a first driving signal to correspondingly control the first switch of the nth switching power conversion unit, and the high level and the low level of the first driving signal output by the nth driving circuit are respectively the voltage of the anode end of the nth bootstrap capacitor and the voltage of the nth node.

In some embodiments, the input terminal of the nth driving circuit is further electrically connected to the power supply voltage and the ground terminal, the output terminal of the nth driving circuit further outputs a second driving signal to correspondingly control the second switch of the nth switching power conversion unit, and a high level and a low level of the second driving signal are respectively the voltage of the power supply voltage and the voltage of the ground terminal. The input end of the nth driving circuit is further electrically connected to the power supply voltage and the ground end, the output end of the nth driving circuit further outputs a second driving signal to correspondingly control the second switch of the nth switching power conversion unit, and the high level and the low level of the second driving signal are respectively the voltage of the power supply voltage and the voltage of the ground end.

In the embodiment of the present disclosure shown in fig. 8A, N is equal to 2. The supply voltage Vcc is connected to the positive terminal of the 2 nd bootstrap capacitor C52 via the 2 nd bootstrap diode D62, and the negative terminal of the bootstrap capacitor C52 is connected to the 2 nd node P2. When the second switch M22 is turned on, the node P2 is shorted to the power ground, the bootstrap diode D62 is turned on, and the positive terminal voltage Vcc1 of the bootstrap capacitor C52 is pulled up to Vcc (ignoring the diode turn-on voltage drop). When the second switch M22 is turned off, the node P2 floats to the ground, and the positive terminal voltage Vcc1 of the bootstrap capacitor C52 can be used to power the driving circuit of the first switch M12, thereby raising the potential of the driving of the first switch M12. The positive terminal voltage Vcc1 of the bootstrap capacitor C52 is connected to the positive terminal of the 1 st bootstrap capacitor C51 via the 1 st bootstrap diode D61, and the negative terminal of the bootstrap capacitor C51 is connected to the 1 st node P1. When the first switch M12 is turned on, the node P1 is shorted with the node P2, the bootstrap diode D61 is turned on, and the positive terminal voltage Vcc2 of the bootstrap capacitor C51 is pulled up to Vcc 1. When the first switch M12 is turned off, the positive terminal voltage Vcc2 of the bootstrap capacitor C51 is used to power the driving of the first switch M11, thereby raising the potential of the driving of the first switch M11. Therefore, each phase of power conversion circuit can realize the bootstrap power supply function only by utilizing one bootstrap diode and one bootstrap capacitor, the circuit is simple and low in cost, the availability of the circuit is greatly improved, and the miniaturization of the power conversion system is facilitated. In addition, the driving circuit IC1 receives the control signal PWM1, is connected to the bootstrap capacitor C51 and the power supply voltage Vcc, and outputs the first driving signal Dri-M11 and the second driving signal Dri-M21 according to the positive terminal voltage Vcc2 of the bootstrap capacitor C51 and the power supply voltage Vcc, respectively, to control the corresponding first switch M11 and the second switch M21. The driving circuit IC2 receives the control signal PWM2, is connected to the bootstrap capacitor C52 and the power supply voltage Vcc, and outputs the first driving signal Dri-M12 and the second driving signal Dri-M22 respectively according to the positive terminal voltage Vcc1 of the bootstrap capacitor C52 and the power supply voltage Vcc, so as to control the corresponding first switch M12 and the second switch M22. The timings of the first driving signals Dri-M11 and Dri-M12 correspond to the timings of the control signals PWM1 and PWM2, respectively, the high level and the low level of the first driving signal Dri-M11 are the positive terminal voltage of the 1 st bootstrap capacitor C51 and the voltage of the 1 st node P1, respectively, and the high level and the low level of the first driving signal Dri-M12 are the positive terminal voltage of the 2 nd bootstrap capacitor C52 and the voltage of the 2 nd node P2, respectively. In some embodiments, the driving circuit (IC1, IC2) is further connected to the controller 10 for receiving and outputting the driving signal according to the control signal (PWM1, PWM 2). In this embodiment, a first switch M11 (upper switch), a first switch M21 (middle switch), and a second switch M22 (lower switch) are electrically connected in series, one end of the upper switch is electrically connected to the positive input terminal, and the other end of the lower switch is electrically connected to the negative input terminal; the voltage at the positive terminal of the 2 nd bootstrap capacitor C52 is used for supplying power to the control signal of the middle switch, and the voltage at the positive terminal of the 1 st bootstrap capacitor C51 is used for supplying power to the control signal of the upper switch.

In the embodiment of the present disclosure shown in fig. 8B, N is equal to 3. The supply voltage Vcc is connected to the positive terminal of the 3 rd bootstrap capacitor C53 via the 3 rd bootstrap diode D63, and the negative terminal of the bootstrap capacitor C53 is connected to the 3 rd node P3. When the second switch M23 is turned on, the node P3 is shorted to the power ground, the bootstrap diode D63 is turned on, and the positive terminal voltage Vcc1 of the bootstrap capacitor C53 is pulled up to Vcc. When the second switch M23 is turned off, the node P3 floats to the ground, and the positive terminal voltage Vcc1 of the bootstrap capacitor C53 is used to power the driving of the first switch M13, thereby raising the potential of the driving of the first switch M13. The positive terminal voltage Vcc1 of the bootstrap capacitor C53 is connected to the positive terminal of the 2 nd bootstrap capacitor C52 via the 2 nd bootstrap diode D62, and the negative terminal of the bootstrap capacitor C52 is connected to the 2 nd node P2. When the first switch M13 is turned on, the node P2 is shorted with the node P3, the bootstrap diode D62 is turned on, and the positive terminal voltage Vcc2 of the bootstrap capacitor C52 is pulled up to Vcc 1. When the first switch M13 is turned off, the positive terminal voltage Vcc2 of the bootstrap capacitor C52 is used to power the driving of the first switch M12, thereby raising the potential of the driving of the first switch M12. The positive terminal voltage Vcc2 of the bootstrap capacitor C52 is connected to the positive terminal of the 1 st bootstrap capacitor C51 via the 1 st bootstrap diode D61, and the negative terminal of the bootstrap capacitor C51 is connected to the 1 st node P1. When the first switch M12 is turned on, the node P1 is shorted with the node P2, the bootstrap diode D61 is turned on, and the positive terminal voltage Vcc3 of the bootstrap capacitor C51 is pulled up to Vcc 2. When the first switch M12 is turned off, the positive terminal voltage Vcc3 of the bootstrap capacitor C51 is used to power the driving of the first switch M11, thereby raising the potential of the driving of the first switch M11. In addition, the driving circuit IC1 receives the control signal PWM1, is connected to the bootstrap capacitor C51 and the power supply voltage Vcc, and outputs two driving signals Dri-M11 and Dri-M21 according to the voltage Vcc3 and the power supply voltage Vcc on the bootstrap capacitor C51, respectively, so as to control the corresponding first switch M11 and second switch M21. The driving circuit IC2 receives the control signal PWM2, is connected to the bootstrap capacitor C52 and the power supply voltage Vcc, and outputs two driving signals Dri-M12 and Dri-M22 according to the positive terminal voltage Vcc2 of the bootstrap capacitor C52 and the power supply voltage Vcc, respectively, to control the corresponding first switch M12 and second switch M22. The driving circuit IC3 receives the control signal PWM1, is connected to the bootstrap capacitor C53 and the power supply voltage Vcc, and outputs two driving signals Dri-M13 and Dri-M23 according to the positive terminal voltage Vcc1 of the bootstrap capacitor C53 and the power supply voltage Vcc, respectively, to control the corresponding first switch M13 and second switch M23. Furthermore, the timing of the first driving signals Dri-M11 and Dri-M13 corresponds to the timing of the control signal PWM1, the timing of the first driving signal Dri-M12 corresponds to the timing of the control signal PWM2, the high level and the low level of the first driving signal Dri-M11 are the positive terminal voltage of the 1 st bootstrap capacitor C51 and the voltage of the 1 st node P1, respectively, the high level and the low level of the first driving signal Dri-M12 are the positive terminal voltage of the 2 nd bootstrap capacitor C52 and the voltage of the 2 nd node P2, respectively, and the high level and the low level of the first driving signal Dri-M13 are the positive terminal voltage of the 3 rd bootstrap capacitor C53 and the voltage of the 3 rd node P3, respectively. In some embodiments, the driving circuit (IC1, IC2, IC3) is further connected to the controller 10 for receiving and outputting the driving signal according to the control signal (PWM1, PWM 2).

Fig. 8C is a schematic circuit structure diagram of a second power conversion circuit, a bootstrap power circuit, and a driving circuit according to a fifteenth embodiment of the disclosure. The bootstrap power circuit and the driving circuit shown in fig. 8C can be applied to the second power conversion circuit. As shown in fig. 8C, the second power conversion circuit includes N switching legs, where each switching leg includes a first switch, a second switch, and a third switch coupled in series, and N is greater than or equal to 2. Each switch bridge arm comprises 2 nodes which are respectively a first node and a second node, wherein the 1 st node is positioned between the first switch and the second switch in the switch bridge arm, and the 2 nd node is positioned between the second switch and the third switch in the switch bridge arm. Taking the first switch leg as an example, the first switch leg is formed by coupling a first switch S11, a second switch S21 and a third switch SR2 in series, a1 st node SWE is located between the first switch S11 and the second switch S21, and a2 nd node SWH is located between the second switch S21 and the third switch SR 2. Each switch bridge arm corresponds to a bootstrap circuit, each bootstrap power supply circuit includes 2 bootstrap capacitors and 2 bootstrap diodes, wherein these two bootstrap diodes are coupled in series in proper order, the negative pole end of the 1 st bootstrap capacitor is electrically connected to the 1 st node of this switch branch, the positive pole end of the 1 st bootstrap capacitor is electrically connected to the negative pole of the 1 st bootstrap diode, the positive pole end of the 1 st bootstrap diode is electrically connected to the positive pole end of the 2 nd bootstrap capacitor, the negative pole end of the 2 nd bootstrap capacitor is electrically connected to the 2 nd node of this switch branch, the positive pole end of the 2 nd bootstrap capacitor is electrically connected to the negative pole of the 2 nd bootstrap diode, the positive pole of the 2 nd bootstrap diode receives supply voltage Vcc.

Referring to fig. 8C again, each switch bridge arm corresponds to 2 driving circuits, the 2 driving circuits are respectively connected to the corresponding bootstrap capacitors, and respectively supply power to the first switch, the second switch, and the third switch of the corresponding switch branch according to the voltage of the positive electrode of the bootstrap capacitor and the power supply voltage Vcc, and output the first driving signal, the second driving signal, and the third driving signal to control the first switch, the second switch, and the third switch of the corresponding switch bridge arm. Therefore, the bootstrap power supply function can be realized and the switch is correspondingly controlled through the bootstrap power supply circuit and the driving circuit, the applicability of the power supply conversion circuit is greatly improved, the miniaturization of a power supply conversion system product is also facilitated, and the bootstrap power supply circuit is simple in structure and low in cost.

In some embodiments, the input end of the 1 st driving circuit corresponding to each switch bridge arm is electrically connected to the cathode of the 1 st bootstrap diode and the 1 st node, the output end of the 1 st driving circuit outputs a first driving signal to correspondingly control the corresponding first switch, and the high level and the low level of the first driving signal output by the 1 st driving circuit are the voltage of the anode terminal of the 1 st bootstrap capacitor and the voltage of the 1 st node, respectively. The input end of the 2 nd driving circuit is electrically connected to the cathode of the 2 nd bootstrap diode and the 2 nd node, the output end of the 2 nd driving circuit outputs a second driving signal and a third driving signal to correspondingly control the corresponding second switch and the third switch, and the high level and the low level of the second driving signal output by the 2 nd driving circuit are respectively the anode terminal voltage of the 2 nd bootstrap capacitor and the voltage of the 2 nd node.

In the embodiment of the present disclosure shown in fig. 8C, N is equal to 2. The second power conversion circuit comprises two bridge arms, two bootstrap power circuits and four driving circuits. To explain with the first switch leg, a bootstrap power supply circuit and two driving circuits corresponding thereto, the power supply voltage VCC is connected to the positive terminal of the 2 nd bootstrap capacitor C14 through the 2 nd bootstrap diode D11, and the negative terminal of the bootstrap capacitor C14 is connected to the 2 nd node SWH of the switch leg. When the third switch SR2 in the first switch leg is turned on, the 2 nd node SWH is shorted to the power ground, the bootstrap diode D11 is turned on, and the positive terminal voltage Vcc11 of the bootstrap capacitor C14 is raised to Vcc (ignoring diode turn-on voltage drop). When the third switch SR2 is turned off, the 2 nd node SWH floats to the ground, and the positive terminal voltage Vcc11 of the bootstrap capacitor C14 can be used to power the driving circuit of the second switch S21, thereby raising the driving potential of the second switch S21. The positive terminal voltage Vcc11 of the bootstrap capacitor C14 is connected to the positive terminal of the 1 st bootstrap capacitor C15 through the 1 st bootstrap diode D12, and the negative terminal of the bootstrap capacitor C15 is connected to the 1 st node SWE of the switch branch. When the second switch S21 is turned on, the 1 st node SWE is shorted with the 2 nd node SWH, the bootstrap diode D12 is turned on, and the positive terminal voltage Vcc12 of the bootstrap capacitor C15 is pulled up to Vcc 11. When the second switch S21 is turned off, the positive terminal voltage Vcc12 of the bootstrap capacitor C15 is used to power the driving of the first switch S11, thereby raising the potential of the driving of the first switch S11. Therefore, each switch bridge arm can realize the bootstrap power supply function by utilizing two bootstrap diodes and two bootstrap capacitors, the circuit is simple and low in cost, the availability of the bootstrap diode is greatly improved, and the miniaturization of a power conversion system is facilitated. In the embodiment, a first switch S11 (an upper switch), a second switch S21 (a middle switch) and a third switch SR2 (a lower switch) on a switch bridge arm are electrically connected in series, one end of the upper switch is electrically connected with an input positive terminal, and the other end of the lower switch is electrically connected with an input negative terminal; the voltage at the positive terminal of the 2 nd bootstrap capacitor C14 is used for supplying power to the control signal of the middle switch, and the voltage at the positive terminal of the 1 st bootstrap capacitor C15 is used for supplying power to the control signal of the upper switch. Similarly, a first switch S12 (upper switch), a second switch S22 (middle switch) and a third switch SR1 (lower switch) on the other switch bridge arm are electrically connected in series, one end of the upper switch is electrically connected with the positive input terminal, and the other end of the lower switch is electrically connected with the negative input terminal; the voltage at the positive terminal of the 2 nd bootstrap capacitor C24 is used for supplying power to the control signal of the middle switch, and the voltage at the positive terminal of the 1 st bootstrap capacitor C25 is used for supplying power to the control signal of the upper switch.

In addition, the driving circuit IC11 receives the control signal PWM1, is connected to the bootstrap capacitor C15 and the power supply voltage Vcc, and outputs the first driving signal Dri-S11 according to the positive terminal voltage Vcc12 of the bootstrap capacitor C15 and the power supply voltage Vcc, so as to control the corresponding first switch S11. The driving circuit IC12 receives the control signal PWM2, is connected to the bootstrap capacitor C14 and the power supply voltage Vcc, and outputs the second driving signal Dri-S21 and the third driving signal Dri-SR2 respectively according to the positive terminal voltage Vcc11 of the bootstrap capacitor C14 and the power supply voltage Vcc, so as to control the corresponding second switch S21 and third switch SR 2. The timings of the first driving signal Dri-S11 and the second driving signal Dri-S21 correspond to the timings of the control signals PWM1 and PWM2, respectively, the timing of the third driving signal Dri-SR2 is complementary to the control signal PWM2, the high level and the low level of the first driving signal Dri-S11 are the positive terminal voltage of the 1 st bootstrap capacitor C15 and the voltage of the 1 st node SWE, respectively, and the high level and the low level of the second driving signal Dri-S21 are the positive terminal voltage of the 2 nd bootstrap capacitor C4 and the voltage of the 2 nd node SWH, respectively.

In some embodiments, the driving circuit (IC11, IC12) is further connected to the controller 10 for receiving and outputting the driving signal according to the control signal (PWM1, PWM 2). Similarly, the working principle of the bootstrap power circuit and the driving circuit corresponding to the second switch branch is the same as that of the first switch branch, and the description is omitted again. Similarly, when the power conversion circuit includes N switching branches, the same method can be used for bootstrap power supply and driving signal output. In addition, the bootstrap power supply and driving method is not limited to the circuit topologies shown in fig. 8A, fig. 8B and fig. 8C, as long as there are 3 switches electrically connected in series in the circuit topology, the 3 switches are respectively an upper switch, a middle switch and a lower switch, and the 3 switches are electrically connected to the input end of the power conversion circuit, wherein the lower switch and the middle switch are not in a conducting state at the same time, and the middle switch and the upper switch are not in a conducting state at the same time, the 3 switches electrically connected in series can adopt the bootstrap power supply and driving circuit disclosed in the present invention.

In the embodiments, the switches may be MOS, SiC or GaN, and further, the lower tube in the embodiments may be replaced by freewheeling diodes, and the positive electrodes of the freewheeling diodes are electrically connected to the negative output terminal. For example, the second switches M21, M22 in fig. 2A, 4A, 5A, 7B, and 8A may be replaced with freewheeling diodes, and the second switches M21, M22, and M23 in fig. 6A and 8B may be replaced with freewheeling diodes. In another embodiment, the diodes in the bootstrap circuits may be replaced by controllable switches, such as D61 and D62 in fig. 8A, and D61, D62 and D63 in fig. 8B, one by one, where the first ends of the controllable switches correspond to the anodes of the bootstrap diodes, the second ends of the controllable switches correspond to the cathodes of the bootstrap diodes, and the controllable switches may be MOS, SiC or GaN.

In addition, because the switching power conversion unit of the power conversion circuit is provided with the inductor, in order to realize the ultra-thinness of the power conversion system product, the magnetic core component wound by the inductor can be designed, so that the thickness of the substrate of the magnetic core component can be reduced only, and the height of the power conversion system product is further reduced. The following is an example of an embodiment of the magnetic core assembly in which N is 2.

Fig. 9A is a schematic view of a partial structure of a magnetic core assembly according to a preferred embodiment of the present disclosure, fig. 9B is a schematic view of an ac magnetic flux direction of the magnetic core assembly of fig. 9A, and fig. 9C is a waveform diagram of an ac magnetic flux in the magnetic core assembly of fig. 9A in one switching period. Please refer to fig. 9A and fig. 2A and 2B. Two inductors L1 and L2 of the two switching power conversion units are wound on the same magnetic core assembly, the magnetic core assembly includes two substrates, two winding posts and two side posts, the two winding posts and the two side posts are disposed on at least one substrate and located between the two substrates, the magnetic core assembly is formed by assembling an upper magnetic core and a lower magnetic core, and the following magnetic core 4 is taken as an example to explain the structure and the winding direction of the magnetic core. The lower core 4 includes two winding

posts

41 and 42, two

side posts

43 and 44, and a substrate 45, wherein the winding

posts

41 and 42 and the side posts 43 and 44 are disposed on the substrate 45, the two

side posts

43 and 44 are disposed on two opposite sides of the winding

posts

41 and 42, respectively, taking the view angle of the front view 9A as an example, the winding post 41 is disposed on the front side of the winding post 42, and the two

side posts

43 and 44 are disposed on the left and right sides of the winding

posts

41 and 42, respectively. In some embodiments, the center of the winding post 41 is equidistant from the center of the winding post 42 and the vertical distance between the two

side posts

43 and 44. In one embodiment, the air gaps may be formed only in the side posts 43 and 44, and the two air gaps may be approximately the same length; it is also possible to provide air gaps on the two

side legs

43 and 44 and the two winding

legs

41 and 42, respectively, and the air gaps on the two side legs are substantially the same length, the air gaps on the two winding legs are substantially the same length, and the air gap between the side legs is equal to or greater than the air gap between the winding legs.

Taking the example of fig. 2A in which the windings of the two inductors L1 and L2 are wound around the two winding

legs

41 and 42, respectively, the dc current directions are the same as those of the

windings

41 and 42 as shown in fig. 9A. As shown in fig. 9C, the duty ratio D of M11 and M12 is equal to 25%, and is 180 degrees out of phase, and the interval from time t0 to time t4 is a switching period Ts. In the interval from time t0 to time t1, the first switch M11 is turned on, the alternating-current magnetic flux Φ ac1 flowing through the spool 41 linearly rises, the alternating-current magnetic flux Φ ac1 is equal to 3 Φ at time t1, the direction of the alternating-current magnetic flux Φ ac2 flowing through the spool 42 is opposite to the direction of the alternating-current magnetic flux Φ ac1, and the alternating-current magnetic flux Φ ac2 is equal to- Φ at time t 1. Therefore, one third of the alternating magnetic flux flowing through the leg 41 flows into the leg 42, and as shown in fig. 9B, since the leg 41 is located at the center position between the two

side legs

43 and 44, that is, the center of the leg 41 is equal to the vertical distance between the two

side legs

43 and 44, the alternating magnetic flux Φ ac1 flows to the leg 42, the side leg 43, and the side leg 44, respectively, to be the same. From this, at time t1, ac magnetic flux Φ ac1 flowing through the leg 41 flows to the leg 42 and the two

side legs

43 and 44, respectively. Similarly, the ac magnetic flux at the time t0, t2, t3 and t4 can be derived, and therefore, the description thereof is omitted here. The foregoing magnetic flux variation Φ is merely illustrative and does not represent a specific magnitude. As can be seen from the above, since ac magnetic flux Φ ac1 flowing through winding post 41 is distributed in three directions of side post 43, side post 44, and winding post 42, ac magnetic flux can be shared equally or approximately shared, so as to reduce the thickness of substrate 45 and reduce the height of power conversion system product, thereby implementing ultra-thinning and improving its applicability, and also greatly improving the power density of power conversion system and reducing vertical thermal resistance. Note that, in this embodiment of the core assembly, the duty ratio D is not limited to 25%, and is applicable to a range of duty ratios of 15% or more and 35% or less, and the ac magnetic flux equalizing effect is best when the duty ratio D is 25%. In addition, the upper core of the core assembly may have the same structure as the lower core 4, or may be an I-type core, and the winding directions of the inductors L1 and L2 are not limited to the examples of the drawings, as long as the ac magnetic flux can be shared to reduce the thickness of the substrate.

Fig. 9D shows a schematic diagram of a partial structure of a magnetic core assembly in another embodiment, fig. 9E shows a schematic diagram of an ac magnetic flux direction of the magnetic core assembly in fig. 9D, and fig. 9F shows another preferred embodiment. Referring to fig. 2A and 2B, two inductors L1 and L2 of two switching power conversion units are wound on the same magnetic core assembly, the magnetic core assembly includes two substrates, two winding posts and a side post, the two winding posts and the side post are disposed on at least one substrate and located between the two substrates, the magnetic core assembly is formed by connecting an upper magnetic core and a lower magnetic core, and the following magnetic cores are taken as examples to describe the structure and winding direction of the magnetic core. The lower core 4 includes two winding

posts

41 and 42, a side post 43 and a substrate 45, wherein the winding

posts

41 and 42 and the side post 43 are disposed on the substrate 45 and located at one side of the substrate 45, and taking the view angle of the front view 9D as an example, the winding post 41 is disposed at the front side of the winding post 42, and the side post 43 is located at the other side of the substrate 45. In some embodiments, the center of the winding post 41 is equidistant from the center of the winding post 42 and the vertical distance of the side posts. In one embodiment, the air gap may be formed only on the side pillar 43, or the air gaps may be formed on the side pillar 43 and the two winding

pillars

41 and 42, respectively, and the lengths of the air gaps on the two winding pillars are substantially the same, and the side pillar air gap is greater than or equal to the winding pillar air gap.

Taking the example where the windings of the two inductors L1 and L2 are wound around the two winding

legs

41 and 42, respectively, the dc current directions are the same as the dc magnetic fluxes on the winding

legs

41 and 42 as shown in fig. 9D. Referring to fig. 9C, one third of the alternating magnetic flux Φ ac1 flowing through the leg 41 flows into the leg 42, and two thirds of the alternating magnetic flux Φ ac1 flows into the side leg 43 as shown in fig. 9E. Although in this embodiment, the ac magnetic flux Φ ac1 flowing through the winding leg 41 cannot be equally distributed in both directions toward the side leg and the winding leg 42, the thickness of the substrate 45 may be reduced to some extent, thereby reducing the height of the power conversion system product. Other embodiments can refer to fig. 9A, and are not repeated herein.

In another embodiment, the side post 42 in fig. 9D can also be divided into two

side posts

43a and 43b, which are disposed on the same side of the two winding

posts

41 and 42, and the ac magnetic flux flowing through the winding post 41 flows to the side post 43b and the winding post 42, and the ac magnetic flux flowing through the winding post 42 also flows to the side post 43a and the winding post 41. Other features of this embodiment can refer to the embodiment corresponding to fig. 9D, and are not described herein again.

Fig. 10A is a partial structural view of a magnetic core assembly according to another preferred embodiment of the present disclosure, fig. 10B is a schematic view of an ac magnetic flux direction of the magnetic core assembly of fig. 10A, and fig. 10C is a waveform diagram of an ac magnetic flux in the magnetic core assembly of fig. 10A in one switching cycle. Please refer to fig. 10A and fig. 2A and 2B. Two inductances L1 and L2 of the two switching power conversion units are wound on the same magnetic core assembly, the magnetic core assembly comprises two substrates, four winding posts and auxiliary post units, the four winding posts and the auxiliary post units are arranged on at least one substrate and located between the two substrates, a winding of the inductance L1 is wound on the at least two winding posts, a winding of the inductance L2 is wound on the at least two winding posts, and a connecting line of central points of the four winding posts is quadrilateral. In this embodiment, the auxiliary pillar unit includes two central pillars (i.e., auxiliary pillars). The core assembly is formed by assembling an upper core and a lower core, and the following core 5 is an example to explain the core structure and the winding direction. The lower magnetic core 5 includes a first winding post 51, a second winding post 52, a third winding post 53, a fourth winding post 54, a first center pillar 55, a second center pillar 56 and a substrate 57, wherein the first winding post 51, the second winding post 52, the third winding post 53, the fourth winding post 54, the first center pillar 55 and the second center pillar 56 are disposed on the substrate 57, the first center pillar 55 is located between the first winding post 51 and the fourth winding post 54, and the second center pillar 56 is located between the second winding post 52 and the third winding post 53. The connecting lines of the center points of the first winding post 51, the second winding post 52, the third winding post 53 and the fourth winding post 54 are quadrilateral, the first winding post 51 and the third winding post 53 are located on a first diagonal of the quadrilateral, and the second winding post 52 and the fourth winding post 54 are located on a second diagonal of the quadrilateral.

Taking the example where the winding of the inductor L1 is wound around the first winding leg 51 and the third winding leg 53, and the winding of the inductor L2 is wound around the second winding leg 52 and the fourth winding leg 54, the direction of the dc current is as shown in fig. 10A, and the dc magnetic fluxes on the four winding legs (51, 52, 53, 54) are superimposed on the two center legs (55, 56). As shown in fig. 10B and 10C, the duty ratio D is equal to 50%, and the interval from time t0 to time t2 is a switching period Ts. In the interval from time t0 to time t1, the first switch M11 is turned on, and the alternating current magnetic flux Φ ac3 flowing through the first and

third spools

51 and 53 linearly increases, and the alternating current magnetic flux Φ ac4 flowing through the second and

fourth spools

52 and 54 linearly decreases. Taking time t1 as an example, one-half of the ac magnetic flux flowing through the first winding leg 51 flows into the second winding leg 52, the remaining one-half flows into the fourth winding leg 54, one-half of the ac magnetic flux flowing through the third winding leg 53 flows into the second winding leg 52, and the remaining one-half flows into the fourth winding leg 54. In the interval from time t1 to time t2, the alternating current magnetic flux Φ ac4 of the second and

fourth spools

52 and 54 linearly increases, and the alternating current magnetic flux Φ ac3 flowing through the first and

third spools

51 and 53 linearly decreases. Taking time t2 as an example, one-half of the ac magnetic flux flowing through the second winding leg 52 flows into the first winding leg 51, the remaining one-half flows into the third winding leg 53, one-half of the ac magnetic flux flowing through the fourth winding leg 54 flows into the first winding leg 51, and the remaining one-half flows into the third winding leg 53. As can be seen from the above, the ac magnetic flux flowing through each winding post (51, 52, 53, 54) respectively flows to two adjacent winding posts, so that the ac magnetic flux is shared, the thickness of the substrate 57 can be reduced, and the height of the power conversion system product can be reduced, thereby implementing ultra-thinning and improving the applicability thereof, and also greatly improving the power density of the power conversion system and reducing the vertical thermal resistance. Note that, in this embodiment of the core assembly, the method of equally distributing the ac magnetic fluxes is not limited to the case where the duty ratio D is 50%, and in the range where the duty ratio D is equal to or greater than 30% and equal to or less than 70%, the ac magnetic fluxes flowing out of the winding legs flow to two adjacent winding legs and one adjacent center leg, respectively, to form 3 closed magnetic flux circuits, but only when D is 50%, the ac magnetic flux equal-distributing effect is optimal when the duty ratio D is equal to 50%. In addition, the upper core of the core assembly may have the same structure as the lower core 5, or may be an I-type core, and the winding directions of the inductors L1 and L2 are not limited to the examples of the drawings, as long as the ac magnetic flux can be shared to reduce the thickness of the substrate.

In another exemplary embodiment with a duty cycle D of 50%, the two center pillars of fig. 10A may be combined into a center pillar 65, the cross section of which is rectangular, as shown in fig. 10D, the

pillars

61, 62, 63, 64 correspond to the

pillars

51, 52, 53, 54 of fig. 10A, respectively, and the center pillar 65 is located at the center of the combination of the

pillars

61 and 62 and the combination of the

pillars

63 and 64. In some embodiments, the winding direction is shown in fig. 10E, (fig. 10E is a top view of fig. 10D), a winding counterclockwise passes around the winding

posts

62 and 64 and the center post 65, and forms an inductor with the winding

posts

61 and 63; the other winding passes counterclockwise around the winding

legs

61 and 63 and the center leg 65, and forms another inductance with the winding

legs

62 and 64. In other embodiments, the winding direction is as shown in fig. 10F (fig. 10F is a top view of fig. 10D), and a winding of an inductor passes through the outer edge side of the winding post 64 after passing through the winding post 61 counterclockwise, and then is wound on the winding post 63 counterclockwise to form an inductor with the winding

posts

61 and 63; the winding of the other inductor passes through the outer edge side of the winding post 61 after passing around the winding post 62 counterclockwise, and then forms another inductor with the winding

posts

62 and 64 after being wound around the winding post 64 counterclockwise. Further, in some embodiments, the center pillar 65 having a rectangular cross section may be replaced with a center pillar 75 having a square cross section, as shown in fig. 10G. The center posts 65 may also be replaced by center posts 85 with a circular or oval cross section, as shown in fig. 10H, the winding direction thereof may refer to the winding manner shown in fig. 10E and 10F, and these embodiments may all realize the same direction of the dc magnetic flux on the winding posts, and when the typical duty ratio D is 50%, the ac magnetic flux flowing through any winding post averagely flows to two adjacent winding posts (refer to fig. 10B), so as to realize the even sharing of the ac magnetic flux, thereby reducing the thickness of the magnetic core substrates.

In another exemplary embodiment with a duty ratio D of 50%, the magnetic core structure of fig. 10I may also be implemented, where the lower magnetic core includes four winding pillars (91, 92, 93, 94), an auxiliary pillar unit and a substrate 97, and the auxiliary pillar unit includes two

side pillars

95 and 96, which are located at two sides of a quadrangle formed by the winding pillars. In this embodiment, the winding manner of the winding may be as shown in fig. 10A, and the dc magnetic fluxes on the winding legs are in the same direction, and when the typical duty ratio D is 50%, the ac magnetic flux flowing through any winding leg averagely flows to two adjacent winding legs (see fig. 10B), so as to achieve the uniform sharing of the ac magnetic flux, thereby reducing the thickness of the magnetic core substrates. In other embodiments, the side post 95 may be located at a side of a line connecting the winding post 91 and the winding post 94, and correspondingly, the side post 96 may be located at a side of a line connecting the winding post 92 and the winding post 97.

In the embodiments shown in fig. 10D, 10G, 10H, and 10I, the method of equally distributing the ac magnetic flux is not limited to the case where the duty ratio D is 50%, and the ac magnetic flux flowing from the winding posts flows to the two adjacent winding posts, the adjacent center posts, or the side posts to form the closed magnetic path in the range of 30% to 70%, but the ac magnetic flux is most effectively shared when the duty ratio D is 50%, and similarly, in the embodiments shown in fig. 10D, 10G, 10H, and 10I, when the ac magnetic flux is not shared, a part of the ac magnetic flux flows through the auxiliary posts. In this embodiment, the air gap may be formed only on the auxiliary column unit, and if the auxiliary column unit includes two auxiliary columns, the lengths of the air gaps on the two auxiliary columns are substantially the same; the auxiliary pole unit and the four winding poles may be respectively provided with air gaps, the lengths of the air gaps on the auxiliary pole unit are substantially the same, the lengths of the air gaps on the four winding poles are substantially the same, and the air gap on the auxiliary pole is greater than or equal to the air gap on the winding pole. The upper magnetic core may have the same structure as the lower magnetic core or an I-shaped magnetic core, and the winding direction of the inductor is not limited thereto, and it is within the scope of the present invention to reduce the thickness of the magnetic core substrate by sharing the ac magnetic flux. In the embodiments of the magnetic core assemblies, the illustration is only an example, and the winding posts, the side posts or the auxiliary posts may be disposed to be attached to the edge of the substrate, or may have a certain distance from the edge of the substrate. Not limited thereto.

Relative terms, such as "upper" or "lower," may be used in the above embodiments to describe one element's relative relationship to another element of an icon. It will be understood that if the device illustrated in the drawings is turned over with its top and bottom reversed, elements described as "top" will be termed "bottom".

The ripple frequency of the current on the input side of the power conversion circuit shown in fig. 2A and 2B is the switching frequency fs, and the ripple frequency is not increased and the amplitude of the ripple is not reduced because 2 switching power conversion units are cascaded. Therefore, a large-sized filter device needs to be placed at the input end of the circuit to filter out the current ripple, but the large-sized filter device increases the size of the power conversion system.

For high power applications, in one embodiment, the invention may use a plurality of power conversion circuits connected in parallel (interleaved) to increase the loading capacity of the power conversion system. The power conversion system includes X power conversion circuits, which are respectively a1 and a2 … … AX as shown in fig. 11A and fig. 11B and fig. 11C, or B1 and B2 … … BX as shown in fig. 11C, where X is an integer greater than 1. The input ends of the X power conversion circuits are all connected in parallel, and the output ends of the X power conversion circuits are all connected in parallel. The circuit structure of each power conversion circuit can refer to the above embodiments, and is not described herein again. In addition, the power conversion system further includes a controller 100.

Fig. 11A and 11B illustrate an embodiment in which a plurality of first power conversion circuits are connected in parallel alternately. In the embodiment shown in fig. 11A, X is an odd number, the controller 100 outputs X sets of control signals (PWM11, PWM12), (PWM21, PWM22) … … (PWMX1, PWMX2), and the control signals PWM11, PWM21 … … PWMX1 are sequentially staggered by 360/X. The control signals PWM11 and PWM12 are used to control the power conversion circuit a1, the control signals PWM21 and PWM22 are used to control the power conversion circuit a2, and so on, the control signals PWMX1 and PWMX2 are used to control the power conversion circuit AX, and the detailed control method thereof can refer to fig. 2A, fig. 3A and fig. 3B, which are not described herein again. In the embodiment shown in FIG. 11B, X is even, the controller 100 outputs Y sets of control signals (PWM11, PWM12), (PWM21, PWM22) … … (PWMY1, PWMY2), where Y is equal to X/2, and the control signals PWM11, PWM21 … … PWMY1 are sequentially staggered by 360/Y (i.e., 720/X). The control signals PWM11 and PWM12 are used to control the power conversion circuits a1 and a (Y +1), the control signals PWM21 and PWM22 are used to control the power conversion circuits a2 and a (Y +2), and so on, the control signals PWMY1 and PWMY2 are used to control the power conversion circuits AY and AX. Fig. 11C illustrates an embodiment of a plurality of second power conversion circuits connected in parallel. In the embodiment shown in fig. 11C, the controller 100 outputs X sets of control signals (PWM11, PWM12), (PWM21, PWM22) … … (PWMX1, PWMX2), and the control signals PWM11, PWM21 … … PWMX1 are sequentially out-of-phase 360/2X. In the embodiment of fig. 11C, the control signals PWM11 and PWM12 are used to control the power conversion circuit B1, the control signals PWM21 and PWM22 are used to control the power conversion circuit B2, and so on, the control signals PWMX1 and PWMX2 are used to control the power conversion circuit BX, so that the staggered parallel connection of the X power conversion circuits is realized by the phase error of the pulse control signal, and the input side current ripple of the power conversion system is reduced, and at the same time, a small-sized filter device can be used to reduce the size of the power conversion system.

Taking the power conversion system including two first power conversion circuits (i.e. X equals 2), the two first power conversion circuits are connected in parallel alternately. As shown in fig. 12, the two first power conversion circuits a1 and a2 are connected in parallel at the input end and the output end, respectively, and the input capacitors at the input end may be at least one input capacitor Cin as shown in fig. 12, or may share one set of input capacitors. The output capacitors at the output terminals may share one set of output capacitors Co as shown in fig. 12, or at least one output capacitor may be provided. The power conversion system further includes a controller 100 outputting a set of PWM pulse control signals PWM11 and PWM 12. In the 1 st power conversion circuit a1, two ends of the energy storage device Cba are respectively a first node SWAa and a second node SWBa, and the first switch M11a of the 1 st switching power conversion unit is controlled by the control signal PWM11, the control signal of the second switch M21a of the 1 st switching power conversion unit is complementary to the control signal PWM11, the first switch M12a of the 2 nd switching power conversion unit is controlled by the control signal PWM12, and the control signal of the second switch M22a of the 2 nd switching power conversion unit is complementary to the control signal PWM 35 12. In the 2 nd power conversion circuit a2, two ends of the energy storage device Cbb are respectively a first node SWAb and a second node SWBb, and the first switch M11b of the 1 st switching power conversion unit is controlled by the control signal PWM12, the control signal of the second switch M21b of the 1 st switching power conversion unit is complementary to the control signal PWM12, the first switch M12b of the 2 nd switching power conversion unit is controlled by the control signal PWM11, and the control signal of the second switch M22b of the 2 nd switching power conversion unit is complementary to the control signal PWM 11.

For the two interleaved parallel power conversion circuits a1 and a2 shown in fig. 12, the method of implementing the pre-charging circuit can be as shown in fig. 13A, and two identical pre-charging circuits are used to pre-charge two energy storage devices Cba, Cbb, respectively, and each pre-charging circuit can be as shown in fig. 5A or fig. 5B, and will not be described in detail here. Taking the power supply circuit as an example of a relaxation circuit, the winding T3a of each pre-charge circuit is coupled to the magnetic element T1 in the forward direction, the turn ratio of the winding T3a to the magnetic element T1 is 1:2, and the output of the pre-charge circuit is electrically connected to the first node SWAa and the second node SWBa to pre-charge the energy storage device Cba. Winding T3b is positively coupled to magnetic element T1 with a 1:2 turns ratio of winding T3b to magnetic element T1, and the output of the precharge circuit is electrically connected to first and second nodes SWAb and SWBb for precharging energy storage device Cbb. When the voltages at the two ends of the energy storage devices Cba and Cbb are charged to Vin/2, the two switching power conversion units can enter an output voltage soft start state.

Since the voltages at the two ends of the energy storage devices Cba and Cbb in the two power conversion circuits a1 and a2 are the same in the steady state, in another embodiment of the present invention, one pre-charge circuit may be used to pre-charge the energy storage devices Cba and Cbb in the two switching power conversion units. As shown in fig. 13B, a winding T3 in the pre-charge circuit is coupled to a magnetic element T1 of the power circuit 22 in a forward direction, and a turn ratio of the winding T3 to the magnetic element T1 is 1:2, a voltage VC3 across a capacitor C3 pre-charges the energy storage device Cba through diodes D71 and D72, a voltage VC3 across the capacitor C3 pre-charges the energy storage device Cbb through diodes D73 and D74, and when voltages across the energy storage devices Cba and Cbb are charged to Vin/2, the two power conversion circuits can enter an output voltage soft start state. The pre-charging circuit is simple in implementation method, the structure of the transformer is simplified, and pre-charging of the energy storage devices in the two power conversion circuits which are connected in parallel in a staggered mode can be achieved only by adding four diodes.

The bootstrap power supply circuit adopted by the interleaved parallel two power conversion circuits a1 and a2 is shown in fig. 14. In this embodiment, for two power conversion circuits, two bootstrap power supply circuits are used to respectively supply power to the driving circuits of the two power conversion circuits, and each bootstrap circuit can be as shown in fig. 8A or fig. 8C, and will not be described in detail here. In the power conversion circuit a1 and the bootstrap power circuit corresponding thereto, when the second switch M22a is turned on, Vcc1a is Vcc, and when the second switch M22a is turned off, the node voltage Vcc1a supplies power to the corresponding driving circuit to drive the first switch M12a to be turned on. When the first switch M12a is turned on, the nodes SWAa and SWCa are shorted, at this time, Vcc2a is Vcc1a, and when the first switch M12a is turned off, the node voltage Vcc2a supplies power to the corresponding driving circuit, driving the first switch M11a to be turned on. Similarly, the working principle of the power conversion circuit a2 and the bootstrap power circuit corresponding thereto can be derived, and will not be described herein again.

Taking the power conversion system including two second power conversion circuits (i.e. X equals to 2) as an example, the two second power conversion circuits are connected in parallel in an interleaved manner. As shown in fig. 15, in the 1 st power conversion circuit B1, the two ends of the energy storage device Cb10c are respectively a first node SWEc and a second node SWFc, and the two ends of the energy storage device Cb11c are respectively a first node SWGc and a second node SWHc; in the 2 nd power conversion circuit B2, two ends of the energy storage device Cb10d are respectively a first node SWEd and a second node SWFd, and two ends of the energy storage device Cb11d are respectively a first node SWGd and a second node SWHd.

For the two interleaved power conversion circuits B1 and B2 in parallel as shown in fig. 15, the precharge circuit thereof can be referred to as shown in fig. 16. Taking the power supply circuit of fig. 16 as an example of an anti-relaxation circuit, fig. 16 employs two identical pre-charge circuits, wherein the winding T3c is coupled to the magnetic element T1 in the forward direction, and the turn ratio of the winding T3c to the magnetic element T1 is 1:2, the output of the pre-charge circuit is electrically connected to two sets of isolation diodes D71c/D72c and D73c/D74c, wherein the output of the isolation diodes D71c/D72c is electrically connected to the 1 st node SWEc and the 2 nd node SWFc respectively, so as to pre-charge the energy storage device Cb10 c; the outputs of the other set of isolation diodes D73c/D74c are electrically connected to node 1 SWGc and node 2 SWHc, respectively, to precharge the energy storage device Cb11 c. The other winding T3D is coupled to the magnetic element T1 in the forward direction, the turn ratio of the winding T3D to the magnetic element T1 is 1:2, and the output of the pre-charging circuit is electrically connected with two sets of isolation diodes D71D/D72D and D73D/D74D to pre-charge the energy storage device Cb 10D; the outputs of one set of isolation diodes D71D/D72D are electrically connected to the 1 st node SWEd and the 2 nd node SWFd, respectively, and the outputs of the other set of isolation diodes D73D/D74D are electrically connected to the 1 st node SWGd and the 2 nd node SWHd, respectively, to precharge the energy storage device Cb 11D. After the voltages at the two terminals of the energy storage devices Cb10c, Cb11c, Cb10d and Cb11d are charged to Vin/2, the two switching power conversion units can enter a soft start state.

For the two interleaved power conversion circuits B1 and B2 shown in fig. 15, in one embodiment, the precharge circuit comprises 4 precharge circuits shown in fig. 5C, i.e., four windings are coupled to the magnetic element in the forward direction respectively, the turns ratios of the four windings to the magnetic element are all 1:2, and the outputs of the four precharge circuits are respectively precharged by four energy storage devices. In another embodiment, the precharge circuit can be implemented with reference to the precharge circuit architecture shown in fig. 5D, with the difference that: the power supply switching circuit comprises four groups of isolation diodes, namely, one winding is positively coupled to a magnetic element, the turn ratio of the winding to the magnetic element is 1:2, the output of the isolation diodes is electrically connected with the four groups of isolation diodes, the output of the isolation diodes in the four groups of isolation diodes is used for pre-charging four energy storage devices respectively, and when the voltage at two ends of the energy storage devices is charged to Vin/2, the two power supply switching circuits can enter an output voltage soft start state. The pre-charging circuit can be flexibly matched through the isolation diodes with different winding numbers and different group numbers, and can meet the pre-charging requirement of the energy storage device.

For the case that the power conversion system includes X second power conversion circuits B1-BX connected in parallel in an interleaved manner, the bootstrap power supply circuit of each power conversion circuit may adopt the bootstrap power supply circuit shown in fig. 8C, and details thereof are not repeated herein.

In another embodiment, in order to satisfy the requirement of large current output, the precharge circuit and the bootstrap power supply circuit may be implemented according to the precharge circuit of fig. 13A or fig. 13B and the bootstrap power supply circuit of fig. 14, and are not described herein again. In addition, when the number of the power conversion circuits is larger or even odd, the corresponding precharge circuit and the bootstrap power circuit can be derived according to the foregoing embodiments, and will not be described herein again. In another embodiment, the diodes in the precharge circuits, the clamp circuits and the bootstrap circuits in fig. 13A, 13B and 14 can be replaced by controllable switches.

In summary, the present disclosure provides a power conversion system, in which a power conversion circuit of the power conversion system is a multi-phase buck converter with an expandable duty ratio, and compared with a conventional buck converter, the power conversion system can effectively increase the duty ratio under the same input/output condition, and accordingly reduce the corresponding jump voltage variation of a switch during on or off periods, thereby reducing the switching loss and increasing the efficiency. In addition, the energy storage device in the power conversion circuit is precharged by the precharge circuit, so that the power conversion circuit enters a soft start state to reduce the voltage stress borne by a switch in the power conversion circuit, and therefore, a low-voltage-resistant switch device can be selected, the cost is effectively reduced, the low-voltage-resistant switch device has low on-resistance, the power conversion efficiency can be improved, and the loss is reduced. In addition, clamping circuits are arranged at two ends of a switch in the power conversion circuit to clamp peak voltages at two ends of the switch, so that the switch is protected, peak energy can be absorbed and fed back, loss of the peak energy is reduced, and efficiency of the power conversion circuit is improved. Moreover, the bootstrap power supply circuit and the driving circuit are arranged to realize the bootstrap power supply function and correspondingly control the switch, so that the applicability of the power conversion circuit is greatly improved, the miniaturization of a power conversion system product is facilitated, and the bootstrap power supply circuit is simple in structure and low in cost. Furthermore, through the special structure of the magnetic core component in the power conversion system, the alternating current magnetic flux flowing through the winding posts can be uniformly shared, and further the thickness of the substrate of the magnetic core component is reduced, so that the height of the power conversion system product is reduced, the ultra-thinning is realized, the applicability is improved, and meanwhile, the power density of the power conversion system is greatly improved, and the vertical thermal resistance is reduced. In addition, for high-power application, the invention can adopt a mode of connecting a plurality of power conversion circuits in parallel in a staggered mode to increase the loading capacity of the power conversion system, and utilizes the staggered phase among control signals to realize the staggered parallel connection of the plurality of power conversion circuits and reduce the input side current ripple of the power conversion system, and simultaneously can adopt a small-size filter device to reduce the size of the power conversion system.

It should be noted that the above-mentioned embodiments illustrate only preferred embodiments of the disclosure, and the disclosure is not limited to the described embodiments, as the scope of the disclosure is determined by the claims. And that this disclosure will be modified by those skilled in the art as deemed to be within the scope and spirit of the appended claims.

Claims

1. A power conversion system, comprising:

x power conversion circuits, wherein X is an integer greater than or equal to 1, each of the power conversion circuits comprising:

an input terminal for receiving an input voltage;

an output terminal for outputting an output voltage;

n cascaded switching power conversion units, wherein each switching power conversion unit comprises a first switch and a second switch, the second switch is coupled with the first switch in series, the second switch is grounded, the first switch of the 1 st switching power conversion unit is coupled with the input end, the first switches of the rest switching power conversion units are sequentially coupled with the first switch of the previous switching power conversion unit in series, and N is a positive integer greater than or equal to 2; and

n nodes, wherein the nth node is located between the first switch of the nth switching power converting unit and the first switch of the (N +1) th switching power converting unit, the nth node is located between the first switch and the second switch of the nth switching power converting unit, and N is a positive integer greater than or equal to 1 and less than N;

x bootstrap power supply circuits respectively electrically connected to the X power conversion circuits, wherein in any one of the bootstrap power supply circuits and the corresponding power conversion circuit thereof, the bootstrap power supply circuit includes N bootstrap capacitors and N bootstrap switches, the N bootstrap switches are sequentially coupled in series, a negative terminal of an nth bootstrap capacitor is electrically connected to an nth node, a positive terminal of an nth bootstrap capacitor is electrically connected to a second terminal of an nth bootstrap switch, and a first terminal of an nth bootstrap switch receives a supply voltage; and

the bootstrap power supply circuit comprises at least N drive circuits, wherein each bootstrap power supply circuit is electrically connected with the corresponding N drive circuits, in any bootstrap power supply circuit and the corresponding N drive circuits thereof, the drive circuit is connected with the corresponding bootstrap capacitor, and respectively supplies power for the drive of the corresponding first switch and the second switch according to the voltage of the positive terminal of the bootstrap capacitor and the power supply voltage, and outputs a first drive signal and a second drive signal to control the corresponding first switch and the second switch.

2. The power conversion system of claim 1, wherein the number of the at least N driving circuits is X X N, and each bootstrap power supply circuit is electrically connected to the corresponding N driving circuits.

3. The power conversion system according to claim 1, wherein in any one of the bootstrap power supply circuits and the power conversion circuit corresponding thereto, a voltage of a positive terminal of an nth bootstrap capacitor is used for supplying power for driving the first switch of an nth switching power conversion unit, and a voltage of a positive terminal of an nth bootstrap capacitor is used for supplying power for driving the first switch of an nth switching power conversion unit.

4. The power conversion system of claims 1-3, further comprising a controller, wherein the controller outputs at least one set of control signals to the at least N driving circuits, each set of control signals comprising two control signals, the two control signals being sequentially 180 degrees out of phase.

5. The power conversion system of claim 4, wherein in any of the power conversion circuits, an odd number of the switching power conversion units are controlled by one of the control signals, and an even number of the switching power conversion units are controlled by the other control signal.

6. The power conversion system of claim 4, wherein when X is greater than 1 and is odd, the controller outputs X sets of control signals, and corresponding ones of the X sets of control signals are sequentially out of phase by 360/X.

7. The power conversion system of claim 4, wherein when X is greater than 1 and is even, the controller outputs X/2 sets of control signals, and the corresponding control signals in the X/2 sets of control signals are sequentially out of phase 720/X.

8. The power conversion system as claimed in claim 1, wherein in any one of the power conversion circuit and its corresponding bootstrap power supply circuit and N of the driving circuits, an input terminal of an nth driving circuit is electrically connected to a second terminal of an nth bootstrap switch and an nth node, an output terminal of the nth driving circuit outputs the first driving signal to correspondingly control the first switch of the nth switching power conversion unit, and a high level and a low level of the first driving signal are a positive terminal voltage of an nth bootstrap capacitor and a voltage of an nth node, respectively.

9. The power conversion system according to claim 8, wherein in any one of the power conversion circuit and its corresponding bootstrap power supply circuit and N of the driving circuits, an input terminal of an nth driving circuit is further electrically connected to the power supply voltage and a ground terminal, an output terminal of the nth driving circuit further outputs the second driving signal to correspondingly control the second switch of the nth switching power conversion unit, and a high level and a low level of the second driving signal are voltages of the power supply voltage and the ground terminal, respectively.

10. The power conversion system as claimed in claim 1, wherein in any one of the power conversion circuit and its corresponding bootstrap power supply circuit and N of the driving circuits, an input terminal of an nth of the driving circuits is electrically connected to a second terminal of an nth of the bootstrap switches and an nth of the nodes, an output terminal of an nth of the driving circuits outputs the first driving signal to correspondingly control the first switch of an nth of the switching power conversion units, and a high level and a low level of the first driving signal are a positive terminal voltage of an nth of the bootstrap capacitors and a voltage of an nth of the nodes, respectively.

11. The power conversion system according to claim 10, wherein in any of the power conversion circuits and its corresponding bootstrap power supply circuit and N of the driving circuits, an input terminal of an nth of the driving circuits is further electrically connected to the power supply voltage and a ground terminal, an output terminal of an nth of the driving circuits further outputs the second driving signal to correspondingly control the second switch of an nth of the switching power conversion units, and a high level and a low level of the second driving signal are voltages of the power supply voltage and the ground terminal, respectively.

12. The power conversion system of claim 1, wherein in any of the power conversion circuits, the N switching power conversion units further comprise N energy storage devices and N inductors, wherein the nth inductor is connected in series between the nth energy storage device and the output terminal, and the nth inductor is connected in series between the nth node and the output terminal.

13. A power conversion system, comprising:

a power conversion circuit, comprising:

an input terminal for receiving an input voltage;

an output terminal for outputting an output voltage; and

at least one switch bridge arm, wherein each switch bridge arm comprises an upper switch, a middle switch, a lower switch and two nodes, the upper switch, the middle switch and the lower switch are sequentially connected in series between the positive end and the negative end of the input end, the 1 st node is positioned between the upper switch and the middle switch, and the 2 nd node is positioned between the middle switch and the lower switch;

a bootstrap power supply circuit, comprising two bootstrap capacitors and two bootstrap switches, wherein the first end of the 2 nd bootstrap switch receives a power supply voltage, the second end of the 2 nd bootstrap switch is electrically connected to the first end of the 1 st bootstrap switch, the second end of the 1 st bootstrap switch is electrically connected to the positive terminal of the 1 st bootstrap capacitor, the negative terminal of the 1 st bootstrap capacitor is electrically connected to the 1 st node, the negative terminal of the 2 nd bootstrap capacitor is electrically connected to the 2 nd node, and the positive terminal of the 2 nd bootstrap capacitor is electrically connected to the second end of the 2 nd bootstrap switch; and

and the driving circuit unit is electrically connected with the bootstrap power supply circuit and used for outputting a driving signal to control the switch in the at least one switch bridge arm according to the voltages of the two nodes, the power supply voltage and at least one control signal.

14. The power conversion system of claim 13, wherein the lower switch and the middle switch are not in a conducting state at the same time, and the middle switch and the upper switch are not in a conducting state at the same time.

15. The power conversion system of claim 13, wherein the at least one control signal comprises control signals of the upper switch, the middle switch, and the lower switch, a voltage of a positive terminal of a1 st bootstrap capacitor is used for supplying power to the control signal of the upper switch, a voltage of a positive terminal of a2 nd bootstrap capacitor is used for supplying power to the control signal of the middle switch, and the supply voltage is used for supplying power to the control signal of the lower switch.

16. The power conversion system of claim 15, wherein the driving circuit unit comprises a first driving circuit, an input terminal of the first driving circuit is electrically connected to the second terminal of the 1 st bootstrap switch and the 1 st node and receives a control signal of the upper switch, an output terminal of the first driving circuit outputs a driving signal of the upper switch to correspondingly control the on/off of the upper switch, wherein a high level and a low level of the driving signal of the upper switch are respectively a positive terminal voltage of the 1 st bootstrap capacitor and a voltage of the 1 st node, and a control timing of the driving signal of the upper switch and the control signal of the upper switch is the same.

17. The power conversion system of claim 16, wherein the driving circuit unit further comprises a second driving circuit, an input terminal of the second driving circuit is electrically connected to the second terminal of the 2 nd bootstrap switch and the 2 nd node and receives a control signal of the middle switch, and an output terminal of the second driving circuit outputs a driving signal of the middle switch to correspondingly control the on/off of the middle switch, wherein a high level and a low level of the driving signal of the middle switch are respectively a positive terminal voltage of the 2 nd bootstrap capacitor and a voltage of the 2 nd node, and a control timing of the driving signal of the middle switch and the control signal of the middle switch is the same.

18. The power conversion system according to claim 17, wherein the input terminal of the second driving circuit is further electrically connected to the power supply voltage and a ground terminal and receives the control signal of the lower switch, the output terminal of the second driving circuit further outputs the driving signal of the lower switch to correspondingly control the on/off of the lower switch, and the high level and the low level of the driving signal of the lower switch are the voltage of the power supply voltage and the voltage of the ground terminal, respectively.

19. The power conversion system of claim 18, wherein the driving signal of the upper switch and the driving signal of the middle switch are out of phase by 180 degrees, and the lower switch and the middle switch are complementarily turned on.

20. The power conversion system of claim 13, wherein the power conversion circuit comprises two switching legs electrically connected in parallel, and the power conversion circuit further comprises two energy storage devices, wherein the 1 st energy storage device is electrically connected between the 1 st node of the 1 st switching leg and the 2 nd node of the 2 nd switching leg, and the 2 nd energy storage device is electrically connected between the 1 st node of the 2 nd switching leg and the 2 nd node of the 1 st switching leg.